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INA301
SBOS713A – SEPTEMBER 2015 – REVISED FEBRUARY 2016
INA301 36-V, High-Speed, Zero-Drift, Voltage-Output,
Current-Shunt Monitor with High-Speed, Overcurrent Comparator
1 Features
3 Description
•
•
•
The INA301 includes both a high common-mode,
current-sensing
amplifier
and
a
high-speed
comparator configured to detect overcurrent
conditions through measuring the voltage developed
across a current-sensing or current-shunt resistor and
comparing that voltage to a defined threshold limit.
The device features an adjustable limit threshold
range that is set using a single external limit-setting
resistor. This current-shunt monitor can measure
differential voltage signals on common-mode voltages
that can vary from 0 V up to 36 V, independent of the
supply voltage.
1
•
•
•
•
•
Wide Common-Mode Input Range: 0 V to 36 V
Dual Output: Amplifier and Comparator Output
High Accuracy Amplifier:
– Offset Voltage: 35 µV (Max)
– Offset Voltage Drift: 0.5 µV/°C (Max)
– Gain Error: 0.1% (Max)
– Gain Error Drift: 10 ppm/°C
Available Amplifier Gains:
– INA301A1: 20 V/V
– INA301A2: 50 V/V
– INA301A3: 100 V/V
Programmable Alert Threshold Set Through a
Single Resistor
Total Alert Response Time: 1 µs
Open-Drain Output With Latching Mode
Package: VSSOP-8
2 Applications
•
•
•
•
•
•
Overcurrent Protection
Power-Supply Protection
Circuit Breakers
Computers and Servers
Telecom Equipment
Battery Management
The open-drain alert output can be configured to
operate in either a transparent mode where the
output status follows the input state or in a latched
mode where the alert output is cleared when the latch
is reset. The device alert response time is under 1 µs,
allowing for quick detection of overcurrent events.
This device operates from a single 2.7-V to 5.5-V
supply, drawing a maximum supply current of 700 µA.
The device is specified over the extended operating
temperature range (–40°C to +125°C), and is
available in an 8-pin VSSOP package.
Device Information
PART NUMBER
INA301
PACKAGE
VSSOP (8)
BODY SIZE
3.00 mm × 3.00 mm
Typical Application
2.7 V to 5.5 V
CBYPASS
0.1 PF
RPULL-UP
10 k
Supply
(0 V to 36 V)
VS
IN+
+
INA301
Microcontroller
OUT
ADC
ALERT
INLoad
GPIO
RESET
GND
GPIO
LIMIT
DAC
RLIMIT
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
INA301
SBOS713A – SEPTEMBER 2015 – REVISED FEBRUARY 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 13
7.1 Overview ................................................................. 13
7.2 Functional Block Diagram ....................................... 13
7.3 Feature Description................................................. 14
7.4 Device Functional Modes........................................ 19
8
Applications and Implementation ...................... 21
8.1 Application Information .......................................... 21
8.2 Typical Application .................................................. 21
9 Power Supply Recommendations...................... 23
10 Layout................................................................... 23
10.1 Layout Guidelines ................................................. 23
10.2 Layout Example .................................................... 23
11 Device and Documentation Support ................. 24
11.1
11.2
11.3
11.4
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
24
24
24
24
12 Mechanical, Packaging, and Orderable
Information ........................................................... 24
4 Revision History
Changes from Original (September 2015) to Revision A
•
2
Page
Released to production .......................................................................................................................................................... 1
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SBOS713A – SEPTEMBER 2015 – REVISED FEBRUARY 2016
5 Pin Configuration and Functions
DGK Package
8-Pin VSSOP
Top View
VS
1
8
IN+
OUT
2
7
IN-
LIMIT
3
6
ALERT
GND
4
5
RESET
Pin Functions
PIN
NO.
NAME
I/O
DESCRIPTION
1
VS
Analog
2
OUT
Analog output
Power supply, 2.7 V to 5.5 V
3
LIMIT
Analog input
4
GND
Analog
5
RESET
Digital input
6
ALERT
Digital output
Overlimit alert, active-low, open-drain output
7
IN–
Analog input
Connect to load side of the shunt resistor
8
IN+
Analog input
Connect to supply side of the shunt resistor
Output voltage
Alert threshold limit input; see the Setting The Current-Limit Threshold section for
details on setting the limit threshold
Ground
Transparent or latch mode selection input
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SBOS713A – SEPTEMBER 2015 – REVISED FEBRUARY 2016
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
Supply voltage, VS
Differential (VIN+) – (VIN–)
(2)
MAX
UNIT
6
V
–40
40
Common-mode (3)
GND – 0.3
40
Analog input
LIMIT pin
GND – 0.3
(VS) + 0.3
V
Analog output
OUT pin
GND – 0.3
(VS) + 0.3
V
Digital input
RESET pin
GND – 0.3
(VS) + 0.3
V
Digital output
ALERT pin
GND – 0.3
6
V
150
°C
150
°C
Analog inputs (IN+, IN–)
Junction temperature, TJ
Storage temperature, Tstg
(1)
(2)
(3)
–65
V
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
VIN+ and VIN– are the voltages at the IN+ and IN– pins, respectively.
Input voltage can exceed the voltage shown without causing damage to the device if the current at that pin is limited to 5 mA.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±1000
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
VCM
Common-mode input voltage
VS
Operating supply voltage
TA
Operating free-air temperature
NOM
MAX
UNIT
12
V
5
V
–40
125
°C
6.4 Thermal Information
INA301
THERMAL METRIC (1)
DGK (MSOP)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
161.5
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
62.3
°C/W
RθJB
Junction-to-board thermal resistance
81.4
°C/W
ψJT
Junction-to-top characterization parameter
6.8
°C/W
ψJB
Junction-to-board characterization parameter
80
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
N/A
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
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SBOS713A – SEPTEMBER 2015 – REVISED FEBRUARY 2016
6.5 Electrical Characteristics
at TA = 25°C, VSENSE = VIN+ – VIN– = 10 mV, VS = 5 V, VIN+ = 12 V, and VLIMIT = 2 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
INPUT
VCM
Common-mode input voltage range
VIN
Differential input voltage range
CMR
Common-mode rejection
0
36
VIN = VIN+ – VIN–, INA301A1
0
250
VIN = VIN+ – VIN–, INA301A2
0
100
VIN = VIN+ – VIN–, INA301A3
0
50
INA301A1, VIN+ = 0 V to 36 V,
TA = –40ºC to +125ºC
100
110
INA301A2, VIN+ = 0 V to 36 V,
TA = –40ºC to +125ºC
106
118
INA301A3, VIN+ = 0 V to 36 V,
TA = –40ºC to +125ºC
110
120
V
mV
dB
INA301A1
±25
±125
INA301A2
±15
±50
INA301A3
±10
±35
TA= –40ºC to +125ºC
0.1
0.5
µV/°C
±0.1
±10
µV/V
VOS
Offset voltage, RTI (1)
dVOS/dT
Offset voltage drift, RTI (1)
PSRR
Power-supply rejection ratio
VS = 2.7 V to 5.5 V, VIN+ = 12 V,
TA = –40ºC to +125ºC
IB
Input bias current
IB+, IB–
120
µA
IOS
Input offset current
VSENSE = 0 mV
±0.1
µA
µV
OUTPUT
INA301A1
G
Gain
Gain error
20
INA301A2
50
INA301A3
100
V/V
INA301A1, VOUT = 0.5 V to VS – 0.5 V
±0.03%
±0.1%
INA301A2, VOUT = 0.5 V to VS – 0.5 V
±0.05%
±0.15%
INA301A3, VOUT = 0.5 V to VS – 0.5 V
±0.11%
±0.2%
3
10
TA= –40ºC to 125ºC
Nonlinearity error
VOUT = 0.5 V to VS – 0.5 V
Maximum capacitive load
No sustained oscillation
ppm/°C
±0.01%
500
pF
VOLTAGE OUTPUT
Swing to VS power-supply rail
RL = 10 kΩ to GND,
TA = –40ºC to +125ºC
VS – 0.05
VS – 0.1
Swing to GND
RL = 10 kΩ to GND,
TA = –40ºC to +125ºC
VGND + 20
VGND + 30
V
mV
FREQUENCY RESPONSE
BW
Bandwidth
SR
INA301A1
550
INA301A2
500
INA301A3
450
Slew rate
kHz
4
V/µs
30
nV/√Hz
NOISE, RTI (1)
Voltage noise density
(1)
RTI = referred-to-input.
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SBOS713A – SEPTEMBER 2015 – REVISED FEBRUARY 2016
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Electrical Characteristics (continued)
at TA = 25°C, VSENSE = VIN+ – VIN– = 10 mV, VS = 5 V, VIN+ = 12 V, and VLIMIT = 2 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
COMPARATOR
tp
Total alert propagation delay
Input overdrive = 1 mV
Slew-rate-limited tp
VOUT step = 0.5 V to 4.5 V, VLIMIT = 4 V
ILIMIT
Limit threshold output current
VOS
Comparator offset voltage
HYS
Hysteresis
TA = 25ºC
79.7
TA = –40ºC to +125ºC
79.2
0.75
1
1
1.5
80
80.3
80.8
INA301A1
1
INA301A2
1
4
INA301A3
1.5
4.5
INA301A1
20
INA301A2
50
INA301A3
100
µs
µA
3.5
mV
mV
VIH
High-level input voltage
1.4
6
VIL
Low-level input voltage
0
0.4
V
V
VOL
Alert low-level output voltage
IOL = 3 mA
70
300
mV
ALERT pin leakage input current
VOH = 3.3 V
0.1
1
µA
Digital leakage input current
0 ≤ VIN ≤ VS
1
µA
POWER SUPPLY
VS
IQ
Operating supply range
Quiescent current
TA = –40ºC to +125ºC
2.7
VSENSE = 0 mV, TA = 25ºC
5.5
500
TA = –40ºC to +125ºC
650
700
V
µA
TEMPERATURE RANGE
Specified range
6
–40
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125
°C
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SBOS713A – SEPTEMBER 2015 – REVISED FEBRUARY 2016
6.6 Typical Characteristics
Input Offset Voltage (PV)
100
80
60
40
20
0
-20
-40
-60
-100
-80
Population
100
80
60
40
20
0
-20
-40
-60
-80
-100
Population
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
Input Offset Voltage (PV)
Figure 1. Input Offset Voltage Distribution (INA301A1)
Figure 2. Input Offset Voltage Distribution (INA301A2)
60
INA301A1
INA301A2
INA301A3
Population
Offset Voltage (PV)
40
20
100
80
60
40
20
0
-20
-40
-60
-80
-100
0
-20
-50
-25
0
25
50
75
Temperature (qC)
100
125
150
Input Offset Voltage (PV)
Figure 4. Input Offset Voltage vs Temperature
5
4
3
2
1
0
-1
-2
-3
-4
-5
10
8
6
4
2
0
-2
-4
-6
-8
-10
Population
Population
Figure 3. Input Offset Voltage Distribution (INA301A3)
CMRR (PV/V)
CMRR (PV/V)
Figure 5. Common-Mode Rejection Ratio Distribution
(INA301A1)
Figure 6. Common-Mode Rejection Ratio Distribution
(INA301A2)
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Typical Characteristics (continued)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
3
2.5
Population
CMRR (Pv/v)
2
INA301A1
INA301A2
INA301A3
1.5
1
0.5
0
-1
-50
3
2.5
2
1.5
1
0
0.5
-1
-0.5
-1.5
-2
-3
-2.5
-0.5
-25
0
25
50
75
Temperature (qC)
100
125
150
CMRR (PV/V)
Figure 7. Common-Mode Rejection Ratio Distribution
(INA301A3)
Figure 8. Common-Mode Rejection Ratio vs Temperature
140
INA301A1
INA301A2
INA301A3
Population
CMRR (dB)
120
100
0.1
0.08
0.06
0.04
0.02
0
1M
-0.02
100k
-0.04
1k
10k
Frequency (Hz)
-0.06
100
-0.1
60
10
-0.08
80
Gain Error (%)
Figure 10. Gain Error Distribution (INA301A1)
Gain Error (%)
0.2
0.16
0.12
0.08
0.04
0
-0.04
-0.08
-0.12
Gain Error (%)
Figure 11. Gain Error Distribution (INA301A2)
8
-0.16
-0.2
Population
0.1
0.08
0.06
0.04
0.02
0
-0.02
-0.04
-0.06
-0.08
-0.1
Population
Figure 9. Common-Mode Rejection Ratio vs Frequency
Figure 12. Gain Error Distribution (INA301A3)
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Typical Characteristics (continued)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
0.5
50
INA301A1
INA301A2
INA301A3
0.4
0.3
40
30
0.1
Gain (dB)
Gain Error (%)
0.2
0
-0.1
-0.2
20
10
0
-0.3
INA301A1
INA301A2
INA301A3
-10
-0.4
-0.5
-50
-20
-25
0
25
50
75
Temperature (qC)
100
125
150
1
10
Figure 13. Gain Error vs Temperature
1k
10k
Frequency (Hz)
100k
1M
10M
Figure 14. Gain vs Frequency
VS
140
Output Voltage Swing (V)
120
100
PSRR (dB)
100
80
60
VS - 1
VS - 2
GND + 3
GND + 2
125ºC
25ºC
-40ºC
GND + 1
40
GND
20
1
10
100
1k
10k
Frequency (Hz)
100k
1M
0
10M
4
6
8
10
12
14
Output Current (mA)
Figure 16. Output Voltage Swing vs Output Current
Figure 15. Power-Supply Rejection Ratio vs Frequency
150
250
200
120
Input Bias Current (PA)
Input Bias Current (PA)
2
150
100
50
90
60
30
0
-50
0
0
5
10
15
20
25
30
Common-Mode Voltage (V)
35
40
Figure 17. Input Bias Current vs Common-Mode Voltage
(VS = 5 V)
0
5
10
15
20
25
30
Common-Mode Voltage (V)
35
40
Figure 18. Input Bias Current vs Common-Mode Voltage
(VS = 0 V)
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Typical Characteristics (continued)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
145
600
140
Quiescent Current (PA)
Input Bias Current (PA)
550
135
130
125
120
115
110
500
450
400
350
105
100
-50
-25
0
25
50
75
Temperature (qC)
100
125
300
2.7
150
Figure 19. Input Bias Current vs Temperature
3.6
3.9 4.2 4.5 4.8
Supply Voltage (V)
5.1
5.4
5.7
Input-Referred Voltage Noise (nV/—Hz)
35
520
Quiescent Current (PA)
3.3
Figure 20. Quiescent Current vs Supply Voltage
540
500
480
460
440
420
-50
3
30
25
20
15
10
INA301A1
INA301A2
INA301A3
5
0
-25
0
25
50
75
Temperature (qC)
100
125
150
1
100
1k
10k
Frequency (Hz)
1M
Input
Output
Output (1 V/div)
Referred-to-Input Voltage Noise
(200 nV/div)
100k
Figure 22. Input-Referred Voltage Noise vs Frequency
Input (200 mV/div)
Figure 21. Quiescent Current vs Temperature
10
Time (1 s/div)
Time (1 Ps/div)
Figure 23. 0.1-Hz to 10-Hz Voltage Noise
(Referred-to-Input)
10
Figure 24. Voltage Output Rising Step Response
(4-VPP Output Step)
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Typical Characteristics (continued)
VOUT (60 mV/div)
Output (1 V/div)
Input
Output
Common-Mode Voltage (10 V/div)
Input (200 mV/div)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
VCM
VOUT
Time (1 Ps/div)
Time (2 Ps/div)
Figure 25. Voltage Output Falling Step Response
(4-VPP Output Step)
Figure 26. Common-Mode Voltage Transient Response
80.8
Voltage (2 V/div)
Limit Current Source (PA)
80.6
VSUPPLY
VOUT
80.4
80.2
80
79.8
79.6
79.4
79.2
-50
Time (5 Ps/div)
0
25
50
75
Temperature (qC)
100
125
150
Figure 28. Limit Current Source vs Temperature
VIN * 20 V/V
Alert
VLIMIT
Voltage (0.5 V/div)
Figure 27. Start-Up Response
Voltage (0.5 V/div)
-25
VIN * 50 V/V
Alert
VLIMIT
Time (200 ns/div)
Time (200 ns/div)
Figure 29. Total Propagation Delay (INA301A1)
Figure 30. Total Propagation Delay (INA301A2)
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Typical Characteristics (continued)
at TA = 25°C, VS = 5 V, VIN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)
1,000
Propagation Delay (ns)
Voltage (0.5 V/div)
800
VIN * 100 V/V
Alert
VLIMIT
600
400
200
0
-50
Time (200 ns/div)
0
25
50
75
Temperature (qC)
100
125
150
Figure 32. Comparator Propagation Delay vs Temperature
(VOD = 1 mV)
Figure 31. Total Propagation Delay (INA301A3)
120
120
100
100
80
80
Hysteresis (mV)
Low-Level Output Voltage (mV)
-25
60
40
INA301A1
INA301A2
INA301A3
60
40
20
20
0
-50
0
0
0.5
1
1.5
2
2.5
3
3.5
4
Low-Level Output Current (mA)
4.5
5
Figure 33. Comparator Alert VOL vs IOL
-25
0
25
50
75
Temperature (qC)
100
125
150
Figure 34. Hysteresis vs Temperature
Voltage (2 V/div)
Reset
Alert
Time (2 Ps/div)
Figure 35. Comparator Reset Response
12
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7 Detailed Description
7.1 Overview
The INA301 is a 36-V common-mode, zero-drift topology, current-sensing amplifier that can be used in both lowside and high-side configurations. These specially-designed, current-sensing amplifiers are able to accurately
measure voltages developed across current-sensing resistors (also known as current-shunt resistors) on
common-mode voltages that far exceed the supply voltage powering the device. Current can be measured on
input voltage rails as high as 36 V, and the device can be powered from supply voltages as low as 2.7 V. The
device can also withstand the full 36-V common-mode voltage at the input pins when the supply voltage is
removed without causing damage.
The zero-drift topology enables high-precision measurements with maximum input offset voltages as low as
35 μV with a temperature contribution of only 0.5 μV/°C over the full temperature range of –40°C to +125°C. The
low total offset voltage of the INA301 enables smaller current-sense resistor values to be used, and allows for a
more efficient system operation without sacrificing measurement accuracy resulting from the smaller input signal.
The INA301 uses a single external resistor to allow for a simple method of setting the corresponding current
threshold level for the device to use for out-of-range comparison. Combining the precision measurement of the
current-sense amplifier and the on-board comparator enables an all-in-one overcurrent detection device. This
combination creates a highly-accurate solution that is capable of fast detection of out-of-range conditions and
allows the system to take corrective actions to prevent potential component or system-wide damage.
7.2 Functional Block Diagram
2.7 V to 5.5 V
CBYPASS
0.1 PF
Power Supply
(0 V to 36 V)
VS
IN+
INA301
+
RPULL-UP
10k
OUT
Gain = 20, 50,
100
INLoad
ALERT
+
RESET
LIMIT
GND
RSET
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7.3 Feature Description
7.3.1 Alert Output
The device ALERT pin is an active-low, open-drain output that is designed to be pulled low when the input
conditions are detected to be out-of-range. This open-drain output pin is recommended to include a 10-kΩ, pullup
resistor to the supply voltage. This open-drain pin can be pulled up to a voltage beyond the supply voltage, VS,
but must not exceed 5.5 V.
Figure 36 shows the alert output response of the internal comparator. When the output voltage of the amplifier is
lower than the voltage developed at the LIMIT pin, the comparator output is in the default high state. When the
amplifier output voltage exceeds the threshold voltage set at the LIMIT pin, the comparator output becomes
active and pulls low. This active low output indicates that the measured signal at the amplifier input has
exceeded the programmed threshold level, indicating an overcurrent or out-of-range condition has occurred.
6
VOUT
VLIMIT
ALERT
5
Voltage (V)
4
3
2
1
0
±1
Time (5 ms/div)
C001
Figure 36. Overcurrent Alert Response
7.3.2 Alert Mode
The device has two output operating modes, transparent and latched, that are selected based on the RESET pin
setting. These modes change how the ALERT pin responds following an alert when the overcurrent condition is
removed.
7.3.2.1 Transparent Output Mode
The device is set to transparent mode when the RESET pin is pulled low, thus allowing the output alert state to
change and follow the input signal with respect to the programmed alert threshold. For example, when the
differential input signal rises above the alert threshold, the alert output pin is pulled low. As soon as the
differential input signal drops below the alert threshold, the output returns to the default high output state. A
common implementation using the device in transparent mode is to connect the ALERT pin to a hardware
interrupt input on a microcontroller. As soon as an overcurrent condition is detected and the ALERT pin is pulled
low, the controller interrupt pin detects the output state change and can begin making changes to the system
operation required to address the overcurrent condition. Under this configuration, the ALERT pin transition from
high to low is captured by the microcontroller so the output can return to the default high state when the
overcurrent event is removed.
7.3.2.2 Latch Output Mode
Some applications do not have the functionality available to continuously monitor the state of the output ALERT
pin to detect an overcurrent condition as described in the Transparent Output Mode section. A typical example of
this application is a system that is only able to poll the ALERT pin state periodically to determine if the system is
functioning correctly. If the device is set to transparent mode in this type of application, the state change of the
ALERT pin can be missed when ALERT is pulled low to indicate an out-of-range event if the out-of-range
condition does not appear during one of these periodic polling events. Latch mode is specifically intended to
accommodate these applications.
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Feature Description (continued)
The device is placed into the corresponding output modes based on the signal connected to RESET, as shown
in Table 1. The difference between latch mode and transparent mode is how the alert output responds when an
overcurrent event ends. In transparent mode (RESET = low), when the differential input signal drops below the
limit threshold level after the ALERT pin asserts because of an overcurrent event, the ALERT pin state returns to
the default high setting to indicate that the overcurrent event has ended.
Table 1. Output Mode Settings
OUTPUT MODE
RESET PIN SETTING
Transparent mode
RESET = low
Latch mode
RESET = high
In latch mode (RESET = high), when an overlimit condition is detected and the ALERT pin is pulled low, the
ALERT pin does not return to the default high state when the differential input signal drops below the alert
threshold level. In order to clear the alert, the RESET pin must be pulled low for at least 100 ns. Pulling the
RESET pin low allows the ALERT pin to return to the default high level provided that the differential input signal
has dropped below the alert threshold. If the input signal is still above the threshold limit when the RESET pin is
pulled low, the ALERT pin remains low. When the alert condition is detected by the system controller, the RESET
pin can be set back to high in order to place the device back in latch mode.
The latch and transparent modes are represented in Figure 37. In Figure 37, when VIN drops back below the
VLIMIT threshold for the first time, the RESET pin is pulled high. With the RESET pin is pulled high, the device is
set to latch mode so that the alert output state does not return high when the input signal drops below the VLIMIT
threshold. Only when the RESET pin is pulled low does the ALERT pin return to the default high level, thus
indicating that the input signal is below the limit threshold. When the input signal drops below the limit threshold
for the second time, the RESET pin is already pulled low. The device is set to transparent mode at this point and
the ALERT pin is pulled back high as soon as the input signal drops below the alert threshold.
VLIMIT
VIN
(VIN+ - VIN-)
0V
Latch Mode
RESET
Transparent Mode
Alert Clears
ALERT
Alert Does Not Clear
Figure 37. Transparent versus Latch Mode
7.3.3 Setting The Current-Limit Threshold
The INA301 determines if an overcurrent event is present by comparing the amplified measured voltage
developed across the current-sensing resistor to the corresponding signal developed at the LIMIT pin. The
threshold voltage for the LIMIT pin can be set using a single external resistor or by connecting an external
voltage source to the LIMIT pin.
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7.3.3.1 Resistor-Controlled Current Limit
The typical approach for setting the limit threshold voltage is to connect a resistor from the LIMIT pin to ground.
The value of this resistor, RLIMIT, is chosen in order to create a corresponding voltage at the LIMIT pin equivalent
to the output voltage, VOUT, when the maximum desired load current is flowing through the current-sensing
resistor. An internal 80-µA current source is connected to the LIMIT pin to create a corresponding voltage used
to compare to the amplifier output voltage, depending on the value of the RLIMIT resistor.
In the equations from Table 2, VTRIP represents the overcurrent threshold that the device is programmed to
monitor for and VLIMIT is the programmed signal set to detect the VTRIP level.
Table 2. Calculating the Limit Threshold Setting Resistor, RLIMIT
PARAMETER
EQUATION
VTRIP
VOUT at the desired current trip value
VLIMIT
Threshold limit voltage
RLIMIT
Calculate the threshold limit-setting resistor
ILOAD × RSENSE x Gain
VLIMIT = VTRIP
ILIMIT × RLIMIT
VLIMIT / ILIMIT
VLIMIT / 80 µA
7.3.3.1.1 Resistor-Controlled Current Limit: Example
For example, if the current level indicating an out-of-range condition is present is 20 A and the current-sense
resistor value is 10 mΩ, then the input threshold signal is 200 mV. The INA301A1 has a gain of 20 so the
resulting output voltage at the 20-A input condition is 4 V. The value for RLIMIT is selected to allow the device to
detect to this 20-A threshold, indicating an overcurrent event has occurred. When the INA301 detects this out-ofrange condition, the ALERT pin asserts and pulls low. For this example, the value of RLIMIT to detect a 4-V level
is calculated to be 50 kΩ, as shown in Table 3.
Table 3. Calculating the Limit Threshold Setting Resistor, RLIMIT: Example
PARAMETER
EQUATION
VTRIP
VOUT at the desired current trip value
VLIMIT
Threshold limit voltage
RLIMIT
Calculate the threshold limit-setting resistor
ILOAD × RSENSE x Gain
20 A x 10 mΩ x 20 V/V = 4 V
VLIMIT = VTRIP
ILIMIT × RLIMIT
VLIMIT / ILIMIT
4 V / 80 µA = 50 kΩ
7.3.3.2 Voltage-Source-Controlled Current Limit
The second method for setting the limit voltage is to connect the LIMIT pin to a programmable digital-to-analog
converter (DAC) or other external voltage source. The benefit of this method is the ability to adjust the currentlimit threshold to account for different threshold voltages that are used for different system operating conditions.
For example, this method can be used in a system that has one current-limit threshold level that must be
monitored during a power-up sequence but different threshold levels that must be monitored during other system
operating modes.
In Table 4, VTRIP represents the overcurrent threshold that the device is programmed to monitor for and VSOURCE
is the programmed signal set to detect the VTRIP level.
Table 4. Calculating the Limit Threshold Voltage Source, VSOURCE
PARAMETER
EQUATION
VTRIP
VOUT at the desired current trip value
ILOAD × RSENSE × Gain
VSOURCE
Program the threshold limit voltage
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7.3.4 Selecting a Current-Sensing Resistor
The device measures the differential voltage developed across a resistor when current flows through the
component to determine if the current being monitored exceeds a defined limit. This resistor is commonly
referred to as a current-sensing resistor or a current-shunt resistor, with each term commonly used
interchangeably. The flexible design of the device allows for measuring a wide differential input signal range
across this current-sensing resistor.
Selecting the value of this current-sensing resistor is based primarily on two factors: the required accuracy of the
current measurement and the allowable power dissipation across the current-sensing resistor. Larger voltages
developed across this resistor allow for more accurate measurements to be made. Amplifiers have fixed internal
errors that are largely dominated by the inherent input offset voltage. When the input signal decreases, these
fixed internal amplifier errors become a larger portion of the measurement and increase the uncertainty in the
measurement accuracy. When the input signal increases, the measurement uncertainty is reduced because the
fixed errors are a smaller percentage of the signal being measured. Therefore, the use of larger value currentsensing resistors inherently improves the measurement accuracy.
However, a system design trade-off must be evaluated through use of larger input signals for improving the
measurement accuracy. Increasing the current sense resistor value results in an increase in power dissipation
across the current-sensing resistor. Increasing the value of the current-shunt resistor increases the differential
voltage developed across the resistor when current passes through the component. This increase in voltage
across the resistor increases the power that the resistor must be able to dissipate. Decreasing the value of the
current-shunt resistor value reduces the power dissipation requirements of the resistor, but increases the
measurement errors resulting from the decreased input signal. Selecting the optimal value for the shunt resistor
requires factoring both the accuracy requirement for the specific application and the allowable power dissipation
of this component.
An increasing number of very low ohmic-value resistors are becoming more widely available with values reaching
down as low as 200 µΩ or lower with power dissipations of up to 5 W that enable large currents to be accurately
monitored with sensing resistors.
7.3.4.1 Selecting a Current-Sensing Resistor: Example
In this example, the trade-offs involved in selecting a current-sensing resistor are discussed. This example
requires 2.5% accuracy for detecting a 10-A overcurrent event where only 250 mW is allowable for the
dissipation across the current-sensing resistor at the full-scale current level. Although the maximum power
dissipation is defined as 250 mW, a lower dissipation is preferred to improve system efficiency. Some initial
assumptions are made that are used in this example: the limit-setting resistor (RLIMIT) is a 1% component and the
maximum tolerance specification for the internal threshold setting current source (0.5%) is used. Given the total
error budget of 2.5%, up to 1% of error is available to be attributed to the measurement error of the device under
these conditions.
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As shown in Table 5, the maximum value calculated for the current-sensing resistor with these requirements is
2.5 mΩ. Although this value satisfies the maximum power dissipation requirement of 250 mW, headroom is
available from the 2.5% maximum total overcurrent detection error to reduce the value of the current-sensing
resistor and reduce the power dissipation further. Selecting a 1.5-mΩ, current-sensing resistor value offers a
good tradeoff for reducing the power dissipation in this scenario by approximately 40% and still remaining within
the accuracy region.
Table 5. Calculating the Current-Sensing Resistor, RSENSE
PARAMETER
EQUATION
VALUE
UNIT
IMAX
Maximum current
10
A
PD_MAX
Maximum allowable power dissipation
250
mW
RSENSE_MAX
Maximum allowable RSENSE
2.5
mΩ
VOS
Offset voltage
150
µV
VOS_ERROR
Initial offset voltage error
EG
Gain error
ERRORTOTAL
Total measurement error
PD_MAX / IMAX2
(VOS / (RSENSE_MAX × IMAX ) × 100
0.6%
0.25%
√(VOS_ERROR2 + EG2)
0.65%
Allowable current threshold accuracy
2.5%
ERRORINITIAL
Initial threshold error
ILIMIT Tolerance + RLIMIT Tolerance
ERRORAVAILABLE
Maximum allowable measurement error
Maximum Error – ERRORINITIAL
2
EG2)
1.5%
1%
VOS_ERROR_MAX
Maximum allowable offset error
√(ERRORAVAILABLE –
VDIFF_MIN
Minimum differential voltage
VOS / VOS_ERROR_MAX (1%)
15
mV
RSENSE_MIN
Minimum sense resistor value
VDIFF_MIN / IMAX
1.5
mΩ
PD_MIN
Lowest possible power dissipation
RSENSE_MIN × IMAX2
150
mW
0.97%
7.3.5 Hysteresis
The on-board comparator in the INA301 is designed to reduce the possibility of oscillations in the alert output
when the measured signal level is near the overlimit threshold level because of noise. When the output voltage
(VOUT) exceeds the voltage developed at the LIMIT pin, the ALERT pin is asserted and pulls low. The output
voltage must drop below the LIMIT pin threshold voltage by the gain-dependent hysteresis level in order for the
ALERT pin to de-assert and return to the nominal high state, as shown in Figure 38.
ALERT
Alert
Output
VOUT
VLIMIT - Hysteresis
VLIMIT
Figure 38. Typical Comparator Hysteresis
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7.4 Device Functional Modes
7.4.1 Input Filtering
External system noise can significantly affect the ability of a comparator to accurately measure and detect
whether input signals exceed the reference threshold levels, thus reliably indicating an overrange condition. The
most obvious effect that external noise can have on the operation of a comparator is to cause a false alert
condition. If a comparator detects a large noise transient coupled into the signal, the device can easily interpret
this transient as an overrange condition.
External filtering can help reduce the amount of noise that reaches the comparator and reduce the likelihood of a
false alert from occurring. The tradeoff to adding this noise filter is that the alert response time is increased
because of the input signal being filtered as well as the noise. Figure 39 shows the implementation of an input
filter for the device.
2.7 V to 5.5 V
CBYPASS
0.1 PF
Supply
(0 V to 36 V)
RPULL-UP
10 k
VS
IN+
RFILTER
”10
CFILTER
INA301
+
OUT
ALERT
INLoad
RESET
LIMIT
GND
RLIMIT
Figure 39. Input Filter
Limiting the amount of input resistance used in this filter is important because this resistance can have a
significant affect on the input signal that reaches the device input pins resulting from the device input bias
currents. A typical system implementation involves placing the current-sensing resistor very near the device so
the traces are very short and the trace impedance is very small. This layout helps reduce the ability of coupling
additional noise into the measurement. Under these conditions, the characteristics of the input bias currents have
minimal affect on device performance.
As illustrated in Figure 40, the input bias currents increase in opposite directions when the differential input
voltage increases. This increase results from the design of the device that allows common-mode input voltages
to far exceed the device supply voltage range. With input filter resistors now placed in series with these unequal
input bias currents, there are unequal voltage drops developed across these input resistors. The difference
between these two drops appears as an added signal that (in this case) subtracts from the voltage developed
across the current-sensing resistor, thus reducing the signal that reaches the device input pins. Smaller value
input resistors reduce this effect of signal attenuation to allow for a more accurate measurement.
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Device Functional Modes (continued)
225
Input Bias Current (µA)
200
175
150
125
100
75
50
25
0
0
50
100
150
200
Differential Input Voltage (mV)
250
C002
Figure 40. Input Bias Current vs Differential Input Voltage
For example, with a differential voltage of 10 mV developed across a current-sensing resistor and using 20-Ω
resistors, the differential signal that actually reaches the device is 9.85 mV. A measurement error of 1.5% is
created as a result of these external input filter resistors. Using 10-Ω input filter resistors instead of the 20-Ω
resistors reduces this added error from 1.5% down to 0.75%.
7.4.2 Using The INA301 with Common-Mode Transients Above 36 V
With a small amount of additional circuitry, the device can be used in circuits subject to transients higher than
36 V. Use only zener diodes or zener-type transient absorbers (sometimes referred to as transzorbs). Any other
type of transient absorber has an unacceptable time delay. Start by adding a pair of resistors, as shown in
Figure 41, as a working impedance for the zener diode. Keeping these resistors as small as possible is best,
preferably 10 Ω or less. Larger values can be used with an additional induced error resulting from a reduced
signal that actually reaches the device input pins. Because this circuit limits only short-term transients, many
applications are satisfied with a 10-Ω resistor along with conventional zener diodes of the lowest power rating
available. This combination uses the least amount of board space. These diodes can be found in packages as
small as SOT-523 or SOD-523.
2.7 V to 5.5 V
CBYPASS
0.1 PF
Supply
(0 V to 36 V)
RPULL-UP
10k
VS
IN+
INA301
+
OUT
RPROTECT
”10
ALERT
INLoad
RESET
LIMIT
GND
RLIMIT
Figure 41. Transient Protection
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8 Applications and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The INA301 is designed to enable easy configuration for detecting overcurrent conditions in an application. This
device is individually targeted towards unidirectional overcurrent detection of a single threshold. However, this
device can also be paired with additional devices and circuitry to create more complex monitoring functional
blocks.
8.2 Typical Application
CBYPASS
0.1 PF
2.7 V to 5.5 V
RPULL-UP
10 k
VS
IN+
+
OUT
IN-
Power Supply
(0 V to 36 V)
OCP+
ALERT
LIMIT
GND
RLIMIT
Current
Output
CBYPASS
0.1 PF
2.7 V to 5.5 V
Load
RPULL-UP
10 k
IN+
+
VS
IN-
OUT
OCP-
ALERT
LIMIT
GND
RLIMIT
Figure 42. Bidirectional Application
8.2.1 Design Requirements
Although the device is only able to measure current through a current-sensing resistor flowing in one direction, a
second INA301 can be used to create a bidirectional monitor.
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Typical Application (continued)
8.2.2 Detailed Design Procedure
With the input pins of a second device reversed across the same current-sensing resistor, the second device is
now able to detect current flowing in the other direction relative to the first device; see Figure 42. The outputs of
each device connect to an AND gate to detect if either of the limit threshold levels are exceeded. As shown in
Table 6, the output of the AND gate is high if neither overcurrent limit thresholds are exceeded. A low output
state of the AND gate indicates that either the positive overcurrent limit or the negative overcurrent limit are
surpassed.
Table 6. Bidirectional Overcurrent Output Status
OCP STATUS
OUTPUT
OCP+
0
OCP–
0
No OCP
1
8.2.3 Application Curve
Input
(5 mV/div)
Alert Output
(1 V/div)
Figure 43 shows two INA301 devices being used in a bidirectional configuration and an output control circuit to
detect if one of the two alerts is exceeded.
Positive Limit
0V
Negtive Limit
Time (5 ms/div)
Figure 43. Bidirectional Application Curve
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9 Power Supply Recommendations
The device input circuitry can accurately measure signals on common-mode voltages beyond the power-supply
voltage, VS. For example, the voltage applied to the VS power-supply pin can be 5 V, whereas the load powersupply voltage being monitored (VCM) can be as high as 36 V. Note also that the device can withstand the full
–0.3 V to 36 V range at the input pins, regardless of whether the device has power applied or not.
Power-supply bypass capacitors are required for stability and must be placed as closely as possible to the supply
and ground pins of the device. A typical value for this supply bypass capacitor is 0.1 µF. Applications with noisy
or high-impedance power supplies can require additional decoupling capacitors to reject power-supply noise.
10 Layout
10.1 Layout Guidelines
•
•
•
Place the power-supply bypass capacitor as closely as possible to the supply and ground pins. The
recommended value of this bypass capacitor is 0.1 µF. Additional decoupling capacitance can be added to
compensate for noisy or high-impedance power supplies.
Make the connection of RLIMIT to the ground pin as direct as possible to limit additional capacitance on this
node. Routing this connection must be limited to the same plane if possible to avoid vias to internal planes. If
the routing can not be made on the same plane and must pass through vias, ensure that a path is routed
from RLIMIT back to the ground pin and that RLIMIT is not simply connected directly to a ground plane.
The open-drain output pin is recommended to be pulled up to the supply voltage rail through a 10-kΩ pullup
resistor.
10.2 Layout Example
RSHU NT
Power
Sup ply
Loa d
5
7
ALE RT RESET
6
IN-
8
Aler t Output
IN+
RPUL L-UP
VIA to
Gro und
Plan e
CBYPASS
1
2
3
4
INA301
VS
OUT
LIMIT
GND
Sup ply
Voltage
VIA to
Gro und
Plan e
RLIMIT
Output Voltage
NOTE: Connect the limit resistor directly to the GND pin.
Figure 44. Recommended Layout
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11 Device and Documentation Support
11.1 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.2 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.3 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.4 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
INA301A1IDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGD6
INA301A1IDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGD6
INA301A2IDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGI6
INA301A2IDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGI6
INA301A3IDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGH6
INA301A3IDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
ZGH6
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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