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LM21212-2
SNVS715B – MARCH 2011 – REVISED JUNE 2019
LM21212-2 2.95-V to 5.5-V, 12-A, Voltage-Mode Synchronous Buck Regulator
With Adjustable Frequency
1 Features
3 Description
•
The LM21212-2 is a monolithic synchronous buck
regulator that is capable of delivering up to 12 A of
continuous output current while producing an output
voltage down to 0.6 V with outstanding efficiency.
The device is optimized to work over an input voltage
range of 2.95 V to 5.5 V, making it suitable for a wide
variety of low voltage systems. The voltage mode
control loop provides high noise immunity and narrow
duty-cycle capability, and it can be compensated to
be stable with any type of output capacitance,
providing maximum flexibility and ease of use.
1
•
•
•
•
•
•
•
•
•
•
•
Integrated 7-mΩ High Side and 4.3-mΩ Low-Side
FET Switches
300-kHz to 1.55-MHz Resistor-Adjustable
Frequency
Adjustable Output Voltage from 0.6 V to VIN
(100% Duty Cycle Capable), ±1% Reference
Input Voltage Range 2.95 V to 5.5 V
Startup Into Prebiased Loads
Output Voltage Tracking Capability
Wide Bandwidth Voltage Loop Error Amplifier
Adjustable Soft-Start With External Capacitor
Precision Enable Pin With Hysteresis
Integrated OVP, OCP, OTP, UVLO, and PowerGood
Thermally Enhanced 20-Pin HTSSOP Exposed
Pad Package
Create a custom design using the LM21212-2 with
the WEBENCH® Power Designer
The LM21212-2 features internal overvoltage
protection (OVP) and overcurrent protection (OCP)
for increased system reliability. A precision enable pin
and integrated UVLO allow turnon of the device to be
tightly controlled and sequenced. Start-up inrush
currents are limited by both an internally fixed and
externally adjustable soft-start circuit. Fault detection
and supply sequencing are possible with the
integrated power good circuit.
Device Information(1)
PART NUMBER
2 Applications
•
•
•
LM21212-2
Broadband, Networking and Wireless
Communications
High-Performance FPGAs, ASICs and
Microprocessors
Simple to Design, High Efficiency Point of Load
Regulation from a 5-V or 3.3-V Bus
PACKAGE
HTSSOP (20)
BODY SIZE (NOM)
6.50 mm × 4.40 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Application Circuit
HTSSOP-20
5,6,7
VIN
CIN
LOUT
PVIN
SW
11-16
VOUT
COUT
RF
4
CC3
AVIN
RFB1
CF
3
LM21212-2
FB
EN
optional
optional
CSS
COMP
18
2 SS/
TRK
CC1 RC1
RFB2
CC2
17
1
FADJ
RADJ
RC2
19
PGOOD
PGND AGND
8,9,10
20
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM21212-2
SNVS715B – MARCH 2011 – REVISED JUNE 2019
www.ti.com
Table of Contents
1
2
3
4
5
6
7
8
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Description, continued ..........................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
5
7.1
7.2
7.3
7.4
7.5
5
5
5
5
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Ratings ...........................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 10
8.1 Overview ................................................................. 10
8.2 Functional Block Diagram ....................................... 10
8.3 Feature Description................................................. 11
8.4 Device Functional Modes........................................ 14
9
Application and Implementation ........................ 15
9.1 Application Information............................................ 15
9.2 Typical Application ................................................. 15
10 Layout................................................................... 27
10.1 Layout Considerations .......................................... 27
10.2 Layout Example .................................................... 27
10.3 Thermal Considerations ........................................ 28
11 Device and Documentation Support ................. 30
11.1
11.2
11.3
11.4
11.5
11.6
Device Support......................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
30
30
30
30
30
31
12 Mechanical, Packaging, and Orderable
Information ........................................................... 31
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (March 2013) to Revision B
•
Editorial changes only, no technical revisions; add links for WEBENCH............................................................................... 1
Changes from Original (March 2013) to Revision A
•
2
Page
Page
Changed layout of National Semiconductor data sheet to TI format...................................................................................... 4
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5 Description, continued
The LM21212-2 is designed to work well in multi-rail power supply architectures. The output voltage of the device
can be configured to track an external voltage rail using the SS/TRK pin. The switching frequency can be
programmed between 300 kHz and 1.55 MHz with an external resistor.
If the output is prebiased at start-up, it does not sink current, allowing the output to smoothly rise past the
prebiased voltage. The regulator is offered in a 20-pin HTSSOP package with an exposed pad that can be
soldered to the PCB, eliminating the need for bulky heat sinks.
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SNVS715B – MARCH 2011 – REVISED JUNE 2019
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6 Pin Configuration and Functions
PWP Package
20-Pin HTSSOP Package
Top View
Top View
20 AGND
FADJ 1
SS/TRK 2
19 FB
18 COMP
EN 3
AVIN 4
17 PGOOD
PVIN 5
16 SW
PVIN 6
EP
15 SW
PVIN 7
14 SW
PGND 8
13 SW
PGND 9
12 SW
PGND 10
11 SW
Pin Descriptions
PIN
NO.
NAME
Description
1
FADJ
Frequency Adjust pin. The switching frequency can be set to a predetermined rate by connecting a resistor
between FADJ and AGND.
2
SS/TRK
3
EN
Active high enable input for the device. If not used, the EN pin can be left open, which will go high due to
an internal current source.
4
AVIN
Analog input voltage supply that generates the internal bias. It is recommended to connect PVIN to AVIN
through a low pass RC filter to minimize the influence of input rail ripple and noise on the analog control
circuitry.
5,6,7
PVIN
Input voltage to the power switches inside the device. These pins should be connected together at the
device. A low ESR input capacitance should be located as close as possible to these pins.
8,9,10
PGND
Power ground pins for the internal power switches.
11-16
SW
17
PGOOD
18
COMP
19
FB
20
AGND
EP
Exposed Pad
4
Soft-start control pin. An internal 2 µA current source charges an external capacitor connected between
this pin and AGND to set the output voltage ramp rate during startup. This pin can also be used to
configure the tracking feature.
Switch node pins. These pins should be tied together locally and connected to the filter inductor.
Open-drain power good indicator.
Compensation pin is connected to the output of the voltage loop error amplifier.
Feedback pin is connected to the inverting input of the voltage loop error amplifier.
Quiet analog ground for the internal reference and bias circuitry.
Exposed metal pad on the underside of the package with an electrical and thermal connection to PGND. It
is recommended to connect this pad to the PC board ground plane in order to improve thermal dissipation.
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7 Specifications
7.1 Absolute Maximum Ratings
See
(1) (2)
PVIN (3), AVIN to GND
−0.3V to +6V
SW (4), EN, FB, COMP, PGOOD, SS/TRK, FADJ to GND
−0.3V to PVIN + 0.3V
−65°C to 150°C
Storage Temperature
Soldering Specification for TSSOP Pb-Free Infrared or Convection (30 sec)
(1)
(2)
(3)
(4)
260°C
Absolute Maximum Ratings indicate limits beyond witch damage to the device may occur. Recommended operating ratings indicate
conditions for which the device is intended to be functional, but do not ensure specific performance limits. For ensured specifications
and test conditions, see the Electrical Characteristics.
If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
The PVIN pin can tolerate transient voltages up to 6.5 V for a period of up to 6ns. These transients can occur during the normal
operation of the device.
The SW pin can tolerate transient voltages up to 9 V for a period of up to 6 ns, and -1 V for a duration of 4 ns. These transients can
occur during the normal operation of the device.
7.2 ESD Ratings
V(ESD)
(1)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
VALUE
UNIT
±2000
V
(1)
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Ratings
PVIN, AVIN to GND
+2.95V to +5.5V
−40°C to +125°C
Junction Temperature
θJA (1)
(1)
24°C/W
Thermal measurements were performed on a 2x2 inch, 4 layer, 2 oz. copper outer layer, 1 oz.copper inner layer board with twelve 8 mil.
vias underneath the EP of the device and an additional sixteen 8 mil. vias under the unexposed package.
7.4 Electrical Characteristics
Unless otherwise stated, the following conditions apply: VPVIN, AVIN = 5V. Limits in standard type are for TJ = 25°C only, limits
in boldface type apply over the junction temperature (TJ) range of −40°C to +125°C. Minimum and maximum limits are
specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C,
and are provided for reference purposes only.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
-1%
0.6
1%
V
SYSTEM
VFB
Feedback pin voltage
VIN = 2.95V to 5.5V
ΔVOUT/ΔIOUT Load Regulation
0.02
%VOUT/
A
ΔVOUT/ΔVIN
0.1
%VOUT/
V
Line Regulation
RDSON HS
High Side Switch On Resistance
ISW = 12A
7.0
9.0
mΩ
RDSON LS
Low Side Switch On Resistance
ISW = 12A
4.3
6.0
mΩ
ICLR
HS Rising Switch Current Limit
17
19
A
ICLF
LS Falling Switch Current Limit
VZX
Zero Cross Voltage
3
12
mV
1.5
3.0
mA
50
70
µA
2.45
2.70
2.95
V
140
200
280
mV
-10
6
20
mV
1.3
1.9
2.5
µA
15
12
-8
IQ
Operating Quiescent Current
ISD
Shutdown Quiescent Current
VEN = 0V
VUVLO
AVIN Undervoltage Lockout
AVIN Rising
VUVLOHYS
AVIN Undervoltage Lockout Hysteresis
VTRACKOS
SS/TRACK PIN accuracy (VSS - VFB)
ISS
0 < VTRACK < 0.55V
Soft-Start Pin Source Current
A
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Electrical Characteristics (continued)
Unless otherwise stated, the following conditions apply: VPVIN, AVIN = 5V. Limits in standard type are for TJ = 25°C only, limits
in boldface type apply over the junction temperature (TJ) range of −40°C to +125°C. Minimum and maximum limits are
specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C,
and are provided for reference purposes only.
PARAMETER
tINTSS
Internal Soft-Start Ramp to Vref
tRESETSS
Device Reset to Soft-Start Ramp
TEST CONDITIONS
CSS = 0
MIN
TYP
MAX
UNIT
350
500
675
µs
50
110
200
µs
OSCILLATOR
fRNG
FADJ Frequency Range
fSW
Switching Frequency
1550
kHz
RADJ = 22.6kΩ
1400
300
1550
1700
kHz
RADJ = 95.3kΩ
465
500
535
tHSBLANK
HS OCP Blanking Time
Rising edge of SW to ICLR comparison
55
ns
tLSBLANK
LS OCP Blanking Time
Falling edge of SW to ICLF comparison
400
ns
tZXBLANK
Zero Cross Blanking Time
Falling edge of SW to VZX comparison
120
ns
Minimum HS on-time
140
ns
PWM Ramp p-p Voltage
0.8
V
95
dBV/V
11
MHz
tMINON
ΔVramp
ERROR AMPLIFIER
VOL
Error Amplifier Open Loop Voltage Gain
GBW
Error Amplifier Gain-Bandwidth Product
IFB
Feedback Pin Bias Current
ICOMP = -65µA to 1mA
VFB = 0.6V
1
nA
ICOMPSRC
COMP Output Source Current
1
mA
ICOMPSINK
COMP Output Sink Current
65
µA
POWERGOOD
VOVP
VOVPHYS
VUVP
VUVPHYS
Overvoltage Protection Rising Threshold
VFB Rising
Overvoltage Protection Hysteresis
VFB Falling
Undervoltage Protection Rising Threshold VFB Rising
Undervoltage Protection Hysteresis
105
112.5
82
90
120
2
VFB Falling
%VFB
%VFB
97
%VFB
2.5
%VFB
tPGDGL
PGOOD Deglitch Low (OVP/UVP
Condition Duration to PGOOD Falling)
15
µs
tPGDGH
PGOOD Deglitch High (minimum low
pulse)
12
µs
RPGOOD
PGOOD Pulldown Resistance
IPGOODLEAK
PGOOD Leakage Current
10
VPGOOD = 5V
20
40
1
Ω
nA
LOGIC
VIHSYNC
SYNC Pin Logic High
VILSYNC
SYNC Pin Logic Low
VIHENR
EN Pin Rising Threshold
VENHYS
EN Pin Hysteresis
IEN
EN Pin Pullup Current
2.0
V
0.8
VEN Rising
VEN = 0V
V
1.20
1.35
1.45
V
50
110
180
mV
2
µA
165
°C
10
°C
THERMAL SHUTDOWN
TTHERMSD
Thermal Shutdown
TTHERMSDHYS Thermal Shutdown Hysteresis
6
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7.5 Typical Characteristics
Unless otherwise specified: VVIN = 5 V, VOUT = 1.2 V, L= 0.56 µH (1.8-mΩ RDCR), CSS = 33 nF, fSW = 500 kHz (RADJ= 95.3 kΩ),
TA = 25°C for efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
96
94
VOUT = 3.3
VOUT = 1.2
98
96
92
EFFICIENCY (%)
EFFICIENCY (%)
100
FSW = 500kHz
FSW = 1MHz
FSW = 1.5MHz
90
88
86
84
94
92
90
88
86
84
82
82
80
80
0
2
4
6
8
10
OUTPUT CURRENT(A)
12
0
Figure 1. Efficiency
12
Figure 2. Efficiency
100
1.5
98
1.4
IPVIN+ IAVIN(mA)
EFFICIENCY (%)
2
4
6
8
10
OUTPUT CURRENT(A)
96
94
92
VIN = 3.3V
VIN = 4.0V
VIN = 5.0V
VIN = 5.5V
90
0
FSW= 300 KHz
1.2
1.1
2
4
6
8
10
OUTPUT CURRENT(A)
VOUT = 2.5 V
1.3
12
1.0
3.0
Inductor P/N
Ser2010-102MLD
3.5
4.0
4.5
5.0
INPUT VOLTAGE (V)
5.5
Figure 3. Efficiency
Figure 4. Non-Switching IQTOTAL vs VIN
0.180
0.602
0.172
1.14
0.164
1.11
0.156
1.08
0.148
1.05
0.140
1.02
0.132
0.99
0.124
0.96
0.116
0.93
0.108
0.601
0.100
0.90
-40 -20 0 20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
VFB(V)
IAVIN(mA)
1.17
IAVIN
IPVIN
IPVIN(mA)
1.20
0.600
0.599
0.598
Figure 5. Non-Switching IAVIN and IPVIN vs Temperature
-40 -20 0 20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
Figure 6. VFB vs Temperature
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Typical Characteristics (continued)
Unless otherwise specified: VVIN = 5 V, VOUT = 1.2 V, L= 0.56 µH (1.8-mΩ RDCR), CSS = 33 nF, fSW = 500 kHz (RADJ= 95.3 kΩ),
TA = 25°C for efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
2.78
1.36
144
2.76
270
1.35
136
2.74
255
1.34
128
2.72
240
1.33
120
2.70
225
1.32
112
2.68
210
1.31
104
2.66
195
1.30
96
2.64
180
1.29
88
2.62
165
1.28
80
2.60
150
-40 -20 0 20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
Figure 7. Enable Threshold and Hysteresis vs Temperature
V UVLO
V UVLOHYS
300
285
VUVLOHYS(mV)
VIHENR(V)
VUVLO(V)
2.80
152
VENHYS(V)
160
V IHENR
V ENHYS
1.37
-40 -20 0 20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
Figure 8. UVLO Threshold and Hysteresis vs Temperature
58
0.68
56
52
50
48
46
0.64
0.62
0.60
0.58
44
0.57
42
0.54
40
0.52
-40 -20 0 20 40 60 80 100 120
JUNCTION TEMPERATURE(°C)
Figure 9. Enable Low Current vs Temperature
0.50
-40 -20 0 20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
Figure 10. OVP/UVP Threshold vs Temperature
160
10
9
152
LOW SIDE
HIGH SIDE
8
148
RDSON(m )
MINIMUM ON-TIME (nS)
156
144
140
136
132
7
6
5
4
128
3
124
2
120
-40 -20 0 20 40 60 80 100 120
JUNCTION TEMPERATURE (°C)
-40 -20 0 20 40 60 80 100 120
JUNCTION TEMPERATURE(°C)
Figure 11. Minimum On-Time vs Temperature
8
VUVP
VOVP
0.66
54
VOVP,VUVP(V)
SHUTDOWN CURRENT ISD(μA)
60
Figure 12. FET Resistance vs Temperature
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Typical Characteristics (continued)
Unless otherwise specified: VVIN = 5 V, VOUT = 1.2 V, L= 0.56 µH (1.8-mΩ RDCR), CSS = 33 nF, fSW = 500 kHz (RADJ= 95.3 kΩ),
TA = 25°C for efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
17.5
CURRENT LIMIT ICLR(A)
17.4
VOUT (500 mV/Div)
17.3
17.2
17.1
VPGOOD (5V/Div)
17.0
16.9
16.8
VENABLE (5V/Div)
16.7
16.6
16.5
-40 -20 0 20 40 60 80 100 120
AMBIENT TEMPERATURE (°C)
2 ms/DIV
Figure 14. Start-up With Prebiased Output
Figure 13. Peak Current Limit vs Temperature
VOUT (500 mV/Div)
VOUT (500 mV/Div)
VPGOOD (5V/Div)
VTRACK (500 mV/Div)
VENABLE (5V/Div)
VPGOOD (5V/Div)
IOUT (10A/Div)
IOUT (10A/Div)
200 µs/DIV
Figure 15. Start-up With SS/TRK Open Circuit
200 ms/DIV
Figure 16. Start-up With Applied Track Signal
VPGOOD (5V/Div)
VOUT (1V/Div)
IL (10A/Div)
10 µs/DIV
Figure 17. Output Overcurrent Condition
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8 Detailed Description
8.1 Overview
The LM21212-2 switching regulator features all of the functions necessary to implement an efficient low voltage
buck regulator using a minimum number of external components. This easy-to-use regulator features two
integrated switches and is capable of supplying up to 12 A of continuous output current. The regulator utilizes
voltage mode control with trailing edge modulation to optimize stability and transient response over the entire
output voltage range. The device can operate at high switching frequency allowing use of a small inductor while
still achieving high efficiency. The precision internal voltage reference allows the output to be set as low as 0.6 V.
Fault protection features include: current limiting, thermal shutdown, overvoltage protection, and shutdown
capability. The device is available in the 20-pin HTSSOP package featuring an exposed pad to aid thermal
dissipation. The LM21212-2 can be used in numerous applications to efficiently step-down from a 5-V or 3.3-V
bus.
8.2 Functional Block Diagram
Ilimit high
FADJ
VREF
AVIN
PVIN
Over
temp
+
-
PVIN
UVLO
2.7V
+
-
SD
OR
Driver
Precision
enable
AVIN
1.35V
+
-
EN
Control
Logic
PWM
comparator
AVIN
OSC
RAMP
+
-
Zero-cross
+
-
PWM
SW
INT
SS
PVIN
+
SS/TRK
0.6V
EA
Driver
FB
OVP
COMP
0.68V
0.54V
+
-
Ilimit low
OR
Powerbad
+
-
PGND
UVP
AGND
10
PGOOD
OR
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8.3 Feature Description
8.3.1 Precision Enable
The enable (EN) pin allows the output of the device to be enabled or disabled with an external control signal.
This pin is a precision analog input that enables the device when the voltage exceeds 1.35V (typical). The EN pin
has 110 mV of hysteresis and will disable the output when the enable voltage falls below 1.24 V (typical). If the
EN pin is not used, it can be left open, and will be pulled high by an internal 2-µA current source. Since the
enable pin has a precise turn-on threshold it can be used along with an external resistor divider network from VIN
to configure the device to turn on at a precise input voltage.
8.3.2 UVLO
The LM21212-2 has a built-in undervoltage lockout protection
the input voltage reaches 2.7V (typical). The UVLO threshold
from responding to power-on glitches during start up. If desired
using the precision enable pin and a resistor divider network
design guide.
circuit that keeps the device from switching until
has 200 mV of hysteresis that keeps the device
the turnon point of the supply can be changed by
connected to VIN as shown in Figure 23 in the
8.3.3 Current Limit
The LM21212-2 has current limit protection to avoid dangerous current levels on the power FETs and inductor. A
current limit condition is met when the current through the high side FET exceeds the rising current limit level
(ICLR). The control circuitry will respond to this event by turning off the high side FET and turning on the low side
FET. This forces a negative voltage on the inductor, thereby causing the inductor current to decrease. The highside FET does not conduct again until the lower current limit level (ICLF) is sensed on the low side FET. At this
point, the device resumes normal switching.
A current limit condition will cause the internal soft-start voltage to ramp downward. After the internal soft-start
ramps below the feedback (FB) pin voltage, (nominally 0.6 V), FB begins to ramp downward, as well. This
voltage foldback limits the power consumption in the device, thereby protecting the device from continuously
supplying power to the load under a condition that does not fall within the device SOA. After the current limit
condition is cleared, the internal soft-start voltage will ramp up again. Figure 18 shows current limit behavior with
VSS, VFB, VOUT and VSW.
8.3.4 Short-Circuit Protection
In the unfortunate event that the output is shorted with a low impedance to ground, the LM21212-2 will limit the
current into the short by resetting the device. A short-circuit condition is sensed by a current-limit condition
coinciding with a voltage on the FB pin that is lower than 100 mV. When this condition occurs, the device will
begin its reset sequence, turning off both power FETs and discharging the soft-start capacitor after tRESETSS
(nominally 110 µs). The device will then attempt to restart. If the short-circuit condition still exists, it will reset
again, and repeat until the short-circuit is cleared. The reset prevents excess current flowing through the FETs in
a highly inefficient manner, potentially causing thermal damage to the device or the bus supply.
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Feature Description (continued)
Iclr
IL
Iclf
VSS
VFB
100 mV
VOUT
VSW
CURRENT LIMIT
SHORT-CIRCUIT
REMOVED
SHORT-CIRCUIT
Figure 18. Current Limit Conditions
8.3.5 Thermal Protection
Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event that the maximum
junction temperature is exceeded. When activated, typically at 165°C, the LM21212-2 tri-states the power FETs
and resets soft start. After the junction cools to approximately 155°C, the device starts up using the normal start
up routine. This feature is provided to prevent catastrophic failures from accidental device overheating. Note that
thermal limit will not stop the die from operating above the specified operating maximum temperature,125°C. The
die should be kept under 125°C to ensure correct operation.
8.3.6 Power-Good Flag
The PGOOD pin provides the user with a way to monitor the status of the LM21212-2. In order to use the
PGOOD pin, the application must provide a pullup resistor to a desired DC voltage (in other words, VIN). PGOOD
responds to a fault condition by pulling the PGOOD pin low with the open-drain output. PGOOD will pull low on
the following conditions – 1) VFB moves above or below the VOVP or VUVP, respectively 2) The enable pin is
brought below the enable threshold 3) The device enters a prebiased output condition (VFB>VSS).
Figure 19 shows the conditions that will cause PGOOD to fall.
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Feature Description (continued)
tRESETSS
Vss
0.6V
Vovp
VOVPHYS
VFB
Vuvp
VUVPHYS
VEN
VPGOOD
VSW
OVP
UVP
DISABLE
tPGDGL
PRE-BIASED
STARTUP
tPGDGH
Figure 19. PGOOD Conditions
8.3.7 Light Load Operation
The LM21212-2 offers increased efficiency when operating at light loads. Whenever the load current is reduced
to a point where the peak-to-peak inductor ripple current is greater than two times the load current, the device
will enter the diode emulation mode preventing significant negative inductor current. The output current at which
this occurs is the critical conduction boundary and can be calculated by Equation 1:
IBOUNDARY =
(VIN ± VOUT) x D
2 x L x fSW
(1)
It can be seen that in diode emulation mode, whenever the inductor current reaches zero the SW node becomes
high impedance. Ringing will occur on this pin as a result of the LC tank circuit formed by the inductor and the
parasitic capacitance at the node. If this ringing is of concern an additional RC snubber circuit can be added from
the switch node to ground.
At very light loads, usually below 500 mA, several pulses may be skipped in between switching cycles, effectively
reducing the switching frequency and further improving light-load efficiency.
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8.4 Device Functional Modes
Switchnode Voltage
Several diagrams are shown in Figure 20 illustrating continuous conduction mode (CCM), discontinuous
conduction mode (DCM), and the boundary condition.
Continuous Conduction Mode (CCM)
VIN
Time (s)
Inductor Current
Continuous Conduction Mode (CCM)
IAVERAGE
Inductor Current
Time (s)
DCM - CCM Boundary
IAVERAGE
Switchnode Voltage
Time (s)
Discontinuous Conduction Mode (DCM)
VIN
Inductor Current
Time (s)
Discontinuous Conduction Mode (DCM)
IPeak
Time (s)
Figure 20. Modes Of Operation for LM21212-2
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The LM21212-2 switching regulator features all of the functions necessary to implement an efficient low voltage
buck regulator using a minimum number of external components. This easy-to-use regulator features two
integrated switches and is capable of supplying up to 12 A of continuous output current. The regulator utilizes
voltage mode control with trailing edge modulation to optimize stability and transient response over the entire
output voltage range. The device can operate at high switching frequency allowing use of a small inductor while
still achieving high efficiency.
9.2 Typical Application
9.2.1 Typical Application 1
HTSSOP-20
5,6,7
VIN
3
RF
4
CIN1 CIN2 CIN3
CF
PVIN
LO
SW
CSS
VOUT
CC3
EN
RFB1
RC2
AVIN
LM21212-2
2
11-16
SS/
TRK
FB
COMP
CO1 CO2 CO3
19
18
CC1 RC1
RFB2
CC2
VIN
1
17
FADJ
RADJ
PGOOD
PGND AGND
8,9,10
RPGOOD
20
Figure 21. Typical Application Schematic 1
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Typical Application (continued)
9.2.1.1 Design Requirements
Table 1. Bill Of Materials (VIN = 3.3 V - 5.5 V, VOUT = 1.2 V, IOUT = 12 A, FSW = 500 kHz)
ID
DESCRIPTION
VENDOR
PART NUMBER
QUANTITY
CF
CAP, CERM, 1 uF, 10V, +/-10%,
X7R, 0603
MuRata
GRM188R71A105KA61D
1
CIN1, CIN2, CIN3, CO1,
CO2, CO3
CAP, CERM, 100 uF, 6.3V, +/-20%,
X5R, 1206
MuRata
GRM31CR60J107ME39L
6
CC1
CAP, CERM, 1800 pF, 50V, +/-5%,
C0G/NP0, 0603
TDK
C1608C0G1H182J
1
CC2
CAP, CERM, 68 pF, 50V, +/-5%,
C0G/NP0, 0603
TDK
C1608C0G1H680J
1
CC3
CAP, CERM, 820 pF, 50V, +/-5%,
C0G/NP0, 0603
TDK
C1608C0G1H821J
1
CSS
CAP, CERM, 0.033 uF, 16V, +/-10%,
X7R, 0603
MuRata
GRM188R71C333KA01D
1
LO
Inductor, Shielded Drum Core,
Powdered Iron, 560nH, 27.5A, 0.0018
ohm, SMD
Vishay-Dale
IHLP4040DZERR56M01
1
RF
RES, 1.0 ohm, 5%, 0.1W, 0603
Vishay-Dale
CRCW06031R00JNEA
1
RC1
RES, 9.31 kohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW06039K31FKEA
1
RC2
RES, 165 ohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW0603165RFKEA
1
RFB1, RFB2, RPGOOD
RES, 10 kohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW060310K0FKEA
3
RADJ
RES, 95.3 kohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW060395K3FKEA
1
9.2.1.2 Detailed Design Procedure
9.2.1.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM21212-2 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
9.2.1.2.2 Output Voltage
The first step in designing the LM21212-2 application is setting the output voltage. This is done by using a
voltage divider between VOUT and AGND, with the middle node connected to VFB. When operating under steadystate conditions, the LM21212-2 will force VOUT such that VFB is driven to 0.6 V.
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VOUT
LM21212-2
RFB1
0.6V
FB
RFB2
Figure 22. Setting VOUT
A good starting point for the lower feedback resistor, RFB2, is 10kΩ. RFB1 can then be calculated the following
equation:
VOUT =
RFB1 + RFB2
0.6V
RFB2
(2)
9.2.1.2.3 Precision Enable
The enable (EN) pin of the LM21212-2 allows the output to be toggled on and off. This pin is a precision analog
input. When the voltage exceeds 1.35V, the controller will try to regulate the output voltage as long as the input
voltage has exceeded the UVLO voltage of 2.70V. There is an internal current source connected to EN so if
enable is not used, the device will turn on automatically. If EN is not toggled directly the device can be
preprogrammed to turn on at a certain input voltage higher than the UVLO voltage. This can be done with an
external resistor divider from AVIN to EN and EN to AGND as shown below in Figure 23.
Input Power
Supply
RA
AVIN
LM21212-2
EN
VOUT
RB
Figure 23. Enable Start-up Through VIN
The resistor values of RA and RB can be relatively sized to allow EN to reach the enable threshold voltage
depending on the input supply voltage. With the enable current source accounted for, the equation solving for RA
is shown below:
RB VPVIN - 1.35V
RA =
1.35V - IENRB
(3)
In the Equation 3 RA is the resistor from VIN to enable, RB is the resistor from enable to ground, IEN is the internal
enable pullup current (2 µA) and 1.35 V is the fixed precision enable threshold voltage. Typical values for RB
range from 10 kΩ to 100 kΩ.
9.2.1.2.4 Soft Start
When EN has exceeded 1.35 V, and both PVIN and AVIN have exceeded the UVLO threshold, the LM21212-2
begins charging the output linearly to the voltage level dictated by the feedback resistor network. The LM21212-2
has a user adjustable soft-start circuit to lengthen the charging time of the output set by a capacitor from the soft
start pin to ground. After enable exceeds 1.35 V, an internal 2-µA current source begins to charge the soft start
capacitor. This allows the user to limit inrush currents due to a high output capacitance and not cause an over
current condition. Adding a soft-start capacitor can also reduce the stress on the input rail. Larger capacitor
values will result in longer start-up times. Use the equation below to approximate the size of the soft-start
capacitor:
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tSS x ISS
= CSS
0.6V
where
•
•
ISSis nominally 2 µA
tSS is the desired start-up time
(4)
If VIN is higher than the UVLO level and enable is toggled high the soft start sequence will begin. There is a small
delay between enable transitioning high and the beginning of the soft-start sequence. This delay allows the
LM21212-2 to initialize its internal circuitry. Once the output has charged to 90% of the nominal output voltage
the power-good flag will transition high. This behavior is illustrated in Figure 24.
Voltage
90% VOUT
(VUVP)
VOUT
Enable
Delay
(tRESETSS)
0V
VEN
VPGOOD
Soft Start Time (tss)
Time
Figure 24. Soft Start Timing
As shown above, the size of the capacitor is influenced by the nominal feedback voltage level 0.6 V, the soft-start
charging current ISS (2 µA), and the desired soft start time. If no soft-start capacitor is used then the LM21212-2
defaults to a minimum startup time of 500 µs. The LM21212-2 will not start up faster than 500 µs. When enable
is cycled or the device enters UVLO, the charge developed on the soft-start capacitor is discharged to reset the
startup process. This also happens when the device enters short circuit mode from an overcurrent event.
9.2.1.2.5 Resistor-Adjustable Frequency
The frequency adjust (FADJ) pin allows the LM21212-2 to be programmed to a predetermined switching
frequency between 300 kHz to 1.55 MHz by connecting a resistor between FADJ and AGND. To determine the
resistor (RADJ) value for a desired frequency, the following equation can be used:
54680
RADJ =
- 13.15
fSW
where
•
•
RADJ is resistance in kΩ
fSW is frequency in kHz
(5)
The desired frequency must fall within the operational frequency range, 300 kHz to 1550 kHz, and a
corresponding resistor must be used for normal operation.
9.2.1.2.6 Inductor Selection
The inductor (L) used in the application will influence the ripple current and the efficiency of the system. The first
selection criteria is to define a ripple current, ΔIL. In a buck converter, it is typically selected to run between 20%
to 30% of the maximum output current. Figure 25 shows the ripple current in a standard buck converter operating
in continuous conduction mode. Larger ripple current results in a smaller inductance value, which will lead to
lower inductor series resistance, and improved efficiency. However, larger ripple current will also cause the
device to operate in discontinuous conduction mode at a higher average output current.
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VSW
VIN
Time
IL
IL AVG = IOUT
'IL
Time
Figure 25. Switch And Inductor Current Waveforms
Once the ripple current has been determined, the appropriate inductor size can be calculated using the following
equation:
L=
(VIN ± VOUT) D
üIL fSW
(6)
9.2.1.2.7 Output Capacitor Selection
The output capacitor, COUT, filters the inductor ripple current and provides a source of charge for transient load
conditions. A wide range of output capacitors may be used with the LM21212-2 that provide various advantages.
The best performance is typically obtained using ceramic, SP or OSCON type chemistries. Typical trade-offs are
that the ceramic capacitor provides extremely low ESR to reduce the output ripple voltage and noise spikes,
while the SP and OSCON capacitors provide a large bulk capacitance in a small volume for transient loading
conditions.
When selecting the value for the output capacitor, the two performance characteristics to consider are the output
voltage ripple and transient response. The output voltage ripple can be approximated by using the following
formula:
'VOUT
'IL x RESR +
1
8 x fSW x COUT
where
•
•
•
•
ΔVOUT (V) is the amount of peak to peak voltage ripple at the power supply output
RESR (Ω) is the series resistance of the output capacitor
fSW (Hz) is the switching frequency
COUT (F) is the output capacitance used in the design
(7)
The amount of output ripple that can be tolerated is application specific; however a general recommendation is to
keep the output ripple less than 1% of the rated output voltage. Keep in mind ceramic capacitors are sometimes
preferred because they have very low ESR; however, depending on package and voltage rating of the capacitor
the value of the capacitance can drop significantly with applied voltage. The output capacitor selection will also
affect the output voltage droop during a load transient. The peak droop on the output voltage during a load
transient is dependent on many factors; however, an approximation of the transient droop ignoring loop
bandwidth can be obtained using the following equation:
VDROOP = 'IOUTSTEP x RESR +
L x 'IOUTSTEP2
COUT x (VIN - VOUT)
where
•
•
•
COUT (F) is the minimum required output capacitance
L (H) is the value of the inductor
VDROOP (V) is the output voltage drop ignoring loop bandwidth considerations
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•
•
•
•
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ΔIOUTSTEP (A) is the load step change
RESR (Ω) is the output capacitor ESR
VIN (V) is the input voltage
VOUT (V) is the set regulator output voltage
(8)
Both the tolerance and voltage coefficient of the capacitor must be examined when designing for a specific
output ripple or transient droop target.
9.2.1.2.8 Input Capacitor Selection
Quality input capacitors are necessary to limit the ripple voltage at the PVIN pin while supplying most of the
switch current during the on-time. Additionally, they help minimize input voltage droop in an output current
transient condition. In general, it is recommended to use a ceramic capacitor for the input as it provides both a
low impedance and small footprint. Use of a high-grade dielectric for the ceramic capacitor, such as X5R or X7R,
will provide improved performance over temperature and also minimize the DC voltage derating that occurs with
Y5V capacitors. The input capacitors should be placed as close as possible to the PVIN and PGND pins.
Non-ceramic input capacitors should be selected for RMS current rating and minimum ripple voltage. A good
approximation for the required ripple current rating is given by the relationship:
IIN-RMS = IOUT D(1 - D)
(9)
As indicated by the RMS ripple current equation, highest requirement for RMS current rating occurs at 50% duty
cycle. For this case, the RMS ripple current rating of the input capacitor should be greater than half the output
current. For best performance, place low ESR ceramic capacitors in parallel with higher capacitance capacitors
to provide the best input filtering for the device.
When operating at low input voltages (3.3 V or lower), additional capacitance may be necessary to protect from
triggering an under-voltage condition on an output current transient. This will depend on the impedance between
the input voltage supply and the LM21212-2, as well as the magnitude and slew rate of the output transient.
The AVIN pin requires a 1-µF ceramic capacitor to AGND and a 1-Ω resistor to PVIN. This RC network filter
inherent noise on PVIN from the sensitive analog circuitry connected to AVIN.
9.2.1.2.9 Control Loop Compensation
The LM21212-2 incorporates a high bandwidth amplifier between the FB and COMP pins to allow the user to
design a compensation network that matches the application. This section will walk through the various steps in
obtaining the open loop transfer function.
There are three main blocks of a voltage mode buck switcher that the power supply designer must consider
when designing the control system; the power train, modulator, and the compensated error amplifier. A closed
loop diagram is shown in Figure 26.
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PWM Modulator
Power Train
VIN
RDCR
DRIVER
LOUT
VOUT
SW
RESR
RO
COUT
PWM
+
Error Amplifier and Compensation
COMP
+
EA
-
CC1
RC1
0.6V
FB
RFB1
RC2 C
C3
RFB2
CC2
Figure 26. Loop Diagram
The power train consists of the output inductor (L) with DCR (DC resistance RDCR), output capacitor (C0) with
ESR (effective series resistance RESR), and load resistance (Ro). The error amplifier (EA) constantly forces FB to
0.6 V. The passive compensation components around the error amplifier help maintain system stability. The
modulator creates the duty cycle by comparing the error amplifier signal with an internally generated ramp set at
the switching frequency.
There are three transfer functions that must be taken into consideration when obtaining the total open loop
transfer function; COMP to SW (modulator) , SW to VOUT (power train), and VOUT to COMP (error amplifier). The
COMP to SW transfer function is simply the gain of the PWM modulator.
GPWM =
Vin
ÂVramp
where
•
ΔVRAMP is the oscillator peak-to-peak ramp voltage (nominally 0.8 V)
(10)
The SW-to-COMP transfer function includes the output inductor, output capacitor, and output load resistance.
The inductor and capacitor create two complex poles at a frequency described by:
fLC =
1
2S
RO + RDCR
LOUTCOUT(RO + RESR)
(11)
In addition to two complex poles, a left half plane zero is created by the output capacitor ESR located at a
frequency described by:
fESR =
1
2SCOUTRES
(12)
A Bode plot showing the power train response is shown in Figure 27
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60
0
-40
40
GAIN (dB)
-120
0
-160
-20
-200
PHASE (°)
-80
20
-240
-40
-280
-60
-80
100
GAIN
PHASE
1k
10k 100k
1M
FREQUENCY (HZ)
-320
-360
10M
Figure 27. Power Train Bode Plot
The complex poles created by the output inductor and capacitor cause a 180° phase shift at the resonant
frequency as seen in Figure 27. The phase is boosted back up to –90° because of the output capacitor ESR
zero. The 180° phase shift must be compensated out and phase boosted through the error amplifier to stabilize
the closed loop response. The compensation network shown around the error amplifier in Figure 26 creates two
poles, two zeros and a pole at the origin. Placing these poles and zeros at the correct frequencies will stabilize
the closed loop response. The Compensated Error Amplifier transfer function is:
s
s
+1
+1
2SfZ1
2SfZ2
GEA = Km
s
s
s
+1
+1
2SfP1
2SfP2
(13)
The pole located at the origin gives high open loop gain at DC, translating into improved load regulation
accuracy. This pole occurs at a very low frequency due to the limited gain of the error amplifier; however, it can
be approximated at DC for the purposes of compensation. The other two poles and two zeros can be located
accordingly to stabilize the voltage mode loop depending on the power stage complex poles and Q. Figure 28 is
an illustration of what the Error Amplifier Compensation transfer function will look like.
GAIN
PHASE
90
80
45
60
0
40
-45
20
-90
0
-135
-20
100
PHASE (°)
GAIN (dB)
100
-180
1k
10k 100k 1M
FREQUENCY (Hz)
10M
Figure 28. Type 3 Compensation Network Bode Plot
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As seen in Figure 28, the two zeros (fLC/2, fLC) in the comensation network give a phase boost. This will cancel
out the effects of the phase loss from the output filter. The compensation network also adds two poles to the
system. One pole should be located at the zero caused by the output capacitor ESR (fESR) and the other pole
should be at half the switching frequency (fSW/2) to roll off the high frequency response. The dependancy of the
pole and zero locations on the compensation components is described below.
fLC
1
fZ1 = 2 = 2SR C
C1 C1
1
fZ2 = fLC = 2S(R + R )C
C1
FB1 C3
fP1 = fESR =
fP2 =
1
2SRC2CC3
fsw
CC1 + CC2
= 2SR C C
2
C1 C1 C2
(14)
An example of the step-by-step procedure to generate compensation component values using the typical
application setup (see Figure 21) is given. The parameters needed for the compensation values are given in the
table below.
PARAMETER
VALUE
VIN
5V
VOUT
1.2 V
IOUT
12 A
fCROSSOVER
100 kHz
L
0.56 µH
RDCR
1.8 mΩ
CO
150 µF
RESR
1 mΩ
ΔVRAMP
0.8 V
fSW
500 kHz
where ΔVRAMP is the oscillator peak-to-peak ramp voltage (nominally 0.8V), and fCROSSOVER is the frequency at
which the open-loop gain is a magnitude of 1. It is recommended that the fcrossover not exceed one-fifth of the
switching frequency. The output capacitance, CO, depends on capacitor chemistry and bias voltage. For MultiLayer Ceramic Capacitors (MLCC), the total capacitance will degrade as the DC bias voltage is increased.
Measuring the actual capacitance value for the output capacitors at the output voltage is recommended to
accurately calculate the compensation network. The example given here is the total output capacitance using the
three MLCC output capacitors biased at 1.2V, as seen in the typical application schematic, . Note that it is more
conservative, from a stability standpoint, to err on the side of a smaller output capacitance value in the
compensation calculations rather than a larger, as this will result in a lower bandwidth but increased phase
margin.
First, a the value of RFB1 should be chosen. A typical value is 10 kΩ. From this, the value of RC1 can be
calculated to set the mid-band gain so that the desired crossover frequency is achieved:
RC1 =
fCROSSOVER 'VRAMP
fLC
VIN
RFB1
100 kHz 0.8 V
10 k:
17.4 kHz 5.0 V
= 9.2 k:
=
(15)
Next, the value of CC1 can be calculated by placing a zero at half of the LC double pole frequency (fLC):
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CC1 =
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1
SfLCRC1
= 1.99 nF
(16)
Now the value of CC2 can be calculated to place a pole at half of the switching frequency (fSW):
CC2 =
CC1
SfSWRC1 CC1 -1
= 71 pF
(17)
RC2 can then be calculated to set the second zero at the LC double pole frequency:
RFB1 fLC
RC2 =
fESR - fLC
= 166:
(18)
Last, CC3 can be calculated to place a pole at the same frequency as the zero created by the output capacitor
ESR:
1
CC3 =
2SfESRRC2
= 898 pF
(19)
An illustration of the total loop response can be seen in Figure 29.
GAIN
PHASE
150
160
140
120
GAIN (dB)
100
100
80
60
50
40
20
0
PHASE MARGIN (°)
200
0
-20
-50
-40
10
100
1k
10k 100k
FREQUENCY (Hz)
1M
Figure 29. Loop Response
It is important to verify the stability by either observing the load transient response or by using a network
analyzer. A phase margin between 45° and 70° is usually desired for voltage mode systems. Excessive phase
margin can cause slow system response to load transients and low phase margin may cause an oscillatory load
transient response. If the load step response peak deviation is larger than desired, increasing fCROSSOVER and
recalculating the compensation components may help but usually at the expense of phase margin.
24
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9.2.1.3 Application Curves
0.10
0.04
0.08
û OUTPUT VOLTAGE (%)
û OUTPUT VOLTAGE (%)
0.03
0.02
0.01
0.00
-0.01
-0.02
VIN = 3.3V
VIN = 5.0V
-0.03
2
4
6
8
10
OUTPUT CURRENT (A)
0.04
0.02
0.00
-0.02
-0.04
-0.06
IOUT = 0A
IOUT = 12A
-0.08
-0.04
0
0.06
12
-0.10
3.0
Figure 30. Load Regulation
3.5
4.0
4.5
5.0
INPUT VOLTAGE (V)
5.5
Figure 31. Line Regulation
VOUT (50 mV/Div)
VOUT (10 mV/Div)
IOUT (5A/Div)
100 µs/DIV
Figure 32. Load Transient Response (FSW = 650 kHz)
2 µs/DIV
Figure 33. Output Voltage Ripple
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9.2.2 Typical Application Schematic 2
HTSSOP-20
LO
5,6,7
VIN
PVIN
CIN1
RF
REN1
3
4
CF
11-16
SW
CC3
EN
RFB1
RC2
AVIN
LM21212-2
REN2
2 SS/
TRK
CO1 CO2
19
FB
CSS
VOUT
COMP
18
CC1 RC1
RFB2
CC2
VIN
1
FADJ
RADJ
PGOOD
17
RPGOOD
PGND AGND
8,9,10
20
Figure 34. Typical Application Schematic 2
9.2.2.1 Design Requirements
Table 2. Bill Of Materials (VIN = 4 V- 5.5 V, VOUT = 0.9 V, IOUT = 8 A, FSW = 1 MHz)
ID
DESCRIPTION
VENDOR
PART NUMBER
QUANTITY
CF
CAP, CERM, 1 uF, 10V, +/-10%,
X7R, 0603
MuRata
GRM188R71A105KA61D
1
CIN1, CO1, CO2
CAP, CERM, 100 uF, 6.3V, +/-20%,
X5R, 1206
MuRata
GRM31CR60J107ME39L
3
CC1
CAP, CERM, 1800 pF, 50V, +/-5%,
C0G/NP0, 0603
MuRata
GRM1885C1H182JA01D
1
CC2
CAP, CERM, 68 pF, 50V, +/-5%,
C0G/NP0, 0603
TDK
C1608C0G1H680J
1
CC3
CAP, CERM, 470 pF, 50V, +/-5%,
C0G/NP0, 0603
TDK
C1608C0G1H471J
1
CSS
CAP, CERM, 0.033 uF, 16V, +/-10%,
X7R, 0603
MuRata
GRM188R71C333KA01D
1
LO
Inductor, Shielded Drum Core,
Superflux, 240nH, 20A, 0.001 ohm,
SMD
Wurth Elektronik
744314024
1
RF
RES, 1.0 ohm, 5%, 0.1W, 0603
Vishay-Dale
CRCW06031R00JNEA
1
RC1
RES, 4.87 kohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW06034K87FKEA
1
RC2
RES, 210 ohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW0603210RFKEA
1
REN1, RFB1, RPGOOD
RES, 10k ohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW060310K0FKEA
3
REN2
RES, 19.6 kohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW060319K6FKEA
1
RFB2
RES, 20.0 kohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW060320K0FKEA
1
RADJ
RES, 41.2 kohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW060341K2FKEA
1
9.2.2.2 Detailed Design Procedure
See Detailed Design Procedure
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10 Layout
10.1 Layout Considerations
PC board layout is an important part of DC/DC converter design. Poor board layout can disrupt the performance
of a DC/DC converter and surrounding circuitry by contributing to EMI, ground bounce, and resistive voltage loss
in the traces. These can send erroneous signals to the DC/DC converter resulting in poor regulation or instability.
Good layout can be implemented by following a few simple design rules.
1. Minimize area of switched current loops. In a buck regulator there are two loops where currents are switched
at high slew rates. The first loop starts from the input capacitor, to the regulator PVIN pin, to the regulator
SW pin, to the inductor then out to the output capacitor and load. The second loop starts from the output
capacitor ground, to the regulator GND pins, to the inductor and then out to the load (see Figure 35). To
minimize both loop areas, the input capacitor must be placed as close as possible to the VIN pin. Grounding
for both the input and output capacitor must be close. Ideally, a ground plane must be placed on the top
layer that connects the PGND pins, the exposed pad (EP) of the device, and the ground connections of the
input and output capacitors in a small area near pins 10 and 11 of the device. The inductor must be placed
as close as possible to the SW pin and output capacitor.
2. Minimize the copper area of the switch node. The six SW pins must be routed on a single top plane to the
pad of the inductor. The inductor must be placed as close as possible to the switch pins of the device with a
wide trace to minimize conductive losses. The inductor can be placed on the bottom side of the PCB relative
to the LM21212-2, but care must be taken to not allow any coupling of the magnetic field of the inductor into
the sensitive feedback or compensation traces.
3. Have a solid ground plane between PGND, the EP and the input and output cap. ground connections. The
ground connections for the AGND, compensation, feedback, and soft-start components must be physically
isolated (located near pins 1 and 20) from the power ground plane but a separate ground connection is not
necessary. If not properly handled, poor grounding can result in degraded load regulation or erratic switching
behavior.
4. Carefully route the connection from the VOUT signal to the compensation network. This node is high
impedance and can be susceptible to noise coupling. The trace must be routed away from the SW pin and
inductor to avoid contaminating the feedback signal with switch noise. Additionally,feedback resistors RFB1
and RFB2 must be located near the device to minimize the trace length to FB between these resistors.
5. Make input and output bus connections as wide as possible. This reduces any voltage drops on the input or
output of the converter and can improve efficiency. Voltage accuracy at the load is important so make sure
feedback voltage sense is made at the load. Doing so will correct for voltage drops at the load and provide
the best output accuracy.
6. Provide adequate device heatsinking. For most 12A designs a four layer board is recommended. Use as
many vias as possible to connect the EP to the power plane heatsink. The vias located underneath the EP
will wick solder into them if they are not filled. Complete solder coverage of the EP to the board is required to
achieve the θJA values described in the previous section. Either an adequate amount of solder must be
applied to the EP pad to fill the vias, or the vias must be filled during manufacturing. See the Thermal
Considerations section to ensure enough copper heatsinking area is used to keep the junction temperature
below 125°C.
10.2 Layout Example
LM21212-2
L
VOUT
SW
PVIN
VIN
CIN
COUT
PGND
LOOP1
LOOP2
Figure 35. Schematic of LM21212-2 Highlighting Layout Sensitive Nodes
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10.3 Thermal Considerations
The thermal characteristics of the LM21212-2 are specified using the parameter θJA, which relates the junction
temperature to the ambient temperature. Although the value of θJA is dependant on many variables, it still can be
used to approximate the operating junction temperature of the device.
To obtain an estimate of the device junction temperature, one may use the following relationship:
TJ = PD TJA + TA
where
•
•
•
TJ is the junction temperature in °C
θJA is the junction to ambient thermal resistance for the LM21212-2
TA is the ambient temperature in °C
(20)
and
PD = PIN (1 - Efficiency) - IOUT2 RDCR
where
•
•
PIN is the input power in Watts (PIN = VIN x IIN)
IOUT is the output load current in A
(21)
It is important to always keep the operating junction temperature (TJ) below 125°C for reliable operation. If the
junction temperature exceeds 165°C the device will cycle in and out of thermal shutdown. If thermal shutdown
occurs it is a sign of inadequate heatsinking or excessive power dissipation in the device.
Figure 36, shown below, provides a better approximation of the θJA for a given PCB copper area. The PCB used
in this test consisted of 4 layers: 1 oz. copper was used for the internal layers while the external layers were
plated to 2 oz. copper weight. To provide an optimal thermal connection, a 3 × 4 array of 8 mil. vias under the
thermal pad were used, and an additional sixteen 8 mil. vias under the rest of the device were used to connect
the 4 layers.
THERMAL RESISTANCE ( JA)
30
28
26
24
22
20
18
16
14
12
10
2
3
4
5
6
7
8
2
BOARD AREA (in )
9
10
Figure 36. Thermal Resistance vs PCB Area (4-Layer Board)
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Thermal Considerations (continued)
Figure 37 shows a plot of the maximum ambient temperature vs output current for the typical application circuit
shown in , assuming a θJA value of 24°C/W.
MAX. AMBIENT TEMPERATURE (°C)
125
115
105
95
85
75
0
2
4
6
IOUT(A)
8
10
12
Figure 37. Maximum Ambient Temperature vs Output Current (0 LFM)
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
11.1.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM21212-2 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
E2E is a trademark of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
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11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM21212MH-2/NOPB
ACTIVE
HTSSOP
PWP
20
73
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
LM21212
MH-2
LM21212MHE-2/NOPB
ACTIVE
HTSSOP
PWP
20
250
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
LM21212
MH-2
LM21212MHX-2/NOPB
ACTIVE
HTSSOP
PWP
20
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
LM21212
MH-2
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of