LM25115A
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SNVS501B – FEBRUARY 2007 – REVISED APRIL 2013
LM25115A Secondary Side Post Regulator/DC-DC Converter with Power-Up/Power-Down
Tracking
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FEATURES
DESCRIPTION
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The LM25115A controller contains all of the features
necessary to produce multiple tracking outputs using
the Secondary Side Post Regulation (SSPR)
technique. The SSPR technique develops a highly
efficient and well regulated auxiliary output from the
secondary side switching waveform of an isolated
power converter. LM25115A can be also used as a
standalone DC/DC synchronous buck controller
(Refer to Standalone DC/DC Synchronous Buck
Mode section). Regulation of the auxiliary output
voltage is achieved by leading edge pulse width
modulation (PWM) of the main channel duty cycle.
Leading edge modulation is compatible with either
current mode or voltage mode control of the main
output. The LM25115A drives external high-side and
low-side NMOS power switches configured as a
synchronous buck regulator. A current sense
amplifier provides overload protection and operates
over a wide common mode input range. Additional
features include a low dropout (LDO) bias regulator,
error amplifier, precision reference, adaptive dead
time control of the gate signals and thermal
shutdown.
1
2
Power-up/Power-down Tracking
Self-synchronization to Main Channel Output
Leading Edge Pulse Width Modulation
Valley Current Mode Control
Standalone DC/DC Synchronous Buck Mode
Operates from AC or DC Input up to 42V
Wide 4.5V to 30V Bias Supply Range
Wide 0.75V to 13.5V Output Range.
Top and Bottom Gate Drivers Sink 2.5A Peak
Adaptive Gate Driver Dead-time Control
Wide Bandwidth Error Amplifier (4MHz)
Programmable Soft-start
Thermal Shutdown Protection
TSSOP-16 package
Typical Application Circuit
Phase Signal
Main
Output
3.3V
FEEDBACK
INPUT
Main Converter
PWM Controller
+12V
Main
3.3V
SYNC
VCC
HB
LM25115A
VBIAS
HO
RAMP
HS
TRK/SS
RS
Auxiliary
Output
2.5V
LO
CO
COMP
CS
VOUT
FB
PGND AGND
Figure 1. Simplified Multiple Output Power Converter Utilizing SSPR Technique
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM25115A
SNVS501B – FEBRUARY 2007 – REVISED APRIL 2013
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Connection Diagram
1
2
3
4
5
6
7
8
CS
VBIAS
VOUT
HB
AGND
HO
CO
HS
COMP
FB
VCC
LO
TRK/SS
PGND
RAMP
SYNC
16
15
14
13
12
11
10
9
Figure 2. 16-Lead TSSOP
See Package Numbers PW0016A
2
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PIN DESCRIPTIONS
Pin
Name
Description
Application Information
1
CS
Current Sense amplifier positive input
A low inductance current sense resistor is connected between CS
and VOUT. Current limiting occurs when the differential voltage
between CS and VOUT exceeds 45mV (typical).
2
VOUT
Current sense amplifier negative input
Connected directly to the output voltage. The current sense
amplifier operates over a voltage range from 0V to 13.5V at the
VOUT pin.
3
AGND
Analog ground
Connect directly to the power ground pin (PGND).
4
CO
Current limit output
For normal current limit operation, connect the CO pin to the
COMP pin through a diode. CO pin is connected to ground through
a resistor in series with a capacitor to provide adequate control
loop compensation for the current limit gm amplifier. Leave this pin
open to disable the current limit function.
5
COMP
Compensation. Error amplifier output
COMP pin pull-up is provided by an internal 300uA current source.
6
FB
Feedback. Error amplifier inverting input
Connected to the regulated output through the feedback resistor
divider and compensation components. The non-inverting input of
the error amplifier is internally connected to the SS pin.
7
TRK/SS
Tracking/Soft-start control
Non-inverting input to error amp with 15 µA pull-up current source.
Can be used with capacitor for soft-start or tied to external divider
of a master output for tracking. TRK/SS is the reference input to the
amplifier when the voltage applied to the pin is < 0.75V. For higher
inputs, the internal reference controls the amplifier.
8
RAMP
PWM Ramp signal
An external capacitor connected to this pin sets the ramp slope for
the voltage mode PWM. The RAMP capacitor is charged with a
current that is proportional to current into the SYNC pin. The
capacitor is discharged at the end of every cycle by an internal
MOSFET.
9
SYNC
Synchronization input
A low impedance current input pin. The current into this pin sets the
RAMP capacitor charge current and the frequency of an internal
oscillator that provides a clock for the free-run (DC input) mode .
10
PGND
Power Ground
Connect directly to the analog ground pin (AGND).
11
LO
Low-side gate driver output
Connect to the gate of the low-side synchronous MOSFET through
a short low inductance path.
12
VCC
Output of bias regulator
Nominal 7V output from the internal LDO bias regulator. Locally
decouple to PGND using a low ESR/ESL capacitor located as
close to controller as possible.
13
HS
High-side MOSFET source connection
Connect to negative terminal of the bootstrap capacitor and the
source terminal of the high-side MOSFET.
14
HO
High-side gate driver output
Connect to the gate of high-side MOSFET through a short low
inductance path.
15
HB
High-side gate driver bootstrap rail
Connect to the cathode of the bootstrap diode and the positive
terminal of the bootstrap capacitor. The bootstrap capacitor
supplies current to charge the high-side MOSFET gate and should
be placed as close to controller as possible.
16
VBIAS
Supply Bias Input
Input to the LDO bias regulator and current sense amplifier that
powers internal blocks. Input range of VBIAS is 4.5V to 30V.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS (1) (2)
VBIAS to GND
–0.3V to 32V
VCC to GND
–0.3V to 9V
HS to GND
–1V to 45V
VOUT, CS to GND
– 0.3V to 15V
−0.3V to 7.0V
All other inputs to GND
Storage Temperature Range
–55°C to +150°C
Junction Temperature
+150°C
ESD Rating
HBM (3)
(1)
(2)
(3)
2 kV
Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Operating Ratings are conditions under
which operation of the device is specified. Operating Ratings do not imply ensured performance limits. For ensured performance limits
and associated test conditions, see the Electrical Characteristics tables.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin.
OPERATING RATINGS
VBIAS supply voltage
5V to 30V
VCC supply voltage
5V to 7.5V
HS voltage
0V to 42V
HB voltage
VCC + HS
Operating Junction Temperature
–40°C to +125°C
TYPICAL OPERATING CONDITIONS
Max
Units
Supply Voltage, VBIAS
Parameter
Min
4.5
30
V
Supply Voltage, VCC
4.5
7
V
Supply voltage bypass, CVBIAS
0.1
1
Reference bypass capacitor, CVCC
0.1
1
HB-HS bootstrap capacitor
Typ
µF
10
µF
0.047
µF
SYNC Current Range (VCC = 4.5V)
50
150
µA
RAMP Saw Tooth Amplitude
1
1.75
V
0.75
13.5
V
VOUT regulation voltage (VBIAS min = 3V + VOUT)
ELECTRICAL CHARACTERISTICS (1)
Unless otherwise specified, TJ = –40°C to +125°C, VBIAS = 12V, No Load on LO or HO.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
4
mA
VBIAS SUPPLY
Ibias
VBIAS Supply Current
FSYNC = 200kHz
VCC LOW DROPOUT BIAS REGULATOR
VccReg
VCC Regulation
VCC open circuit. Outputs not switching
VCC Current Limit
( (2))
VCC Under-voltage Lockout Voltage
Positive going VCC
VCC Under-voltage Hysteresis
(1)
(2)
4
6.65
7
7.15
40
4
0.2
0.25
V
mA
4.5
V
0.3
V
Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are specified through correlation
using Statistical Quality Control (SQC) methods. Limits are used to calculate TI’s Average Outgoing Quality Level (AOQL).
Device thermal limitations may limit usable range.
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ELECTRICAL CHARACTERISTICS(1) (continued)
Unless otherwise specified, TJ = –40°C to +125°C, VBIAS = 12V, No Load on LO or HO.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
15
20
µA
TRACK / SOFT-START
SS Pull-up Source
10
SS Discharge Impedance
Ω
140
ERROR AMPLIFIER and FEEDBACK REFERENCE
VREF
GBW
Vio
FB Reference Voltage
Measured at FB pin
FB Input Bias Current
FB = 2V
.737
.750
.763
V
0.2
0.5
µA
COMP Source Current
300
µA
Open Loop Voltage Gain
60
dB
Gain Bandwidth Product
4
MHz
Input Offset Voltage
22
mV
COMP Offset
Threshold for VHO = high RAMP = CS =
VOUT = 0V
2
V
RAMP Offset
Threshold for VHO = high COMP = 1.5V,
CS = VOUT = 0V
1.0
V
CURRENT SENSE AMPLIFIER
Current Sense Amplifier Headroom
Headroom = Vbias – Vout
Vbias= 4.5 V and Vout= 1.5 V
3
Current Sense Amplifier Gain
V
16
V/V
Output DC Offset
1.27
V
Amplifier Bandwidth
500
kHz
Slow ILIMIT Amp Transconductance
5
mA / V
Overall Transconductance
90
CURRENT LIMIT
VCL = VCS - VVOUT
VOUT = 6V and CO/COMP = 1.5V
39
45
51
mV
Slow ILimit Foldback
VCL = VCS - VVOUT
VOUT = 0V and CO/COMP = 1.5V
34
39
46
mV
Fast ILimit Pull-Down Current
Vds = 2V
45
mA
60
mV
VOUT = 6V
VCL = VCS - VVOUT to cause LO to
shutoff
-17
mV
Fast ILimit Threshold
VCLNEG
mA / V
Slow ILimit Threshold
Negative Current Limit
CO Clamp Voltage
5.5
ICO Pull-Up Current
6
6.5
V
15
µA
2.5
kΩ
RAMP GENERATOR
SYNC Input Impedance
SYNC Threshold
End of cycle detection threshold
Free Run Mode Peak Threshold
RAMP peak voltage with dc current
applied to SYNC.
Current Mirror Gain
Ratio of RAMP charge current to SYNC
input current.
Discharge Impedance
20
2.7
µA
2.35
V
3.3
A/A
Ω
100
LOW-SIDE GATE DRIVER
VOLL
LO Low-state Output Voltage
ILO = 100mA
0.15
0.5
V
VOHL
LO High-state Output Voltage
ILO = -100mA, VOHL = VCC -VLO
0.35
0.8
V
LO Rise Time
CLOAD = 1000pF
15
ns
LO Fall Time
CLOAD = 1000pF
12
ns
IOHL
Peak LO Source Current
VLO = 0V
2
A
IOLL
Peak LO Sink Current
VLO = 12V
2.5
A
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ELECTRICAL CHARACTERISTICS(1) (continued)
Unless otherwise specified, TJ = –40°C to +125°C, VBIAS = 12V, No Load on LO or HO.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
V
HIGH-SIDE GATE DRIVER
VOLH
HO Low-state Output Voltage
IHO = 100mA
0.15
0.5
VOHH
HO High-state Output Voltage
IHO = -100mA, VOHH = VHB –VHO
0.35
0.8
HO Rise Time
CLOAD = 1000pF
15
ns
V
HO High-side Fall Time
CLOAD = 1000pF
12
ns
IOHH
Peak HO Source Current
VHO = 0V
2
A
IOLH
Peak HO Sink Current
VHO = 12V
2.5
A
LO Fall to HO Rise Delay
CLOAD = 0
40
ns
HO Fall to LO Rise Delay
CLOAD = 0
50
ns
SYNC Fall to HO Fall Delay
CLOAD = 0
120
ns
SYNC Rise to LO Fall Delay
CLOAD = 0
80
ns
165
°C
25
°C
SWITCHING CHARACTERISITCS
THERMAL SHUTDOWN
TSD
Thermal Shutdown Temp.
150
Thermal Shutdown Hysteresis
THERMAL RESISTANCE
6
θJA
Junction to Ambient
PW Package
125
°C/W
θJA
Junction to Ambient
SDA Package
32
°C/W
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency
vs.
Load Current and Vphase
(VOUT = 2.5V)
VCC Regulator Start-up Characteristics, VCC
vs.
VBIAS
100
16
Vphase = 6V
98
14
VBIAS
12
94 Vphase = 8V
10
VCC (V)
EFFICIENCY (%)
96
92
90
Vphase = 12V
8
VCC
6
88
86
4
84
2
82
0
1
2
3
4
5
6
0
7
0
2
4
6
8
10
12
14
16
LOAD (A)
VBIAS (V)
Figure 3.
Figure 4.
Current Value (CV)
vs.
Current Limit (VCL)
Current Sense Amplifier Gain and Phase
vs.
Frequency
2.5
25
VOUT = 6V
2
5
Gain
-10
20
Offset 1.27V
0.5
0
-20 -10
0
10
20
30
40
50
60
15
-25
10
-40
5
-55
0
-70
100
VCL (mV)
1K
10K
100K
PHASE (o)
1
GAIN (dB)
CV (V)
Phase
16 V/V
1.5
1M
FREQUENCY (Hz)
Figure 5.
Figure 6.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Current Error Amplifier Transconductance
Overall Current Amplifier Transconductance
300
700
200
600
VOUT = 6V
VOUT = 6V
ICO (PA)
100
ICO (PA)
VOUT = 0V
500
5.9 mA/V
0
400
88 mA/V
300
-100
99 mA/V
200
-200
100
-300
1.96 1.98
2
0
30 32
2.1
2.02 2.04 2.06 2.08
34
36
38
40
42
44
46 48
VCS - VOUT (mV)
CV (V)
Figure 7.
Figure 8.
Common Mode Output Voltage
vs.
Positive Current Limit
Common Mode Output Voltage
vs.
Negative Current Limit (Room Temp)
12
14
o
-40 C
12
10
8
125oC
8
VOUT (V)
VOUT (V)
10
6
6
4
4
2
2
27oC
0
0
10
20
30
40
0
-17.8 -17.6 -17.4 -17.2 -17 -16.8 -16.6 -16.4 -16.2
50
VCL (mV)
VCL (mV)
Figure 9.
Figure 10.
VCC Load Regulation to Current Limit
8
7
VCC (V)
6
5
4
3
2
1
0
0
5
10 15 20 25 30 35 40 45 50
ICC (mA)
Figure 11.
8
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BLOCK DIAGRAM
VCC
VBIAS
7V LDO
REGULATOR
VCC
LOGIC
UVLO
7V
THERMAL
LIMIT
SYNC
HB
I SYNC
15 PA
VCC
2.5k
CLK
2.5k
VCC
I SYNC x 3
RAMP
R
Q
S
Q
LEVEL
SHIFT
HO
DRIVER
HS
0.7V
BUFFER
CLK
CRMIX
FB
100k
PWM
COMPARATOR
VCC
40k
VCC
VCC
LO
DRIVER
ERROR AMP
(Sink Only)
15 PA
ADAPTIVE
DEAD TIME
DELAY
55k
300 PA
TRK/SS
1V
NEGATIVE
CURRENT
DETECTOR
PGND
0.75V
AGND
ENABLE
ILIMIT SLOW
Gm = 5 mA/V
COMP
CURRENT SENSE AMP
Gain = 16
CS
VBIAS
CV
ILIMIT FAST
2V
1.27V
VCC
VOUT
2.35V
15 PA
CO
VINT
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DETAILED OPERATING DESCRIPTION
The LM25115A controller contains all of the features necessary to implement multiple output power converters
utilizing the Secondary Side Post Regulation (SSPR) technique. The SSPR technique develops a highly efficient
and well regulated auxiliary output from the secondary side switching waveform of an isolated power converter.
Regulation of the auxiliary output voltage is achieved by leading edge pulse width modulation (PWM) of the main
channel duty cycle. Leading edge modulation is compatible with either current mode or voltage mode control of
the main output. The LM25115A drives external high-side and low-side NMOS power switches configured as a
synchronous buck regulator. A current sense amplifier provides overload protection and operates over a wide
common mode input range from 0V to 13.5V. Additional features include a low dropout (LDO) bias regulator,
error amplifier, precision reference, adaptive dead time control of the gate driver signals and thermal shutdown.
Low Drop-Out Bias Regulator (VCC)
The LM25115A contains an internal LDO regulator that operates over an input supply range from 4.5V to 30V.
The output of the regulator at the VCC pin is nominally regulated at 7V and is internally current limited to 40mA.
VCC is the main supply to the internal logic, PWM controller, and gate driver circuits. When power is applied to
the VBIAS pin, the regulator is enabled and sources current into an external capacitor connected to the VCC pin.
The recommended output capacitor range for the VCC regulator is 0.1uF to 100uF. When the voltage at the VCC
pin reaches the VCC under-voltage lockout threshold of 4.25V, the controller is enabled. The controller is
disabled if VCC falls below 4.0V (250mV hysteresis). In applications where an appropriate regulated dc bias
supply is available, the LM25115A controller can be powered directly through the VCC pin instead of the VBIAS
pin. In this configuration, it is recommended that the VCC and the VBIAS pins be connected together such that
the external bias voltage is applied to both pins. The allowable VCC range when biased from an external supply
is 4.5V to 7V.
Synchronization (SYNC) and Feed-Forward (RAMP)
The pulsing “phase signal” from the main converter synchronizes the PWM ramp and gate drive outputs of the
LM25115A. The phase signal is the square wave output from the transformer secondary winding before
rectification (Figure 1). A resistor connected from the phase signal to the low impedance SYNC pin produces a
square wave current (ISYNC) as shown in Figure 12. A current comparator at the SYNC input monitors ISYNC
relative to an internal 15µA reference. When ISYNC exceeds 15µA, the internal clock signal (CLK) is reset and the
capacitor connected to the RAMP begins to charge. The current source that charges the RAMP capacitor is
equal to 3 times the ISYNC current. The falling edge of the phase signal sets the CLK signal and discharges the
RAMP capacitor until the next rising edge of the phase signal. The RAMP capacitor is discharged to ground by a
low impedance (100Ω) n-channel MOSFET. The input impedance at SYNC pin is 2.5kΩ which is normally much
smaller than the external SYNC pin resistance.
The RAMP and SYNC functions illustrated in Figure 12 provide line voltage feed-forward to improve the
regulation of the auxiliary output when the input voltage of the main converter changes. Varying the input voltage
to the main converter produces proportional variations in amplitude of the phase signal. The main channel PWM
controller adjusts the pulse width of the phase signal to maintain constant volt*seconds and a regulated main
output as shown in Figure 13. The variation of the phase signal amplitude and duration are reflected in the slope
and duty cycle of the RAMP signal of the LM25115A (ISYNC α phase signal amplitude). As a result, the duty cycle
of the LM25115A is automatically adjusted to regulate the auxiliary output voltage with virtually no change in the
PWM threshold voltage. Transient line regulation is improved because the PWM duty cycle of the auxiliary
converter is immediately corrected, independent of the delays of the voltage regulation loop.
10
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Phase
Signal
RSYNC
SYNC
CLK
15 PA
Isync
2.5k
2.5k
Isync x 3
RAMP
BUFFER
CRAMP
CLK
Figure 12. Line Feed-Forward Diagram
12V
Phase signal
6V
Main Output = 3.3V
RAMP pin
PWM Threshold
12V
HS pin
6V
Secondary Output = 2.5V
Figure 13. Line Feed-Forward Waveforms
The recommended SYNC input current range is 50µA to 150µA. The SYNC pin resistor (RSYNC) should be
selected to set the SYNC current (ISYNC) to 150µA with the maximum phase signal amplitude, VPHASE(max). This
will ensure that ISYNC stays within the recommended range over a 3:1 change in phase signal amplitude. The
SYNC pin resistor is therefore:
RSYNC = (VPHASE(max) / 150µA) - 2.5kΩ
Once ISYNC has been established by selecting RSYNC, the RAMP signal slope/amplitude may be programmed by
selecting the proper RAMP pin capacitor value. The RAMP signal slope should be selected to provide adequate
slope compensation for the Valley current mode control scheme (Please refer to the Valley Current Mode Control
section). The recommended peak amplitude of the ramp waveform is 1.75V.
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Error Amplifier and Soft-Start (FB, CO, COMP & TRK/SS)
An internal wide bandwidth error amplifier is provided within the LM25115A for voltage feedback to the PWM
controller. The amplifier’s inverting input is connected to the FB pin. The output of the auxiliary converter is
regulated by connecting a voltage setting resistor divider between the output and the FB pin. Loop compensation
networks are connected between the FB pin and the error amplifier output (COMP). The amplifier has two noninverting inputs. The first non-inverting input connects to a 0.75V bandgap reference while the second noninverting input connects to the TRK/SS pin and it has 15 µA pull-up current source. The TRK/SS pin can be tied
to an external resistor divider from the master output for tracking, or it can be tied to a capacitor for soft-start .
TRK/SS is the reference input to the amplifier when the voltage applied to the pin is < 0.75V. For higher inputs,
the internal reference controls the amplifier. When the VCC voltage is below the UVLO threshold, the TRK/SS
pin is discharged to ground. When VCC rises and exceeds the positive going UVLO threshold (4.25V), the
TRK/SS pin is released and allowed to rise. If an external capacitor is connected to the TRK/SS pin, it will be
charged by the internal 15uA pull-up current source to gradually increase the non-inverting input of the error
amplifier to 0.75V. During start-up, the output of the LM25115A converter will follow the following equation:
VOUT(t) = VOUT(final) x15 µA x t /(.75 Vx Css )
where
•
•
Css = external Soft-Start capacitor
VOUT(final) = regulator output set point
Pull-up current for the error amplifier output is provided by an internal 300µA current source. The PWM threshold
signal at the COMP pin can be controlled by either the open drain error amplifier or the open drain current
amplifier connected through the CO pin to COMP. Since the internal error amplifier is configured as an open
drain output it can be disabled by connecting FB to ground. The current sense amplifier and current limiting
function will be described in a later section.
Power-Up/Power-Down Tracking
The LM25115A can track the output of a master power supply during soft start by connecting a resistor divider to
the TRACK pin (Figure 14). Therefore, the output voltage slew rate of the LM25115A will be controlled by the
master supply for loads that require precise sequencing. In order to track properly the output voltage of the
LM25115A must be lower than the output voltage of the master supply.
One way to use the tracking feature is to design the tracking resistor divider so that the master supply output
voltage (VOUT1) and the LM25115A output voltage (VOUT2) both rise together and reach their target values at
the same time. For this case, the equation governing the values of the tracking divider resistors RT1 and RT2 is:
RT1 =
0.8 x RT2
VOUT1 - 0.8
A value of 10kΩ (1%) is recommended for RT2 as a good compromise between high precision and low quiescent
current through the divider. If the master supply voltage was 3.3V and the LM25115A output voltage was 2.5 V,
then the value of RT1 needed to give the two supplies identical soft start times would be 2.94 kΩ (1%). The
timing diagram and waveforms for the equal soft start time configuration are shown in Figure 15 and Figure 16.
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3.3V
Master
Power
Supply
V
OUT1
R
V
T2
V
SS
VOUT1
OUT2
2.5V
TRACK
LM5115A
R
T1
R
VOUT2
FB2
FB
V
FB
R FB1
Figure 14.
Figure 15.
Vphase = 10V
CH1 = Master output, 1V/Div
CH2 = COMP, 5/Div
CH3 = Iout, 1A/Div
CH4 = SSPR Output (Slave), 1V/Div
Horizontal Resolution= 200 µs/Div
Figure 16. Tracking with Equal Soft Start Time
Alternatively, the tracking feature can be used to create equal slew rates between the output voltages of the
master supply and the LM25115A. This method ensures that the output voltage of the LM25115A always
reaches regulation before the output voltage of the master supply. In this case, the tracking resistors can be
determined based on the following equation:
RT1 =
0.8 x RT2
VOUT2 - 0.8
Again, a value of 10kΩ 1% is recommended for RT2. For the case of VOUT1 = 3.3V and VOUT2 = 2.5V, RT1
should be 4.32 kΩ 1%. The timing diagram and the waveforms for equal slew rates configuration are shown in
Figure 17 and Figure 18.
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3.3V
2.5V
VOUT1
2.5V
VOUT2
Figure 17.
Vphase = 10V
CH1 = Master output, 1V/Div
CH2 = COMP, 5/Div
CH3 = Iout, 1A/Div
CH4 = SSPR Output (Slave), 1V/Div
Horizontal Resolution= 200 µs/Div
Figure 18. Tracking with Equal Slew Rate
Leading Edge Pulse Width Modulation
Unlike conventional voltage mode controllers, the LM25115A implements leading edge pulse width modulation. A
current source equal to 3 times the ISYNC current is used to charge the capacitor connected to the RAMP pin as
shown in Figure 19. The ramp signal and the output of the error amplifier (COMP) are combined through a
resistor network to produce a voltage ramp with variable dc offset (CRMIX in Figure 19). The high-side MOSFET
which drives the HS pin is held in the off state at the beginning of the phase signal. When the voltage of CRMIX
exceeds the internal threshold voltage CV, the PWM comparator turns on the high-side MOSFET. The HS pin
rises and the MOSFET delivers current from the main converter phase signal to the output of the auxiliary
regulator. The PWM cycle ends when the phase signal falls and power is no longer supplied to the drain of the
high-side MOSFET.
Leading edge modulation of the auxiliary PWM controller is required if the main converter uses peak current
mode control. If trailing edge modulation were used, the additional load on the transformer secondary from the
auxiliary channel would be drawn only during the first portion of the phase signal pulse. Referring to Figure 20,
the turn-off of the high- side MOSFET of the auxiliary regulator would create a non-monotonic negative step in
the transformer current. This negative current step would produce instability in a peak current mode controller.
With leading edge modulation, the additional load presented by the auxiliary regulator on the transformer
secondary will be present during the latter portion of the phase signal. This positive step in the phase signal
current can be accommodated by a peak current mode controller without instability.
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Isync x 3
0.7V
RAMP
Phase or CLK
BUFFER
CLK
CRAMP
55k
RAMP
CRMIX
PWM
CV
40k
COMP
CV
ERROR
AMP
FB
CRMIX
100k
HS
TRK/SS
Leading Edge
Modulation
0.75V
Figure 19. Synchronization and Leading Edge Modulation
Main
PWM
Main
PWM
Auxilary
PWM
Trailing Edge
Modulation
Auxilary
PWM
Leading Edge
Modulation
Peak Current
Threshold
Peak Current
Threshold
Transformer
Current
Transformer
Current
Figure 20. Leading versus Trailing Edge Modulation
Valley Current Mode Control
The LM25115A controller uniquely utilizes the elements and benefits of valley current mode control in
conjunction with leading edge modulation to correct changes in output voltage due to line and load transients.
Contrary to peak current mode control, valley current mode control turns on the high-side MOSFET when the
inductor valley current reaches a programmable threshold. This programmable threshold (CRMIX) is the sum of
the output of voltage error amplifier and the RAMP signal generated at the RAMP pin. Valley current mode
control experiences sub-harmonic oscillation when the duty ratio, D, is less than or equal to 50%. Therefore,
adequate slope compensation is needed for the proper operation across the full range of the duty ratio. The
RAMP signal is proportional to the input voltage and it provides the required slope compensation for the valley
current mode scheme. The desired RAMP pin capacitance can be calculated from the following equation:
CRAMP = (0.05 x L) /(RSYNC x RSENSE)
where
•
•
•
L is the power inductor
RSYNC is the SYNC pin resistor
RSENSE is the current sense resistor
The current sense amplifier shown in Figure 21 monitors the inductor current as it flows through a sense resistor
connected between CS and VOUT. The voltage gain of the sense amplifier is nominally equal to 16. The current
sense output signal is shifted by 1.27V to produce the internal CV reference signal. The CV signal is applied to
the negative input of the PWM comparator and compared to CRMIX as illustrated in Figure 21. Therefore when
CRMIX exceeds the PWM threshold (CV), the PWM comparator turns on the high-side MOSFET. Insure that the
Vbias voltage is at least 3V above the regulated output voltage (VOUT) to provide enough headroom for the
current sense amplifier.
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Valley current mode control improves the control loop stability and bandwidth. It also eliminates the R-C lead
network in the feedback path that is normally required with voltage mode control (Figure 22). Eliminating the lead
network not only simplifies the compensation, but also reduces sensitivity to output noise that could pass through
the lead network to the error amplifier.
The design of the voltage feedback path through the error amp begins with the selection of R1 and R2 in
Figure 22 to set the regulated output voltage. The steady state output voltage after soft-start is determined by the
following equation:
VOUT(final) = 0.75V x (1+R1/R2)
The parallel impedance of the R1, R2 resistor divider should be approximately 2kΩ (between 0.5kΩ and 5kΩ).
Lower resistance values may not be properly driven by the error amplifier output and higher feedback resistances
can introduce noise sensitivity. The next step in the design process is selection of R3, which sets the ac gain of
the error amplifier.
The capacitor C1 is connected in series with R3 to increase the dc gain of the voltage regulation loop and
improve output voltage accuracy. The corner frequency set by R3 x C1 should be less than 1/10th of the crossover frequency of the overall converter such that capacitor C1 does not add phase lag at the crossover
frequency.
Negative Current
Comparator
Low Side
Enable
1V
CS
1.27V
VOUT
ILIMIT SLOW
Gm = 5 mA/V
Vbias
ILIMIT FAST
CO
2V
VCL
AV=16
2.35V
Current
Sense Amp
PWM
Comparator
CV
to PWM
Latch
CRMIX
FB
TRK/SS
0.75V
ERROR
AMP
COMP
Figure 21. Current Sensing and Limiting
VOUT
No Lead
Network
Required
R1
FB
PWM
ERROR
AMP
CV
40k
TRK/SS
0.75V
100k
COMP
R3
C1
R2
Figure 22. Voltage Sensing and Feedback
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Current Limiting (CS, CO and VOUT)
Current limiting is implemented through the current sense amplifier as illustrated in Figure 21. The current sense
amplifier monitors the inductor current that flows through a sense resistor connected between CS and VOUT.
The voltage gain of the current sense amplifier is nominally equal to 16. The output of current sense signal is
shifted by 1.27V to produce the internal CV reference signal. The CV signal drives two current limit amplifiers.
Both of the current limit amplifiers have open drain (sink only) output stages which are connected to the CO pin.
The CO pin is typically connected to the COMP pin through a diode (the cathode is connected to the CO pin and
the anode is connected to the COMP pin). The slow current limit amplifier has a nominal transconductance of 5
mA/V and provides constant current mode operation at the desired current limit set point. The fast current limit
amplifier has nominal current pull-down capability of 100mA and provides protection against fast over-current
conditions. During normal operation, the voltage error amplifier controls the COMP pin voltage which adjusts the
PWM duty cycle by varying the internal CRMIX level. However when the current sense input voltage, VCL,
exceeds 45mV, the slow current limit amplifier gradually pulls down on COMP through the CO pin. Pulling COMP
low reduces the CRMIX signal and thereby reducing the operating duty cycle. By controlling the operating duty
cycle, the slow current limit amplifier will force a constant current mode of operation at the desired current limit
set point (Figure 23). A resistor in series with a capacitor are connected from the CO Pin to ground to provide
adequate control loop compensation for the slow current limit (Figure 21). The desired current limit set point,
ILimit, can be programmed by selecting the proper current sense resistor, RSENSE,using the following equation:
RSENSE = 0.045 V/ ILimit
In the event that the current sense input voltage, VCL, exceeds 60mV, the fast current limit amplifier will pull down
hard on COMP through the CO pin. This will reduce the CRMIX signal to a voltage below the CV signal level.
Therefore, the PWM comparator will inhibit output pulses. Once the fault condition is removed, the fast current
limit amplifier will release COMP. Therefore, the CRMIX signal will increase to a normal operating threshold and
the switching will resume (Figure 24). A current limit fold-back feature is provided by the LM25115A to reduce the
peak output current delivered to a shorted load. When the common mode input voltage to the current sense
amplifier (CS and VOUT pins) falls below 2V, the current limit threshold is reduced from the normal level. At
common mode voltages > 2V, the current limit threshold is nominally 45mV. When VOUT is reduced to 0 V the
current limit threshold drops to 39mV to reduce stress on the inductor and power MOSFETs.
Vphase=10V
CH1 = CO, 5V/Div
CH2 = COMP, 5/Div
CH3 = Iout, 5A/Div
CH4 = SSPR Switch Signal, 5V/Div
Horizontal Resolution= 2 µs/Div
Figure 23. SSPR Steady State Current Limit (Output Shorted)
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Vphase=10V
CH1 = CO, 5V/Div
CH2 = COMP, 5/Div
CH3 = Iout, 10A/Div
CH4 = SSPR Switch Signal, 4V/Div
Horizontal Resolution= 20 µs/Div
Figure 24. SSPR Short Circuit Transient (No-Load to Short-Circuit)
Negative Current Limit
Under certain conditions synchronous buck regulators are capable of sinking current from the output capacitors.
This energy is stored in the inductor and returned to the input source. The LM25115A detects this current
reversal by detecting a negative voltage being developed across the current sense resistor. The intent of this
negative current comparator is to protect the low-side MOSFET from excessive currents. Excessive negative
current can also lead to a large positive voltage spike on the HS pin at the turn-off of the low-side MOSFET. This
voltage spike may damage the chip if its magnitude exceeds the maximum voltage rating of the part. The
negative current comparator threshold is sufficiently negative to allow inductor current to reverse at no load or
light load conditions. It is not intended to support discontinuous conduction mode with diode emulation by the
low-side MOSFET. The negative current comparator shown in Figure 21 monitors the CV signal and compares
this signal to a fixed 1V threshold. This corresponds to a negative VCL voltage between CS and VOUT of -17mV.
The negative current limit comparator turns off the low-side MOSFET for the remainder of the cycle when the VCL
input falls below this threshold.
Gate Driver Outputs (HO & LO)
The LM25115A provides two gate driver outputs, the floating high-side gate driver HO and the synchronous
rectifier low-side driver LO. The low-side driver is powered directly by the VCC regulator. The high-side gate
driver is powered from a bootstrap capacitor connected between HB and HS. An external diode connected
between VCC and HB charges the bootstrap capacitor when the HS is low. When the high-side MOSFET is
turned on, HB rises with HS to a peak voltage equal to VCC + VHS - VD where VD is the forward drop of the
external bootstrap diode. Both output drivers have adaptive dead-time control to avoid shoot through currents.
The adaptive dead-time control circuit monitors the state of each driver to ensure that one MOSFET is turned off
before the other is turned on. The HB and VCC capacitors should be placed close to the pins of the LM25115A
to minimize voltage transients due to parasitic inductances and the high peak output currents of the drivers. The
recommended range of the HB capacitor is 0.047µF to 0.22µF.
Both drivers are controlled by the PWM logic signal from the PWM latch. When the phase signal is low, the
outputs are held in the reset state with the low-side MOSFET on and the high-side MOSFET off. When the phase
signal switches to the high state, the PWM latch reset signal is de-asserted. The high-side MOSFET remains off
until the PWM latch is set by the PWM comparator (CRMIX > CV as shown in Figure 19). When the PWM latch
is set, the LO driver turns off the low-side MOSFET and the HO driver turns on the high-side MOSFET. The highside pulse is terminated when the phase signal falls and SYNC input comparator resets the PWM latch.
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Thermal Protection
Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event the maximum junction
temperature limit is exceeded. When activated, typically at 165 degrees Celsius, the controller is forced into a low
power standby state with the output drivers and the bias regulator disabled. The device will restart when the
junction temperature falls below the thermal shutdown hysteresis, which is typically 25 degrees. The thermal
protection feature is provided to prevent catastrophic failures from accidental device overheating.
Standalone DC/DC Synchronous Buck Mode
The LM25115A can be configured as a standalone DC/DC synchronous buck controller. In this mode the
LM25115A uses leading edge modulation in conjunction with valley current mode control to control the
synchronous buck power stage. The internal oscillator within the LM25115A sets the clock frequency for the high
and low-side drivers of the external synchronous buck power MOSFETs . The clock frequency in the
synchronous buck mode is programmed by the SYNC pin resistor and RAMP pin capacitor. Connecting a
resistor between a dc bias supply and the SYNC pin produces a current, ISYNC, which sets the charging current of
the RAMP pin capacitor. The RAMP capacitor is charged until its voltage reaches the peak ramp threshold of
2.25V. The RAMP capacitor is then discharged for 300ns before beginning a new PWM cycle. The 300ns reset
time of the RAMP pin sets the minimum off-time of the PWM controller in this mode. The internal clock frequency
in the synchronous buck mode is set by ISYNC, the ramp capacitor, the peak ramp threshold, and the 300ns
deadtime.
FCLK ≊ 1 / ((CRAMP x 2.25V) / (ISYNC x 3) + 300ns)
See the LM5115 dc evaluation board application note (AN-1367, literature number SNVA106) for more details on
the synchronous buck mode. Please note that LM25115A is similar to LM5115 except for the tracking feature.
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Application Circuit
(Inputs from LM5025 Forward Active Clamp Converter, 36V to 78V)
Figure 25. LM25115A Secondary Side Post Regulator
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REVISION HISTORY
Changes from Revision A (March 2013) to Revision B
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 19
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM25115AMT/NOPB
ACTIVE
TSSOP
PW
16
92
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L25115
AMT
LM25115AMTX/NOPB
ACTIVE
TSSOP
PW
16
2500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L25115
AMT
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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