Sample &
Buy
Product
Folder
Support &
Community
Tools &
Software
Technical
Documents
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
LM26001/-Q1 1.5-A Switching Regulator With High-Efficiency Sleep Mode
1 Features
3 Description
•
The LM26001 is a switching regulator designed for
the high-efficiency requirements of applications with
standby modes. The device features a low-current
sleep mode to maintain efficiency under light-load
conditions and current-mode control for accurate
regulation over a wide input voltage range. Quiescent
current is reduced to 10 µA typically in shutdown
mode and less than 40 µA in sleep mode. Forced
PWM mode is also available to disable sleep mode.
1
•
•
•
•
•
•
•
•
•
•
•
•
•
•
LM26001-Q1 is an Automotive-Grade Product that
is AEC-Q100 Grade 1 Qualified (–40°C to +125°C
Operating Junction Temperature)
High-Efficiency Sleep Mode
40-µA Typical Iq in Sleep Mode
10-µA Typical Iq in Shutdown Mode
3.0-V Minimum Input Voltage
4.0-V to 38-V Continuous Input Range
1.5% Reference Accuracy
Cycle-by-Cycle Current Limit
Adjustable Frequency (150 kHz to 500 kHz)
Synchronizable to an External Clock
Power Good Flag
Forced PWM Function
Adjustable Soft-Start
HTSSOP-16 Exposed Pad Package
Thermal Shut Down
The LM26001 can deliver up to 1.5 A of continuous
load current with a fixed current limit, through the
internal N-channel switch. The part has a wide input
voltage range of 4.0 V to 38 V and can operate with
input voltages as low as 3 V during line transients.
Operating frequency is adjustable from 150 kHz to
500 kHz with a single resistor and can be
synchronized to an external clock.
Other features include Power Good, adjustable softstart, enable pin, input undervoltage protection, and
an internal bootstrap diode for reduced component
count.
2 Applications
•
•
•
•
•
Device Information(1)
Automotive Telematics
Navigation Systems
In-Dash Instrumentation
Battery-Powered Applications
Standby Power for Home Gateways/Set-top
Boxes
PART NUMBER
LM26001
LM26001-Q1
PACKAGE
BODY SIZE (NOM)
HTSSOP (16)
5.00 mm x 4.40 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
Typical Application Circuit
VIN
1
2
C1
3
4
R4
EN
VDD
11
5
SYNC
VIN
VBIAS
VIN
SW
PGOOD
SW
EN
BOOT
LM26001
SYNC
R3
L
14
+
C4
D1
R1
C6
7
FB
SS
COMP
FREQ
VDD
FPWM
GND
C5
10
VOUT
15
9
R6
12
16
EP
6
13
C8
R2
8
C3
R5
17
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
4
4
4
5
5
5
8
Absolute Maximum Ratings ......................................
ESD Ratings - LM26001 ...........................................
ESD Ratings - LM26001-Q1 .....................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 11
7.1 Overview ................................................................. 11
7.2 Functional Block Diagram ....................................... 11
7.3 Feature Description................................................. 12
7.4 Device Functional Modes........................................ 16
8
Applications and Implementation ...................... 17
8.1 Application Information............................................ 17
8.2 Typical Application .................................................. 17
9 Power Supply Recommendations...................... 23
10 Layout................................................................... 23
10.1 Layout Guidelines ................................................. 23
10.2 Layout Example .................................................... 24
10.3 Thermal Considerations and TSD......................... 24
11 Device and Documentation Support ................. 25
11.1
11.2
11.3
11.4
11.5
Documentation Support ........................................
Related Links ........................................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
25
25
25
25
25
12 Mechanical, Packaging, and Orderable
Information ........................................................... 25
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision H (November 2014) to Revision I
Page
•
Update made to the Power Dissipation description in Section 6.1 ....................................................................................... 4
•
Changed ESD Ratings to ± and moved storage temp to Absolute Maximum Ratings .......................................................... 4
Changes from Revision G (April 2013) to Revision H
•
Page
Added Pin Configuration and Functions section, Handling Rating table, Feature Description section, Device
Functional Modes, Application and Implementation section, Power Supply Recommendations section, Layout
section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information
section ................................................................................................................................................................................... 1
Changes from Revision F (April 2013) to Revision G
•
2
Page
Changed layout of National Data Sheet to TI format ........................................................................................................... 24
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
5 Pin Configuration and Functions
16-Pin
HTSSOP Package
Top View
VIN 1
16 SW
VIN 2
15 SW
PGOOD 3
14 BOOT
EN 4
13 VDD
SS 5
12 VBIAS
COMP 6
FB 7
11 SYNC
10 FPWM
9 FREQ
GND 8
Exposed Pad
Connect to GND
Pin Functions
PIN
I/O
DESCRIPTION
NO.
NAME
1
VIN
A
Power supply input
2
VIN
A
Power supply input
3
PGOOD
O
Power Good pin. An open-drain output which goes high when the output voltage is greater than
92% of nominal.
4
EN
I
Enable is an analog level input pin. When pulled below 0.8 V, the device enters shutdown mode.
5
SS
A
Soft-start pin. Connect a capacitor from this pin to GND to set the soft-start time.
6
COMP
A
Compensation pin. Connect to a resistor capacitor pair to compensate the control loop.
7
FB
A
Feedback pin. Connect to a resistor divider between Vout and GND to set output voltage.
8
GND
G
Ground
9
FREQ
A
Frequency adjust pin. Connect a resistor from this pin to GND to set the operating frequency.
10
FPWM
I
FPWM is a logic level input pin. For normal operation, connect to GND. When pulled high, sleep
mode operation is disabled.
11
SYNC
I
Frequency synchronization pin. Connect to an external clock signal for synchronized operation.
SYNC must be pulled low for non-synchronized operation.
12
VBIAS
A
Connect to an external 3-V or greater supply to bypass the internal regulator for improved efficiency.
If not used, VBIAS should be tied to GND.
13
VDD
A
The output of the internal regulator. Bypass with a minimum 1.0-µF capacitor.
14
BOOT
A
Bootstrap capacitor pin. Connect a 0.1-µF minimum ceramic capacitor from this pin to SW to
generate the gate drive bootstrap voltage.
15
SW
A
Switch pin. The source of the internal N-channel switch.
16
SW
A
Switch pin. The source of the internal N-channel switch.
EP
EP
G
Exposed Pad thermal connection. Connect to GND.
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
3
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
6 Specifications
6.1 Absolute Maximum Ratings (1) (2)
Voltages
from the
indicated
pins to
GND
MIN
MAX
UNIT
VIN
–0.3
40
V
SW (3)
–0.5
40
V
VDD
–0.3
7
V
VBIAS
–0.3
10
V
FB
–0.3
6
V
SW-0.3
SW+7
V
PGOOD
–0.3
7
V
FREQ
–0.3
7
V
SYNC
–0.3
7
V
EN
–0.3
40
V
FPWM
–0.3
y7
V
SS
–0.3
7
V
2.6
W
215
°C
220
°C
150
°C
BOOT
Power Dissipation (4) (5)
Recomme Vapor Phase (70s)
nded Lead
Infrared (15s)
Temperatu
re
Storage
Tstg
temperatur
e
(1)
(2)
(3)
(4)
(5)
–65
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Recommended Operating Conditions indicate
conditions for which the device is intended to be functional, but do not ensure specific performance limits. For ensured specifications
and test conditions, see the Electrical Characteristics.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
The absolute maximum specification applies to DC voltage. An extended negative voltage limit of -2V applies for a pulse of up to 1 µs,
and –1 V for a pulse of up to 20 µs.
The maximum allowable power dissipation is a function of the maximum junction temperature, TJ_MAX, the junction-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature is calculated
using: PD_MAX = (TJ_MAX - TA) /θJA. The maximum power dissipation of 2.6W is determined using TA = 25°C, θJA = 38°C/W, and
TJ_MAX = 125°C. The number stated here reflects the maximum power dissipation for the package and not the device.
For Device Power Dissipation, please refer to section 10.3.
6.2 ESD Ratings - LM26001
VALUE
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins
V(ESD)
Electrostatic discharge
(1)
(1)
(2)
±2
±1
kV
± 200
V
Charged device model (CDM), per JEDEC specification JESD22-C101,
all pins (2)
Machine model
UNIT
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 ESD Ratings - LM26001-Q1
VALUE
Human body model (HBM), per AEC Q100-002 (1)
V(ESD)
Electrostatic discharge
Charged device model (CDM), per AEC
Q100-011
Corner pins (1, 8, 9, and
16)
Other pins
Machine model
(1)
4
UNIT
±2
±1
kV
±1
± 200
V
AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
6.4 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
Operating Junction Temp.
–40
125
°C
Supply Voltage (1)
3.0
38
V
(1)
UNIT
Below 4.0-V input, power dissipation may increase due to increased RDS(ON). Therefore, a minimum input voltage of 4.0 V is required to
operate continuously within specification. A minimum of 3.9 V (typical) is also required for startup.
6.5 Thermal Information
LM26001
THERMAL METRIC
(1)
PWP
UNIT
16 PINS
RθJA
Junction-to-ambient thermal resistance
38.8
RθJC(top)
Junction-to-case (top) thermal resistance
23.0
RθJB
Junction-to-board thermal resistance
16.7
ψJT
Junction-to-top characterization parameter
0.6
ψJB
Junction-to-board characterization parameter
16.4
RθJC(bot)
Junction-to-case (bottom) thermal resistance
1.7
(1)
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
6.6 Electrical Characteristics
Unless otherwise stated, Vin=12 V. Minimum and Maximum limits are ensured through test, design, or statistical correlation.
Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. (1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SYSTEM
ISD (2)
Shutdown Current
EN = 0 V
10.8
EN = 0 V, –40°C ≤ TJ ≤ 125°C
Iq_Sleep_VB (2)
Quiescent Current
Sleep mode, VBIAS = 5 V
38
Sleep mode, VBIAS = 5 V, –40°C
≤ TJ ≤ 125°C
Iq_Sleep_VDD
Quiescent Current
µA
20
µA
70
Sleep mode, VBIAS = GND
75
Sleep mode, VBIAS = GND,
–40°C ≤ TJ ≤ 125°C
µA
125
Iq_PWM_VB
Quiescent Current
PWM mode, VBIAS = 5 V
150
230
µA
Iq_PWM_VDD
Quiescent Current
PWM mode, VBIAS = GND
0.65
0.85
mA
IBIAS_Sleep (2)
Bias Current
Sleep mode, VBIAS = 5 V
33
Sleep mode, VBIAS = 5 V, –40°C
≤ TJ ≤ 125°C
IBIAS_PWM
Bias Current
PWM mode, VBIAS = 5 V
VFB
Feedback Voltage
5 V < Vin < 38 V
5 V < Vin < 38 V, –40°C ≤ TJ ≤
125°C
IFB
FB Bias Current
ΔVOUT/ΔVIN
Vout line regulation
5 V < Vin < 38 V
ΔVOUT/ΔIOUT
Vout load regulation
0.8 V < VCOMP < 1.15 V
VDD
VDD output voltage
7 V < Vin < 35 V, IVDD= 0 mA to 5
mA
(2)
0.5
0.70
1.234
1.2155
mA
V
1.2525
±200
7 V < Vin < 35 V, IVDD= 0 mA to 5
mA, –40°C ≤ TJ ≤ 125°C
(1)
µA
85
0.001
nA
%/V
0.07%
5.95
5.50
V
6.50
All room temperature limits are 100% production tested. All limits at temperature extremes are ensured through correlation using
standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
Iq and ISD specify the current into the VIN pin. IBIAS is the current into the VBIAS pin when the VBIAS voltage is greater than 3 V. All
quiescent current specifications apply to non-switching operation.
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
5
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
Electrical Characteristics (continued)
Unless otherwise stated, Vin=12 V. Minimum and Maximum limits are ensured through test, design, or statistical correlation.
Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.(1)
PARAMETER
TEST CONDITIONS
ISS_Source
Soft-start source current
Vbias_th
VBIAS On Voltage
Specified at IBIAS = 92.5% of full
value
Switch on Resistance
Isw = 1A
MIN
TYP
MAX
2.2
–40°C ≤ TJ ≤ 125°C
1.5
2.64
UNIT
µA
4.6
2.9
3.07
V
SWITCHING
RDS(ON)
Isw_off
Switch off state leakage current
0.12
Vin = 38 V, VSW = 0 V
0.42
0.002
Vin = 38 V, VSW = 0 V, –40°C ≤
TJ ≤ 125°C
fsw
Switching Frequency
VFREQ
FREQ voltage
fSW range
Switching Frequency range
–40°C ≤ TJ ≤ 125°C
VSYNC
Sync pin threshold
SYNC rising
Ω
0.2
Isw = 1A, –40°C ≤ TJ ≤ 125°C
µA
5.0
RFREQ = 62k, 124k, 240k
±10%
1.0
150
V
500
1.2
SYNC rising, –40°C ≤ TJ ≤ 125°C
kHz
V
1.6
SYNC falling
1.1
SYNC falling, –40°C ≤ TJ ≤ 125°C
0.8
Sync pin hysteresis
114
mV
ISYNC
SYNC leakage current
FSYNC_UP
Upper frequency synchronization range As compared to nominal fSW,
–40°C ≤ TJ ≤ 125°C
6
30%
nA
FSYNC_DN
Lower frequency synchronization range As compared to nominal fSW,
–40°C ≤ TJ ≤ 125°C
–20%
TOFFMIN
Minimum Off-time
365
ns
TONMIN
Minimum On-time
155
ns
THSLEEP_HYS
Sleep mode threshold hysteresis
VFB rising, % of THWAKE
THWAKE
Wake up threshold
Measured at falling FB, COMP =
0.6 V
IBOOT
BOOT pin leakage current
BOOT = 16 V, SW = 10 V
101.2%
1.234
V
0.0006
BOOT = 16 V, SW = 10 V, –40°C
≤ TJ ≤ 125°C
µA
5.0
PROTECTION
ILIMPK
Peak Current Limit
2.5
–40°C ≤ TJ ≤ 125°C
1.85
VFB_SC
Short circuit frequency foldback
threshold
Measured at FB falling
F_min_sc
Min Frequency in foldback
VFB < 0.3 V
VTH_PGOOD
Power Good Threshold
Measured at FB, PGOOD rising
A
3.2
0.87
V
71
Measured at FB, PGOOD rising,
–40°C ≤ TJ ≤ 125°C
PGOOD hysteresis
kHz
92%
89%
2%
95%
7%
8%
IPGOOD_HI
PGOOD leakage current
PGOOD = 5 V
0.2
nA
RDS_PGOOD
PGOOD on resistance
PGOOD sink current = 500 µA
64
Ω
6
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
Electrical Characteristics (continued)
Unless otherwise stated, Vin=12 V. Minimum and Maximum limits are ensured through test, design, or statistical correlation.
Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.(1)
PARAMETER
VUVLO
Under-voltage Lock-Out Threshold
TEST CONDITIONS
MIN
Vin falling , shutdown, VDD = VIN
2.60
Vin rising, soft-start, VDD = VIN
Thermal Shutdown Threshold
Thermal resistance
UNIT
V
3.20
3.9
Vin rising, soft-start, VDD = VIN,
–40°C ≤ TJ ≤ 125°C
θJA
MAX
2.9
Vin falling , shutdown, VDD = VIN,
–40°C ≤ TJ ≤ 125°C
TSD
TYP
3.60
Power dissipation = 1W, 0 lfpm air
flow
4.20
160
°C
38
°C/W
1.2
V
LOGIC
VthEN
Enable Threshold voltage
–40°C ≤ TJ ≤ 125°C
0.8
Enable hysteresis
IEN_Source
EN source current
EN = 0 V
VTH_FPWM
FPWM threshold
IFPWM
FPWM leakage current
–40°C ≤ TJ ≤ 125°C
1.4
120
mV
4.5
µA
1.2
V
0.8
FPWM = 5 V
1.6
35
nA
EA
gm
Error amp trans-conductance
670
–40°C ≤ TJ ≤ 125°C
ICOMP
VCOMP
COMP source current
VCOMP = 0.9 V
COMP sink current
VCOMP = 0.9 V
COMP pin voltage range
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
400
µmho
1000
56
µA
56
0.64
µA
1.27
Submit Documentation Feedback
V
7
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
6.7 Typical Characteristics
1.236
1.236
1.234
1.235
VFB (V)
VFB (V)
Unless otherwise specified the following conditions apply: VIN = 12 V, TJ = 25°C.
1.232
1.230
1.234
1.233
1.228
-40 -20
1.232
0
20
40
60
0
80 100 120 140
5
10
15
TEMPERATURE (ºC)
700
35
40
IQ
(VBIAS=0V)
IQ
(VBIAS=0V)
80
30
Figure 2. VFB vs Vin (IDC = 300 mA)
90
600
60
IVBIAS
(VBIAS=5V)
50
CURRENT (PA)
70
CURRENT (PA)
25
VIN (V)
Figure 1. VFB vs Temperature
40
IQ
(VBIAS=5V)
30
20
-40 -20
20
0
20
40
500
IVBIAS
(VBIAS=5V)
400
300
IQ
(VBIAS=5V)
200
100
-40 -20
60 80 100 120 140
0
20
40
60
80 100 120 140
TEMPERATURE (ºC)
TEMPERATURE (ºC)
Figure 3. IQ and IVBIAS vs Temperature (Sleep Mode)
Figure 4. IQ and IVBIAS vs Temperature (PWM Mode)
4.1
3.9
On Threshold
3.7
101
3.5
VIN (V)
SWITCHING FREQUENCY (%)
102
100
3.3
3.1
99
Off Threshold
2.9
2.7
98
-40 -20
8
0
20
40 60
80 100 120 140
2.5
-40 -20
0
20
40
60
80 100 120 140
TEMPERATURE (ºC)
TEMPERATURE (ºC)
Figure 5. Normalized Switching Frequency vs Temperature
(300 kHz)
Figure 6. UVLO Threshold vs Temperature (VDD = VIN)
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
Typical Characteristics (continued)
Unless otherwise specified the following conditions apply: VIN = 12 V, TJ = 25°C.
350
SWITCHING FREQUENCY (kHz)
3.0
ILIM PEAK (A)
2.8
2.6
2.4
2.2
2.0
-40 -20
0
20
40
60
300
250
200
150
100
50
0
0.2
80 100 120 140
0.4
0.6
Figure 7. Peak Current Limit vs Temperature
1.2
Figure 8. Short Circuit Foldback Frequency vs VFB (325 kHz
Nominal)
100
100
5Vout
5Vout
90
EFFICIENCY (%)
90
EFFICIENCY (%)
1.0
VFB (V)
TEMPERATURE (ºC)
3.3Vout
80
70
FPWM
mode
3.3Vout
80
70
FPWM
mode
60
60
50
0.8
1
10
100
1000
10000
50
1
10
100
1000
10000
IDC (mA)
IDC (mA)
Figure 9. Efficiency vs Load Current (330 kHz)
Figure 10. Efficiency vs Load Current (500 kHz)
Vout
1V/Div
Vout
40 mV/Div
PGOOD
5V/Div
Iout
500 mA/Div
SS
1V/Div
EN
10V/Div
1 Ps/DIV
200 Ps/DIV
Figure 11. Startup Waveforms
Figure 12. Load Transient Response
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
9
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
Typical Characteristics (continued)
Unless otherwise specified the following conditions apply: VIN = 12 V, TJ = 25°C.
500
1A 125°C
VIN-VOUT DROPOUT (mV)
450
400
1A 25°C
350
300
1A -40°C
250
200
150
40mA -40°C
100
40mA
25 to 125°C
50
0
3
3.5
4
4.5
VIN (V)
5
5.5
Figure 13. Low Input Voltage Dropout Nominal VOUT = 5 V
10
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
7 Detailed Description
7.1 Overview
The LM26001 is a current mode PWM buck regulator. At the beginning of each clock cycle, the internal high-side
switch turns on, allowing current to ramp up in the inductor. The inductor current is internally monitored during
each switching cycle. A control signal derived from the inductor current is compared to the voltage control signal
at the COMP pin, derived from the feedback voltage. When the inductor current reaches the threshold, the highside switch is turned off and inductor current ramps down. While the switch is off, inductor current is supplied
through the catch diode. This cycle repeats at the next clock cycle. In this way, duty cycle and output voltage are
controlled by regulating inductor current. Current mode control provides superior line and load regulation. Other
benefits include cycle by cycle current limiting and a simplified compensation scheme. Typical PWM waveforms
are shown in Figure 14.
Vout
10 mV/Div
IL
500 mA/Div
ID
1A/Div
VSW
5V/Div
1 Ps/DIV
Figure 14. PWM Waveforms 1-A Load, Vin = 12 V
7.2 Functional Block Diagram
VIN
5 PA
BG
IREF
TSD
SD
on
LDO
UVLO
VDD_low
VDD
qn
EN
Switchover
control
fpwm
VBIAS
LG
FPWM
wake
+
-
BG
VREG
Sleep
Reset
0.6V
Sleep
Set
+
-
FPWM / Sleep
Peak Current
Control
EA
+
FB
+
blanking
COMP
0.9V
+
0.92BG
BOOT
+
V clamp
BG
Sync and
bootstrap
control
fpwm
sleep
frequency
foldback
VIN
I Sense
Corrective
Ramp
ff
qn
+
-
+
PWM
Comp
PG
PWM Control
Logic
SW
sleep
PGOOD
SW
Clock / Sync
2 PA
+
LG
ss end
ff
SS
SS
logic
soft start
+
FREQ
VDD_low
TSD
on
SD
+
SYNC
GND
EP
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
11
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
7.3 Feature Description
7.3.1 Sleep Mode
In light load conditions, the LM26001 automatically switches into sleep mode for improved efficiency. As loading
decreases, the voltage at FB increases and the COMP voltage decreases. When the COMP voltage reaches the
0.6-V (typical) clamp threshold, and the FB voltage rises 1% above nominal, sleep mode is enabled and
switching stops. The regulator remains in sleep mode until the FB voltage falls to the reset threshold, at which
point switching resumes. This 1% FB window limits the corresponding output ripple to approximately 1% of
nominal output voltage. The sleep cycle will repeat until load current is increased. Figure 15 shows typical
switching and output voltage waveforms in sleep mode.
Vout
50 mV/Div
IL
200 mA/Div
VSW
5V/Div
100 Ps/DIV
Figure 15. Sleep Mode Waveforms 25-mA Load, Vin = 12 V
In sleep mode, quiescent current is reduced to less than 40 µA when not switching. The DC sleep mode
threshold can be calculated according to the equation below:
2
ISleep = Imin + 0.13 P Vin - Vout
L
x
fsw x L
D x 2 x (Vin ± Vout)
(1)
Where Imin = Ilim/16 (2.5A/16 typically) and D = duty cycle, defined as (Vout + Vdiode)/Vin.
When load current increases above this limit, the LM26001 is forced back into normal PWM operation. The sleep
mode threshold varies with frequency, inductance, and duty cycle as shown in Figure 16.
250
200
500 kHz
IDC (mA)
150
100
330 kHz
22 PH
15 PH
22 PH
50 15 PH
0
0
10
20
30
40
VIN (V)
Figure 16. Sleep Mode Threshold vs Vin Vout = 3.3 V
Below the sleep threshold, decreasing load current results in longer sleep cycles, which can be quantified as
shown below:
Dwake = Iload/Isleep
(2)
Where Dwake is the percentage of time awake when the load current is below the sleep threshold.
Sleep mode combined with low IQ operation minimizes the input supply current. Input supply current in sleep
mode can be calculated based on the wake duty cycle, as shown below:
12
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
Feature Description (continued)
Iin = Iq + (IQG x Dwake) + (Io x D)
(3)
Where IQG is the gate drive current, calculated as:
IQG = (4.6 x 10-9) x fSW
And Io is the sum of Iload, Ibias, and current through the feedback resistors.
Because this calculation applies only to sleep mode, use the Iq_Sleep_VB and IBIAS_SLEEP values from the Electrical
Characteristics. If VBIAS is connected to ground, use the same equation with Ibias equal to zero and Iq_Sleep_VDD.
7.3.2 FPWM
Pulling the FPWM pin high disables sleep mode and forces the LM26001 to always operate in PWM mode. Light
load efficiency is reduced in PWM mode, but switching frequency remains stable. The FPWM pin can be
connected to the VDD pin to pull it high. In FPWM mode, under light load conditions, the regulator operates in
discontinuous conduction mode (DCM) . In discontinuous conduction mode, current through the inductor starts at
zero and ramps up to its peak, then ramps down to zero again. Until the next cycle, the inductor current remains
at zero. At nominal load currents, in FPWM mode, the device operates in continuous conduction mode, where
positive current always flows in the inductor. Typical discontinuous operation waveforms are shown in Figure 17.
Vout
10 mV/Div
IL
200 mA/Div
VSW
5V/Div
1 Ps/DIV
Figure 17. Discontinuous Mode Waveforms 75-mA Load, Vin = 12 V
At very light load, in FPWM mode, the LM26001 may enter sleep mode. This is to prevent an over-voltage
condition from occurring. However, the FPWM sleep threshold is much lower than in normal operation.
7.3.3 Enable
The LM26001 provides a shutdown function via the EN pin to disable the device when the output voltage does
not need to be maintained. EN is an analog level input with typically 120 mV of hysteresis. The device is active
when the EN pin is above 1.2 V (typical) and in shutdown mode when EN is below this threshold. When EN goes
high, the internal VDD regulator turns on and charges the VDD capacitor. When VDD reaches 3.9 V (typical), the
soft-start pin begins to source current. In shutdown mode, the VDD regulator shuts down and total quiescent
current is reduced to 10 µA (typical). Because the EN pin sources 4.5 µA (typical) of pull-up current, this pin can
be left open for always-on operation. When open, EN will be pulled up to VIN.
If EN is connected to VIN, it must be connected through a 10 kΩ resistor to limit noise spikes. EN can also be
driven externally with a maximum voltage of 38V or VIN + 15V, whichever is lower.
7.3.4 Soft-Start
The soft-start feature provides a controlled output voltage ramp up at startup. This reduces inrush current and
eliminates output overshoot at turn-on. The soft-start pin, SS, must be connected to GND through a capacitor. At
power-on, enable, or UVLO recovery, an internal 2.2 µA (typical) current charges the soft-start capacitor. During
soft-start, the error amplifier output voltage is controlled by both the soft-start voltage and the feedback loop. As
the SS pin voltage ramps up, the duty cycle increases proportional to the soft-start ramp, causing the output
voltage to ramp up. The rate at which the duty cycle increases depends on the capacitance of the soft-start
capacitor. The higher the capacitance, the slower the output voltage ramps up. The soft-start capacitor value can
be calculated with the following equation:
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
13
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
Feature Description (continued)
Css =
Iss x tss
1.234V
(4)
Where tss is the desired soft-start time and Iss is the soft-start source current. During soft-start, current limit and
synchronization remain in effect, while sleep mode and frequency foldback are disabled. Soft-start mode ends
when the SS pin voltage reaches 1.23 V typical. At this point, output voltage control is transferred to the FB pin
and the SS pin is discharged.
7.3.5 Current Limit
The peak current limit is set internally by directly measuring peak inductor current through the internal switch. To
ensure accurate current sensing, VIN should be bypassed with a minimum 1-µF ceramic capacitor placed directly
at the pin.
When the inductor current reaches the current limit threshold, the internal FET turns off immediately allowing
inductor current to ramp down until the next cycle. This reduction in duty cycle corresponds to a reduction in
output voltage.
The current limit comparator is disabled for less than 100 ns at the leading edge for increased immunity to
switching noise.
Because the current limit monitors peak inductor current, the DC load current limit threshold varies with
inductance and frequency. Assuming a minimum current limit of 1.85A, maximum load current can be calculated
as follows:
Iloadmax = 1.85A -
Iripple
2
(5)
Where Iripple is the peak-to-peak inductor ripple current, calculated as shown below:
Iripple =
(Vin ± Vout) x Vout
fsw x L x Vin
(6)
To find the worst case (lowest) current limit threshold, use the maximum input voltage and minimum current limit
specification.
During high over-current conditions, such as output short circuit, the LM26001 employs frequency foldback as a
second level of protection. If the feedback voltage falls below the short circuit threshold of 0.9 V, operating
frequency is reduced, thereby reducing average switch current. This is especially helpful in short circuit
conditions, when inductor current can rise very high during the minimum on-time. Frequency reduction begins at
20% below the nominal frequency setting. The minimum operating frequency in foldback mode is 71 kHz typical.
If the FB voltage falls below the frequency foldback threshold during frequency synchronized operation, the
SYNC function is disabled. Operating frequency versus FB voltage in short circuit conditions is shown in the
Typical Characteristics section.
Under conditions where the on time is close to minimum (less than 200 nsec typically), such as high input
voltage and high switching frequency, the current limit may not function properly. This is because the current limit
circuit cannot reduce the on-time below minimum which prevents entry into frequency foldback mode. There are
two ways to ensure proper current limit and foldback operation under high input voltage conditions. First, the
operating frequency can be reduced to increase the nominal on time. Second, the inductor value can be
increased to slow the current ramp and reduce the peak over-current.
7.3.6 Frequency Adjustment and Synchronization
The switching frequency of the LM26001 can be adjusted between 150 kHz and 500 kHz using a single external
resistor. This resistor is connected from the FREQ pin to ground as shown in the typical application. The resistor
value can be calculated with the following empirically derived equation:
RFREQ = (6.25 x 1010) x fSW-1.042
14
Submit Documentation Feedback
(7)
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
Feature Description (continued)
SWITCHING FREQUENCY (kHz)
600
500
400
300
200
100
0
50
100
150
200
250
300
RFREQ (k:)
Figure 18. Switching Frequency vs RFREQ
The switching frequency can also be synchronized to an external clock signal using the SYNC pin. The SYNC
pin allows the operating frequency to be varied above and below the nominal frequency setting. The adjustment
range is from 30% above nominal to 20% below nominal. External synchronization requires a 1.2V (typical) peak
signal level at the SYNC pin. The FREQ resistor must always be connected to initialize the nominal operating
frequency. The operating frequency is synchronized to the falling edge of the SYNC input. When SYNC goes
low, the high-side switch turns on. This allows any duty cycle to be used for the sync signal when synchronizing
to a frequency higher than nominal. When synchronizing to a lower frequency, however, there is a minimum duty
cycle requirement for the SYNC signal, given in the equation below:
Sync_Dmin t 1 -
fsync
fnom
(8)
Where fnom is the nominal switching frequency set by the FREQ resistor, and fsync is a square wave. If the
SYNC pin is not used, it must be pulled low for normal operation. A 10 kΩ pull-down resistor is recommended to
protect against a missing sync signal. Although the LM26001 is designed to operate at up to 500 kHz, maximum
load current may be limited at higher frequencies due to increased temperature rise. See the Thermal
Considerations and TSD section.
7.3.7 VBIAS
The VBIAS pin is used to bypass the internal regulator which provides the bias voltage to the LM26001. When
the VBIAS pin is connected to a voltage greater than 3 V, the internal regulator automatically switches over to the
VBIAS input. This reduces the current into VIN (Iq) and increases system efficiency. Using the VBIAS pin has the
added benefit of reducing power dissipation within the device.
For most applications where 3 V < Vout < 10V, VBIAS can be connected to Vout. If not used, VBIAS should be
tied to GND.
If VBIAS drops below 2.9 V (typical), the device automatically switches over to supply the internal bias voltage
from Vin.
7.3.8 Low VIN Operation and UVLO
The LM26001 is designed to remain operational during short line transients when input voltage may drop as low
as 3.0 V. Minimum nominal operating input voltage is 4.0 V. Below this voltage, switch RDS(ON) increases, due to
the lower gate drive voltage from VDD. The minimum voltage required at VDD is approximately 3.5 V for normal
operation within specification.
VDD can also be used as a pull-up voltage for functions such as PGOOD and FPWM. Note that if VDD is used
externally, the pin is not recommended for loads greater than 1 mA.
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
15
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
Feature Description (continued)
If the input voltage approaches the nominal output voltage, the duty cycle is maximized to hold up the output
voltage. In this mode of operation, once the duty cycle reaches its maximum, the LM26001 can skip a maximum
of seven off pulses, effectively increasing the duty cycle and thus minimizing the dropout from input to output.
Typical off-pulse skipping waveforms are shown in Figure 19.
Vout
20 mV/Div
IL
100 mA/Div
VSW
2V/Div
4 Ps/DIV
Figure 19. Off-pulse Skipping Waveforms Vin = 3.5 V, Vnom = 3.3 V, fnom = 305 kHz
UVLO is sensed at both VIN and VDD, and is activated when either voltage falls below 2.9 V (typical). Although
VDD is typically less than 200 mV below VIN, it will not discharge through VIN. Therefore when the VIN voltage
drops rapidly, VDD may remain high, especially in sleep mode. For fast line voltage transients, using a larger
capacitor at the VDD pin can help to hold off a UVLO shutdown by extending the VDD discharge time. By
holding up VDD, a larger cap can also reduce the RDS(ON) (and dropout voltage) in low VIN conditions.
Alternately, under heavy loading the VDD voltage can fall several hundred mV below VIN. In this case, UVLO
may be triggered by VDD even though the VIN voltage is above the UVLO threshold.
When UVLO is activated the LM26001 enters a standby state in which VDD remains charged. As input voltage
and VDD voltage rise above 3.9 V (typical) the device will restart from softstart mode.
7.3.9 PGOOD
A power good pin, PGOOD, is available to monitor the output voltage status. The pin is internally connected to
an open-drain MOSFET, which remains open while the output voltage is within operating range. PGOOD goes
low (low impedance to ground) when the output falls below 85% of nominal or EN is pulled low. When the output
voltage returns to within 92% of nominal, as measured at the FB pin, PGOOD returns to a high state. For
improved noise immunity, there is a 5 µs delay between the PGOOD threshold and the PGOOD pin going low.
7.4 Device Functional Modes
The LM26001 has three basic operation mode: Shutdown, Sleep or light load operation and full operation.
The part enters shutdown mode when the EN pin is pulled low. In this mode the converter is disabled and the
quiescent current minimized See Enable for more details.
The part enters sleep mode when the converter is active (EN high) and the output current is low. Sleep mode is
activated as the COMP voltage naturally falls below a typical 0.6V threshold in light load operation. When
operating in sleep mode, the switching events of the converter are reduced in order to lower the current
consumption of the system. Forcing the FPWM pin high will prevent sleep mode operation. For details of
operation in sleep mode as well as entering and exiting sleep mode. See Sleep Mode.
When the part in enabled and the output load is higher, the part will be in full PWM operation.
In addition to these normal functioning modes, the LM26001 has a frequency foldback operating mode which
reduces the operating frequency to protect from short circuits. See Current Limit.
16
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
8 Applications and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM26001 offers efficient step-down function meeting the requirements of automotive applications. Current
mode control ensures smooth and safe operation thanks to its cycle by cycle current limiting function. Its low
minimum ON time allows it to operate over a wide range of output voltages. The following sections detail the
steps required for designing a successful application with the LM26001 from component selection to layout.
8.2 Typical Application
Figure 20 shows a complete typical application schematic. The components have been selected based on the
design criteria given in the following sections.
VIN : 4V - 38V
+ C2
47 PF
50V
C1
3.3 PF
50V
1
2
3
PGOOD
R4
200k
4
EN
11
VDD
5
SYNC
R6
10k
VIN
VBIAS
VIN
SW
PGOOD
SW
EN
BOOT
LM26001
SYNC
R3
120k
1%
SS
FPWM
14
C4
0.1 PF
D1
3A
60V
VOUT : 3.3V
R1
56k
1%
+
C6
100 PF
12 m:
7
FB
COMP
FREQ
10
16
15
9
C5
10 nF
L
22 PH
3.5A
12
VDD
EP
17
GND
6
13
Ref #
C8
4.7 nF
8
C3
10 PF
R5
15k
C9
47 pF
R2
33k
1%
Manufacturer
Part Number
C1
TDK
3225JB1H335K
C2
Panasonic
EEVFK1H470P
C6
Panasonic
EEFVEOK101R
D1
NIEC
NSQ03A06
L1
TDK
SLF12565T220M3R5
Figure 20. Example Circuit 1.5A Max, 305 kHz
8.2.1 Design Requirements
The following parameters are needed to properly design the application and size the components:
PARAMETERS
VALUES
Vout
Output voltage
Vin min
Maximum input voltage
Vin max
Minimum input voltage
Iout max
Maximum output current
Fsw
Switching Frequency
Fbw
Bandwidth of the converter
8.2.2 Detailed Design Procedure
8.2.2.1 Setting Output Voltage
The output voltage is set by the ratio of a voltage divider at the FB pin as shown in the typical application. The
resistor values can be determined by the following equation:
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
17
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
R2 =
www.ti.com
R1
§ Vout -1 ·
© Vfb ¹
(9)
Where Vfb = 1.234V typically.
A maximum value of 150kΩ is recommended for the sum of R1 and R2.
As input voltage decreases towards the nominal output voltage, the LM26001 can skip up to seven off-pulses as
described in the Low VIN Operation and UVLO section. In low output voltage applications, if the on-time reaches
TonMIN, the device will skip on-pulses to maintain regulation. There is no limit to the number of pulses that are
skipped. In this mode of operation, however, output ripple voltage may increase slightly.
8.2.2.2 Inductor
The output inductor should be selected based on inductor ripple current. The amount of inductor ripple current
compared to load current, or ripple content, is defined as Iripple/Iload. Ripple content should be less than 40%.
Inductor ripple current, Iripple, can be calculated as shown below:
Iripple =
(Vin ± Vout) x Vout
fsw x L x Vin
(10)
Larger ripple content increases losses in the inductor and reduces the effective current limit.
Larger inductance values result in lower output ripple voltage and higher efficiency, but a slightly degraded
transient response. Lower inductance values allow for smaller case size, but the increased ripple lowers the
effective current limit threshold.
Remember that inductor value also affects the sleep mode threshold as shown in Figure 16.
When choosing the inductor, the saturation current rating must be higher than the maximum peak inductor
current and the RMS current rating should be higher than the maximum load current. Peak inductor current,
Ipeak, is calculated as:
Ipeak = Iload +
Iripple
2
(11)
For example, at a maximum load of 1.5A and a ripple content of 40%, peak inductor current is equal to 1.8A
which is safely below the minimum current limit of 1.85A. By increasing the inductor size, ripple content and peak
inductor current are lowered, which increases the current limit margin.
The size of the output inductor can also be determined using the desired output ripple voltage, Vrip. The
equation to determine the minimum inductance value based on Vrip is as follows:
LMIN =
(Vin ± Vout) x Vout x Re
Vin x fsw x Vrip
(12)
Where Re is the ESR of the output capacitors, and Vrip is a peak-to-peak value. This equation assumes that the
output capacitors have some amount of ESR. It does not apply to ceramic output capacitors.
If this method is used, ripple content should still be verified to be less than 40%.
8.2.2.3 Output Capacitor
The primary criterion for selecting an output capacitor is equivalent series resistance, or ESR.
ESR (Re) can be selected based on the requirements for output ripple voltage and transient response. Once an
inductor value has been selected, ripple voltage can be calculated for a given Re using the equation above for
Lmin. Lower ESR values result in lower output ripple.
Re can also be calculated from the following equation:
ReMAX =
18
'Vt
'It
Submit Documentation Feedback
(13)
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
Where ΔVt is the allowed voltage excursion during a load transient, and ΔIt is the maximum expected load
transient. If the total ESR is too high, the load transient requirement cannot be met, no matter how large the
output capacitance. If the ESR criteria for ripple voltage and transient excursion cannot be met, more capacitors
should be used in parallel. For non-ceramic capacitors, the minimum output capacitance is of secondary
importance, and is determined only by the load transient requirement.
If there is not enough capacitance, the output voltage excursion will exceed the maximum allowed value even if
the maximum ESR requirement is met. The minimum capacitance is calculated as follows:
2
2
L x §©'Vt - ('Vt) - ('It x Re)
§
©
CMIN =
2
Vout x Re
(14)
It is assumed the total ESR, Re, is no greater than ReMAX. Also, it is assumed that L has already been selected.
Generally speaking, the output capacitance requirement decreases with Re, ΔIt, and L. A typical value greater
than 100 µF works well for most applications.
8.2.2.4 Input Capacitor
In a switching converter, very fast switching pulse currents are drawn from the input rail. Therefore, input
capacitors are required to reduce noise, EMI, and ripple at the input to the LM26001. Capacitors must be
selected that can handle both the maximum ripple RMS current at highest ambient temperature as well as the
maximum input voltage. The equation for calculating the RMS input ripple current is shown below:
Iload x Vout x (Vin ± Vout)
Irms =
Vin
(15)
For noise suppression, a ceramic capacitor in the range of 1.0 µF to 10 µF should be placed as close as possible
to the VIN pin.
A larger, high ESR input capacitor should also be used. This capacitor is recommended for damping input
voltage spikes during power-on and for holding up the input voltage during transients. In low input voltage
applications, line transients may fall below the UVLO threshold if there is not enough input capacitance. Both
tantalum and electrolytic type capacitors are suitable for the bulk capacitor. However, large tantalums may not be
available for high input voltages and their working voltage must be derated by at least 2X.
8.2.2.5 Bootstrap
The drive voltage for the internal switch is supplied via the BOOT pin. This pin must be connected to a ceramic
capacitor, Cboot, from the switch node, shown as C4 in the typical application. The LM26001 provides the VDD
voltage internally, so no external diode is needed. A maximum value of 0.1 uF is recommended for Cboot.
Values smaller than 0.01 uF may result in insufficient hold up time for the drive voltage and increased power
dissipation.
During low Vin operation, when the on-time is extended, the bootstrap capacitor is at risk of discharging. If the
Cboot capacitor is discharged below approximately 2.5V, the LM26001 enters a high frequency re-charge mode.
The Cboot cap is re-charged via the LG synchronous FET shown in the block diagram. Switching returns to
normal when the Cboot cap has been recharged.
8.2.2.6 Catch Diode
When the internal switch is off, output current flows through the catch diode. Alternately, when the switch is on,
the diode sees a reverse voltage equal to Vin. Therefore, the important parameters for selecting the catch diode
are peak current and peak inverse voltage. The average current through the diode is given by:
IDAVE = Iload x (1-D)
(16)
Where D is the duty cycle, defined as Vout/Vin. The catch diode conducts the largest currents during the lowest
duty cycle. Therefore IDAVE should be calculated assuming maximum input voltage. The diode should be rated to
handle this current continuously. For over-current or short circuit conditions, the catch diode should be rated to
handle peak currents equal to the peak current limit.
The peak inverse voltage rating of the diode must be greater than maximum input voltage.
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
19
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
A Schottky diode must be used. It's low forward voltage maximizes efficiency and BOOT voltage, while also
protecting the SW pin against large negative voltage spikes.
8.2.2.7 Compensation
The purpose of loop compensation is to ensure stable operation while maximizing dynamic performance. Stability
can be analyzed with loop gain measurements, while dynamic performance is analyzed with both loop gain and
load transient response. Loop gain is equal to the product of control-output transfer function (power stage) and
the feedback transfer function (the compensation network).
For stability purposes, our target is to have a loop gain slope that is -20dB /decade from a very low frequency to
beyond the crossover frequency. Also, the crossover frequency should not exceed one-fifth of the switching
frequency, i.e. 60 kHz in the case of 300 kHz switching frequency.
For dynamic purposes, the higher the bandwidth, the faster the load transient response. A large DC gain means
high DC regulation accuracy (i.e. DC voltage changes little with load or line variations). To achieve this loop gain,
the compensation components should be set according to the shape of the control-output bode plot. A typical
plot is shown in Figure 21.
fz
fn
0
0
-45
-20
-90
-40
-135
-60
0.01
0.1
1
10
100
PHASE (º)
GAIN (dB)
fp
20
-180
1000
FREQUENCY (kHz)
Figure 21. Control-Output Transfer Function
The control-output transfer function consists of one pole (fp), one zero (fz), and a double pole at fn (half the
switching frequency).
Referring to Figure 21, the following should be done to create a -20dB /decade roll-off of the loop gain:
1. Place a pole at 0 Hz (fpc)
2. Place a zero at fp (fzc)
3. Place a second pole at fz (fpc1)
GAIN (dB)
The resulting feedback (compensation) bode plot is shown in Figure 22. Adding the control-output response to
the feedback response will then result in a nearly continuous –20db/decade slope.
-2
0dB
/de
c
0dB/dec
-2
0dB
/
B
fpc
(0Hz)
fzc
dec
fpc1
FREQUENCY
Figure 22. Feedback Transfer Function
20
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
The control-output corner frequencies can be determined approximately by the following equations:
fz =
1
2S x Re x Co
fp =
(17)
1
10 x S x Ro x Co
+
0.5
2 x S x L x fsw x Co
(18)
fsw
fn =
2
(19)
Where Co is the output capacitance, Ro is the load resistance, Re is the output capacitor ESR, and fsw is the
switching frequency. The effects of slope compensation and current sense gain are included in this equation.
However, the equation is an approximation intended to simplify loop compensation calculations. To derive the
exact transfer function, use 0.2V/V sense amp gain and 36mVp-p slope compensation.
Since fp is determined by the output network, it shifts with loading. Determine the range of frequencies
(fpmin/max) across the expected load range. Then determine the compensation values as described below and
shown in Figure 23.
5
6
SS
COMP
7 FB
C8
C9
R5
R2
R1
C10
To Vout
Figure 23. Compensation Network
1. The compensation network automatically introduces a low frequency pole (fpc), which is close to 0Hz.
2. Once the fp range is determined, R5 should be calculated using:
R5 =
B § R1 + R2
x
gm ©
R2
·
¹
(20)
Where B is the desired feedback gain in v/v between fp and fz, and gm is the transconductance of the error
amplifier. A gain value around 10dB (3.3v/v) is generally a good starting point. Bandwidth increases with
increasing values of R5.
3. Next, place a zero (fzc) near fp using C8. C8 can be determined with the following equation:
C8 =
1
2 x S x fPMAX x R5
(21)
The selected value of C8 should place fzc within a decade above or below fpmax, and not less than fpmin. A
higher C8 value (closer to fpmin) generally provides a more stable loop, but too high a value will slow the
transient response time. Conversely, a smaller C8 value will result in a faster transient response, but lower phase
margin.
4. A second pole (fpc1) can also be placed at fz. This pole can be created with a single capacitor, C9. The
minimum value for this capacitor can be calculated by:
C9 =
1
2 x S x fz x R5
(22)
C9 may not be necessary in all applications. However if the operating frequency is being synchronized below the
nominal frequency, C9 is recommended. Although it is not required for stability, C9 is very helpful in suppressing
noise.
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
21
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
A phase lead capacitor can also be added to increase the phase and gain margins. The phase lead capacitor is
most helpful for high input voltage applications or when synchronizing to a frequency greater than nominal. This
capacitor, shown as C10 in Figure 23, should be placed in parallel with the top feedback resistor, R1. C10
introduces an additional zero and pole to the compensation network. These frequencies can be calculated as
shown below:
fzff =
fpff =
1
2 x S x R1 x C10
(23)
fzff x Vout
Vfb
(24)
A phase lead capacitor will boost loop phase around the region of the zero frequency, fzff. fzff should be placed
somewhat below the fpz1 frequency set by C9. However, if C10 is too large, it will have no effect.
8.2.3 Application Curves
Vout
1V/Div
Vout
40 mV/Div
PGOOD
5V/Div
Iout
500 mA/Div
SS
1V/Div
EN
10V/Div
1 Ps/DIV
200 Ps/DIV
Figure 24. Startup Waveforms
22
Submit Documentation Feedback
Figure 25. Load Transient Response
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
9 Power Supply Recommendations
The LM26001 is designed to operate from various DC power supply including a car battery. If so, VIN input
should be protected from reversal voltage and voltage dump over 48 Volts. The impedance of the input supply
rail should be low enough that the input current transient does not cause drop below VIN UVLO level. If the input
supply is connected by using long wires, additional bulk capacitance may be required in addition to normal input
capacitor.
10 Layout
10.1 Layout Guidelines
Good board layout is critical for switching regulators such as the LM26001. First, the ground plane area must be
sufficient for thermal dissipation purposes, and second, appropriate guidelines must be followed to reduce the
effects of switching noise.
Switch mode converters are very fast switching devices. In such devices, the rapid increase of input current
combined with parasitic trace inductance generates unwanted Ldi/dt noise spikes at the SW node and also at the
VIN node. The magnitude of this noise tends to increase as the output current increases. This parasitic spike
noise may turn into electromagnetic interference (EMI), and can also cause problems in device performance.
Therefore, care must be taken in layout to minimize the effect of this switching noise.
The current sensing circuit in current mode devices can be easily affected by switching noise. This noise can
cause duty cycle jitter which leads to increased spectral noise. Although the LM26001 has 100ns blanking time at
the beginning of every cycle to ignore this noise, some noise may remain after the blanking time. Following the
important guidelines below will help minimize switching noise and its effect on current sensing.
The switch node area should be as small as possible. The catch diode, input capacitors, and output capacitors
should be grounded to a large ground plane, with the bulk input capacitor grounded as close as possible to the
catch diode anode. Additionally, the ground area between the catch diode and bulk input capacitor is very noisy
and should be somewhat isolated from the rest of the ground plane.
A ceramic input capacitor must be connected as close as possible to the VIN pin and grounded close to the GND
pin. Often this capacitor is most easily located on the bottom side of the pcb. If placement close to the GND pin
is not practical, the ceramic input capacitor can also be grounded close to the catch diode ground. The above
layout recommendations are illustrated below in Figure 26.
It is a good practice to connect the EP, GND pin, and small signal components (COMP, FB, FREQ) to a separate
ground plane, shown in Figure 26 as EP GND, and in the schematics as a signal ground symbol. Both the
exposed pad and the GND pin must be connected to ground. This quieter plane should be connected to the high
current ground plane at a quiet location, preferably near the Vout ground as shown by the dashed line in
Figure 26.
The EP GND plane should be made as large as possible, since it is also used for thermal dissipation. Several
vias can be placed directly below the EP to increase heat flow to other layers when they are available. The
recommended via hole diameter is 0.3mm.
The trace from the FB pin to the resistor divider should be short and the entire feedback trace must be kept away
from the inductor and switch node. See AN-1229 SIMPLE SWITCHER® PCB Layout Guidelines, SNVA054, for
more information regarding PCB layout for switching regulators.
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
23
LM26001, LM26001-Q1
SNVS430I – MAY 2006 – REVISED MARCH 2015
www.ti.com
10.2 Layout Example
Vin
Vout
+
GND
SW
EP
GND
Figure 26. Example PCB Layout
10.3 Thermal Considerations and TSD
Although the LM26001 has a built in current limit, at ambient temperatures above 80°C, device temperature rise
may limit the actual maximum load current. Therefore, temperature rise must be taken into consideration to
determine the maximum allowable load current.
Temperature rise is a function of the power dissipation within the device. The following equations can be used to
calculate power dissipation (PD) and temperature rise, where total PD is the sum of FET switching losses, FET
DC losses, drive losses, Iq, and VBIAS losses:
PDTOTAL = PswAC + PswDC + PQG + PIq + PVBIAS
(25)
§ Vin x 10 ·
© 1.33 ¹
-9
PswAC = Vin x Iload x fsw x
(26)
(27)
(28)
(29)
(30)
PswDC = D x Iload2 x (0.2 + 0.00065 x (Tj - 25))
PQG = Vin x 4.6 x 10-9 x fsw
PIq = Vin x Iq
PVBIAS = Vbias x IVBIAS
Given this total power dissipation, junction temperature can be calculated as follows:
Tj = Ta + (PDTOTAL x θJA)
(31)
Where θJA=38°C/W (typically) when using a multi-layer board with a large copper plane area. θJA varies with
board type and metallization area.
To calculate the maximum allowable power dissipation, assume Tj = 125°C. To ensure that junction temperature
does not exceed the maximum operating rating of 125°C, power dissipation should be verified at the maximum
expected operating frequency, maximum ambient temperature, and minimum and maximum input voltage. The
calculated maximum load current is based on continuous operation and may be exceeded during transient
conditions.
If the power dissipation remains above the maximum allowable level, device temperature will continue to rise.
When the junction temperature exceeds its maximum, the LM26001 engages Thermal Shut Down (TSD). In
TSD, the part remains in a shutdown state until the junction temperature falls to within normal operating limits. At
this point, the device restarts in soft-start mode.
24
Submit Documentation Feedback
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001, LM26001-Q1
www.ti.com
SNVS430I – MAY 2006 – REVISED MARCH 2015
11 Device and Documentation Support
11.1 Documentation Support
11.1.1 Related Documentation
AN-1229 SIMPLE SWITCHER® PCB Layout Guidelines, SNVA054
IC Package Thermal Metrics application report, SPRA953
11.2 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 1. Related Links
PARTS
PRODUCT FOLDER
SAMPLE & BUY
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
LM26001
Click here
Click here
Click here
Click here
Click here
LM26001-Q1
Click here
Click here
Click here
Click here
Click here
11.3 Trademarks
All trademarks are the property of their respective owners.
11.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.5 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
Submit Documentation Feedback
25
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM26001MXA/NOPB
ACTIVE
HTSSOP
PWP
16
92
RoHS & Green
Call TI | SN
Level-1-260C-UNLIM
-40 to 125
L26001
MXA
LM26001MXAX/NOPB
ACTIVE
HTSSOP
PWP
16
2500
RoHS & Green
Call TI | SN
Level-1-260C-UNLIM
-40 to 125
L26001
MXA
LM26001QMXA/NOPB
ACTIVE
HTSSOP
PWP
16
92
RoHS & Green
Call TI | SN
Level-1-260C-UNLIM
-40 to 125
L26001
QMXA
LM26001QMXAX/NOPB
ACTIVE
HTSSOP
PWP
16
2500
RoHS & Green
Call TI | SN
Level-1-260C-UNLIM
-40 to 125
L26001
QMXA
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
工商网监
湘ICP备2023018690号
