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LM27341, LM27342, LM27341-Q1, LM27342-Q1
SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
LM2734x and LM2734x-Q1 2-MHz, 1.5-A or 2-A, Wide Input Range, Step-Down, DC-DC
Regulator With Frequency Synchronization
1 Features
3 Description
•
The LM2734x and LM2734x-Q1 regulators are
monolithic, high-frequency, PWM step-down DC-DC
converters in 10-pin WSON and 10-pin MSOPPowerPAD packages. They contain all the active
functions to provide local DC-DC conversion with fast
transient response and accurate regulation in the
smallest possible PCB area.
1
•
•
•
•
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•
•
•
•
•
•
•
2
•
•
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Space-Saving, 3 mm × 3 mm 10-Pin WSON and
MSOP-PowerPAD Packages
Wide Input Voltage Range: 3 V to 20 V
Wide Output Voltage Range: 1 V to 18 V
LM27341 Delivers 1.5-A Maximum Output Current
LM27342 Delivers 2-A Maximum Output Current
High Switching Frequency: 2 MHz
Frequency Synchronization:
1 MHz < fSW < 2.35 MHz
150-mΩ NMOS Switch With Internal Bootstrap
Supply
70-nA Shutdown Current
Internal Voltage Reference Accuracy of 1%
Peak Current-Mode, PWM Operation
Thermal Shutdown
LM27341-Q1 and LM27342-Q1 are AEC-Q100
Grade 1 Qualified and Manufactured on an
Automotive Grade Flow
With a minimum of external components, the
LM2734x and LM2734x-Q1 are easy to use. The
ability to drive 1.5-A or 2-A loads respectively, with an
internal 150-mΩ NMOS switch results in the best
power density available. The world-class control
circuitry allows for on-times as low as 65 ns, thus
supporting exceptionally high frequency conversion.
Switching frequency is internally set to 2 MHz and
synchronizable from 1 to 2.35 MHz, which allows the
use of extremely small surface mount inductors and
chip capacitors. Even though the operating frequency
is very high, efficiencies up to 90% are easy to
achieve. External shutdown is included, which
features an ultra-low shutdown current of 70 nA. The
LM2734x and LM2734x-Q1 use peak current-mode
control and internal compensation to provide highperformance regulation over a wide range of
operating conditions. Additional features include
internal soft-start circuitry to reduce inrush current,
pulse-by-pulse current limit, thermal shutdown, and
output overvoltage protection.
Applications
Local 12-V to Vcore Step-Down Converters
Radio Power Supply
Core Power in HDDs
Set-Top Boxes
Automotive
USB Powered Devices
DSL Modems
Device Information(1)
PART NUMBER
PACKAGE
BODY SIZE (NOM)
MSOP-PowerPAD (10) 4.90 mm × 3.00 mm
LM2734x
LM2734x-Q1
WSON (10)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application Circuit
VIN
PVIN
BOOST
C2
AVIN
L1
C1
VOUT
SW
C3
D1
LM27341/2
ON
OFF
EN
R1
CLK
SYNC
GND/DAP
FB
R2
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM27341, LM27342, LM27341-Q1, LM27342-Q1
SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
5
5
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 12
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
12
13
13
16
8
Application and Implementation ........................ 17
8.1 Application Information............................................ 17
8.2 Typical Applications ................................................ 28
9 Power Supply Recommendations...................... 41
10 Layout................................................................... 41
10.1 Layout Guidelines ................................................. 41
10.2 Layout Example ................................................... 42
11 Device and Documentation Support ................. 43
11.1
11.2
11.3
11.4
11.5
11.6
11.7
11.8
Device Support......................................................
Documentation Support ........................................
Related Links ........................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
43
43
43
43
43
43
44
44
12 Mechanical, Packaging, and Orderable
Information ........................................................... 44
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision E (April 2013) to Revision F
Page
•
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section .................................................................................................. 1
•
Changed values in the Thermal Information table to align with JEDEC standards................................................................ 5
Changes from Revision D (April 2013) to Revision E
•
2
Page
Changed layout of National Semiconductor Data Sheet to TI format .................................................................................... 1
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
5 Pin Configuration and Functions
DSC Package
10-Pin WSON
Top View
DGQ Package
10-Pin MSOP-PowerPAD
Top View
SW
1
10
PVIN
SW
1
10
PVIN
SW
2
9
PVIN
SW
2
9
PVIN
BOOST
3
8
AVIN
BOOST
3
8
AVIN
EN
4
7
GND
EN
4
7
GND
SYNC
5
6
FB
SYNC
5
6
FB
DAP
DAP
Not to scale
Not to scale
Pin Functions
PIN
TYPE (1)
DESCRIPTION
NO.
NAME
1, 2
SW
O
Output switch. Connects to the inductor, catch diode, and bootstrap capacitor.
3
BOOST
I
Boost voltage that drives the internal NMOS control switch. A bootstrap capacitor is connected
between the BOOST and SW pins.
4
EN
I
Enable control input. Logic high enables operation. Do not allow this pin to float or be greater than
VIN + 0.3 V.
5
SYNC
I
Frequency synchronization input. Drive this pin with an external clock or pulse train. Ground it to
use the internal clock.
6
FB
I
Feedback pin. Connect FB to the external resistor divider to set output voltage.
7
GND
G
Signal and power ground pin. Place the bottom resistor of the feedback network as close as
possible to this pin for accurate regulation.
8
AVIN
I
Supply voltage for the control circuitry.
9, 10
PVIN
I
Supply voltage for output power stage. Connect a bypass capacitor to this pin.
DAP
DAP
G
Signal or power ground and thermal connection. Tie this directly to GND (pin 7).
See Application Information regarding optimum thermal layout.
(1)
G = Ground, I = Input, O = Output
Copyright © 2008–2016, Texas Instruments Incorporated
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LM27341, LM27342, LM27341-Q1, LM27342-Q1
SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
www.ti.com
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2)
MIN
MAX
UNIT
AVIN, PVIN
–0.5
24
V
SW voltage
–0.5
24
V
Boost voltage
–0.5
28
V
Boost to SW voltage
–0.5
6
V
FB voltage
–0.5
3
V
SYNC voltage
–0.5
6
V
EN voltage
–0.5
VIN + 0.3
V
Soldering, infrared reflow (5 s)
260
°C
Junction temperature, TJ
150
°C
150
°C
Storage temperature, Tstg
(1)
(2)
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
6.2 ESD Ratings
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
(1) (2)
VALUE
UNIT
±2000
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
Human body model, 1.5 kΩ in series with 100 pF.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted) (1)
MIN
MAX
AVIN, PVIN
3
20
UNIT
V
SW voltage
–0.5
20
V
Boost voltage
V
–0.5
24
Boost to SW voltage
3
5.5
V
Junction temperature
–40
125
°C
(1)
4
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of
device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or
other conditions beyond those indicated in the recommended Operating Ratings is not implied. The recommended Operating Ratings
indicate conditions at which the device is functional and should not be operated beyond such conditions.
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
6.4 Thermal Information
LM2734x, LM2734x-Q1
THERMAL METRIC (1)
DSC (WSON)
DGQ (MSOP-PowerPAD)
10 PINS
10 PINS
UNIT
RθJA
Junction-to-ambient thermal resistance (2)
47.6
49.5
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
36.5
53.6
°C/W
RθJB
Junction-to-board thermal resistance
22.5
33.7
°C/W
ψJT
Junction-to-top characterization parameter
0.4
3.9
°C/W
ψJB
Junction-to-board characterization parameter
22.7
33.4
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
4.7
3.5
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
Thermal shutdown will occur if the junction temperature exceeds 165°C. The maximum power dissipation is a function of TJ(MAX), RθJA
and TA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) – TA)/ RθJA. All numbers apply for
packages soldered directly onto a 3" × 3" PCB with 2 oz. copper on 4 layers in still air.
6.5 Electrical Characteristics
TJ = 25°C, VIN = 12 V, and VBOOST – VSW = 4.3 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
0.99
1
1.01
0.984
1
1.014
UNIT
SYSTEM PARAMETERS
TJ = 0°C to 85°C
VFB
Feedback voltage
ΔVFB/ΔVIN
Feedback voltage line
regulation
IFB
Feedback input bias current
OVP
Overvoltage protection
TJ = –40°C to 125°C
VIN = 3 V to 20 V
TJ = 25°C
Undervoltage hysteresis
SS
100
VFB at which PWM halts
UVLO
VIN falling from UVLO
1.13
TJ = 25°C
TJ = –40°C to 125°C
TJ = –40°C to 125°C
Quiescent current
IBOOST
Boost pin current
2.6
fSW= 2 MHz
V
2.9
0.47
0.3
0.5
IQ = IQ_AVIN + IQ_PVIN
nA
2.75
TJ = 25°C
Soft-start time
IQ
V
20
TJ = –40°C to 125°C
VIN rising until VSW is
switching
Undervoltage lockout
0.003%
V
V
0.6
1
1.5
ms
VFB = 1.1 (not switching)
2.4
mA
VEN = 0 V (shutdown)
70
nA
TJ = 25°C
8.2
TJ = –40°C to 125°C
10
fSW= 1 MHz
4.4
mA
6
OSCILLATOR
TJ = 25°C
2
fSW
Switching frequency
SYNC = GND
VFB_FOLD
FB pin voltage
SYNC input is overridden
0.53
fFOLD_MIN
Frequency foldback
minimum
VFB = 0 V
220
TJ = –40°C to 125°C
1.75
2.3
MHz
V
250
kHz
2.35
MHz
LOGIC INPUTS (EN, SYNC)
fSYNC
SYNC frequency range
1
VIL
EN, SYNC logic low
threshold
Logic falling edge
VIH
EN, SYNC logic high
threshold
Logic rising edge
tSYNC_HIGH
SYNC, time required above
VIH to ensure a logical high
0.4
V
Copyright © 2008–2016, Texas Instruments Incorporated
1.8
100
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Electrical Characteristics (continued)
TJ = 25°C, VIN = 12 V, and VBOOST – VSW = 4.3 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
tSYNC_LOW
SYNC, time required below
VIL to ensure a logical low
ISYNC
SYNC pin current
IEN
MIN
VSYNC < 5 V
MAX
UNIT
100
ns
20
VEN = 3 V
Enable pin current
TYP
VIN = VEN = 20 V
nA
6
15
50
100
µA
INTERNAL MOSFET
TJ = 25°C
150
RDS(ON)
Switch ON-resistance
ICL
Switch current limit
TJ = –40°C to 125°C
DMAX
Maximum duty cycle
SYNC = GND
tMIN
Minimum ON-time
65
ns
ISW
Switch leakage current
40
nA
3.9
V
TJ = –40°C to 125°C
320
LM27342
2.5
4
LM27341
2
3.7
TJ = 25°C
mΩ
A
93%
TJ = –40°C to 125°C
85%
BOOST LDO
VLDO
Boost LDO output voltage
THERMAL
TSHDN
6
Thermal shutdown
temperature
Junction temperature rising
165
Thermal shutdown
hysteresis
Junction temperature falling
15
°C
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
6.6 Typical Characteristics
TA = 25°C, VIN = 12 V, and VBOOST – VSW = 4.3 V (unless otherwise noted)
VOUT = 5 V
fSW = 2 MHz
VOUT = 5 V
Figure 2. Load Transient
See Figure 34
Figure 1. Efficiency vs Load Current
See Figure 34
VOUT = 3.3 V
fSW = 2 MHz
VOUT = 3.3 V
fSW = 2 MHz
Figure 5. Efficiency vs Load Current
See Figure 49
Copyright © 2008–2016, Texas Instruments Incorporated
IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 4. Load Transient
See Figure 40
Figure 3. Efficiency vs Load Current
See Figure 40
VOUT = 1.8 V
IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
VOUT = 1.8 V
IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 6. Load Transient
See Figure 49
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Typical Characteristics (continued)
TA = 25°C, VIN = 12 V, and VBOOST – VSW = 4.3 V (unless otherwise noted)
VIN = 10 V to 15 V
8
VOUT = 3.3 V
No CFF
VIN = 10 V to 15 V
VOUT = 3.3 V
Figure 7. Line Transient
See Figure 43
Figure 8. Line Transient
See Figure 40
Figure 9. Short Circuit
Figure 10. Short-Circuit Release
Figure 11. Soft Start With EN Tied to VIN
Figure 12. Soft Start With EN Tied to VIN
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
Typical Characteristics (continued)
TA = 25°C, VIN = 12 V, and VBOOST – VSW = 4.3 V (unless otherwise noted)
VIN = 12 V
VOUT = 5 V
L = 2.2 µH
COUT = 44 µF
Iout =1 A
VIN = 12 V
VOUT = 3.3 V
Figure 13. Bode Plot
See Figure 34
VIN = 5 V
VOUT = 1.8 V
L = 1.0 µH
COUT = 44 µF
L = 1.5 µH
COUT = 44 µF
Figure 14. Bode Plot
See Figure 40
Iout =1 A
VIN = 5 V
VOUT = 1.2 V
L = 0.56 µH
COUT = 68 µF
Figure 15. Bode Plot
See Figure 49
Figure 16. Bode Plot
See Figure 55
Figure 17. Sync Functionality
Figure 18. Loss of Synchronization
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Iout =1 A
Iout =1 A
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Typical Characteristics (continued)
TA = 25°C, VIN = 12 V, and VBOOST – VSW = 4.3 V (unless otherwise noted)
VSYNC = GND
fSW = 2 MHz
Figure 19. Oscillator Frequency vs Temperature
Figure 20. Oscillator Frequency vs VFB
Figure 21. VFB vs Temperature
Figure 22. VFB vs VIN
VIN = 12 V
Figure 23. Current Limit vs Temperature
10
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Figure 24. RDSON vs Temperature
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
Typical Characteristics (continued)
TA = 25°C, VIN = 12 V, and VBOOST – VSW = 4.3 V (unless otherwise noted)
IQ = IAVIN + IPVIN
Figure 25. IQ (Shutdown) vs Temperature
Copyright © 2008–2016, Texas Instruments Incorporated
Figure 26. IEN vs VEN
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
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7 Detailed Description
7.1 Overview
The LM2734x and LM2734x-Q1 are a constant-frequency, peak current-mode PWM buck regulator IC that
delivers a 1.5-A or 2-A load current. The regulator has a preset switching frequency of 2 MHz. This high
frequency allows the LM2734x and LM2734x-Q1 to operate with small surface-mount capacitors and inductors,
resulting in a DC-DC converter that requires a minimum amount of board space. The LM2734x and LM2734x-Q1
are internally compensated, which reduces design time, and requires few external components.
The following operating description of the LM2734x and LM2734x-Q1 refers to Functional Block Diagram and to
the waveforms in Figure 27. The LM2734x and LM2734x-Q1 supply a regulated output voltage by switching the
internal NMOS switch at a constant frequency and varying the duty cycle. A switching cycle begins at the falling
edge of the reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic
turns on the internal NMOS switch. During this on-time, the SW pin voltage (VSW) swings up to approximately
VIN, and the inductor current (iL) increases with a linear slope. The current-sense amplifier measures iL, which
generates an output proportional to the switch current typically called the sense signal. The sense signal is
summed with the regulator’s corrective ramp and compared to the error amplifier’s output, which is proportional
to the difference between the feedback voltage (VFB) and VREF. When the output of the PWM comparator goes
high, the switch turns off until the next switching cycle begins. During the switch off-time (tOFF), inductor current
discharges through the catch diode D1, which forces the SW pin (VSW) to swing below ground by the forward
voltage (VD1) of the catch diode. The regulator loop adjusts the duty cycle (D) to maintain a constant output
voltage.
VSW
D = tON/TSW
SW Voltage
VIN
tOFF
tON
0
-VD1
Inductor Current
iL
t
TSW
ILPK
IOUT
0
'iL
t
Figure 27. LM2734x Waveforms of SW Pin Voltage and Inductor Current
12
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7.2 Functional Block Diagram
BOOST
D2
LDO
PVIN
C2
Switch
0.15 Ÿ
RSENSE
L
SW
VOUT
iL
+
AVIN
D1
Driver
Current Sense
Amplifier
C3
EN
Under
Voltage
Lockout
PWM Logic
PWM
Comparator
Reset
Pulse
Current
Limit
Thermal
Shutdown
+
+
Error
Signal
ISENSE
+
OVP Comparator
+
Corrective
Ramp
Soft Start
SYNC
Oscillator
1.13V +FB
Internal
Compensation
Error
Amplifier
+
+
R2
GND
+
Freq. Foldback Amplifier
VREF +1.0V
R1
+
0.53V -
Copyright © 2016, Texas Instruments Incorporated
7.3 Feature Description
7.3.1 Boost Function
Capacitor C2 in Functional Block Diagram, commonly referred to as CBOOST, is used to store a voltage VBOOST.
When LM2734x and LM2734x-Q1 start up, an internal LDO charges CBOOST through an internal diode, to a
voltage sufficient to turn the internal NMOS switch on. The gate drive voltage supplied to the internal NMOS
switch is VBOOST – VSW.
During a normal switching cycle, when the internal NMOS control switch is off (tOFF) (see Figure 27), VBOOST
equals VLDO minus the forward voltage of the internal diode (VD2). At the same time the inductor current (iL)
forward biases the catch diode D1 forcing the SW pin to swing below ground by the forward voltage drop of the
catch diode (VD1). Therefore, the voltage stored across CBOOST is calculated with Equation 1 and Equation 2.
VBOOST – VSW = VLDO – VD2 + VD1
VBOOST = VSW + VLDO – VD2 + VD1
(1)
(2)
When the NMOS switch turns on (tON), the switch pin rises to Equation 3.
VSW = VIN – (RDSON × IL)
(3)
Then the D1 undergoes reverse biasing, and forces VBOOST to rise. The voltage at VBOOST is then calculated with
Equation 4.
VBOOST = VIN – (RDSON × IL) + VLDO – VD2 + VD1
(4)
Which is approximately calculated with Equation 5.
VIN + VLDO – 0.4 V
(5)
VBOOST has pulled itself up by its bootstraps, or boosted to a higher voltage.
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Feature Description (continued)
7.3.2 Low Input Voltage Considerations
When the input voltage is below 5 V and the duty cycle is greater than 75%, the gate drive voltage developed
across CBOOST might not be sufficient for proper operation of the NMOS switch. In this case, CBOOST must be
charged through an external Schottky diode attached to a 5-V voltage rail (see Figure 28). This ensures that the
gate drive voltage is high enough for proper operation of the NMOS switch in the triode region. Maintain VBOOST –
VSW less than the 6-V absolute maximum rating.
D2
VIN
PVIN
5V
BOOST
C2
AVIN
L1
C1
VOUT
SW
C3
D1
LM27342
ON
EN
OFF
R1
SYNC
CLK
GND/DAP
FB
R2
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Figure 28. External Diode Charges CBOOST
7.3.3 High Output Voltage Considerations
When the output voltage is greater than 3.3 V, a minimum load current is required to charge CBOOST (see
Figure 29). The minimum load current forward biases the catch diode D1 forcing the SW pin to swing below
ground. This allows CBOOST to charge, ensuring that the gate drive voltage is high enough for proper operation.
The minimum load current depends on many factors including the inductor value.
Figure 29. Minimum Load Current for L = 1.5 µH
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Feature Description (continued)
7.3.4 Frequency Synchronization
The LM2734x and LM2734x-Q1 switching frequency can be synchronized to an external clock, between 1 MHz
and 2.35 MHz, applied at the SYNC pin. At the first rising edge applied to the SYNC pin, the internal oscillator is
overridden and subsequent positive edges initiate switching cycles. If the external SYNC signal is lost during
operation, the LM2734x and LM2734x-Q1 revert to its internal 2-MHz oscillator within 1.5 µs. To disable
frequency synchronization and use the internal 2-MHz oscillator, connect the SYNC pin to GND.
The SYNC pin gives the designer the flexibility to optimize their design. A lower switching frequency can be
chosen for higher efficiency. A higher switching frequency can be chosen to keep EMI out of sensitive ranges
such as the AM radio band. Synchronization can also be used to eliminate beat frequencies generated by the
interaction of multiple switching power converters. Synchronizing multiple switching power converters result in
cleaner power rails.
The selected switching frequency (fSYNC) and the minimum on-time (tMIN) limit the minimum duty cycle (DMIN) of
the device as calculated with Equation 6.
DMIN = tMIN × fSYNC
(6)
Operation below DMIN is not recommended. The LM2734x and LM2734x-Q1 skip pulses to keep the output
voltage in regulation, and the current limit is not ensured. The switching is in phase but no longer at the same
switching frequency as the SYNC signal.
7.3.5 Current Limit
The LM2734x and LM2734x-Q1 use cycle-by-cycle current limiting to protect the output switch. During each
switching cycle, a current limit comparator detects if the output switch current exceeds 2 A minimum (LM27341)
or 2.5 A minimum (LM27342), and turns off the switch until the next switching cycle begins.
7.3.6 Frequency Foldback
The LM2734x and LM2734x-Q1 employ frequency foldback to protect the device from current run-away during
output short-circuit. Once the FB pin voltage falls below regulation, the switch frequency smoothly reduce with
the falling FB voltage until the switch frequency reaches 220 kHz (typical). If the device is synchronized to an
external clock, synchronization is disabled until the FB pin voltage exceeds 0.53 V.
7.3.7 Output Overvoltage Protection
The overvoltage comparator turns off the internal power NFET when the FB pin voltage exceeds the internal
reference voltage by 13% (VFB > 1.13 × VREF). With the power NFET turned off the output voltage decreases
toward the regulation level.
7.3.8 Undervoltage Lockout
Undervoltage lockout (UVLO) prevents the LM2734x and LM2734x-Q1 from operating until the input voltage
exceeds 2.75 V (typical).
The UVLO threshold has approximately 470 mV of hysteresis, so the part operates until VIN drops below
2.28 V (typical). Hysteresis prevents the part from turning off during power up if VIN has finite impedance.
7.3.9 Thermal Shutdown
Thermal shutdown limits total power dissipation by turning off the internal NMOS switch when the IC junction
temperature exceeds 165°C (typical). After thermal shutdown occurs, hysteresis prevents the internal NMOS
switch from turning on until the junction temperature drops to approximately 150°C.
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7.4 Device Functional Modes
7.4.1 Enable Pin and Shutdown Mode
Connect the EN pin to a voltage source greater than 1.8 V to enable operation of the LM2734x and LM2734x-Q1.
Apply a voltage less than 0.4 V to put the part into shutdown mode. In shutdown mode, the quiescent current
drops to typically 70 nA. Switch leakage adds another 40 nA from the input supply. For proper operation, the
LM2734x and LM2734x-Q1 EN pin must never be left floating, and the voltage must never exceed VIN + 0.3 V.
The simplest way to enable the operation of LM2734x and LM2734x-Q1 is to connect the EN pin to AVIN which
allows self start-up of the LM2734x and LM2734x-Q1 when the input voltage is applied.
When the rise time of VIN is longer than the soft-start time of the LM2734x and LM2734x-Q1, this method may
result in an overshoot in output voltage. In such applications, the EN pin voltage can be controlled by a separate
logic signal, or tied to a resistor divider, which reaches 1.8 V after VIN is fully established (see Figure 30). This
minimizes the potential for output voltage overshoot during a slow VIN ramp condition. Use the lowest value of
VIN, seen in your application when calculating the resistor network using Equation 7, to ensure that the 1.8 V
minimum EN threshold is reached.
VIN
PVIN
BOOST
C2
AVIN
L1
C1
VOUT
SW
R3
C3
D1
LM27342
EN
R4
R1
SYNC
CLK
GND/DAP
FB
R2
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Figure 30. Resistor Divider on EN
R3 =
VIN
- 1 x R4
1.8
(7)
7.4.2 Soft-Start Mode
The LM2734x and LM2734x-Q1 have a fixed internal soft-start of 1 ms (typical). During soft start, the error
amplifier’s reference voltage ramps from 0 V to its nominal value of 1 V in approximately 1 ms. This forces the
regulator output to ramp in a controlled fashion, which helps reduce inrush current. Upon soft start, the part is
initially in frequency foldback and the frequency rises as FB rises. The regulator rises gradually to 2 MHz. The
LM2734x and LM2734x-Q1 allows synchronization to an external clock at FB > 0.53 V.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 Inductor Selection
Inductor selection is critical to the performance of the LM2734x and LM2734x-Q1. The selection of the inductor
affects stability, transient response and efficiency. A key factor in inductor selection is determining the ripple
current (ΔiL) (see Figure 27). The ripple current (ΔiL) is important in many ways.
First, by allowing more ripple current, lower inductance values can be used with a corresponding decrease in
physical dimensions and improved transient response. On the other hand, allowing less ripple current increases
the maximum achievable load current and reduce the output voltage ripple (see Output Capacitor for more
details on calculating output voltage ripple). Increasing the maximum load current is achieved by ensuring that
the peak inductor current (ILPK) never exceeds the minimum current limit of 2 A for the LM27341 or 2.5 A for the
LM27342 in Equation 8.
ILPK = IOUT + ΔiL / 2
(8)
Secondly, the slope of the ripple current affects the current control loop. The LM2734x and LM2734x-Q1 have a
fixed slope corrective ramp. When the slope of the current ripple becomes significantly less than the converter’s
corrective ramp, the inductor pole moves from high frequencies to lower frequencies. This negates one
advantage that peak current-mode control has overvoltage-mode control, which is, a single low-frequency pole in
the power stage of the converter. This can reduce the phase margin, crossover frequency and potentially cause
instability in the converter. Contrarily, when the slope of the ripple current becomes significantly greater than the
converter’s corrective ramp, resonant peaking can occur in the control loop. This can also cause instability
(subharmonic oscillation) in the converter. For the power supply designer, this means that for lower switching
frequencies the current ripple must be increased to keep the inductor pole well above crossover. It also means
that for higher switching frequencies the current ripple must be decreased to avoid resonant peaking.
With all these factors, the desired ripple current is selected with Equation 9. The ripple ratio (r) is defined as the
ratio of inductor ripple current (ΔiL) to output current (IOUT), evaluated at maximum load.
r
'iL
IOUT
(9)
A good compromise between physical size, transient response and efficiency is achieved when we set the ripple
ratio between 0.2 and 0.4. The recommended ripple ratio versus duty cycle shown in Figure 31 is based upon
this compromise and control loop optimizations. Note that this is just a guideline. See AN-1197 Selecting
Inductors for Buck Converters for further considerations.
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Application Information (continued)
Figure 31. Recommended Ripple Ratio vs Duty Cycle
The duty cycle (D) can be approximated quickly using the ratio of output voltage (VOUT) to input voltage (VIN) in
Equation 10.
VOUT
VIN
D
(10)
The application's lowest input voltage must be used to calculate the ripple ratio. The catch diode forward voltage
drop (VD1) and the voltage drop across the internal NFET (VDS) must be included to calculate a more accurate
duty cycle. Calculate D by using Equation 11.
VOUT VD1
VIN VD1 VDS
D
(11)
VDS can be approximated with Equation 12.
VDS = IOUT × RDS(ON)
(12)
The diode forward drop (VD1) can range from 0.3 V to 0.5 V depending on the quality of the diode. The lower VD1
is, the higher the operating efficiency of the converter.
Now that the ripple current or ripple ratio is determined, the required inductance is calculated with Equation 13.
L
VOUT VD1
u (1 DMIN)
IOUT u r u fsw
where
•
•
•
DMIN is the duty cycle calculated with the maximum input voltage
fsw is the switching frequency
IOUT is the maximum output current of 2 A
(13)
Using IOUT = 2 A minimizes the inductor's physical size.
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Application Information (continued)
8.1.1.1 Inductor Calculation Example
Operating conditions for the LM27342 are listed in Table 1.
Table 1. Operating Conditions for Inductor Example
OPERATING PARAMETERS
VIN = 7 – 16 V
VOUT = 3.3 V
fSW = 2 MHz
VD1 = 0.5 V
IOUT = 2 A
First the maximum duty cycle is calculated with Equation 14.
DMAX = (VOUT + VD1) / (VIN + VD1 – VDS)
= (3.3 V + 0.5 V) / (7 V + 0.5 V – 0.3 V)
= 0.528
(14)
Using Figure 31 gives us a recommended ripple ratio = 0.4.
Now the minimum duty cycle is calculated with Equation 15.
DMIN = (VOUT + VD1) / (VIN + VD1 – VDS)
= (3.3 V + 0.5 V) / (16 V + 0.5 V – 0.3 V)
= 0.235
(15)
The inductance can now be calculated with Equation 16.
L = (1 – DMIN) × (VOUT + VD1) / (IOUT × r × fsw)
= (1 – 0.235) × (3.3 V + .5 V) / (2 A × 0.4 × 2 MHz)
= 1.817 µH
(16)
This is close to the standard inductance value of 1.8 µH. This leads to a 1% deviation from the recommended
ripple ratio, which is now 0.4038.
Finally, we check that the peak current does not reach the minimum current limit of 2.5 A with Equation 17.
ILPK = IOUT × (1 + r / 2)
= 2 A × (1 + .4038 / 2)
= 2.404 A
(17)
The peak current is less than 2.5 A, so the DC load specification can be met with this ripple ratio. To design for
the LM27341 simply replace IOUT = 1.5 A in the equations for ILPK and see that ILPK does not exceed the
LM27341's current limit of 2 A (minimum).
8.1.2 Inductor Material Selection
When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating.
Inductor saturation results in a sudden reduction in inductance and prevent the regulator from operating correctly.
To prevent the inductor from saturating over the entire –40 °C to 125 °C range, pick an inductor with a saturation
current higher than the upper limit of ICL listed in Electrical Characteristics.
Ferrite core inductors are recommended to reduce AC loss and fringing magnetic flux. The drawback of ferrite
core inductors is their quick saturation characteristic. The current limit circuit has a propagation delay and so is
oftentimes not fast enough to stop a saturated inductor from going above the current limit. This has the potential
to damage the internal switch. To prevent a ferrite core inductor from getting into saturation, the inductor
saturation current rating must be higher than the switch current limit ICL. The LM2734x and LM2734x-Q1 are
quite robust in handling short pulses of current that are a few amps above the current limit. Saturation protection
is provided by a second current limit which is 30% higher than the cycle by cycle current limit. When the
saturation protection is triggered the part turns off the output switch and attempt to soft start. When a
compromise must be made, pick an inductor with a saturation current just above the lower limit of the ICL. Be
sure to validate the short-circuit protection over the intended temperature range.
An inductor's saturation current is usually lower when hot. So consult the inductor vendor if the saturation current
rating is only specified at room temperature.
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Soft saturation inductors such as the iron powder types can also be used. Such inductors do not saturate
suddenly and therefore are safer when there is a severe overload or even shorted output. Their physical sizes
are usually smaller than the Ferrite core inductors. The downside is their fringing flux and higher power
dissipation due to relatively high AC loss, especially at high frequencies.
8.1.3 Input Capacitor
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and Equivalent Series
Inductance (ESL). The recommended input capacitance is 10 µF, although 4.7 µF works well for input voltages
below 6 V. The input voltage rating is specifically stated by the capacitor manufacturer. Make sure to check any
recommended deratings and also verify if there is any significant change in capacitance at the operating input
voltage and the operating temperature. The input capacitor maximum RMS input current rating (IRMS-IN) must be
greater than Equation 18.
IRMS
IN
r2
IOUT u D u (1 D
)
12
where
•
•
•
r is the ripple ratio defined earlier
IOUT is the output current
D is the duty cycle
(18)
Equation 18 shows that maximum RMS capacitor current occurs when D = 0.5. Always calculate the RMS at the
point where the duty cycle, D, is closest to 0.5. The ESL of an input capacitor is usually determined by the
effective cross sectional area of the current path. A large leaded capacitor has high ESL and a 0805 ceramic
chip capacitor has very low ESL. At the operating frequencies of the LM2734x and LM2734x-Q1, certain
capacitors may have an ESL so large that the resulting impedance (2πfL) is higher than that required to provide
stable operation. As a result, TI strongly recommends surface-mount capacitors. Sanyo POSCAP, Tantalum or
Niobium, Panasonic SP or Cornell Dubilier Low ESR are all good choices for input capacitors and have
acceptable ESL. Multilayer ceramic capacitors (MLCC) have very low ESL. For MLCCs, TI recommends using
X7R or X5R dielectrics. Consult the capacitor manufacturer's data sheet to see how rated capacitance varies
over operating conditions.
8.1.4 Output Capacitor
The output capacitor is selected based upon the desired output ripple and transient response. The LM2734x and
LM2734x-Q1 loop compensation is designed for ceramic capacitors. A minimum of 22 µF is required at 2 MHz
(33 µF at 1 MHz) while 47 µF to 100 µF is recommended for improved transient response and higher phase
margin. The output voltage ripple of the converter is calculated with Equation 19.
'VOUT
'iL u (RESR
1
)
8 u fSW u COUT
(19)
When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the
output ripple is approximately sinusoidal and 90° phase shifted from the switching action. Another benefit of
ceramic capacitors is their ability to bypass high-frequency noise. A certain amount of switching edge noise
couples through parasitic capacitances in the inductor to the output. A ceramic capacitor bypasses this noise
while a tantalum does not.
The transient response is determined by the speed of the control loop and the ability of the output capacitor to
provide the initial current of a load transient. Capacitance can be increased significantly with little detriment to the
regulator stability. However, increasing the capacitance provides dimininshing improvement over 100 µF in most
applications, because the bandwidth of the control loop decreases as output capacitance increases. If improved
transient performance is required, add a feedforward capacitor. This becomes especially important for higher
output voltages where the bandwidth of the LM2734x and LM2734x-Q1 is lower (see Feedforward Capacitor
(Optional) and Frequency Synchronization for more information).
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Check the RMS current rating of the capacitor. The RMS current rating of the capacitor chosen must also meet
the following condition with Equation 20.
IRMS
OUT
IOUT u
r
12
where
•
•
IOUT is the output current
r is the ripple ratio
(20)
8.1.5 Catch Diode
The catch diode (D1) conducts during the switch off-time. A Schottky diode is recommended for its fast switching
times and low forward voltage drop. The catch diode must be chosen so that its current rating is greater than
Equation 21.
ID1 = IOUT × (1 - D)
(21)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency, choose a Schottky diode with a low forward voltage drop.
8.1.6 Boost Diode (Optional)
For circuits with input voltages VIN < 5 V and duty cycles (D) > 0.75 V, TI recommends a small-signal Schottky
diode. A good choice is the BAT54 small signal diode. The cathode of the diode is connected to the BOOST pin
and the anode to a 5-V voltage rail.
8.1.7 Boost Capacitor
A ceramic 0.1-µF capacitor with a voltage rating of at least 6.3 V is sufficient. The X7R and X5R MLCCs provide
the best performance.
8.1.8 Output Voltage
The output voltage is set using the following equation where R2 is connected between the FB pin and GND, and
R1 is connected between VOUT and the FB pin in Equation 22. A good starting value for R2 is 1 kΩ.
§ VOUT
·
R1 ¨
1¸ u R2
© VREF
¹
(22)
8.1.9 Feedforward Capacitor (Optional)
A feedforward capacitor (CFF) can improve the transient response of the converter. Place CFF in parallel with R1.
The value of CFF must place a zero in the loop response at, or above, the pole of the output capacitor and RLOAD
as calculated in Equation 23. The CFF capacitor increases the crossover frequency of the design, thus a larger
minimum output capacitance is required for designs using CFF. CFF must only be used with an output capacitance
greater than or equal to 44 µF.
CFF
VOUT u COUT
IOUT u R1
(23)
8.1.10 Calculating Efficiency and Junction Temperature
The complete LM2734x and LM2734x-Q1 DC-DC converter efficiency can be calculated with Equation 24 or
Equation 25.
K
POUT
PIN
(24)
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POUT
POUT PLOSS
(25)
To determine the most significant power losses, see the following equations. Other losses totaling less than 2%
are not discussed.
Power loss (PLOSS) is the sum of two basic types of losses in the converter, switching and conduction.
Conduction losses usually dominate at higher output loads, where as switching losses remain relatively fixed and
dominate at lower output loads. The first step in determining the losses is to calculate the duty cycle (D) with
Equation 26.
D
VOUT VD1
VIN VD1 VDS
(26)
VDS is the voltage drop across the internal NFET when it is on, and is equal to Equation 27.
VDS = IOUT × RDSON
(27)
VD is the forward voltage drop across the Schottky diode. It can be obtained from the Electrical Characteristics
section of the Schottky diode data sheet. If the voltage drop across the inductor (VDCR) is accounted for, the
equation changes to Equation 28.
D
VOUT VD1 VDCR
VIN VD1 VDS
(28)
VDCR usually gives only a minor duty cycle change, and has been omitted in the examples for simplicity.
8.1.10.1 Schottky Diode Conduction Losses
The conduction losses in the free-wheeling Schottky diode are calculated with Equation 29.
PDIODE = VD1 × IOUT (1 – D)
(29)
Often this is the single most significant power loss in the circuit. Care must be taken to choose a Schottky diode
that has a low forward voltage drop.
8.1.10.2 Inductor Conduction Losses
Another significant external power loss is the conduction loss in the output inductor. The equation can be
simplified to Equation 30.
PIND = IOUT2 × RDCR
(30)
8.1.10.3 MOSFET Conduction Losses
The LM2734x and LM2734x-Q1 conduction loss is mainly associated with the internal NFET calculated with
Equation 31.
PCOND = IOUT2 × RDSON × D
(31)
8.1.10.4 MOSFET Switching Losses
Switching losses are also associated with the internal NFET. They occur during the switch on and off transition
periods, where voltages and currents overlap resulting in power loss. The simplest means to determine this loss
is to empirically measuring the rise and fall times (10% to 90%) of the switch at the switch node with
Equation 32, Equation 33, and Equation 34. Typical values are listed in Table 2.
PSWF = 1 / 2 (VIN × IOUT × fSW × tFALL)
PSWR = 1 / 2 (VIN × IOUT × fSW × tRISE)
PSW = PSWF + PSWR
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(32)
(33)
(34)
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Table 2. Typical Rise and Fall Times vs Input Voltage
VIN
tRISE
tFALL
5V
8 ns
8 ns
10 V
9 ns
9 ns
15 V
10 ns
10 ns
8.1.10.5 IC Quiescent Losses
Another loss is the power required for operation of the internal circuitry calculated with Equation 35.
PQ = IQ × VIN
(35)
IQ is the quiescent operating current, and is typically around 2.4 mA.
8.1.10.6 MOSFET Driver Losses
The other operating power that needs calculation is that required to drive the internal NFET with Equation 36.
PBOOST = IBOOST × VBOOST
(36)
VBOOST is normally between 3 VDC and 5 VDC. The IBOOST rms current is dependant on switching frequency fSW.
IBOOST is approximately 8.2 mA at 2 MHz and 4.4 mA at 1 MHz.
8.1.10.7 Total Power Losses
Total power losses is calculated with Equation 37.
PLOSS = PCOND + PSWR + PSWF + PQ + PBOOST + PDIODE + PIND
(37)
Losses internal to the LM2734x and LM2734x-Q1 is calculated with Equation 38.
PINTERNAL = PCOND + PSWR + PSWF + PQ + PBOOST
(38)
8.1.10.8 Efficiency Calculation Example
Operating conditions are listed in Table 3.
Table 3. Operating Conditions for Efficiency Calculation
OPERATING PARAMETERS
VIN = 12 V
VOUT = 3.3 V
IOUT = 2 A
fSW = 2 MHZ
VD1 = 0.5 V
RDCR = 20 mΩ
Internal power losses are calculated with Equation 39 through Equation 43.
PCOND = IOUT2 × RDSON × D
= 22 × 0.15 Ω × 0.314
= 188 mW
PSW = (VIN × IOUT × fSW × tFALL)
= (12 V × 2A × 2 MHz × 10 ns)
= 480 mW
PQ = IQ × VIN
= 2.4 mA × 12 V
= 29 mW
PBOOST = IBOOST × VBOOST
= 8.2 mA × 4.5 V
= 37 mW
PINTERNAL = PCOND + PSW + PQ + PBOOST= 733 mW
(39)
(40)
(41)
(42)
(43)
Total power losses are calculated with Equation 44 through Equation 46.
PDIODE = VD1 × IOUT (1 – D)
= 0.5 V × 2 × (1 – 0.314)
= 686 mW
PIND = IOUT2 × RDCR
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= 22 × 20 mΩ
= 80 mW
PLOSS = PINTERNAL + PDIODE + PIND = 1.499 W
(45)
(46)
The efficiency can now be estimated with Equation 47.
POUT
POUT PLOSS
K
6.6W
6.6W 1.499W
81%
(47)
With this information, we can estimate the junction temperature of the LM2734x and LM2734x-Q1.
8.1.10.9 Calculating Junction Temperature
The thermal definitions are:
• TJ = IC junction temperature
• TA = Ambient temperature
• RθJC = Thermal resistance from IC junction to device case
• RθJA = Thermal resistance from IC junction to ambient air
Figure 32. Cross-Sectional View of Integrated Circuit Mounted on a Printed Circuit Board
Heat in the LM2734x and LM2734x-Q1 due to internal power dissipation is removed through conduction and/or
convection.
8.1.10.9.1 Conduction
Heat transfer occurs through cross sectional areas of material. Depending on the material, the transfer of heat
can be considered to have poor-to-good thermal conductivity properties (insulator versus conductor).
Heat transfer goes from Silicon → Lead Frame → PCB.
8.1.10.9.2 Convection
Heat transfer is by means of airflow. This could be from a fan or natural convection. Natural convection occurs
when air currents rise from the hot device to cooler air.
Thermal impedance is defined with Equation 48.
RT
'T
Power
(48)
Thermal impedance from the silicon junction to the ambient air is defined with Equation 49.
RTJA
24
TJ TA
Power
(49)
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This impedance can vary depending on the thermal properties of the PCB. This includes PCB size, weight of
copper used to route traces , the ground plane, and the number of layers within the PCB. The type and number
of thermal vias can also make a large difference in the thermal impedance. Thermal vias are necessary in most
applications. They conduct heat from the surface of the PCB to the ground plane. Six to nine thermal vias must
be placed under the exposed pad to the ground plane. Placing more than nine thermal vias results in only a
small reduction to RθJA for the same copper area. These vias must have 8-mil holes to avoid wicking solder away
from the DAP. See AN-1187 Leadless Leadframe Package (SNOA401) and AN-1520 A Guide to Board Layout
for Best Thermal Resistance for Exposed Packages (SNVA183) for more information on package thermal
performance. If a compromise for cost needs to be made, the thermal vias for the MSOP-PowerPAD package
can range from 8 mils to 14 mils, increasing the possibility of solder wicking.
To predict the silicon junction temperature for a given application, three methods can be used. The first is useful
before prototyping and the other two can more accurately predict the junction temperature within the application.
8.1.10.9.3 Method 1
The first method predicts the junction temperature by extrapolating a best guess RθJA from the table or graph.
The tables and graph are for natural convection. The internal dissipation can be calculated using the efficiency
calculations. This allows the user to make a rough prediction of the junction temperature in their application.
Methods two and three can later be used to determine the junction temperature more accurately.
Table 4 and Table 5 have values of RθJA for the WSON and the MSOP-PowerPAD packages.
Table 4. RθJA Values for the MSOP-PowerPAD at 1-W Dissipation
NUMBER OF
BOARD LAYERS
SIZE OF BOTTOM LAYER
COPPER CONNECTED TO DAP
SIZE OF TOP LAYER COPPER
CONNECTED TO DAP
NUMBER OF 10 MIL
THERMAL VIAS
RθJA
2
0.25 in2
0.05 in2
8
80.6°C/W
2
0.5625 in
0.05 in2
8
70.9°C/W
2
1 in2
0.05 in2
8
62.1°C/W
2
2
2
2
1.3225 in
0.05 in
8
54.6°C/W
4 (eval board)
3.25 in2
2.25 in2
14
35.3°C/W
Table 5. RθJA Values for the WSON at 1-W Dissipation
NUMBER OF
BOARD LAYERS
SIZE OF BOTTOM LAYER
COPPER CONNECTED TO DAP
SIZE OF TOP LAYER COPPER
CONNECTED TO DAP
NUMBER OF 8 MIL
THERMAL VIAS
RθJA
2
0.25 in2
0.05 in2
8
78°C/W
2
2
2
0.5625 in
0.05 in
8
65.6°C/W
2
1 in2
0.05 in2
8
58.6°C/W
2
1.3225 in2
0.05 in2
8
50°C/W
15
30.7°C/W
4 (eval board)
2
3.25 in
Copyright © 2008–2016, Texas Instruments Incorporated
2
2.25 in
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www.ti.com
Eight thermal vias and natural convection
Figure 33. Estimate of Thermal Resistance vs Ground Copper Area
8.1.10.9.4 Method 2
The second method requires the user to know the thermal impedance of the silicon junction to case. RθJC is
approximately 9.5°C/W for the MSOP-PowerPAD package or 9.1°C/W for the WSON. The case temperature
must be measured on the bottom of the PCB at a thermal through directly under the DAP of the LM2734x and
LM2734x-Q1. The solder resist must be removed from this area for temperature testing. The reading is more
accurate if it is taken midway between pins 2 and 9, where the NMOS switch is placed. Knowing the internal
dissipation from Method 1, calculate the case temperature (TC) with Equation 50 and Equation 51.
RTJC
TJ TC
Power
(50)
(51)
TJ = (RθJC × PLOSS) + TC
8.1.10.9.4.1 Method 2 Example
The operating conditions are the same as the previous efficiency calculation listed in Table 6.
Table 6. Operating Conditions for Efficiency Calculation
OPERATING PARAMETERS
VIN = 12 V
VOUT = 3.3 V
IOUT = 2 A
fSW = 2 MHz
VD1 = 0.5 V
RDCR = 20 mΩ
Internal power losses are calculated with Equation 52 through Equation 56.
PCOND = IOUT2 × RDSON × D
= 22 × 0.15 Ω × 0.314
= 188 mW
PSW = (VIN × IOUT × fSW × tFALL)
= (12 V × 2 A × 2 MHz × 10 ns)
= 480 mW
PQ = IQ × VIN
= 1.5 mA × 12 V
= 29 mW
PBOOST = IBOOST × VBOOST
= 7 mA × 4.5 V
= 37 mW
PINTERNAL = PCOND + PSW + PQ + PBOOST = 733 mW
26
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(52)
(53)
(54)
(55)
(56)
Copyright © 2008–2016, Texas Instruments Incorporated
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
The junction temperature can now be estimated with Equation 57.
TJ = (RθJC × PINTERNAL) + TC
(57)
A TI MSOP-PowerPAD evaluation board was used to determine the TJ of the LM2734x and LM2734x-Q1. The
four layer PCB is constructed using FR4 with 2-oz copper traces. There is a ground plane on the internal layer
directly beneath the device, and a ground plane on the bottom layer. The ground plane is accessed by fourteen
10-mil vias. The board measures 2 in × 2 in (50.8 mm × 50.8 mm). It was placed in a container with no airflow.
The case temperature measured on this LM27342MY Demo Board was 48.7°C. Therefore, TJ is calculated with
Equation 58 and Equation 59.
TJ = (9.5°C/W × 733 mW) + 48.7°C
TJ = 55.66°C
(58)
(59)
To keep the junction temperature below 125°C for this layout, the ambient temperature must stay below
94.33°C as in Equation 60, Equation 61, and Equation 62.
TA_MAX = TJ_MAX – TJ +TA
TA_MAX = 125°C – 55.66°C + 25°C
TA_MAX = 94.33°C
(60)
(61)
(62)
8.1.10.9.5 Method 3
The third method can also give a very accurate estimate of silicon junction temperature. The first step is to
determine RθJA of the application. The LM2734x and LM2734x-Q1 has overtemperature protection circuitry.
When the silicon temperature reaches 165°C, the device stops switching. The protection circuitry has a
hysteresis of 15°C. Once the silicon temperature has decreased to approximately 150°C, the device starts
switching again. Knowing this, the RθJA for any PCB can be characterized during the early stages of the design
by raising the ambient temperature in the given application until the circuit enters thermal shutdown. If the SWpin is monitored, it is obvious when the internal NFET stops switching indicating a junction temperature of 165°C.
We can calculate the internal power dissipation from the above methods. All that is required for calculation is the
estimate of RDSON at 165°C. The value is approximately 0.267 Ω. With this, the junction temperature, and the
ambient temperature, RθJA, can be determined with Equation 63.
RTJA
165qC TA
PINTERNAL
(63)
Once this is determined, the maximum ambient temperature allowed for a desired junction temperature can be
found.
8.1.10.9.5.1 Method 3 Example
The operating conditions are the same as the previous efficiency calculation listed in Table 7.
Table 7. Operating Conditions for Efficiency Calculation
OPERATING PARAMETERS
VIN = 12 V
VOUT = 3.3 V
IOUT = 2A
fSW = 2 MHz
VD1 = 0.5 V
RDCR = 20 mΩ
Internal power losses are calculated with Equation 64 through Equation 68.
PCOND = IOUT2 × RDSON × D
= 22 × 0.267 Ω x .314
= 335 mW
PSW = (VIN × IOUT × fSW × tFALL)
= (12 V × 2 A × 2 MHz × 10 nS)
= 480 mW
PQ = IQ × VIN
= 1.5 mA × 12 V
= 29 mW
PBOOST = IBOOST × VBOOST
= 7 mA × 4.5 V
Copyright © 2008–2016, Texas Instruments Incorporated
(64)
(65)
(66)
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= 37 mW
PINTERNAL = PCOND + PSW + PQ + PBOOST = 881 mW
(67)
(68)
A TI MSOP-PowerPAD evaluation board was used to determine the RθJA of the board. The four-layer PCB is
constructed using FR4 with 2oz copper traces. There is a ground plane on the internal layer directly beneath the
device, and a ground plane on the bottom layer. The ground plane is accessed by fourteen 10-mil vias. The
board measures 2 in × 2 in (50.8 mm × 50.8 mm). It was placed in an oven with no forced airflow.
The ambient temperature was raised to 132°C, and at that temperature, the device went into thermal shutdown.
RθJA can be calculated with Equation 69.
RTJA
165qC 132qC
0.881 W
37.46 qC / W
(69)
To keep the junction temperature below 125°C for this layout, the ambient temperature must stay below 92°C as
in Equation 70, Equation 71, and Equation 72.
TA_MAX = TJ_MAX – (RθJA × PINTERNAL)
TA_MAX = 125°C – (37.46°C/W × 0.881 W)
TA_MAX = 92°C
(70)
(71)
(72)
This calculation of the maximum ambient temperature is only 2.3°C different from the calculation using method 2.
The methods described above to find the junction temperature in the MSOP-PowerPAD package can also be
used to calculate the junction temperature in the WSON package. The 10-pin WSON package has a
RθJC = 9.1°C/W, while RθJA can vary depending on the layout. RθJA can be calculated in the same manner as
described in method 3.
8.2 Typical Applications
8.2.1 LM2734x Configuration From VIN = 7 V to 16 V, VOUT = 5 V For Full Load at 2 MHz
V IN
PVIN
BOOST
C2
AVIN
L1
C1
SW
VOUT
C3
D1
LM27341/2
C4
ON
EN
OFF
R1
FB
SYNC
CLK
2 MHz
GND/DAP
R2
C5
Copyright © 2016, Texas Instruments Incorporated
VIN = 7 V to 16 V, VOUT = 5 V
fSW = 2 MHz
IOUT = Full load
Figure 34. LM2734x Configuration From VIN = 7 V to 16 V, VOUT = 5 V For Full Load at 2-MHz Schematic
28
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
Typical Applications (continued)
8.2.1.1 Design Requirements
Create 5-V output at full-rated load for VIN range of 7 V to 16 V with switching frequency FSW = 2 MHz using
external synchronization.
8.2.1.2 Detailed Design Procedure
The device must be able to operate at any voltage within the recommended operating range. The load current
must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle
the full expected load current as well as the peak current generated during load transients and start-up. The
inrush current at start-up depends on the output capacitor selection.
Table 8 lists the bill of materials for VIN = 7 V to 16 V, VOUT = 5 V for full load at 2 MHz. See Figure 34.
Table 8. Bill of Materials
PART NAME
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
Buck regulator
U1
1.5-A or 2-A Buck regulator
LM2734x and LM2734x-Q1
TI
CPVIN
C1
10 µF
GRM32DR71E106KA12L
Murata
CBOOST
C2
0.1 µF
GRM188R71C104KA01D
Murata
COUT
C3
22 µF
C3225X7R1C226K
TDK
COUT
C4
22 µF
C3225X7R1C226K
TDK
CFF
C5
0.18 µF
0603ZC184KAT2A
AVX
Catch diode
D1
Schottky diode, Vf = 0.32 V
CMS06
Toshiba
Inductor
L1
2.2 µH
CDRHD5D28RHPNP
Sumida
Feedback resistor
R1
560 Ω
CRCW0603560RFKEA
Vishay
Feedback resistor
R2
140 Ω
CRCW0603140RFKEA
Vishay
8.2.1.3 Application Curves
Figure 35. LM27342 Efficiency vs Load Current
Copyright © 2008–2016, Texas Instruments Incorporated
IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 36. Transient Response
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
www.ti.com
8.2.2 LM2734x Configuration From VIN = 7 V to 16 V, VOUT = 5 V For Full Load at 1 MHz
VIN
PVIN
BOOST
C2
AVIN
L1
C1
VOUT
SW
D1
LM27341/2
C3
C4
ON
OFF
EN
R1
CLK
1 MHz
SYNC
FB
GND / DAP
R2
C5
Copyright © 2016, Texas Instruments Incorporated
VIN = 7 V to 16 V, VOUT = 5 V
fSW = 1 MHz
IOUT = Full load
Figure 37. LM2734x Configuration From VIN = 7 V to 16 V, VOUT = 5 V For Full Load at 1-MHz Schematic
8.2.2.1 Design Requirements
Create 5-V output at full-rated load for VIN range of 7 V to 16 V with switching frequency FSW = 1 MHz using
external synchronization.
8.2.2.2 Detailed Design Procedure
The device must be able to operate at any voltage within the recommended operating range. The load current
must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle
the full expected load current as well as the peak current generated during load transients and start-up. The
inrush current at start-up depends on the output capacitor selection.
Table 9 lists the bill of materials for VIN = 7 V to 16 V, VOUT = 5 V for full load at 1 MHz. See Figure 37.
Table 9. Bill of Materials
PART NAME
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
Buck regulator
U1
1.5-A or 2-A Buck regulator
LM2734x and LM2734x-Q1
TI
CPVIN
C1
10 µF
GRM32DR71E106KA12L
Murata
CBOOST
C2
0.1 µF
GRM188R71C104KA01D
Murata
COUT
C3
47 µF
GRM32ER61A476KE20L
Murata
COUT
C4
22 µF
C3225X7R1C226K
TDK
CFF
C5
0.27 µF
C0603C274K4RACTU
Kemet
Catch diode
D1
Schottky diode, Vf = 0.32 V
CMS06
Toshiba
Inductor
L1
3.3 µH
CDRH6D26HPNP
Sumida
Feedback resistor
R1
560 Ω
CRCW0603560RFKEA
Vishay
Feedback resistor
R2
140 Ω
CRCW0603140RFKEA
Vishay
30
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LM27341, LM27342, LM27341-Q1, LM27342-Q1
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
8.2.2.3 Application Curves
Figure 38. LM27342 Efficiency vs Load Current
8.2.3
IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 39. Transient Response
LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2 MHz
V IN
PVIN
BOOST
C2
AVIN
L1
C1
SW
VOUT
D1
LM27341/2
C3
C4
ON
EN
OFF
R1
FB
SYNC
CLK
2 MHz
GND/DAP
R2
C5
Copyright © 2016, Texas Instruments Incorporated
VIN = 5 V to 16 V, VOUT = 3.3 V
fSW = 2 MHz
IOUT = Full load
Figure 40. LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2-MHz Schematic
8.2.3.1 Design Requirements
Create 3.3-V output at full-rated load for VIN range of 5 V to 16 V with switching frequency FSW = 2 MHz using
external synchronization.
Copyright © 2008–2016, Texas Instruments Incorporated
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
www.ti.com
8.2.3.2 Detailed Design Procedure
The device must be able to operate at any voltage within the recommended operating range. The load current
must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle
the full expected load current as well as the peak current generated during load transients and start-up. The
inrush current at start-up depends on the output capacitor selection.
Table 10 lists the bill of materials for VIN = 5 V to 16 V, VOUT = 3.3 V for full load at 2 MHz. See Figure 40.
Table 10. Bill of Materials
PART NAME
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
Buck regulator
U1
1.5-A or 2-A Buck regulator
LM2734x and LM2734x-Q1
TI
CPVIN
C1
10 µF
GRM32DR71E106KA12L
Murata
CBOOST
C2
0.1 µF
GRM188R71C104KA01D
Murata
COUT
C3
22 µF
C3225X7R1C226K
TDK
COUT
C4
22 µF
C3225X7R1C226K
TDK
CFF
C5
0.18 µF
0603ZC184KAT2A
AVX
Catch diode
D1
Schottky diode, Vf = 0.32 V
CMS06
Toshiba
Inductor
L1
1.5 µH
CDRH5D18BHPNP
Sumida
Feedback resistor
R1
430 Ω
CRCW0603430RFKEA
Vishay
Feedback resistor
R2
187 Ω
CRCW0603187RFKEA
Vishay
8.2.3.3 Application Curves
Figure 41. LM27342 Efficiency vs Load Current
32
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IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 42. Transient Response
Copyright © 2008–2016, Texas Instruments Incorporated
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LM27341, LM27342, LM27341-Q1, LM27342-Q1
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
8.2.4 LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2 MHz
With SYNC = GND
BOOST
PVIN
VIN
C2
AVIN
L1
C1
SW
EN
VOUT
D1
LM27341/2
C3
C4
R1
SYNC
FB
GND/DAP
R2
Copyright © 2016, Texas Instruments Incorporated
VIN = 5 V to 16 V, VOUT = 3.3 V
fSW = 2 MHz
IOUT = Full load
Figure 43. LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V
For Full Load at 2 MHz With SYNC = GND Schematic
8.2.4.1 Design Requirements
Create 3.3-V output at full-rated load for VIN range of 5 V to 16 V with switching frequency FSW = 2 MHz using
internal oscillator.
8.2.4.2 Detailed Design Procedure
The device must be able to operate at any voltage within the recommended operating range. The load current
must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle
the full expected load current as well as the peak current generated during load transients and start-up. The
inrush current at start-up depends on the output capacitor selection.
Table 11 lists the bill of materials for VIN = 5 V to 16 V, VOUT = 3.3 V for full load at 2 MHz with SYNC = GND.
See Figure 43.
Copyright © 2008–2016, Texas Instruments Incorporated
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Table 11. Bill of Materials
PART NAME
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
Buck regulator
U1
1.5-A or 2-A Buck regulator
LM2734x and LM2734x-Q1
TI
CPVIN
C1
10 µF
GRM32DR71E106KA12L
Murata
CBOOST
C2
0.1 µF
GRM188R71C104KA01D
Murata
COUT
C3
22 µF
C3225X7R1C226K
TDK
COUT
C4
22 µF
C3225X7R1C226K
TDK
Catch diode
D1
Schottky diode, Vf = 0.32 V
CMS06
Toshiba
Inductor
L1
1.5 µH
CDRH5D18BHPNP
Sumida
Feedback resistor
R1
430 Ω
CRCW0603430RFKEA
Vishay
Feedback resistor
R2
187 Ω
CRCW0603187RFKEA
Vishay
8.2.4.3 Application Curves
IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 45. Transient Response
Figure 44. LM27342 Efficiency vs Load Current
8.2.5 LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2 MHz
With SYNC = 1 MHz
VIN
PVIN
BOOST
C2
AVIN
L1
C1
VOUT
SW
D1
LM27341/2
C3
C4
ON
OFF
EN
R1
CLK
1 MHz
SYNC
FB
GND / DAP
R2
C5
Copyright © 2016, Texas Instruments Incorporated
VIN = 5 V to 16 V, VOUT = 3.3 V
fSW = 1 MHz
IOUT = Full load
Figure 46. LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V
For Full Load at 2 MHz With SYNC = 1-MHz Schematic
34
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
8.2.5.1 Design Requirements
Create 1.8-V output at full-rated load for VIN range of 5 V to 16 V with switching frequency FSW = 1 MHz using
external synchronization.
8.2.5.2 Detailed Design Procedure
The device must be able to operate at any voltage within the recommended operating range. The load current
must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle
the full expected load current as well as the peak current generated during load transients and start-up. The
inrush current at start-up depends on the output capacitor selection.
Table 12 lists the bill of materials for VIN = 5 V to 16 V, VOUT = 3.3 V for full load at 2 MHz with SYNC = 1 MHz.
See Figure 46.
Table 12. Bill of Materials
PART NAME
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
Buck regulator
U1
1.5-A or 2-A Buck regulator
LM2734x and LM2734x-Q1
TI
CPVIN
C1
10 µF
GRM32DR71E106KA12L
Murata
CBOOST
C2
0.1 µF
GRM188R71C104KA01D
Murata
COUT
C3
47 µF
GRM32ER61A476KE20L
Murata
COUT
C4
22 µF
C3225X7R1C226K
TDK
CFF
C5
0.27 µF
C0603C274K4RACTU
Kemet
Catch diode
D1
Schottky diode, Vf = 0.32 V
CMS06
Toshiba
Inductor
L1
2.7 µH
CDRH5D18BHPNP
Sumida
Feedback resistor
R1
430 Ω
CRCW0603430RFKEA
Vishay
Feedback resistor
R2
187 Ω
CRCW0603187RFKEA
Vishay
8.2.5.3 Application Curves
Figure 47. LM27342 Efficiency vs Load Current
Copyright © 2008–2016, Texas Instruments Incorporated
IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 48. Transient Response
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8.2.6 LM2734x Configuration From VIN = 3.3 V to 16 V, VOUT = 1.8 V For Full Load at 2 MHz
With SYNC = 1 GND
BOOST
PVIN
VIN
C2
AVIN
L1
C1
SW
EN
VOUT
C3
D1
LM27341/2
C4
R1
SYNC
FB
GND/DAP
R2
Copyright © 2016, Texas Instruments Incorporated
VIN = 3.3 V to 16 V, VOUT = 1.8 V
fSW = 2 MHz
IOUT = Full load
Figure 49. LM2734x Configuration From VIN = 3.3 V to 16 V, VOUT = 1.8 V
For Full Load at 2 MHz With SYNC = GND Schematic
8.2.6.1 Design Requirements
Create 1.8-V output at full-rated load for VIN range of 3.3 V to 16 V with switching frequency FSW = 2 MHz using
internal oscillator.
8.2.6.2 Detailed Design Procedure
The device must be able to operate at any voltage within the recommended operating range. The load current
must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle
the full expected load current as well as the peak current generated during load transients and start-up. The
inrush current at start-up depends on the output capacitor selection.
Table 13 lists the bill of materials for VIN = 3.3 V to 16 V, VOUT = 1.8 V for full load at 2 MHz with SYNC = GND.
See Figure 49.
Table 13. Bill of Materials
PART NAME
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
Buck regulator
U1
1.5-A or 2-A Buck regulator
LM2734x and LM2734x-Q1
TI
CPVIN
C1
10 µF
GRM32DR71E106KA12L
Murata
CBOOST
C2
0.1 µF
GRM188R71C104KA01D
Murata
COUT
C3
22 µF
C3225X7R1C226K
TDK
COUT
C4
22 µF
C3225X7R1C226K
TDK
Catch diode
D1
Schottky diode, Vf = 0.32 V
CMS06
Toshiba
Inductor
L1
1 µH
CDRH5D18BHPNP
Sumida
Feedback resistor
R1
12 kΩ
CRCW060312K0FKEA
Vishay
Feedback resistor
R2
15 kΩ
CRCW060315K0FKEA
Vishay
36
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
8.2.6.3 Application Curves
IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 51. Transient Response
Figure 50. LM27342 Efficiency vs Load Current
8.2.7 LM2734x Configuration From VIN = 3.3 V to 16 V, VOUT = 1.8 V For Full Load at 2 MHz
With SYNC = 1 MHz
VIN
PVIN
BOOST
C2
AVIN
L1
C1
VOUT
SW
D1
LM27341/2
C3
C4
ON
OFF
EN
R1
CLK
1 MHz
SYNC
FB
GND / DAP
R2
C5
Copyright © 2016, Texas Instruments Incorporated
VIN = 3.3 V to 16 V, VOUT = 1.8 V
fSW = 1 MHz
IOUT = Full load
Figure 52. LM2734x Configuration From VIN = 3.3 V to 16 V, VOUT = 1.8 V
For Full Load at 2 MHz With SYNC = 1-MHz Schematic
8.2.7.1 Design Requirements
Create 1.8-V output at full-rated load for VIN range of 3.3 V to 16 V with switching frequency FSW = 1 MHz using
external synchronization.
8.2.7.2 Detailed Design Procedure
The device must be able to operate at any voltage within the recommended operating range. The load current
must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle
the full expected load current as well as the peak current generated during load transients and start-up. The
inrush current at start-up depends on the output capacitor selection.
Table 14 lists the bill of materials for VIN = 3.3 V to 16 V, VOUT = 1.8 V for full load at 2 MHz with SYNC = 1 MHz.
See Figure 52.
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Table 14. Bill of Materials
PART NAME
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
Buck regulator
U1
1.5-A or 2-A Buck regulator
LM2734x and LM2734x-Q1
TI
CPVIN
C1
10 µF
GRM32DR71E106KA12L
Murata
CBOOST
C2
0.1 µF
GRM188R71C104KA01D
Murata
COUT
C3
22 µF
C3225X7R1C226K
TDK
COUT
C4
22 µF
C3225X7R1C226K
TDK
CFF
C5
3.9 nF
GRM188R71H392KA01D
Murata
Catch diode
D1
Schottky diode, Vf = 0.32 V
CMS06
Toshiba
Inductor
L1
1.8 µH
CDRH5D18BHPNP
Sumida
Feedback resistor
R1
12 kΩ
CRCW060312K0FKEA
Vishay
Feedback resistor
R2
15 kΩ
CRCW060315K0FKEA
Vishay
8.2.7.3 Application Curves
Figure 53. LM27342 Efficiency vs Load Current
38
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IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 54. Transient Response
Copyright © 2008–2016, Texas Instruments Incorporated
Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1
LM27341, LM27342, LM27341-Q1, LM27342-Q1
www.ti.com
SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
8.2.8 LM2734x Configuration From VIN = 3.3 V to 9 V, VOUT = 1.2 V For Full Load at 2 MHz
With SYNC = 2 MHz
V IN
PVIN
BOOST
C2
AVIN
L1
C1
SW
VOUT
C3
D1
LM27341/2
C4
ON
EN
OFF
R1
FB
SYNC
CLK
2 MHz
GND/DAP
R2
C5
Copyright © 2016, Texas Instruments Incorporated
VIN = 3.3 V to 9 V, VOUT = 1.2 V
fSW = 2 MHz
IOUT = Full load
Figure 55. LM2734x Configuration From VIN = 3.3 V to 9 V, VOUT = 1.2 V
For Full Load at 2 MHz With SYNC = 2-MHz Schematic
8.2.8.1 Design Requirements
Create 1.2-V output at full-rated load for VIN range of 3.3 V to 9 V with switching frequency FSW = 2 MHz using
external synchronization.
8.2.8.2 Detailed Design Procedure
The device must be able to operate at any voltage within the recommended operating range. The load current
must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle
the full expected load current as well as the peak current generated during load transients and start-up. The
inrush current at start-up depends on the output capacitor selection.
Table 15 lists the bill of materials for VIN = 3.3 V to 9 V, VOUT = 1.2 V for full load at 2 MHz with SYNC = 2 MHz.
See Figure 55.
Table 15. Bill of Materials
PART NAME
PART ID
PART VALUE
PART NUMBER
MANUFACTURER
Buck regulator
U1
1.5-A or 2-A Buck regulator
LM2734x and LM2734x-Q1
TI
CPVIN
C1
10 µF
GRM32DR71E106KA12L
Murata
CBOOST
C2
0.1 µF
GRM188R71C104KA01D
Murata
COUT
C3
47 µF
GRM32ER61A476KE20L
Murata
COUT
C4
22 µF
C3225X7R1C226K
TDK
CFF
C5
Not mounted
Catch diode
D1
Schottky diode, Vf = 0.32 V
CMS06
Toshiba
Inductor
L1
0.56 µH
CDRH2D18/HPNP
Sumida
Feedback resistor
R1
1.02 kΩ
CRCW06031K02FKEA
Vishay
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Table 15. Bill of Materials (continued)
PART NAME
Feedback resistor
PART ID
R2
PART VALUE
5.1 kΩ
PART NUMBER
CRCW06035K10FKEA
MANUFACTURER
Vishay
8.2.8.3 Application Curves
Figure 56. LM27342 Efficiency vs Load Current
40
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IOUT = 100 mA – 2 A at Slewrate = 2 A / µs
Figure 57. Transient Response
Copyright © 2008–2016, Texas Instruments Incorporated
Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1
LM27341, LM27342, LM27341-Q1, LM27342-Q1
www.ti.com
SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
9 Power Supply Recommendations
The input voltage is rated as 3 V to 18 V; however, take care in certain circuit configurations (for example,
VBOOST derived from VIN where the requirement that VBOOST – VSW < 5.5 V must be observed). Also, for the best
efficiency, VBOOST must be at least 2.5 V above VSW. The voltage on the enable pin must not exceed VIN by more
than 0.3 V.
10 Layout
10.1 Layout Guidelines
10.1.1 Compact Layout
The performance of any switching converter depends as much upon the layout of the PCB as the component
selection. The following guidelines help the user design a circuit with maximum rejection of outside EMI and
minimum generation of unwanted EMI.
Parasitic inductance can be reduced by keeping the power path components close together and keeping the
area of the loops small, on which high currents travel. Short, thick traces or copper pours (shapes) are best. In
particular, the switch node (where L1, D1, and the SW pin connect) must be just large enough to connect all
three components without excessive heating from the current it carries. The LM2734x and LM2734x-Q1 operate
in two distinct cycles (see Figure 27) whose high current paths are shown in Figure 58.
+
-
Figure 58. Buck Converter Current Loops
The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop
represents the high current path during the off-time.
10.1.2 Ground Plane and Shape Routing
The diagram of Figure 58 is also useful for analyzing the flow of continuous current versus the flow of pulsating
currents. The circuit paths with current flow during both the on-time and off-time are considered to be continuous
current, while those that carry current during the on-time or off-time only are pulsating currents. Preference in
routing must be given to the pulsating current paths, as these are the portions of the circuit most likely to emit
EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just like
any other circuit path. The path between the input source and the input capacitor and the path between the catch
diode and the load are examples of continuous current paths. In contrast, the path between the catch diode and
the input capacitor carries a large pulsating current. This path must be routed with a short, thick shape,
preferably on the component side of the PCB. Multiple vias in parallel must be used right at the pad of the input
capacitor to connect the component side shapes to the ground plane. A second pulsating current loop that is
often ignored is the gate drive loop formed by the SW and BOOST pins and boost capacitor CBOOST. To minimize
this loop and the EMI it generates, keep CBOOST close to the SW and BOOST pins.
10.1.3 FB Loop
The FB pin is a high-impedance input, and the loop created by R2, the FB pin and ground must be made as
small as possible to maximize noise rejection. R2 must therefore be placed as close as possible to the FB and
GND pins of the IC.
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Layout Guidelines (continued)
10.1.4 PCB Summary
1. Minimize the parasitic inductance by keeping the power path components close together and keeping the
area of the high-current loops small.
2. The most important consideration when completing the layout is the close coupling of the GND connections
of the CIN capacitor and the catch diode D1. These ground connections must be immediately adjacent, with
multiple vias in parallel at the pad of the input capacitor connected to GND. Place CIN and D1 as close to the
IC as possible.
3. Next in importance is the location of the GND connection of the COUT capacitor, which must be near the GND
connections of CIN and D1.
4. There must be a continuous ground plane on the copper layer directly beneath the converter. This reduces
parasitic inductance and EMI.
5. The FB pin is a high impedance node and care must be taken to make the FB trace short to avoid noise
pickup and inaccurate regulation. The feedback resistors must be placed as close as possible to the IC, with
the GND of R2 placed as close as possible to the GND of the IC. The VOUT trace to R1 must be routed away
from the inductor and any other traces that are switching.
6. High AC currents flow through the VIN, SW and VOUT traces, so they must be as short and wide as possible.
However, making the traces wide increases radiated noise, so the layout designer must make this trade-off.
Radiated noise can be decreased by choosing a shielded inductor.
The remaining components must also be placed as close as possible to the IC. See AN-1229 SIMPLE
SWITCHER® PCB Layout Guidelines (SNVA054) for further considerations and the LM27342 demo board as an
example of a four-layer layout.
10.2 Layout Example
Figure 59. Top Layer and Overlay
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LM27341, LM27342, LM27341-Q1, LM27342-Q1
www.ti.com
SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
• AN-1197 Selecting Inductors for Buck Converters (SNVA038)
• AN-1187 Leadless Leadframe Package (SNOA401)
• AN-1520 A Guide to Board Layout for Best Thermal Resistance for Exposed Packages (SNVA183)
• AN-1229 SIMPLE SWITCHER® PCB Layout Guidelines (SNVA054)
11.3 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 16. Related Links
PARTS
PRODUCT FOLDER
SAMPLE & BUY
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
LM27341
Click here
Click here
Click here
Click here
Click here
LM27342
Click here
Click here
Click here
Click here
Click here
LM27341-Q1
Click here
Click here
Click here
Click here
Click here
LM27342-Q1
Click here
Click here
Click here
Click here
Click here
11.4 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.5 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.6 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
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SNVS497F – NOVEMBER 2008 – REVISED SEPTEMBER 2016
www.ti.com
11.7 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.8 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
44
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM27341MY/NOPB
ACTIVE
HVSSOP
DGQ
10
1000
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SSCB
LM27341QMY/NOPB
ACTIVE
HVSSOP
DGQ
10
1000
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SSJB
LM27341QMYX/NOPB
ACTIVE
HVSSOP
DGQ
10
3500
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SSJB
LM27341SD/NOPB
ACTIVE
WSON
DSC
10
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L231B
LM27342MY/NOPB
ACTIVE
HVSSOP
DGQ
10
1000
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SSCA
LM27342MYX/NOPB
ACTIVE
HVSSOP
DGQ
10
3500
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SSCA
LM27342QMY/NOPB
ACTIVE
HVSSOP
DGQ
10
1000
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SSJA
LM27342QMYX/NOPB
ACTIVE
HVSSOP
DGQ
10
3500
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SSJA
LM27342SD/NOPB
ACTIVE
WSON
DSC
10
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L231A
LM27342SDX/NOPB
ACTIVE
WSON
DSC
10
4500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L231A
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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