LM2742
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SNVS266C – MARCH 2004 – REVISED MARCH 2013
LM2742 N-Channel FET Synchronous Buck Regulator Controller for Low Output Voltages
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FEATURES
DESCRIPTION
•
•
•
The LM2742 is a high-speed, synchronous, switching
regulator controller. It is intended to control currents
of 0.7A to 20A with up to 95% conversion efficiencies.
Power up and down sequencing is achieved with the
power-good flag, adjustable soft-start and output
enable features. The LM2742 operates from a lowcurrent 5V bias and can convert from a 1V to 16V
power rail. The part utilizes a fixed-frequency,
voltage-mode, PWM control architecture and the
switching frequency is adjustable from 50kHz to
2MHz by setting the value of an external resistor.
Current limit is achieved by monitoring the voltage
drop across the on-resistance of the low-side
MOSFET, which enables on-times on the order of
40ns, one of the best in the industry. The wide range
of operating frequencies gives the power supply
designer the flexibility to fine-tune component size,
cost, noise and efficiency. The adaptive, nonoverlapping MOSFET gate-drivers and high-side
bootstrap structure helps to further maximize
efficiency. The high-side power FET drain voltage can
be from 1V to 16V and the output voltage is
adjustable down to 0.6V.
1
2
•
•
•
•
•
•
Input Power from 1V to 16V
Output Voltage Adjustable down to 0.6V
Power Good Flag, Adjustable Soft-start and
Output Enable for Easy Power Sequencing
Reference Accuracy: 1.5% (0°C–125°C)
Current Limit Without Sense Resistor
Soft Start
Switching Frequency from 50 kHz to 2 MHz
40ns Typical Minimum On-time
TSSOP-14 Package
APPLICATIONS
•
•
•
•
•
•
POL Power Supply Modules
Cable Modems
Set-Top Boxes/ Home Gateways
DDR Core Power
High-Efficiency Distributed Power
Local Regulation of Core Power
TYPICAL APPLICATION
+5V
0.1P
RIN
10:
CIN
2.2 PF
RFADJ
Q1
VCC
HG
SD
BOOT
LM2742
CSS
12n
LG
SS
PGND
SGND
PGND
EAO
2.2k
CIN1,2
10 PF
6.3V
Si4884DY
1.5 PH
6.1A, 9.6 m:
RCS
ISEN
PWGD
FREQ
63.4k
VIN = 3.3V
CBOOT
D1
Q2
VO = 1.2V@5A
L1
Si4884DY
Rfb2
10k
+
CO1,2
2200 PF
6.3V, 2.8A
FB
Rfb1
10k
CC1
CC2
180p
2.2p
RC1
392k
Figure 1. Typical Application Circuit
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2004–2013, Texas Instruments Incorporated
LM2742
SNVS266C – MARCH 2004 – REVISED MARCH 2013
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
CONNECTION DIAGRAM
1
2
4
5
6
7
HG
LG
PGND
SGND
VCC
PWGD
ISEN
PGND
LM2742
3
BOOT
SD
FREQ
FB
SS
EAO
14
13
12
11
10
9
8
Figure 2. 14-Lead Plastic TSSOP
θJA = 155°C/W
PIN DESCRIPTIONS
BOOT (Pin 1) - Supply rail for the N-channel MOSFET gate drive. The voltage should be at least one gate
threshold above the regulator input voltage to properly turn on the high-side N-FET.
LG (Pin 2) - Gate drive for the low-side N-channel MOSFET. This signal is interlocked with HG to avoid shootthrough problems
PGND (Pins 3, 13) - Ground for FET drive circuitry. It should be connected to system ground.
SGND (Pin 4) - Ground for signal level circuitry. It should be connected to system ground.
VCC (Pin 5) - Supply rail for the controller.
PWGD (Pin 6) - Power Good. This is an open drain output. The pin is pulled low when the chip is in UVP, OVP,
or UVLO mode. During normal operation, this pin is connected to VCC or other voltage source through a pull-up
resistor.
ISEN (Pin 7) - Current limit threshold setting. This sources a fixed 50µA current. A resistor of appropriate value
should be connected between this pin and the drain of the low-side FET.
EAO (Pin 8) - Output of the error amplifier. The voltage level on this pin is compared with an internally generated
ramp signal to determine the duty cycle. This pin is necessary for compensating the control loop.
SS (Pin 9) - Soft start pin. A capacitor connected between this pin and ground sets the speed at which the output
voltage ramps up. Larger capacitor value results in slower output voltage ramp but also lower inrush current.
FB (Pin 10) - This is the inverting input of the error amplifier, which is used for sensing the output voltage and
compensating the control loop.
FREQ (Pin 11) - The switching frequency is set by connecting a resistor between this pin and ground.
SD (Pin 12) - IC Logic Shutdown. When this pin is pulled low the chip turns off both the high side and low side
switches. While this pin is low, the IC will not start up. An internal 20µA pull-up connects this pin to VCC. For a
device which turns on the low side switch during shutdown, see the pin compatible LM2737.
HG (Pin 14) - Gate drive for the high-side N-channel MOSFET. This signal is interlocked with LG to avoid shootthrough problems.
2
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ABSOLUTE MAXIMUM RATINGS
(1)
If Military/Aerospace specified devices are required, contact the Texas Instruments Semiconductor Sales Office/
Distributors for availability and specifications.
VCC
7V
BOOTV
LG and HG to GND
21V
(2)
-2V to 21V
Junction Temperature
150°C
Storage Temperature
−65°C to 150°C
Soldering Information Lead
Temperature (soldering, 10sec)
260°C
Infrared or Convection (20sec)
235°C
ESD Rating
(1)
(2)
2 kV
Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating ratings indicate conditions for
which the device operates correctly. Operating Ratings do not imply ensured performance limits.
The LG and HG pin can have -2V to -0.5V applied for a maximum duty cycle of 10% with a maximum period of 1 second. There is no
duty cycle or maximum period limitation for a LG and HG pin voltage range of -0.5V to 21V.
RECOMMENDED OPERATING CONDITIONS
Supply Voltage (VCC)
4.5V to 5.5V
−40°C to +125°C
Junction Temperature Range
Thermal Resistance (θJA)
155°C/W
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ELECTRICAL CHARACTERISTICS
VCC = 5V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA=TJ=+25°C. Limits appearing in
boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are specified by design,
test, or statistical analysis.
Symbol
VFB_ADJ
VON
Parameter
FB Pin Voltage
UVLO Thresholds
Min
Typ
Max
VCC = 4.5V, 0°C to +125°C
Conditions
0.591
0.6
0.609
VCC = 5V, 0°C to +125°C
0.591
0.6
0.609
VCC = 5.5V, 0°C to +125°C
0.591
0.6
0.609
VCC = 4.5V, −40°C to +125°C
0.589
0.6
0.609
VCC = 5V, −40°C to +125°C
0.589
0.6
0.609
VCC = 5.5V, −40°C to +125°C
0.589
0.6
0.609
Rising
Falling
4.2
3.6
Units
V
V
SD = 5V, FB = 0.55V
Fsw = 600kHz
1
1.5
2
SD = 5V, FB = 0.65V
Fsw = 600kHz
0.8
1.7
2.2
Shutdown VCC Current
SD = 0V
0.15
0.4
0.7
tPWGD1
PWGD Pin Response Time
FB Voltage Going Up
6
tPWGD2
PWGD Pin Response Time
FB Voltage Going Down
6
µs
20
µA
IQ-V5
ISD
ISS-ON
ISS-OC
ISEN-TH
Operating VCC Current
mA
SD Pin Internal Pull-up Current
SS Pin Source Current
SS Voltage = 2.5V
0°C to +125°C
-40°C to +125°C
SS Pin Sink Current During Over
Current
SS Voltage = 2.5V
ISEN Pin Source Current Trip Point
0°C to +125°C
-40°C to +125°C
8
5
11
11
µs
15
15
95
35
28
50
50
mA
µA
µA
65
65
µA
ERROR AMPLIFIER
GBW
G
Error Amplifier Unity Gain
Bandwidth
5
MHz
Error Amplifier DC Gain
60
dB
SR
Error Amplifier Slew Rate
IFB
FB Pin Bias Current
FB = 0.55V
FB = 0.65V
6
IEAO
EAO Pin Current Sourcing and
Sinking
VEAO = 2.5, FB = 0.55V
VEAO = 2.5, FB = 0.65V
2.8
0.8
mA
VEA
Error Amplifier Maximum Swing
Minimum
Maximum
1.2
3.2
V
BOOT Pin Quiescent Current
BOOT = 12V, EN = 0
0°C to +125°C
-40°C to +125°C
95
95
0
0
15
30
V/µA
100
155
nA
GATE DRIVE
IQ-BOOT
4
160
215
µA
RDS1
Top FET Driver Pull-Up ON
resistance
BOOT-SW = 5V at 350mA
3
Ω
RDS2
Top FET Driver Pull-Down ON
resistance
BOOT-SW = 5V at 350mA
2
Ω
RDS3
Bottom FET Driver Pull-Up ON
resistance
BOOT-SW = 5V at 350mA
3
Ω
RDS4
Bottom FET Driver Pull-Down ON
resistance
BOOT-SW = 5V at 350mA
2
Ω
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SNVS266C – MARCH 2004 – REVISED MARCH 2013
ELECTRICAL CHARACTERISTICS (continued)
VCC = 5V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA=TJ=+25°C. Limits appearing in
boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are specified by design,
test, or statistical analysis.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
OSCILLATOR
RFADJ = 590kΩ
50
RFADJ = 88.7kΩ
fOSC
D
ton-min
PWM Frequency
Max Duty Cycle
300
RFADJ = 42.2kΩ, 0°C to +125°C
500
600
700
RFADJ = 42.2kΩ, -40°C to +125°C
490
600
700
kHz
RFADJ = 17.4kΩ
1400
RFADJ = 11.3kΩ
2000
fPWM = 300kHz
fPWM = 600kHz
90
88
%
40
ns
Minimum on-time
LOGIC INPUTS AND OUTPUTS
VSD-IH
SD Pin Logic High Trip Point
VSD-IL
SD Pin Logic Low Trip Point
0°C to +125°C
-40°C to +125°C
1.3
1.25
1.6
1.6
PWGD Pin Trip Points
FB Voltage Going Down
0°C to +125°C
-40°C to +125°C
0.413
0.410
0.430
0.430
0.446
0.446
V
FB Voltage Going Up
0°C to +125°C
-40°C to +125°C
0.691
0.688
0.710
0.710
0.734
0.734
V
VPWGD-TH-LO
VPWGD-TH-HI
VPWGD-HYS
PWGD Pin Trip Points
PWGD Hysteresis
2.6
FB Voltage Going Down FB Voltage Going
Up
3.5
V
35
110
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V
mV
5
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SNVS266C – MARCH 2004 – REVISED MARCH 2013
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency (VO = 1.5V)
FSW = 300kHz, TA = 25°C
Efficiency (VO = 3.3V)
FSW = 300kHz, TA = 25°C
100
100
Vin = 5V
Vin = 3.3V
90
90
EFFICIENCY (%)
EFFICIENCY (%)
80
Vin = 5V
70
60
50
Vin = 12V
40
80
70
50
40
30
30
0.1
20
0.2
1
3
5
9
7
0.5
2
4
6
8
10
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Figure 3.
Figure 4.
VCC Operating Current vs Temperature
FSW = 600kHz, No-Load
Bootpin Current vs Temperature for BOOTV = 12V
FSW = 600kHz, Si4826DY FET, No-Load
30.3
1.64
1.62
30.1
Without
Bootstrap
(Vboot = 12V)
1.6
BOOT PIN CURRENT (mA)
OPEARTING CURRENT(mA)
Vin = 12V
60
1.58
1.56
With
Bootstrap
(Vboot = 5V)
1.54
1.52
1.5
29.9
29.7
29.5
29.3
29.1
1.48
28.9
1.46
0
20
35
55
75
95
115
AMBIENT TEMPERATURE ( C)
Figure 6.
Bootpin Current vs Temperature with 5V Bootstrap
FSW = 600kHz, Si4826DY FET, No-Load
PWM Frequency vs Temperature
for RFADJ = 43.2kΩ
8.6
630
8.4
628
8.2
626
PWM FREQUENCY (kHz)
BOOT PIN CURRENT (mA)
Figure 5.
8
7.8
7.6
7.4
7.2
624
622
620
618
616
614
7
612
0 10 20 25 35 45 55 65 75 85 95 105115125
0 10 20 25 35 45 55 65 75 85 95 105115 125
o
AMBIENT TEMPERATURE ( C)
AMBIENT TEMPERATURE (oC)
Figure 7.
6
0 10 20 25 35 45 55 65 75 85 95105115 125
AMBIENT TEMPERATURE (oC)
o
Figure 8.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
RFADJ vs PWM Frequency
(in 100 to 800kHz range), TA = 25°C
RFADJ vs PWM Frequency
(in 900 to 2000kHz range), TA = 25°C
500
30
400
RF-ADJ (k:)
RF-ADJ (k:)
25
300
200
20
15
100
0
100 150 200 250 300 350 400450 500 600 700 800
PWM FREQUENCY (kHz)
10
9001000110012001300140015001600170018001900
PWM FREQUENCY (kHz)
Figure 9.
Figure 10.
VCC Operating Current Plus Boot Current vs
PWM Frequency (Si4826DY FET, TA = 25°C)
Switch Waveforms (HG Falling)
VIN = 5V, VO = 1.8V
IO = 3A, CSS = 10nF, FSW = 600kHz
40
VCC PLUS BOOT CURRENT
35
30
25
20
15
10
5
0
100 300 500 700 9001100 13001500 17001900
PWM FREQUENCY (kHz)
Figure 11.
Figure 12.
Switch Waveforms (HG Rising)
VIN = 5V, VO = 1.8V
IO = 3A, FSW = 600kHz
Start-Up (No-Load)
VIN = 10V, VO = 1.2V
CSS = 10nF, FSW = 300kHz
Figure 13.
Figure 14.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Start-Up (Full-Load)
VIN = 10V, VO = 1.2V
IO = 10A, CSS = 10nF, FSW = 300kHz
Start Up (No-Load, 10x CSS)
VIN = 10V, VO = 1.2V
CSS = 100nF, FSW = 300kHz
Figure 15.
Figure 16.
Start Up (Full Load, 10x CSS)
VIN = 10V, VO = 1.2V
IO = 10A, CSS = 100nF, FSW = 300kHz
Start Up (Into 1.2V Pre-Bias)
VIN = 12V, VO = 2.5V
No Load, No Soft Start Capacitor, FSW = 300kHz
VO = 2.5
VO
Pre-bias = 1.2V
2.0V
VCSS
2A/div
IIN
VSD
5.0V
20 Ps/DIV
Figure 17.
Figure 18.
Start Up (Into 1.2V Pre-Bias)
VIN = 12V, VO = 2.5V
No Load, CSS = 10nF, FSW = 300kHz
Shutdown
VIN = 12V, VO = 1.2V
IO = 10A, CSS = 10nF, FSW = 300kHz
VO
VO
Pre-bias = 1.2V
500 mV
VCSS
2.0V
VCSS
2.0V
IIN
500 mA/div
1A/div
IIN
VSD
5.0V
5.0V
VSD
8
400 Ps/DIV
40 Ps/DIV
Figure 19.
Figure 20.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Shutdown (No Load)
VIN = 12V, VO = 1.2V
IO = 10A, CSS = 10nF
FSW = 300kHz
Load Transient Response (IO = 0 to 4A)
VIN = 12V, VO = 1.2V
FSW = 300kHz
VO
500 mV
VCSS
2.0V
IIN
1A/div
VSD
5.0V
40 Ps/DIV
Figure 21.
Figure 22.
Load Transient Response (IO = 4 to 0A)
VIN = 12V, VO = 1.2V
FSW = 300kHz
Line Transient Response (VIN =5V to 12V)
VO = 1.2V, IO = 5A
FSW = 300kHz
Figure 23.
Figure 24.
Line Transient Response (VIN =12V to 5V)
VO = 1.2V, IO = 5A
FSW = 300kHz
Line Transient Response
VO = 1.2V, IO = 5A
FSW = 300kHz
Figure 25.
Figure 26.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
VIN
Peak Current During Current Limit
= 12V, VO = 3.3V, ILIM = 4A, FSW = 300kHz, L = 15 µH
Peak Current During Current Limit
VIN = 12V, VO = 3.3V, ILIM = 4A, FSW = 300kHz, L = 15 µH
2V/DIV
VO
VO
2V/DIV
2A/DIV
IL
2A/DIV
2V/DIV
IL
2V/DIV
VCSS
VCSS
10V/DIV
10V/DIV
VSW
VSW
10
4 Ps/DIV
200 Ps/DIV
Figure 27.
Figure 28.
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BLOCK DIAGRAM
FREQ
VCC
CLOCK &
RAMP
UVLO
SD
PGND
PGND
SGND
20 PA
off
LOGIC
BOOT
10 Ps
DELAY
HG
PWGD
off
SYNCHRONOUS
DRIVER LOGIC
3.05V
10 PA
HIGH
LOW
LG
0.708V
tol.=+/-2%
0.42V
tol.=+/-2%
hyst.=12%
S
OUTPUT CLAMP
HI: 3.25V
LO: 1.25V
SS
3.25V
1.25V
oc
95 PA
R
R>S
SS
CMP
PWM
50 PA
BG =
0.6V
ISEN
ILIM
EA
oc
FB
EAO
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APPLICATION INFORMATION
THEORY OF OPERATION
The LM2742 is a voltage-mode, high-speed synchronous buck regulator with a PWM control scheme. It is
designed for use in set-top boxes, thin clients, DSL/Cable modems, and other applications that require high
efficiency buck converters. It has power good (PWRGD), and output shutdown (SD). Current limit is achieved by
sensing the voltage VDS across the low side FET. During current limit the high side gate is turned off and the low
side gate turned on. The soft start capacitor is discharged by a 95µA source (reducing the maximum duty cycle)
until the current is under control.
START UP
When VCC exceeds 4.2V and the shutdown pin SD sees a logic high the soft start capacitor begins charging
through an internal fixed 10µA source. During this time the output of the error amplifier is allowed to rise with the
voltage of the soft start capacitor. This capacitor, CSS, determines soft start time, and can be determined
approximately by:
Css
t ss
2.5 u 105
(1)
An application for a microprocessor might need a delay of 3ms, in which case CSS would be 12nF. For a different
device, a 100ms delay might be more appropriate, in which case CSS would be 400nF. (390 10%) During soft
start the PWRGD flag is forced low and is released when the voltage reaches a set value. At this point this chip
enters normal operation mode and the Power Good flag is released.
Since the output is floating when the LM2742 is turned off, it is possible that the output capacitor may be
precharged to some positive value. During start-up, the LM2742 operates fully synchronous and will discharge
the output capacitor to some extent depending on the output voltage, soft start capacitance, and the size of the
output capacitor.
NORMAL OPERATION
While in normal operation mode, the LM2742 regulates the output voltage by controlling the duty cycle of the
high side and low side FETs. The equation governing output voltage is:
VO = 0.6 x (RFB1 + RFB2) / RFB1
(2)
The PWM frequency is adjustable between 50kHz and 2MHz and is set by an external resistor, RFADJ, between
the FREQ pin and ground. The resistance needed for a desired frequency is approximately:
1.0526
RFADJ
§ 20500 ·
¨
¸
¨ freq ªkHz º ¸
¬
¼¹
©
k:
(3)
MOSFET GATE DRIVERS
The LM2742 has two gate drivers designed for driving N-channel MOSFETs in a synchronous mode. Power for
the drivers is supplied through the BOOT pin. For the high side gate (HG) to fully turn on the top FET, the BOOT
voltage must be at least one VGS(th) greater than Vin. (BOOT ≥ 2*Vin) This voltage can be supplied by a
separate, higher voltage source, or supplied from a local charge pump structure. In a system such as a desktop
computer, both 5V and 12V are usually available. Hence if Vin was 5V, the 12V supply could be used for BOOT.
12V is more than 2*Vin, so the HG would operate correctly. For a BOOT of 12V, the initial gate charging current
is 2A, and the initial gate discharging current is typically 6A.
12
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D1
5V
+
VCC
BOOT
CBOOT
HG
LM2742
+
LG
Figure 29. BOOT Supplied by Charge Pump
In a system without a separate, higher voltage, a charge pump (bootstrap) can be built using a diode and small
capacitor, Figure 29. The capacitor serves to maintain enough voltage between the top FET gate and source to
control the device even when the top FET is on and its source has risen up to the input voltage level.
The LM2742 gate drives use a BiCMOS design. Unlike some other bipolar control ICs, the gate drivers have railto-rail swing, ensuring no spurious turn-on due to capacitive coupling.
POWER GOOD SIGNAL
The power good signal is the or-gated flag representing over-voltage and under-voltage protection. If the output
voltage is 18% over it's nominal value, VFB = 0.7V, or falls 30% below that value, VFB = 0.41V, the power good
flag goes low. It will return to a logic high whenever the feedback pin voltage is between 70% and 118% of 0.6V.
The power good pin is an open drain output that can be pulled up to logic voltages of 5V or less with a 10kΩ
resistor.
UVLO
The 4.2V turn-on threshold on VCC has a built in hysteresis of 0.6V. Therefore, if VCC drops below 3.6V, the chip
enters UVLO mode. UVLO consists of turning off the top FET, turning off the bottom FET, and remaining in that
condition until VCC rises above 4.2V. As with shutdown, the soft start capacitor is discharged through a FET,
ensuring that the next start-up will be smooth.
CURRENT LIMIT
Current limit is realized by sensing the voltage across the low side FET while it is on. The RDSON of the FET is a
known value, hence the current through the FET can be determined as:
VDS = I * RDSON
(4)
The current through the low side FET while it is on is also the falling portion of the triangle wave inductor current.
The current limit threshold is determined by an external resistor, RCS, connected between the switch node and
the ISEN pin. A constant current of 50 µA is forced through RCS, causing a fixed voltage drop. This fixed voltage is
compared against VDS and if the latter is higher, the current limit of the chip has been reached. RCS can be found
by using the following equation:
RCS = RDSON(LOW) * ILIM/50µA
(5)
For example, a conservative 15A current limit in a 10A design with a minimum RDSON of 10mΩ would require a
3.3kΩ resistor. Because current sensing is done across the low side FET, no minimum high side on-time is
necessary. In the current limit mode the LM2727/37 will turn the high side off and the keep low side on for as
long as necessary. The LM2727/37 enters current limit mode if the inductor current exceeds the current limit
threshold at the point where the high side FET turns off and the low side FET turns on. (The point of peak
inductor current. See Figure 30.) Note that in normal operation mode the high side FET always turns on at the
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beginning of a clock cycle. In current limit mode, by contrast, the high side FET on pulse is skipped. This causes
inductor current to fall. Unlike a normal operation switching cycle, however, in a current limit mode switching
cycle the high side FET will turn on as soon as inductor current has fallen to the current limit threshold. The
LM2727/37 will continue to skip high side FET pulses until the inductor current peak is below the current limit
threshold, at which point the system resumes normal operation.
Normal Operation
Current Limit
ILIM
IL
D
Figure 30. Current Limit Threshold
Unlike a high side FET current sensing scheme, which limits the peaks of inductor current, low side current
sensing is only allowed to limit the current during the converter off-time, when inductor current is falling.
Therefore in a typical current limit plot the valleys are normally well defined, but the peaks are variable, according
to the duty cycle. The PWM error amplifier and comparator control the off pulse of the high side FET, even
during current limit mode, meaning that peak inductor current can exceed the current limit threshold. Assuming
that the output inductor does not saturate, the maximum peak inductor current during current limit mode can be
calculated with the following equation:
IPK-CL
ILIM
TOSC
200 ns
VIN
VO
L
(6)
Where TOSC is the inverse of switching frequency fOSC. The 200ns term represents the minimum off-time of the
duty cycle, which ensures enough time for correct operation of the current sensing circuitry. See the plots entitled
Peak Current During Current Limit in the Typical Performance Characteristics section.
In order to minimize the time period in which peak inductor current exceeds the current limit threshold, the IC
also discharges the soft start capacitor through a fixed 95 µA source. The output of the LM2727/37 internal error
amplifier is limited by the voltage on the soft start capacitor. Hence, discharging the soft start capacitor reduces
the maximum duty cycle D of the controller. During severe current limit this reduction in duty cycle will reduce the
output voltage if the current limit conditions last for an extended time. Output inductor current will be reduced in
turn to a flat level equal to the current limit threshold. The third benefit of the soft start capacitor discharge is a
smooth, controlled ramp of output voltage when the current limit condition is cleared. During the first few
nanoseconds after the low side gate turns on, the low side FET body diode conducts. This causes an additional
0.7V drop in VDS. The range of VDS is normally much lower. For example, if RDSON were 10mΩ and the current
through the FET was 10A, VDS would be 0.1V. The current limit would see 0.7V as a 70A current and enter
current limit immediately. Hence current limit is masked during the time it takes for the high side switch to turn off
and the low side switch to turn on.
SHUT DOWN
If the shutdown pin SD is pulled low, the LM2742 discharges the soft start capacitor through a MOSFET switch.
The high side and low side switches are turned off. The LM2742 remains in this state until SD is released.
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DESIGN CONSIDERATIONS
The following is a design procedure for all the components needed to create the circuit shown in Figure 32 in the
Example Circuits section, a 5V in to 1.2V out converter, capable of delivering 10A with an efficiency of 85%. The
switching frequency is 300kHz. The same procedures can be followed to create many other designs with varying
input voltages, output voltages, and output currents.
Input Capacitor
The input capacitors in a Buck switching converter are subjected to high stress due to the input current
waveform, which is a square wave. Hence input caps are selected for their ripple current capability and their
ability to withstand the heat generated as that ripple current runs through their ESR. Input rms ripple current is
approximately:
png
(7)
The power dissipated by each input capacitor is:
PD
2
Irms
ESR
rip
n
2
(8)
Here, n is the number of capacitors, and indicates that power loss in each cap decreases rapidly as the number
of input caps increase. The worst-case ripple for a Buck converter occurs during full load, when the duty cycle D
= 50%.
In the 5V to 1.2V case, D = 1.2/5 = 0.24. With a 10A maximum load the ripple current is 4.3A. The Sanyo
10MV5600AX aluminum electrolytic capacitor has a ripple current rating of 2.35A, up to 105°C. Two such
capacitors make a conservative design that allows for unequal current sharing between individual caps. Each
capacitor has a maximum ESR of 18mΩ at 100 kHz. Power loss in each device is then 0.05W, and total loss is
0.1W. Other possibilities for input and output capacitors include MLCC, tantalum, OSCON, SP, and POSCAPS.
Input Inductor
The input inductor serves two basic purposes. First, in high power applications, the input inductor helps insulate
the input power supply from switching noise. This is especially important if other switching converters draw
current from the same supply. Noise at high frequency, such as that developed by the LM2742 at 1MHz
operation, could pass through the input stage of a slower converter, contaminating and possibly interfering with
its operation.
An input inductor also helps shield the LM2742 from high frequency noise generated by other switching
converters. The second purpose of the input inductor is to limit the input current slew rate. During a change from
no-load to full-load, the input inductor sees the highest voltage change across it, equal to the full load current
times the input capacitor ESR. This value divided by the maximum allowable input current slew rate gives the
minimum input inductance:
Lin
'V
§ di ·
¨ dt ¸
© ¹max
(9)
In the case of a desktop computer system, the input current slew rate is the system power supply or "silver box"
output current slew rate, which is typically about 0.1A/µs. Total input capacitor ESR is 9mΩ, hence ΔV is
10*0.009 = 90 mV, and the minimum inductance required is 0.9µH. The input inductor should be rated to handle
the DC input current, which is approximated by:
I IN
DC
IO D
K
(10)
In this case IIN-DC is about 2.8A. One possible choice is the TDK SLF12575T-1R2N8R2, a 1.2µH device that can
handle 8.2Arms, and has a DCR of 7mΩ.
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Output Inductor
The output inductor forms the first half of the power stage in a Buck converter. It is responsible for smoothing the
square wave created by the switching action and for controlling the output current ripple. (ΔIo) The inductance is
chosen by selecting between tradeoffs in output ripple, efficiency, and response time. The smaller the output
inductor, the more quickly the converter can respond to transients in the load current. If the inductor value is
increased, the ripple through the output capacitor is reduced and thus the output ripple is reduced. As shown in
the efficiency calculations, a smaller inductor requires a higher switching frequency to maintain the same level of
output current ripple. An increase in frequency can mean increasing loss in the FETs due to the charging and
discharging of the gates. Generally the switching frequency is chosen so that conduction loss outweighs
switching loss. The equation for output inductor selection is:
(11)
A good range for ΔIo is 25 to 50% of the output current. In the past, 30% was considered a maximum value for
output currents higher than about 2Amps, but as output capacitor technology improves the ripple current can be
allowed to increase. Plugging in the values for output current ripple, input voltage, output voltage, switching
frequency, and assuming a 40% peak-to-peak output current ripple yields an inductance of 1.5µH. The output
inductor must be rated to handle the peak current (also equal to the peak switch current), which is (Io + 0.5*ΔIo).
This is 12A for a 10A design. The Coilcraft D05022-152HC is 1.5µH, is rated to 15Arms, and has a DCR of 4mΩ.
Output Capacitor
The output capacitor forms the second half of the power stage of a Buck switching converter. It is used to control
the output voltage ripple (ΔVo) and to supply load current during fast load transients.
In this example the output current is 10A and the expected type of capacitor is an aluminum electrolytic, as with
the input capacitors. (Other possibilities include ceramic, tantalum, and solid electrolyte capacitors, however the
ceramic type often do not have the large capacitance needed to supply current for load transients, and tantalums
tend to be more expensive than aluminum electrolytic.) Aluminum capacitors tend to have very high capacitance
and fairly low ESR, meaning that the ESR zero, which affects system stability, will be much lower than the
switching frequency. The large capacitance means that at switching frequency, the ESR is dominant, hence the
type and number of output capacitors is selected on the basis of ESR. One simple formula to find the maximum
ESR based on the desired output voltage ripple, ΔVo and the designed output current ripple, ΔIo, is:
ESRMAX
'Vo
'Io
(12)
In this example, in order to maintain a 2% peak-to-peak output voltage ripple and a 40% peak-to-peak inductor
current ripple, the required maximum ESR is 6mΩ. Three Sanyo 10MV5600AX capacitors in parallel will give an
equivalent ESR of 6mΩ. The total bulk capacitance of 16.8mF is enough to supply even severe load transients.
Using the same capacitors for both input and output also keeps the bill of materials simple.
MOSFETS
MOSFETS are a critical part of any switching controller and have a direct impact on the system efficiency. In this
case the target efficiency is 85% and this is the variable that will determine which devices are acceptable. Loss
from the capacitors, inductors, and the LM2742 is detailed in the Efficiency section, and come to about 0.54W.
To meet the target efficiency, this leaves 1.45W for the FET conduction loss, gate charging loss, and switching
loss. Switching loss is particularly difficult to estimate because it depends on many factors. When the load
current is more than about 1 or 2 amps, conduction losses outweigh the switching and gate charging losses. This
allows FET selection based on the RDSON of the FET. Adding the FET switching and gate-charging losses to the
equation leaves 1.2W for conduction losses. The equation for conduction loss is:
PCnd = D(I2o * RDSON *k) + (1-D)(I2o * RDSON *k)
(13)
The factor k is a constant which is added to account for the increasing RDSON of a FET due to heating. Here, k =
1.3. The Si4442DY has a typical RDSON of 4.1mΩ. When plugged into the equation for PCND the result is a loss of
0.533W. If this design were for a 5V to 2.5V circuit, an equal number of FETs on the high and low sides would be
the best solution. With the duty cycle D = 0.24, it becomes apparent that the low side FET carries the load
current 76% of the time. Adding a second FET in parallel to the bottom FET could improve the efficiency by
lowering the effective RDSON. The lower the duty cycle, the more effective a second or even third FET can be. For
a minimal increase in gate charging loss (0.054W) the decrease in conduction loss is 0.15W. What was an 85%
design improves to 86% for the added cost of one SO-8 MOSFET.
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Control Loop Components
The circuit is this design example and the others shown in the Example Circuits section have been compensated
to improve their DC gain and bandwidth. The result of this compensation is better line and load transient
responses. For the LM2742, the top feedback divider resistor, Rfb2, is also a part of the compensation. For the
10A, 5V to 1.2V design, the values are:
Cc1 = 4.7pF 10%, Cc2 = 1nF 10%, Rc = 229kΩ 1%. These values give a phase margin of 63° and a bandwidth
of 29.3kHz.
Support Capacitors and Resistors
The Cinx capacitors are high frequency bypass devices, designed to filter harmonics of the switching frequency
and input noise. Two 1µF ceramic capacitors with a sufficient voltage rating (10V for the Circuit of Figure 32) will
work well in almost any case.
RIN and CIN are standard filter components designed to ensure smooth DC voltage for the chip supply.
Depending on noise, RIN should be 10 to 100Ω, and CIN should be between 0.1 and 2.2 µF. CBOOT is the
bootstrap capacitor, and should be 0.1µF. (In the case of a separate, higher supply to the BOOT pin, this 0.1µF
cap can be used to bypass the supply.) Using a Schottky device for the bootstrap diode allows the minimum drop
for both high and low side drivers. The On Semiconductor BAT54 or MBR0520 work well.
Rp is a standard pull-up resistor for the open-drain power good signal, and should be 10kΩ. If this feature is not
necessary, it can be omitted.
RCS is the resistor used to set the current limit. Since the design calls for a peak current magnitude (Io + 0.5 *
ΔIo) of 12A, a safe setting would be 15A. (This is well below the saturation current of the output inductor, which is
25A.) Following the equation from the Current Limit section, use a 3.3kΩ resistor.
RFADJ is used to set the switching frequency of the chip. Following the equation in the Theory of Operation
section, the closest 1% tolerance resistor to obtain fSW = 300kHz is 88.7kΩ.
CSS depends on the users requirements. Based on the equation for CSS in the Theory of Operation section, for a
3ms delay, a 12nF capacitor will suffice.
EFFICIENCY CALCULATIONS
A reasonable estimation of the efficiency of a switching controller can be obtained by adding together the loss is
each current carrying element and using the equation:
K
Po
Po
Ptotal
(14)
loss
The following shows an efficiency calculation to complement the Circuit of Figure 32. Output power for this circuit
is 1.2V x 10A = 12W.
Chip Operating Loss
PIQ = IQ-VCC *VCC
(15)
2mA x 5V = 0.01W
FET Gate Charging Loss
PGC = n * VCC * QGS * fOSC
(16)
The value n is the total number of FETs used. The Si4442DY has a typical total gate charge, QGS, of 36nC and
an rds-on of 4.1mΩ. For a single FET on top and bottom: 2*5*36E-9*300,000 = 0.108W
FET Switching Loss
PSW = 0.5 * Vin * IO * (tr + tf)* fOSC
(17)
The Si4442DY has a typical rise time tr and fall time tf of 11 and 47ns, respectively. 0.5*5*10*58E-9*300,000 =
0.435W
FET Conduction Loss
PCn = 0.533W
(18)
Input Capacitor Loss
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PCin
I INrms
2
Irms
rip
www.ti.com
ESR
n
rip
IO
(19)
D 1 D
(20)
4.282*0.018/2 = 0.164W
Input Inductor Loss
PLin = I2in * DCRinput-L
I D
IIN O
Kest ' d
(21)
(22)
2
2.82 *0.007 = 0.055W
Output Inductor Loss
PLout = I2o * DCRoutput-L
(23)
2
10 *0.004 = 0.4W
System Efficiency
12
22 1.7
87.5%
(24)
Example Circuits
+5V
RIN
10
CIN
2.2P
RFADJ
0.1P
CINX
1 PF,
25V
Q1
VCC
HG
SD
BOOT
2.7 PH
14.4A, 4.5 m:
RCS
ISEN
PWGD
LM2742
FREQ
88.7k
1.2 PH
8.2A, 6.9 m:
CBOOT
D1
SS
LG
Vo = 3.3V@10A
L1
1.8k
Q2
Rfb2
PGND
49.9k
CSS
12n
SGND
PGND
EAO
VIN = 12V
LIN
+ CIN1,2
2 x 10 PF
25V, 3.3A
Rc2
+ Co1-4
4 x 100 PF
10V, 55 m:
Cc3
FB
8.45k
Cc1
Cc2
270p
6.8p
470p
Rfb1
11k
Rc1
143.3k
Figure 31. 5V-16V to 3.3V, 10A, 300kHz
This circuit and the one featured on the front page have been designed to deliver high current and high efficiency
in a small package, both in area and in height The tallest component in this circuit is the inductor L1, which is
6mm tall. The compensation has been designed to tolerate input voltages from 5 to 16V.
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1.2 PH
8.2A, 6.9 m:
CBOOT
D1
RIN
10
VIN = 5V
LIN
CINx1, 2 + CIN1,2
2x1 PF
2 x 5600 PF
10V
10V, 2.35A
0.1P
Q1
CIN
VCC
2.2P
SD
HG
BOOT
PWGD
RFADJ
ISEN
LM2742
FREQ
88.7k
CSS
12n
1.5 PH
15A, 4 m:
RCS
LG
SS
PGND
SGND
PGND
EAO
Vo = 1.2V@10A
L1
1.5k
Q2
+ Co1-3
3 x 5600 PF
10V, 3.1A
18 m:
Rfb2
4.99k
FB
Rfb1
4.99k
Cc1
Cc2
Rc1
4.7p
229k
270p
Figure 32. 5V to 1.2V, 10A, 300kHz
This circuit design, detailed in the Design Considerations section, uses inexpensive aluminum capacitors and offthe-shelf inductors. It can deliver 10A at better than 85% efficiency. Large bulk capacitance on input and output
ensure stable operation.
+12V
VIN = 5V
Cc
0.1P
RIN
10
CIN
VCC
2.2P
SD
RFADJ
HG
BOOT
LM2742
FREQ
43.2k
Css
12n
Rcs
ISEN
PWGD
LG
SS
PGND
SGND
PGND
EAO
+
Q1/Q2
2.7k
CIN1
100 PF
10V, 1.9A
2.2 PH
6.1A, 12 m:
Vo = 1.8V@3A
L1
Rfb2
4.99k
+
Co1
1 x 220 PF
4V, 55 m:
FB
Rfb1
Cc1
Cc2
560p
10p
2.49k
RC1
51.1k
Figure 33. 5V to 1.8V, 3A, 600kHz
The example circuit of Figure 33 has been designed for minimum component count and overall solution size. A
switching frequency of 600kHz allows the use of small input/output capacitors and a small inductor. The
availability of separate 5V and 12V supplies (such as those available from desk-top computer supplies) and the
low current further reduce component count. Using the 12V supply to power the MOSFET drivers eliminates the
bootstrap diode, D1. At low currents, smaller FETs or dual FETs are often the most efficient solutions. Here, the
Si4826DY, an asymmetric dual FET in an SO-8 package, yields 92% efficiency at a load of 2A.
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+5V
1 PH
4.5A, 7.5 m:
CBOOT
D1
VIN = 3.3V
RIN
10
LIN
0.1P
CIN
VCC
2.2P
SD
BOOT
PWGD
RFADJ
Q1
HG
Rcs
1 PH
11A, 3.7 m:
3.3k
L1
ISEN
FREQ
LG
Css
12n
SS
PGND
SGND
PGND
EAO
+ CIN1
1 x 5600 PF
10V, 2.35A
Vo = 0.8V@5A
+ Co1,2
2 x 4700 PF
16V, 2.8A
Rfb2
Q2
49.9k
CINx
1 PF
10V
4.99k
FB
Rfb1
Cc1
14.9k
Cc2
Rc1
4.7p
147k
680p
Figure 34. 3.3V to 0.8V, 5A, 500kHz
The circuit of Figure 34 demonstrates the LM2742 delivering a low output voltage at high efficiency (87%). A
separate 5V supply is required to run the chip, however the input voltage can be as low as 2.2
+5V
D1
1 PH
6.4A, 7.3 m:
CBOOT
RIN
10
0.1P
HG
VCC
CIN
2.2P
SD
BOOT
PWGD
RFADJ
LG
SS
PGND
SGND
PGND
EAO
Vo = 1.8V@1A
Rfb2
10k
Rc2
66.5
Cc1
Cc2
1 x 15 PF
25V, 3.3A
L1
1.5k
FB
Css
+
Q1/Q2
3.3 PH
4.1A, 17.4 m:
ISEN
LM2742
FREQ
17.4k
Rcs
VIN = 5 to 15V
LIN
Cc3
680p
+
Co1
1 x 15 PF
25V, 3.1 m:
Rfb1
4.99k
Rc1
22p
39n
680p
10.7k
1 PF
6.4A, 7.3 m:
+5V
CBOOT
D1
RIN
10
CIN
2.2P
VIN = 5 to 15V
LIN
+ CIN1
1 x 15 PF
25V, 3.3A
0.1P
SD
BOOT
PWGD
RFADJ
LM2742
FREQ
17.4k
Q1/Q2
4.7 PF
3.4A, 26 m:
HG
VCC
ISEN
LG
SS
PGND
SGND
PGND
EAO
Rcs
Rfb2
10k
Rc2
FB
54.9
Cc1
Cc2
1n
27p
Vo = 3.3V@1A
L1
1.5k
+ Co1
1 x 15 PF
25V, 3.1 m:
Cc3
820p
Rfb1
2.21k
Rc1
12.1k
Figure 35. 1.8V and 3.3V, 1A, 1.4MHz, Simultaneous
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The circuits in Figure 35 are intended for ADSL applications, where the high switching frequency keeps noise out
of the data transmission range. In this design, the 1.8 and 3.3V outputs come up simultaneously by using the
same softstart capacitor. Because two current sources now charge the same capacitor, the capacitance must be
doubled to achieve the same softstart time. (Here, 40nF is used to achieve a 5ms softstart time.) A common
softstart capacitor means that, should one circuit enter current limit, the other circuit will also enter current limit.
The additional compensation components Rc2 and Cc3 are needed for the low ESR, all ceramic output
capacitors, and the wide (3x) range of Vin.
To 2nd LM27x7
VIN = 11 to 13V
+5V
LM78L05
1 PH, 6.4A
7.3 m:
CBOOT
D1
0.1P
CIN
VCC
2.2P
SD
LM2742
FREQ
32.5k
Css
12n
Rcs
680 PF
16V, 1.54A
Vo = 3.3V@3A
ISEN
PWGD
RFADJ
+ CIN1
CINx
10 PF
16V
Q1/Q2
4.2 PH, 5.5A
15 m:
HG
BOOT
2k
LG
SS
PGND
SGND
PGND
EAO
VIN = 11 to 13V
LIN
+ Co1,2
2 x 680 PF
16V,
1.54A
Rfb2
10k
RC2
CC3
Cox
10 PF
25V
FB
2.37k
CC1
CC2
Rfb1
4.7n
2.21k
RC1
8.2p
52.3k
1n
Figure 36. 12V Unregulated to 3.3V, 3A, 750kHz
This circuit shows the LM2742 paired with a cost effective solution to provide the 5V chip power supply, using no
extra components other than the LM78L05 regulator itself. The input voltage comes from a 'brick' power supply
which does not regulate the 12V line tightly. Additional, inexpensive 10uF ceramic capacitors (Cinx and Cox)
help isolate devices with sensitive databands, such as DSL and cable modems, from switching noise and
harmonics.
+5V (low current source)
CBOOT
D1
VIN = 12V
0.1P
CIN
2.2P
RFADJ
HG
VCC
SD
BOOT
PWGD
ISEN
LM2742
FREQ
267k
CSS
12n
PGND
SGND
PGND
EAO
Vo = 5V@1.8A
L1
LG
SS
CINX + CIN1
680 PF
10 PF
16V
16V
1.54A
Q1
47 PH, 2.7A
53 m:
D2
RFB2
10k
RC2
CC3
+ Co1,2
2 x 680 PF
16V
26 m:
Cox
10 PF
6.3V
FB
750
CC1
CC2
3.9n
56p
22n
RFB1
1.37k
RC1
61.9k
Figure 37. 12V to 5V, 1.8A, 100kHz
In situations where low cost is very important, the LM2742 can also be used as an asynchronous controller, as
shown in the above circuit. Although a a schottky diode in place of the bottom FET will not be as efficient, it will
cost much less than the FET. The 5V at low current needed to run the LM2742 could come from a zener diode or
inexpensive regulator, such as the one shown in Figure 36. Because the LM2742 senses current in the low side
MOSFET, the current limit feature will not function in an asynchronous design. The ISEN pin should be left open
in this case.
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Table 1. Bill of Materials for Typical Application Circuit (Figure 1)
ID
U1
Part Number
Type
Synchronous
Controller
LM2742
Q1, Q2
Si4884DY
L1
RLF7030T-1R5N6R1
N-MOSFET
Cin1, Cin2
C2012X5R1J106M
MLCC
Capacitor
Cinx
C3216X7R1E105K
Co1, Co2
6MV2200WG
Cboot
VJ1206X104XXA
Inductor
Size
Parameters
Qty.
Vendor
TSSOP-14
TSSOP-14
1
NSC
SO-8
30V, 13mΩ, 15nC
1
Vishay
7.1x7.1x3.2mm
1.5µH, 6.1A 9.6mΩ
1
TDK
0805
10µF 6.3V
2
TDK
1206
1µF, 25V
1
TDK
10mm D 20mm H
2200µF 6.3V125mΩ
2
Sanyo
Capacitor
1206
0.1µF, 25V
1
Vishay
AL-E
Cin
C3216X7R1E225K
Capacitor
1206
2.2µF, 25V
1
TDK
Css
VJ1206X123KXX
Capacitor
1206
12nF, 25V
1
Vishay
Cc1
VJ1206A2R2KXX
Capacitor
1206
2.2pF 10%
1
Vishay
Cc2
VJ1206A181KXX
Capacitor
1206
180pF 10%
1
Vishay
Rin
CRCW1206100J
Resistor
1206
10Ω 5%
1
Vishay
Rfadj
CRCW12066342F
Resistor
1206
63.4kΩ 1%
1
Vishay
Rc1
CRCW12063923F
Resistor
1206
392kΩ 1%
1
Vishay
Rfb1
CRCW12061002F
Resistor
1206
10kΩ 1%
1
Vishay
Rfb2
CRCW12061002F
Resistor
1206
10kΩ 1%
1
Vishay
Rcs
CRCW1206222J
Resistor
1206
2.2kΩ 5%
1
Vishay
Table 2. Bill of Materials for Circuit of Figure 31
(Identical to BOM for 1.5V except as noted below)
ID
Part Number
Size
Parameters
Qty.
Vendor
L1
RLF12560T-2R7N110
Inductor
Type
12.5x12.8x6mm
2.7µH, 14.4A 4.5mΩ
1
TDK
Co1, Co2,
Co3, Co4
10TPB100M
POSCAP
7.3x4.3x2.8mm
100µF 10V 1.9Arms
4
Sanyo
Cc1
VJ1206A6R8KXX
Capacitor
1206
6.8pF 10%
1
Vishay
Cc2
VJ1206A271KXX
Capacitor
1206
270pF 10%
1
Vishay
Cc3
VJ1206A471KXX
Capacitor
1206
470pF 10%
1
Vishay
Rc2
CRCW12068451F
Resistor
1206
8.45kΩ 1%
1
Vishay
Rfb1
CRCW12061102F
Resistor
1206
11kΩ 1%
1
Vishay
Qty.
Vendor
1
NSC
Table 3. Bill of Materials for Circuit of Figure 32
ID
22
Part Number
Type
Synchronous
Controller
Size
Parameters
U1
LM2742
Q1
Si4442DY
N-MOSFET
SO-8
30V, 4.1mΩ, @ 4.5V, 36nC
1
Vishay
Q2
Si4442DY
N-MOSFET
SO-8
30V, 4.1mΩ, @ 4.5V, 36nC
1
Vishay
D1
BAT-54
SOT-23
30V
1
Vishay
Lin
SLF12575T-1R2N8R2
Inductor
12.5x12.5x7.5mm
12µH, 8.2A, 6.9mΩ
1
Coilcraft
L1
D05022-152HC
Inductor
22.35x16.26x8mm
1.5µH, 15A,4mΩ
1
Coilcraft
16mm D 25mm H
5600µF10V 2.35Arms
2
Sanyo
Schottky Diode
Cin1, Cin2
10MV5600AX
Aluminum
Electrolytic
Cinx
TSSOP-14
C3216X7R1E105K
Capacitor
1206
1µF, 25V
1
TDK
Co1, Co2,
Co3
10MV5600AX
Aluminum
Electrolytic
16mm D 25mm H
5600µF10V 2.35Arms
2
Sanyo
Cboot
VJ1206X104XXA
Capacitor
1206
0.1µF, 25V
1
Vishay
Cin
C3216X7R1E225K
Capacitor
1206
2.2µF, 25V
1
TDK
Css
VJ1206X123KXX
Capacitor
1206
12nF, 25V
1
Vishay
Cc1
VJ1206A4R7KXX
Capacitor
1206
4.7pF 10%
1
Vishay
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Table 3. Bill of Materials for Circuit of Figure 32 (continued)
ID
Part Number
Size
Parameters
Qty.
Vendor
Cc2
VJ1206A102KXX
Capacitor
Type
1206
1nF 10%
1
Vishay
Rin
CRCW1206100J
Resistor
1206
10Ω 5%
1
Vishay
Rfadj
CRCW12068872F
Resistor
1206
88.7kΩ 1%
1
Vishay
Rc1
CRCW12062293F
Resistor
1206
229kΩ 1%
1
Vishay
Rfb1
CRCW12064991F
Resistor
1206
4.99kΩ 1%
1
Vishay
Rfb2
CRCW12064991F
Resistor
1206
4.99kΩ 1%
1
Vishay
Rcs
CRCW1206152J
Resistor
1206
1.5kΩ 5%
1
Vishay
Qty.
Vendor
1
NSC
30V, 24mΩ/ 8nC
Top 16.5mΩ/ 15nC
1
Vishay
Table 4. Bill of Materials for Circuit of Figure 33
ID
Part Number
U1
LM2742
Type
Q1/Q2
Si4826DY
L1
DO3316P-222
Inductor
12.95x9.4x 5.21mm
2.2µH, 6.1A, 12mΩ
1
Coilcraft
Cin1
10TPB100ML
POSCAP
7.3x4.3x3.1mm
100µF 10V 1.9Arms
1
Sanyo
Co1
4TPB220ML
POSCAP
7.3x4.3x3.1mm
220µF 4V 1.9Arms
1
Sanyo
Cc
C3216X7R1E105K
Capacitor
1206
1µF, 25V
1
TDK
Cin
C3216X7R1E225K
Capacitor
1206
2.2µF, 25V
1
TDK
Css
VJ1206X123KXX
Capacitor
1206
12nF, 25V
1
Vishay
Cc1
VJ1206A100KXX
Capacitor
1206
10pF 10%
1
Vishay
Cc2
VJ1206A561KXX
Capacitor
1206
560pF 10%
1
Vishay
Rin
CRCW1206100J
Resistor
1206
10Ω 5%
1
Vishay
Rfadj
CRCW12064222F
Resistor
1206
42.2kΩ 1%
1
Vishay
Rc1
CRCW12065112F
Resistor
1206
51.1kΩ 1%
1
Vishay
Rfb1
CRCW12062491F
Resistor
1206
2.49kΩ 1%
1
Vishay
Rfb2
CRCW12064991F
Resistor
1206
4.99kΩ 1%
1
Vishay
Rcs
CRCW1206272J
Resistor
1206
2.7kΩ 5%
1
Vishay
Qty.
Vendor
1
NSC
Synchronous
Controller
Asymetric Dual
N-MOSFET
Size
Parameters
TSSOP-14
SO-8
Table 5. Bill of Materials for Circuit of Figure 34
ID
Part Number
U1
LM2742
Type
Q1
Si4884DY
N-MOSFET
SO-8
30V, 13.5mΩ, @ 4.5V
15.3nC
1
Vishay
Q2
Si4884DY
N-MOSFET
SO-8
30V, 13.5mΩ, @ 4.5V
15.3nC
1
Vishay
SOT-23
30V
1
Vishay
7.29x7.29 3.51mm
1µH, 11A 3.7mΩ
1
Pulse
12x12x4.5 mm
1µH, 11A, 3.7mΩ
1
Pulse
16mm D 25mm H
5600µF 10V 2.35Arms
1
Sanyo
Synchronous
Controller
D1
BAT-54
Lin
P1166.102T
Schottky Diode
Inductor
Inductor
L1
P1168.102T
Cin1
10MV5600AX
Aluminum
Electrolytic
Size
Parameters
TSSOP-14
Cinx
C3216X7R1E105K
Capacitor
1206
1µF, 25V
1
TDK
Co1, Co2,
Co3
16MV4700WX
Aluminum
Electrolytic
12.5mm D 30mm H
4700µF 16V 2.8Arms
2
Sanyo
Cboot
VJ1206X104XXA
Capacitor
1206
0.1µF, 25V
1
Vishay
Cin
C3216X7R1E225K
Capacitor
1206
2.2µF, 25V
1
TDK
Css
VJ1206X123KXX
Capacitor
1206
12nF, 25V
1
Vishay
Cc1
VJ1206A4R7KXX
Capacitor
1206
4.7pF 10%
1
Vishay
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LM2742
SNVS266C – MARCH 2004 – REVISED MARCH 2013
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Table 5. Bill of Materials for Circuit of Figure 34 (continued)
ID
Part Number
Size
Parameters
Qty.
Vendor
Cc2
VJ1206A681KXX
Capacitor
Type
1206
680pF 10%
1
Vishay
Rin
CRCW1206100J
Resistor
1206
10Ω 5%
1
Vishay
Rfadj
CRCW12064992F
Resistor
1206
49.9kΩ 1%
1
Vishay
Rc1
CRCW12061473F
Resistor
1206
147kΩ 1%
1
Vishay
Rfb1
CRCW12061492F
Resistor
1206
14.9kΩ 1%
1
Vishay
Rfb2
CRCW12064991F
Resistor
1206
4.99kΩ 1%
1
Vishay
Rcs
CRCW1206332J
Resistor
1206
3.3kΩ 5%
1
Vishay
Qty.
Vendor
1
NSC
30V, 24mΩ/ 8nC
Top 16.5mΩ/ 15nC
1
Vishay
Table 6. Bill of Materials for Circuit of Figure 35
ID
Part Number
U1
LM2742
Type
Q1/Q2
Si4826DY
Assymetric Dual
N-MOSFET
Schottky Diode
Synchronous
Controller
Size
Parameters
TSSOP-14
SO-8
D1
BAT-54
SOT-23
30V
1
Vishay
Lin
RLF7030T-1R0N64
Inductor
6.8x7.1x3.2mm
1µH, 6.4A, 7.3mΩ
1
TDK
Inductor
L1
RLF7030T-3R3M4R1
6.8x7.1x3.2mm
3.3µH, 4.1A, 17.4mΩ
1
TDK
Cin1
C4532X5R1E156M
MLCC
1812
15µF 25V 3.3Arms
1
Sanyo
Co1
C4532X5R1E156M
MLCC
1812
15µF 25V 3.3Arms
1
Sanyo
Cboot
VJ1206X104XXA
Capacitor
1206
0.1µF, 25V
1
TDK
Cin
C3216X7R1E225K
Capacitor
1206
2.2µF, 25V
1
TDK
Css
VJ1206X393KXX
Capacitor
1206
39nF, 25V
1
Vishay
Cc1
VJ1206A220KXX
Capacitor
1206
22pF 10%
1
Vishay
Cc2
VJ1206A681KXX
Capacitor
1206
680pF 10%
1
Vishay
Cc3
VJ1206A681KXX
Capacitor
1206
680pF 10%
1
Vishay
Rin
CRCW1206100J
Resistor
1206
10Ω 5%
1
Vishay
Rfadj
CRCW12061742F
Resistor
1206
17.4kΩ 1%
1
Vishay
Rc1
CRCW12061072F
Resistor
1206
10.7kΩ 1%
1
Vishay
Rc2
CRCW120666R5F
Resistor
1206
66.5Ω 1%
1
Vishay
Rfb1
CRCW12064991F
Resistor
1206
4.99kΩ 1%
1
Vishay
Rfb2
CRCW12061002F
Resistor
1206
10kΩ 1%
1
Vishay
Rcs
CRCW1206152J
Resistor
1206
1.5kΩ 5%
1
Vishay
Vendor
Table 7. Bill of Materials for 3.3V Circuit of Figure 35
(Identical to BOM for 1.8V except as noted below)
24
ID
Part Number
L1
RLF7030T-4R7M3R4
Cc1
VJ1206A270KXX
Cc2
Cc3
Rc1
Type
Size
Parameters
Qty.
6.8x7.1x 3.2mm
4.7µH, 3.4A, 26mΩ
1
TDK
Capacitor
1206
27pF 10%
1
Vishay
VJ1206X102KXX
Capacitor
1206
1nF 10%
1
Vishay
VJ1206A821KXX
Capacitor
1206
820pF 10%
1
Vishay
CRCW12061212F
Resistor
1206
12.1kΩ 1%
1
Vishay
Inductor
Rc2
CRCW12054R9F
Resistor
1206
54.9Ω 1%
1
Vishay
Rfb1
CRCW12062211F
Resistor
1206
2.21kΩ 1%
1
Vishay
Rfb2
CRCW12061002F
Resistor
1206
10kΩ 1%
1
Vishay
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LM2742
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SNVS266C – MARCH 2004 – REVISED MARCH 2013
Table 8. Bill of Materials for Circuit of Figure 36
ID
Part Number
U1
LM2742
Synchronous
Controller
Type
U2
LM78L05
Voltage Regulator
SO-8
Q1/Q2
Si4826DY
Assymetric Dual NMOSFET
SO-8
Schottky Diode
Size
Parameters
TSSOP-14
Qty.
Vendor
1
NSC
1
NSC
30V, 24mΩ/ 8nC
Top 16.5mΩ/ 15nC
1
Vishay
D1
BAT-54
SOT-23
30V
1
Vishay
Lin
RLF7030T-1R0N64
Inductor
6.8x7.1x3.2mm
1µH, 6.4A, 7.3mΩ
1
TDK
L1
SLF12565T-4R2N5R5
Inductor
12.5x12.5x6.5mm
4.2µH, 5.5A, 15mΩ
1
TDK
Sanyo
Cin1
16MV680WG
D: 10mm L: 12.5mm
680µF 16V 3.4Arms
1
Cinx
C3216X5R1C106M
Al-E
MLCC
1210
10µF 16V 3.4Arms
1
TDK
Co1 Co2
16MV680WG
MLCC
1812
15µF 25V 3.3Arms
1
Sanyo
Cox
C3216X5R10J06M
MLCC
1206
10µF 6.3V 2.7A
TDK
Cboot
VJ1206X104XXA
Capacitor
1206
0.1µF, 25V
1
Cin
C3216X7R1E225K
Capacitor
1206
2.2µF, 25V
1
Vishay
TDK
Css
VJ1206X123KXX
Capacitor
1206
12nF, 25V
1
Vishay
Cc1
VJ1206A8R2KXX
Capacitor
1206
8.2pF 10%
1
Vishay
Cc2
VJ1206X102KXX
Capacitor
1206
1nF 10%
1
Vishay
Cc3
VJ1206X472KXX
Capacitor
1206
4.7nF 10%
1
Vishay
Rfadj
CRCW12063252F
Resistor
1206
32.5kΩ 1%
1
Vishay
Rc1
CRCW12065232F
Resistor
1206
52.3kΩ 1%
1
Vishay
Rc2
CRCW120662371F
Resistor
1206
2.37Ω 1%
1
Vishay
Rfb1
CRCW12062211F
Resistor
1206
2.21kΩ 1%
1
Vishay
Rfb2
CRCW12061002F
Resistor
1206
10kΩ 1%
1
Vishay
Rcs
CRCW1206202J
Resistor
1206
2kΩ 5%
1
Vishay
Qty.
Vendor
Table 9. Bill of Materials for Circuit of Figure 37
ID
Part Number
Type
Size
Synchronous Controller
TSSOP-14
U1
LM2742
Q1
Si4894DY
D2
MBRS330T3
L1
SLF12565T-470M2R4
D1
MBR0520
Cin1
16MV680WG
Cinx
C3216X5R1C106M
Co1, Co2
16MV680WG
Cox
C3216X5R10J06M
Cboot
Parameters
1
NSC
N-MOSFET
SO-8
30V, 15mΩ, 11.5nC
1
Vishay
Schottky Diode
SO-8
30V, 3A
1
ON
TDK
Inductor
12.5x12.8x 4.7mm
47µH, 2.7A 53mΩ
1
Schottky Diode
1812
20V 0.5A
1
ON
Al-E
1206
680µF, 16V, 1.54Arms
1
Sanyo
MLCC
1206
10µF, 16V, 3.4Arms
1
TDK
Sanyo
Al-E
D: 10mm L: 12.5mm
680µF 16V 26mΩ
2
MLCC
1206
10µF, 6.3V 2.7A
1
TDK
VJ1206X104XXA
Capacitor
1206
0.1µF, 25V
1
Vishay
Cin
C3216X7R1E225K
Capacitor
1206
2.2µF, 25V
1
TDK
Css
VJ1206X123KXX
Capacitor
1206
12nF, 25V
1
Vishay
Cc1
VJ1206A561KXX
Capacitor
1206
56pF 10%
1
Vishay
Cc2
VJ1206X392KXX
Capacitor
1206
3.9nF 10%
1
Vishay
Cc3
VJ1206X223KXX
Capacitor
1206
22nF 10%
1
Vishay
Rfadj
CRCW12062673F
Resistor
1206
267kΩ 1%
1
Vishay
Rc1
CRCW12066192F
Resistor
1206
61.9kΩ 1%
1
Vishay
Rc2
CRCW12067503F
Resistor
1206
750kΩ 1%
1
Vishay
Rfb1
CRCW12061371F
Resistor
1206
1.37kΩ 1%
1
Vishay
Rfb2
CRCW12061002F
Resistor
1206
10kΩ 1%
1
Vishay
Rcs
CRCW1206122F
Resistor
1206
1.2kΩ 5%
1
Vishay
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LM2742
SNVS266C – MARCH 2004 – REVISED MARCH 2013
www.ti.com
REVISION HISTORY
Changes from Revision B (March 2013) to Revision C
•
26
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 25
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM2742MTC/NOPB
ACTIVE
TSSOP
PW
14
94
RoHS & Green
NIPDAU | SN
Level-1-260C-UNLIM
-40 to 125
2742
MTC
LM2742MTCX/NOPB
ACTIVE
TSSOP
PW
14
2500
RoHS & Green
NIPDAU | SN
Level-1-260C-UNLIM
-40 to 125
2742
MTC
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
