LM3075EVAL

User's Guide SNVA130A – January 2006 – Revised April 2013 AN-1411 LM3075 Evaluation Board 1 Introduction The LM3075 is a current mode synchronous buck controller. Use of synchronous rectification and pulseskipping operation at light load achieves high efficiency over a wide load range. Current mode control assures excellent line and load regulation and a wide loop bandwidth for fast response to load transients. It is achieved by sensing across the high side NFET, eliminating the need for a sense resistor. The switching frequency is selectable to 200 kHz or 300 kHz. Protection Features include over-voltage protection (OVP), under-voltage protection (UVP), thermal protection, and positive and negative peak current limiting. This evaluation board generates 5V at 5A from a wide input voltage range of 8V to 25V utilizing the LM3075. The switching frequency is set at 300 kHz. This document contains the demo board schematic, Bill of Materials, circuit design description, and PCB layout guide. Performance data and typical waveforms are also provided. For detailed information about the LM3075, please see LM3075 High Efficiency, Synchronous Current Mode Buck Controller (SNVS398). Figure 1. LM3075 Evaluation Board Top View All trademarks are the property of their respective owners. SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated AN-1411 LM3075 Evaluation Board 1 Electrical Specifications 2 www.ti.com Electrical Specifications Description Input Voltage VIN Min Typical Max Unit 8 - 25 V A Output Current IOUT 0 - 5 Output Voltage VOUT 4.80 5 5.20 V Ripple and Noise Irr(peak-peak) - - 50 mV +7% % 7 A 5.25 V Load Transient Load Step 5A Voltage Deviation Current Limit -7% ILIM 6 6.5 Startup Overshoot Undershoot Switching Frequency 3 4.75 fsw V 300 kHz Evaluation Board Setup Figure 2. LM3075 Evaluation Board Setup 2 AN-1411 LM3075 Evaluation Board SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Quick Setup Table www.ti.com Figure 3. LM3075 Evaluation Board Bottom View 4 Quick Setup Table Item Description Notes 1 DC Input Voltage range 8V ~ 25V 2 DC load current 0A ~ 5A, +5V 3 EN Float , Enable default EN = GND, Disable 4 FPWM FPWM float, skip mode available default FPWM = GND, skip mode disable 5 Frequency selection Rfs1 = 0, Rfs2 = open, the switching frequency is 300 kHz Rfs2 = open, Rfs2 = 0, FS connects to GND, the switching frequency is 200 kHz at typical In this design, the Vlin5 is chosen to drive the bootstrap circuit. The inductor and capacitor were chosen for operation at a 300 kHz switching frequency. Connecting the FS pin to Vlin5 through the zero ohm resistor Rfs1 sets the switching frequency to 300kHz. Grounding the FS pin (open Rfs1 and set Rfs2 to 0 Ohms) programs the regulator to 200 kHz. This evaluation board leaves options for input capacitors Cin1/Cin2, inductor L1, and output capacitors Cout1/Cout2. It provides the flexibility to evaluate the LM3075 in several different configurations. 5 Component Selection Detailed design procedures can be found in the LM3075 datasheet (SNVS398). In this document, we will give brief instructions on the circuit design and describe the procedures used in testing the board. SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated AN-1411 LM3075 Evaluation Board 3 Component Selection 5.1 www.ti.com Output Capacitor (Cout1, Cout2) and Inductor (L1) The first priority in component selection is given to the output capacitors as there is often less flexibility in this choice than in other parameters. These capacitors must meet requirements for output voltage rating, high ripple current rating, low ESR and surface mount design. Sanyo POSCAPs and Panasonic SP-CAPs are good choices. Here we selected the SP-CAP UE series. The typical ESR is 12 mΩ and the ripple current rating is 3.3A at 100 KHz/ 20°C. The allowable steady-state ripple voltage is 50 mV peak to peak maximum. In order to ensure reasonable design margins 40 mV is used. The allowed output voltage excursion during a load transient is: 'V TRANS = (7% - 4% ) x 5 V - 40 mV 2 = 130 mV (1) The maximum allowed total combined ESR is: RESR(MAX) = 130mV = 26 m: 5A (2) The minimum inductance value is determined as follows: L MIN = 25 V - 5.0 V 5.0 V x 12 m: x 300 kHz x 25 V 40 mV = 4 PH (3) A 7.8 µH inductor from TDK (RLF12545-7R8N5R4) is chosen. The RMS current rating is 5.4A that should prove adequate. The actual selection process usually involves several iterations of all of the above steps, from ripple voltage selection, to capacitor selection, to inductance calculations. Both the highest and the lowest input and output voltages and load transient requirements should be considered. The inductor ripple current can be determined as follows: 'IL = 5.0 V 25 V - 5.0 V x = 1.7A 300 kHz x 7.8 PH 25 V (4) Given a maximum load current of 5A, the ripple content is 1.7A / 5A = 34%. In general, the ripple current is designed to be between 20% and 40% of the full load output current although less than 50% is fine. Larger ripple current causes excessive losses in the inductor, higher RMS switch currents and overall lower efficiency. On the plus side, transient response will generally be better with higher ripple current. The corresponding minimum capacitance is calculated as follows: 5.6 PH x 130 mV C OUT = (160 mV)2- (5A x 12 m:) 2 2 5 x (12 m:) = 114 PF. (5) The output capacitance should be chosen to be slightly larger than the calculated value as capacitance and ESR vary somewhat with temperature. Here we would likely be fine choosing 180µF. 5.2 Input Capacitor (Cin1, Cin2) The input capacitor is selected to handle both the maximum ripple RMS current at highest ambient temperature as well as the maximum input voltage 25V. The maximum total input ripple RMS current for duty cycles under 50% is: Iin_rms = 5A 0.42 x (1 - 0.42) = 2.46A (6) Choose input capacitors that can handle 2.46A RMS of ripple current at the highest expected ambient temperature. Actually, there are several types of capacitor to choose from. The OS-CON series from Sanyo and TR3 series from Vishay are good choices with low ESR, high ripple current rating and small size. In this design, we selected the Vishay TR3 series 22 µF 35V. 4 AN-1411 LM3075 Evaluation Board SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Component Selection www.ti.com 5.3 MOSFET (Q1, Q2) In this design, VLIN5 is chosen to drive the bootstrap circuit. The top FET starts to turn on when the input voltage exceeds the threshold voltage of the UVLO, which has a minimum threshold of 3.8V. In this case VLIN5 will be approximately 3.8V when the LM3975’s UVLO clears. Thus the bias voltage to the top FET driver is about 3V after the bootstrap diode. So the gate threshold voltage of the top FET should be less than 3V (maximum). Vishay’s Si4840 is chosen for Q1 and Q2. The Si4840 has a maximum Rds(on) of 12 mΩ at VGS = 4.5V. The maximum gate threshold voltage is 3V. The total gate charge is 28 nC at maximum. Some FETs have on-resistance specified at 2.7V and would be potentially good choices as well, though slightly more expensive than the 4.5V specified parts. 5.4 Output Voltage Setting (Rfb1, Rfb2) The output voltage is set by the ratio of Rfb1 and Rfb2. The resistor values can be determined by the following equation: Rfb2 = Rfb1 V § OU T - 1· © VFB ¹ (7) Where VFB is the typical value of feedback pin voltage and VOUT is the nominal output voltage. Although increasing the value of Rfb1 and Rfb2 increases efficiency, this also decreases accuracy due to bias currents at the VFB input. Therefore, a maximum value is recommended for Rfb2 of about 15 kΩ . Rfb1 is simply calculated as 5 kΩ using the above equation. 5.5 Current Limit Setting (RLIM) With an external resistor connected between the ILIM pin and the CSH pin the LM3075’s internal 10µA current sink on the ILIM pin produces a voltage across the resistor to serve as the reference voltage for current limit. Adding a 10 nF capacitor across this resistor filters unwanted noise that could improperly trip the current limit comparator. The current limit resistor, RLIM, is calculated as follows: R 5.6 LI M §6.5A + 1 x 2A· x 12 m: 2 © ¹ = 10 P A (8) Switching Noise Reduction (Rboot) Rboot is added in series with the CBOOT pin to slow down the gate drive (HDRV) rise time. A slower drain current transition time will reduce shoot through currents and ringing on the switch node caused by parasitic inductance in the power path. Usually a 3.3 to 5.1 ohm resistor is sufficient to suppress excessive noise. It is important to note that the addition of these resistors does increase the power loss in the system and thus decrease the efficiency. It is therefore important to choose the size of the resistor carefully since the top FET switching losses will increase with higher resistance values. 5.7 Soft Start (Css, Cff) Capacitor Cff is used to add a little phase lead into the feedback loop. Add this cap if excessive overshoot is observed during start up or an improvement in transient response is required. SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated AN-1411 LM3075 Evaluation Board 5 PCB Layout Guide 6 www.ti.com PCB Layout Guide The layout should begin with the placement of power path components such as Cin1, Cin2, Q1, Q2, L1, Cout1, Cout2 and current sense resistor. The input capacitor Cout1 and Cout2 should always be placed as close as possible to the current sense resistor or the drain of the top FET. The power MOSFETs, Q1and Q2, should be located close to the inductor, L1. The source of Q2 should be as close as possible to the ground of input capacitor. This helps to keep large pulse currents out of the ground plane. The decoupling capacitor Cout3 should be located close to the output terminal. The objective is to minimize the loop area of power path, and minimize the stray inductance. The controller should be close to Q1 and Q2 in order to keep the gate drive paths as short as possible. The control circuits should be placed in the quiet area out of the power path loop. AGND and PGND should be directly connected with a wide trace. Connect PGND to the main ground plane with a via close to the pin. The bootstrap circuit Dboot and Cboot should be located adjacent to the SW and Cboot pins. All of the power source pins Vin, Vlin5 and Vdd should be well bypassed, respectively by Cbyp, Clin5 and Cvdd. In this design, the current limit is configured to sense the Vds of top switching FET, Q1. This configuration has the advantages of saving PCB area while lowering power loss and cost. However, the connection approach from CSH / CSL to the Drain/Source of the FETs needs to be thought out carefully. Poor layout will result in unwanted noise spikes at the source node of switching FET Q1. In general, the connection should be immediately adjacent to the Drain and Source pins respectively. The two traces should be placed in parallel. Figure 4. LM3075 Evaluation Board Top Layer Layout 6 AN-1411 LM3075 Evaluation Board Figure 5. LM3075 Evaluation Board Bottom Layer Layout SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Evaluation Board Schematic www.ti.com 7 Evaluation Board Schematic EXT Rfs2 Rfs1 Rvdd 1 F 4.7 13 Rpgood 100k VLIN5 U1 VDD Cvdd PGOOD Css 10 nF LDRV 12 Q2 Si4840 Cc1 22 nF Rc1 10k Cout2 150 F Cff 47 nF FB COMP AGND PGND 10 11 VIN +8V ± 25V GND SS Cc2 470 pF + Cin1 22 F 18 Q1 HDRV 15 Si4840 17 CSL Dboot BAT54C 14 Rboot L1 CBOOT Cboot 7.8 H 4.7 16 0.1 F SW 4.7 F PGOOD 19 ILIM CSH LM3075 1 Cvlin5 VDD VLIN5 * VIN Cbyp 1 F Cin2 22 F VDD EN FPWM FS Clim 10 nF 2 3 5 EN EXT 7 FPWM 4 FS + Cout1 + 150 F Rfb1 15.0k VOUT +5V/5A Cout3 0.1 F GND 6 *: Not Insert Rfb2 4.99k Figure 6. LM3075 Evaluation Board Schematic 8 Bill of Materials Table 1. Bill of Materials Designation Description Size Manufacture Part # Vendor Cbyp Cap Ceramic X7R 1µF 50V 1206 C3216X7R1H105K TDK Cin1, Cin2 Cap Tantalum 22µF 35V 2815 293D226X0035E2T Vishay Cin2 Cap Tantalum 22µF 35V 2815 293D226X0035E2T Vishay Cilim Cap MLCC 10nF 25V X7R 0805 VJ0805Y103KXX Vishay Cboot Cap MLCC 0.1µF 25V X7R 0805 VJ0805Y104KXX Vishay Cc2 Cap MLCC 470pF 25V X7R 0805 VJ0805Y471KXX Vishay Cc1 Cap MLCC 22nF 25V X7R 0805 VJ0805Y223KXX Vishay Vishay Css Cap MLCC 10nF 25V X7R 0805 VJ0805Y103KXX Cvdd Cap MLCC 4.7µF 10V X7R 1206 C3216X7R1C475K TDK Cvlin5 Cap MLCC 1µF 10V X7R 0805 VJ0805Y105KXQ Vishay Cout3 Cap MLCC 0.1µF 25V X7R 0805 VJ0805Y104KXX Vishay Cout1 Cap POSCAP 150µF 6.3V D2E 6TPE150MI Sanyo Cout2 Cap POSCAP 150µF 6.3V D2E 6TPE150MI Sanyo Cff Cap MLCC 0.047µF 25V X7R 0805 VJ0805Y473KXX Vishay Rilim Resistor Chip 9.09kΩ J 0805 CRCW08059091J Vishay Rfb2 Resistor Chip 4.99kΩ F 0805 CRCCW08054991F Vishay Rfb1 Resistor Chip 15.0kΩ F 0805 CRCW08051502F Vishay Rboot Resistor Chip 1Ω J 0805 CRCW08051R00J Vishay Rc1 Resistor Chip 10kΩ J 0805 CRCW0805103J Vishay Rvdd Resistor Chip 4.7Ω J 0805 CRCW08054R7J Vishay Rpgood Resistor Chip 100kΩ J 0805 CRCW0805105J Vishay Rfs1 Resistor Chip 0 J 1206 CRCW12060R0J Vishay SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated AN-1411 LM3075 Evaluation Board 7 Typical Performance and Waveforms www.ti.com Table 1. Bill of Materials (continued) 9 Designation Description Size Manufacture Part # Vendor Q1 FET N 40V 12A SO-8 Si4840 Vishay Vishay Q2 FET N 40V 12A SO-8 Si4840 L1 Inductor 7.8µH 5.4A 12.5x12.5x6 RLF12545-7R8N5R4PF TDK Dboot Diode Shottky 30V 0.5A SOT-23 BAT54C Vishay U1 IC LM3075 TSSOP-20 LM3075 Texas Instruments PCB LM3075 Demoboard Aug-05 Texas Instruments Typical Performance and Waveforms 97 INPUT VOLATGE (V) 95 VIN = 8V 93 91 VIN = 12V 89 VIN = 25V 87 85 0 1 2 3 4 5 OUTPUT CURRENT (A) Figure 7. Efficiency 8 AN-1411 LM3075 Evaluation Board SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Typical Performance and Waveforms www.ti.com OUTPUT VOLTAGE (V) 4.960 4.955 4.950 VIN = 25V 4.945 VIN = 8V VIN = 12V 4.940 0 1 2 3 4 5 6 OUTPUT CURRENT (A) Figure 8. Output Regulation Figure 9. Ripple and Noise SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated AN-1411 LM3075 Evaluation Board 9 Typical Performance and Waveforms www.ti.com Figure 10. Ripple and Noise Figure 11. Startup 10 AN-1411 LM3075 Evaluation Board SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Typical Performance and Waveforms www.ti.com Figure 12. Startup Figure 13. Load Transient Response SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated AN-1411 LM3075 Evaluation Board 11 Typical Performance and Waveforms www.ti.com Figure 14. Load Transient Response 12 AN-1411 LM3075 Evaluation Board SNVA130A – January 2006 – Revised April 2013 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. Buyers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. All semiconductor products (also referred to herein as “components”) are sold subject to TI’s terms and conditions of sale supplied at the time of order acknowledgment. TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms and conditions of sale of semiconductor products. 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