LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
LM3209-G3 Seamless-Transition Buck-Boost Converter for Battery-Powered 3G/4G RF
Power Amplifiers
Check for Samples: LM3209-G3
FEATURES
APPLICATIONS
•
•
•
•
•
1
2
•
•
•
•
•
•
•
•
•
•
Operates From a Single Li-Ion Cell: 2.7V to
5.5V
Adjustable Output Voltage: 0.6V to 4.2V
1A Maximum Load Capability for VIN ≥ 3.2V,
VOUT = 3.6V
2.4 MHz (typ.) Switching Frequency
Seamless Buck-Boost Mode Transition
Fast Output Voltage Transition: 0.8V to 4.0V in
20 µs
High-Efficiency: 95% typ. at 3.7 VIN, 3.5 VOUT, at
300 mA
Cycle-by-cycle Over-Current Limit
Output Over-Voltage Clamp
Internal Compensation
12-bump DSBGA Package
Battery-Powered 3G/4G RF PAs
Cellular Phones
Portable Hard Disk Drives
PDAs
DESCRIPTION
The LM3209-G3 is buck-boost DC/DC converter
designed to generate output voltages above or below
a given input voltage. It is particularly suitable for
single-cell Li-ion batteries for portable applications.
The LM3209-G3 operates at a 2.4 MHz typical
switching frequency in full synchronous operation
providing seamless transitions between buck and
boost operating modes.
The power converter topology needs only one
external inductor and two capacitors. Five internal
power switches enable high overall efficiency.
The LM3209-G3 is internally compensated for buck
and boost modes of operation, thus providing an
optimal transient response.
The LM3209-G3 is available in an 12-bump lead-free
DSBGA package of size 2.0 mm x 2.5 mm x 0.6 mm.
TYPICAL APPLICATION CIRCUIT
2.2 PH
SW1
SW2
VIN: 2.7V to 5.5V
VOUT: 0.6V to 4.2V
PVIN
VOUT
PVIN
FB
RF PA
LM3209-G3
VCON
EN
+
4.7 PF
10 PF
SGND
PGND
DAC
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2009–2013, Texas Instruments Incorporated
LM3209-G3
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
www.ti.com
CONNECTION DIAGRAMS AND PACKAGE MARK INFORMATION
Figure 1. 12–Bump Thin DSBGA Package, Large Bump
A1
A2
A3
A3
A2
A1
B1
B2
B3
B3
B2
B1
C1
C2
C3
C3
C2
C1
D1
D2
D3
D3
D2
D1
Top View
Bottom View
PIN DESCRIPTIONS
Pin #
Name
A1
NC
Description
This pin is shorted to ground internally. Leave this pin floating.
B1
VCON
C1
FB
Output voltage program pin. Analog voltage from DAC/controller to set VOUT.
D1
VOUT
A2
NC
Supply voltage for analog circuits of LM3209-G3. This pin is connected to PVIN via a 36Ω
resistor internally. Leave this pin floating.
Enable Pin. Pulling this pin higher than 1.2V enables part to function.
Feedback input to inverting input of error amplifier. Connect output voltage directly to this node
at load point.
Regulated output voltage of LM3209-G3. Connect this to a 4.7 µF ceramic output filter
capacitor to GND.
B2
EN
C2
SGND
D2
SW2
Switch pin for Internal Power Switches M3 and M4. Connect inductor between SW1 and SW2.
A3
PVIN
Power MOSFET input and power current input pin. Optional low-pass filtering may help buck
and buck-boost modes for radiated EMI and noise reduction.
B3
PVIN
Power MOSFET input and power current input pin. Optional low-pass filtering may help buck
and buck-boost modes for radiated EMI and noise reduction.
C3
SW1
Switch pin for Internal Power Switches M1 and M2. Connect inductor between SW1 and SW2.
D3
PGND
Signal Ground for analog circuits.
Power Ground for Power MOSFETs and gate drive circuitry.
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
2
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
ABSOLUTE MAXIMUM RATINGS (1) (2) (3)
−0.2V to +6.0V
PVIN pin: Voltage to GND
−0.2V) to (VIN +0.2V) w/6.0V
max.
EN, FB, VCON, VOUT pin: Voltage to GND
−0.2V to +0.2V
PGND to SGND
(PGND −0.2V)
to (PVIN +0.2V) w/6.0V
SW1, SW2
Continuous Power Dissipation (4)
Internally Limited
Junction Temperature (TJ-MAX)
+150°C
−65°C to +150°C
Storage Temperature Range
Maximum Lead Temperature (Soldering 10 sec.)
+260°C
ESD Rating, Human Body Model (5) (6)
(1)
(2)
(3)
(4)
(5)
(6)
2kV
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltages are with respect to the potential at the GND pins.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
Internal thermal shutdown circuitry protects the device from permanent damage. Thermal shutdown engages at TJ = 150°C (typ.) and
disengages at TJ = 120°C (typ.).
The Human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. (MIL-STD-883 3015.7)
Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper ESD
handling procedure can result in damage.
OPERATING RATINGS (1) (2)
Input Voltage Range
2.7V to 5.5V
Recommended Load Current
0 to 650 mA
−30°C to +125°C
Junction Temperature (TJ) Range
Ambient Temperature (TA) Range (3)
(1)
(2)
(3)
−30°C to +85°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltages are with respect to the potential at the GND pins.
In applications where high power dissipation and/or poor package thermal resistance is present, the maximum ambient temperature may
have to be de-rated. Maximum ambient temperature (TA-MAX) is dependent on the maximum operating junction temperature (TJ-MAX-OP =
125°C), the maximum power dissipation of the device in the application (PD-MAX), and the junction-to ambient thermal resistance of the
part/package in the application (θJA), as given by the following equation: TA-MAX = TJ-MAX-OP – (θJA × PD-MAX).
THERMAL PROPERTIES
Junction-to-Ambient Thermal, Resistance (θJA), YZR Package
85°C/W
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
3
LM3209-G3
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
www.ti.com
ELECTRICAL CHARACTERISTICS (1) (2)
Limits in standard typeface are for TA = TJ = 25°C. Limits in boldface type apply over the full operating ambient temperature
range (−30°C ≤ TJ = TA ≤ +85°C). Unless otherwise noted, specifications apply to the LM3209-G3 Typical Application Circuit
with: PVIN = EN = 3.6V.
Symbol
VFB,
VFB,
Min
Typ
Max
Units
min
Min FB voltage
Parameter
VCON = 0.2V
Conditions
0.530
0.600
0.670
V
max
Max FB voltage
VCON = 1.4V
4.130
4.200
4.270
V
(3)
IQ
Quiescent current
No switching ,
VCON = 0.1V, FB = PVIN
0.8
2.0
mA
ISHDN
Shutdown supply current
EN = 0V, VCON = 0V,
SW1 = SW2 = VOUT = 0V
0.02
2
µA
RDSON
PMOS
Buck PMOS switch on resistance
(Small PFET)
M1, ISW1 = 200 mA
415
RDSON
PMOS
Buck PMOS switch on resistance
(Large+Small PFET)
M1, ISW1 = 200 mA
120
140
165
mΩ
RDSON
PMOS
Buck PMOS switch on resistance
during boost operation
M1, ISW1 = 200 mA
80
90
110
mΩ
RDSON
NMOS
Buck and Boost NMOS switch on
resistance
M2, ISW1 = -200 mA
M4, ISW2 = −200 mA
215
230
285
mΩ
RDSON
PMOS
Boost PMOS switch on resistance
(between SW2 and VOUT)
M3, ISW2 = 200 mA,
VOUT = 3.4V
90
105
135
mΩ
RDSON
NMOS
NMOS output switch on
resistance (between SW2 and
VOUT)
M5, ISW2 = 200 mA
VOUT = 0.8V
100
110
135
mΩ
ILIM_L
Input Current Limit (Large)
Open Loop (4)
1500
1700
1900
mA
ILIM_S
Input Current Limit (Small)
Open Loop
(4)
750
850
ISHRT
Output Short Current
FB ≤ 0.35V
FOSC
Internal Oscillator Frequency
Gain
Internal Gain (5)
IEN
EN pin pull down current
5
10
µA
ICON
VCON pin input current
0.02
2
µA
VIH
Logic High Input Threshold for EN
VIL
Logic Low Input Threshold for EN
(1)
(2)
(3)
(4)
(5)
mΩ
mA
850
2.1
0.2V ≤ VCON ≤ 1.4V
2.4
mA
2.7
3
MHz
V/V
1.2
V
0.6
V
All voltages are with respect to the potential at the GND pins.
Min and Max limits are verified by design, test, or statistical analysis.
IQ specified here is when the part is not switching.
Current limit is built-in, fixed, and not adjustable.
To calculate VOUT, use the following equation: VOUT = VCON × 3
Dissipation Rating Table
4
Ambient Temperature
TA = 25°C
TA = 55°C
TA = 85°C
Power Dissipation
1.176 (W)
0.82 (W)
0.47 (W)
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
SYSTEM CHARACTERISTICS
The following spec table entries are specified by design and verifications providing the component values in the typical
application circuit are used (L = 2.2 µH, DCR = 110 mΩ, MIPSZ2520D2R2/FDK; CIN = 10 µF 6.3V, C1608X5R0J106K/TDK;
COUT = 4.7 µF, 6.3V, ECJ1VB0J475K/Panasonic). These parameters are not verified by production testing. Min and Max
limits in apply over the full operating ambient temperature range (−30°C ≤ TA ≤ 85°C) and over the VIN range (= PVIN = EN)
= 2.7V to 5.5V unless otherwise specified.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
TON
Turn on time (time for output to
reach 0V→90% × 3.5V)
EN = L to H, VIN = 3.7V,
IOUT = 0 mA
35
50
µs
TOFF
Turn off time (time output to
reach 3.5V→10% × 3.5V)
EN = H to L, VIN = 3.7V,
IOUT = 0 mA
50
100
µs
IOUT_MAX
Max output current
VIN ≥ 3.2V, VOUT = 4.2V
Buck (% M1 on)
100
10
pF
+70
mV
50
mV
50
mV
Maximum Duty Cycle
CCON
VCON input capacitance
VCON = 1V,
Test frequency = 100 kHz
VCON_LIN
VCON linearity
0.2V ≤ VCON ≤ 1.4V
Ripple voltage
VIN ≥ 3.2V, 0.6 ≤ VOUT ≤ 4.2V,
0 mA ≤ IOUT ≤ 430 mA, TA = 25°C
Ripple voltage in mode transition
VIN = 3.0V to 5.0V,
VIN = TR = TF = 30s
3.3V ≤ VOUT ≤ 4.2V
VOUT
Output Voltage Accuracy
15
VCON = 0.2V, IOUT = 70 mA
0.530
0.600
0.670
VCON = 0.4V, IOUT = 70 mA
1.130
1.200
1.270
VCON = 0.833V, IOUT = 200 mA
2.430
2.50
2.57
VCON = 1.167V, IOUT = 300 mA
3.431
3.50
3.57
VCON = 1.333V, IOUT = 350 mA
3.930
4.000
4.070
VCON = 1.4V, IOUT = 500 mA, VIN ≥ 3.2V
4.13
4.20
4.27
V
VIN = 3.2V to 4.9V, VIN TR = TF = 10 µs,
VOUT = 3.5V
10
mV
Load Regulation
IOUT = 0 mA to 500 mA,
IOUT = TR = TF = 1 µs,
VIN = 3.2V to 4.9V
20
mV
200
mV
VCON transient response
overshoot
VCON transient response rise
time
VCON transient response fall
time
η
−70
%
Line Regulation
ΔVOUT
VCON_TR
mA
50
DMAX
VO_RIPPLE
500
Boost (% M4 on)
Efficiency
VIN = 3.2V to 4.2V,
VOUT = 0.8V to 4.0V,
VCON Tr = Tf = 1 µs,
RLOAD = 11.4Ω
20
µs
50
VIN = 3.7V, VOUT = 1.2V,
IOUT = 70 mA
80
85
VIN = 3.7V, VOUT = 2.5V,
IOUT = 200 mA
90
95
VIN = 3.7V, VOUT = 3.5V,
IOUT = 300 mA
90
95
VIN = 3.7V, VOUT = 4.1V,
IOUT = 350 mA
85
95
%
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
5
LM3209-G3
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
www.ti.com
FUNCTIONAL BLOCK DIAGRAM
PVIN
PVIN
NC
To
Analog Supply
SW1
SW2
VOUT
SMALL
FET
LARGE
FET
M6_g
M6
M3
GATE
DRIVE
CIRCUITS
M1
M2
M5
M4
Error
Amplifier
PWM
CONTROL
LOGIC
+
1.7A
-
+
VCON
-
FB
Input Over Current
Protection
INTERNAL
LOOP
COMPENSATION
M6_g
EN
CLK
PWM RAMP
NC
SGND
6
PGND
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
TYPICAL PERFORMANCE CHARACTERISTICS
(VIN = PVIN = EN = 3.6V and TA = 25°C, unless otherwise noted)
Quiescent Current
vs
Supply Voltage
(VCON = 0.5, PVIN = VOUT = FB, No Switching)
Shutdown Current
vs
Temperature
(VCON = VOUT = SW1 = SW2 = EN = 0V)
Figure 2.
Figure 3.
Closed Loop Supply Current
vs
Output Voltage
(No load)
Switching Frequency
vs
Temperature
(VOUT = 3.5V, IOUT = 300 mA)
Figure 4.
Figure 5.
Current Limit
vs
Temperature
(Large PFET, VOUT = 3.5V)
Current Limit
vs
Temperature
(Small PFET, VOUT = 1.2V)
Figure 6.
Figure 7.
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
7
LM3209-G3
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
www.ti.com
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
(VIN = PVIN = EN = 3.6V and TA = 25°C, unless otherwise noted)
VCON Voltage
vs
Output Voltage
(No load)
Load Capability
vs
Output Voltage
5.00
VIN = 2.7V, 3.6V, 4.2V, 5.5V
OUTPUT VOLTAGE (V)
4.00
3.00
2.00
1.00
0.00
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
VCON VOLTAGE (V)
Figure 8.
Figure 9.
EN High Threshold
vs
Supply Voltage
8
(VIN
Efficiency
vs
Output Voltage
= 3.7V, RLOAD = 15Ω)
Figure 10.
Figure 11.
Efficiency
vs
Output Current
(VOUT = 1.2V)
Efficiency
vs
Output Current
(VOUT = 2.5V)
Figure 12.
Figure 13.
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
(VIN = PVIN = EN = 3.6V and TA = 25°C, unless otherwise noted)
Efficiency
vs
Output Current
(VOUT = 3.5V)
Efficiency
vs
Output Current
(VOUT = 4.1V)
Figure 14.
Figure 15.
RDSON
vs
Temperature
(M1 PFET, ISW = 200 mA)
RDSON
vs
Supply Voltage
(M1 Small PFET, ISW = 200 mA)
Figure 16.
Figure 17.
RDSON
vs
Supply Voltage
(M2, M4 NFET, ISW = -200 mA)
RDSON
vs
Supply Voltage
(M3 and M5 FET, ISW = 200 mA)
Figure 18.
Figure 19.
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
9
LM3209-G3
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
www.ti.com
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
(VIN = PVIN = EN = 3.6V and TA = 25°C, unless otherwise noted)
VCON Transient Response
(VIN = 3.7V, VOUT = 0.8V/4.0V, RLOAD = 15Ω)
Output Voltage Ripple in Buck Mode
(VOUT = 1.2V, IOUT = 70 mA)
2V/DIV
VCON
SW1
5V/DIV
SW2
5V/DIV
1V/DIV
VOUT
VOUT
10 mV/DIV
AC Coupled
IL
200 mA/DIV
1A/DIV
IL
20 μs/DIV
500 ns/DIV
Figure 20.
Figure 21.
Output Voltage Ripple in Buck-Boost Mode
(VIN = 3.598V, VOUT = 3.5V, IOUT = 300 mA)
Output Voltage Ripple in Boost Mode
(VIN = 3.2V, VOUT = 4.0V, IOUT = 350 mA)
SW1
5V/DIV
SW1
5V/DIV
SW2
5V/DIV
SW2
5V/DIV
VOUT
20 mV/DIV
AC Coupled
VOUT
20 mV/DIV
AC Coupled
IL
200 mA/DIV
IL
200 mA/DIV
500 ns/DIV
500 ns/DIV
Figure 22.
Figure 23.
Startup
(VOUT = 0V to 3.5V)
Shutdown
(VOUT = 3.5V to 0V)
EN
4V/DIV
VOUT
1V/DIV
IL
1A/DIV
EN
4V/DIV
VOUT
1V/DIV
10 és/DIV
20 és/DIV
Figure 24.
10
Figure 25.
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
(VIN = PVIN = EN = 3.6V and TA = 25°C, unless otherwise noted)
Time Current Limit
(VOUT = 2.5V to Ground shorted, RLOAD = 15Ω)
SW1
5V/DIV
SW2
5V/DIV
Load Transient
(VIN = 3.3V, VOUT = 4.2V, RLOAD = 11Ω/22Ω)
IOUT
100
mA/DIV
VOUT
50 mV/
DIV
1V/DIV
VOUT
1A/DIV
IL
20 Ps/DIV
2 és/DIV
Figure 26.
Figure 27.
Load Transient
(VIN = 4.2V, VOUT = 3.6V, RLOAD = 11Ω/22Ω)
Line Transient
(VIN = 3.6V-4.2, VOUT = 3.9V, RLOAD = 11.4Ω)
Figure 28.
Figure 29.
Line Transient
(VIN = 2.7V-3.3V, VOUT = 3.9V, RLOAD = 11.4Ω)
Figure 30.
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
11
LM3209-G3
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
www.ti.com
OPERATION DESCRIPTION
The LM3209-G3 buck-boost converter provides high-efficiency, low-noise power for RF power amplifiers (PAs) in
mobile phones, portable communicators and similar battery-powered RF devices. It is designed to allow the RF
PA to operate at maximum efficiency for a wide range of power levels from a single Li-Ion battery cell. The
capability of LM3209-G3 to provide an output voltage lower than as well as higher than the input battery voltage
also enables the PA to operate with high linearity for a wide range of battery voltages thereby extending the
usable voltage range of the battery. The converter feedback loop is internally compensated for both buck and
boost operation and the architecture is such that it provides seamless transition between buck and boost mode of
operation.
The efficiency of LM3209-G3 is typically around 95% for a 300 mA load with 3.5V output, 3.7V input. The
LM3209-G3 has an RDSON management scheme for low as well as high output voltage. This achieves high
efficiency for a wide range of output voltage. The output voltage is dynamically programmable from 0.6V to 4.2V
by adjusting the voltage on the control pin, VCON , without the need for external feedback resistors. The fast
output voltage transient response of LM3209-G3 makes it suitable for adaptively adjusting the PA supply voltage
depending on its transmitting power which prolongs battery life.
Additional features include current overload protection, output over voltage clamp and thermal overload
shutdown.
The LM3209-G3 is constructed using a chip-scale 12-bump DSBGA package that offers the smallest possible
size for space-critical applications such as cell phones where board area is an important design consideration.
Use of high switching frequency (2.4 MHz, typ.) reduces the size of the external components. As shown in
Typical Application Circuit, only three external power components are required for circuit operation. Use of the
DSBGA package requires special design considerations for implementation. (See DSBGA PACKAGE
ASSEMBLY AND USE in the APPLICATION INFORMATION section.) Its fine bump-pitch requires careful board
design and precision assembly equipment. Use of this package is best suited for opaque-case applications,
where its edges are not subjected to high-intensity ambient red or infrared light.
SHUTDOWN MODE
Setting the EN digital pin low (< 0.6V) places the LM3209-G3 in shutdown mode (0.01 µA ISHDN typ.). During
shutdown, the output of LM3209-G3 is pulled to ground enabling complete discharge of the output capacitor.
Setting EN high (>1.2V) enables normal operation. EN should be set low to turn off the LM3209-G3 during
power-up and under voltage conditions when the power supply is less than the 2.7V minimum operating voltage.
VCON,ON
The output is disabled when VCON is below 125 mV (typ.). It is enabled when VCON is above 150 mV (typ.).
The threshold has approximately 25 mV (typ.) of hysteresis.
RDSON MANAGEMENT
The LM3209-G3 has a unique RDSON management function to improve efficiency in low output voltage as well as
high output voltage conditions. When VCON < 0.775V (typ.) the device uses only a small part of the PMOS M1
to minimize drive loss. When VCON > 0.775V, a large PMOS is also used along with the small PMOS. For RF
PAs, the current consumption typically increases with its supply voltage and hence higher supply voltage for the
PA also means higher current delivered to it. Adding a large PMOS for VCON > 1.124V reduces the conduction
losses thereby achieving higher efficiency. The LM3209-G3 can also provide output voltages higher than the
battery voltage. This boost mode of operation is typically used when the battery voltage has discharged to a low
voltage that is not sufficient to provide the required linearity in the PA. A special RDSON management scheme is
designed for operation well into boost mode such that an auxiliary PMOS switch is also turned on along with the
large and small PMOS switches. This effectively reduces the RDSON of M1 to a very low value in order to keep
the efficiency maximized. Since M1 conducts all the time in boost mode, reducing the RDSON of M1 achieves a
significant improvement in efficiency.
12
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
SUPPLY CURRENT LIMIT
A current limit feature allows the LM3209-G3 to protect itself and external components during overload
conditions. In PWM mode, a 1700 mA (typ.) cycle-by-cycle current limit is normally used when VCON is above
0.775V (typ.) and an 850 mA (typ.) limit is used when VCON is below 0.775V (typ.). If an excessive load pulls the
output voltage down to approximately 0.35V, then the device switches to a timed current limit mode. The current
limit in this mode is 850 mA (typ.) independent of the set VCON voltage. In timed current limit mode, the internal
PMOS M1 is turned off after the current limit is hit and the beginning of the next cycle is inhibited for 3.5 µs to
force the inductor current to ramp down to a safe value.
REVERSE CURRENT LIMIT
Since LM3209-G3 features dynamically adjustable output voltage, the inductor current can build up to high
values in either direction depending on the output voltage transient. For a low to high output voltage transient,
the inductor current flows from SW1 pin to SW2 pin, and this current is limited by the current limit feature
monitoring PMOS M1. For a high to low output voltage transient, the inductor current flows from SW2 pin to SW1
pin, and this current needs to be limited to protect the LM3209-G3 as well as the external components. A reverse
current limit feature allows monitoring the reverse inductor current that also flows through NMOS M2. A -1.2A
(typ.) cycle-by-cycle current limit is used to limit the reverse current. When the reverse current hits the reverse
current limit during a PWM cycle, NMOS M2 is turned off and MOSFET M1 and M4 are turned on for the rest of
that switching cycle. This allows the inductor to build current in the opposite direction thereby limiting the reverse
current. It should be noted that the power MOSFET switches M3 and M4 do not have their own current limiting
circuits and are dependent on the current limit operation implemented for power MOSFETs M1 and M2 to protect
them. The implication of this is that any external forcing of voltage/current on SW2 pin or misuse of SW2 pin may
be detrimental to the part and may damage the internal circuits.
DYNAMICALLY ADJUSTABLE OUTPUT VOLTAGE
The LM3209-G3 features dynamically adjustable output voltage which eliminates the need for external feedback
resistors. The output can be set from 0.6V to 4.2V by changing the voltage on the analog VCON pin. This feature
is useful in cell phone RF PA applications where peak power is needed only when the handset is far away from
the base station or when data is being transmitted. In other instances, the transmitting power can be reduced.
Hence the supply voltage to the PA can be reduced, promoting longer battery life. In order to adaptively adjust
the supply voltage to the PA in real time in a cell phone application, the output voltage transition should be fast
enough in order to meet the RF transmit signal specifications. LM3209-G3 offers ultra fast output voltage
transitions without drawing very large currents from the battery supply. With a current limit of 1700 mA (typ.), the
output voltage can transition from 0.8V to 4.0V in less than 20 µs with a load resistance of 11.4Ω.
SEAMLESS MODE TRANSITION
In a typical non-inverting buck-boost converter, all four power switches, M1 through M4, are switched every
cycle. This operation increases MOSFET drive losses and lowers the converter efficiency. The LM3209-G3
switches only two power switches every cycle to improve converter efficiency. Hence it operates either as a buck
converter or a boost converter depending upon the input and output voltage conditions. This creates a boundary
between the buck and boost modes of operation. When the input battery voltage is close to the set output
voltage, the converter automatically switches to four-switch operation seamlessly such that the output voltage
does not see any perturbations at the mode boundary. The excellent mode transition capability of LM3209-G3
enables low noise output with the highest efficiency. Internal feedback loop compensation ensures stable
operation in buck, boost, as well as the buck-boost mode transition region.
VCON OVER-VOLTAGE CLAMP
The LM3209-G3 features an internal clamp on the analog VCON pin voltage to limit the output voltage to a
maximum safe value. The VCON voltage is internally switched to a reference voltage of approximately 1.6V
when the VCON in voltage exceeds 1.6V. This limits the output voltage to approximately 4.8V and protects the
part from over voltage stress.
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
13
LM3209-G3
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
www.ti.com
THERMAL OVERLOAD PROTECTION
The LM3209-G3 has a thermal overload protection function that operates to protect itself from short-term
and over-load conditions. When the junction temperature exceeds approximately 150°C, the device
operation. All power MOSFET switches are turned off. When the temperature drops below 120°C,
operation resumes. Prolonged operation in thermal overload conditions may damage the device
considered bad practice.
14
Submit Documentation Feedback
misuse
inhibits
normal
and is
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
APPLICATION INFORMATION
SETTING THE OUTPUT VOLTAGE
The LM3209-G3 features a pin-controlled variable output voltage which eliminates the need for external feedback
resistors. It can be programmed for an output voltage from 0.6V to 4.2V by setting the voltage on the VCON pin,
as in the following formula.
VOUT = 3 x VCON
(1)
When VCON is between 0.2V and 1.4V, the output voltage will follow this formula.
Internally, VCON is clamped to avoid exceeding the maximum output voltage. When the VCON voltage is greater
than 1.6V, the output voltage is regulated at approximately 4.8V.
OUTPUT CURRENT CAPABILITY
The LM3209-G3 is designed for a maximum load capability of 650 mA when VIN ≥ 3.0V and 500 mA when VIN <
3.0V.
Table 1. Output Voltage vs. Maximum Output Current
VOUT
VIN
Maximum IOUT
4.2V
> 3.0V
650 mA
4.2V
≤ 3.0V
500 mA
3.6V
≥ 3.2V
1000 mA
RECOMMENDED EXTERNAL COMPONENTS
INDUCTOR SELECTION
A 2.2 µH inductor with a saturation current rating over 1900 mA and low inductance drop at the full DC bias
condition is recommended for almost all applications. An inductor with a DC resistance of less than 0.1Ω and
lower ESR should be used to get good efficiency for the entire output current range.
If an inductance with a lower ISAT rating is used in the application, the VCON Transient Response time will be
affected. The rise time of the output voltage will be increased because the inductor will saturate and cannot
charge the output capacitor quickly enough. If a winding type inductor is selected, the efficiency in light load
conditions may be degraded due to higher ESR losses.
Table 2. Suggested Inductors (2.2 µH)
Vendor
Part Number
Dimensions (mm)
ISAT (30%)
IRATING(Δ40°C)
DCR (mΩ)
FDK
MIPSZ2520D2R2 (2.2 µH)
2.5 x 2.0 x 1.0
1.5A
1.1A
110
Murata
LQH2HPN1R0NG0
2.5 x 2.0 x 1.2
1.8A
1.1A
115
Samsung
CIG22H2R2MNE
2.5 x 2.0 x 1.2
1.9A
1.6A
116
INPUT CAPACITOR SELECTION
A ceramic input capacitor of 10 µF, 6.3V or higher is sufficient for most applications. Place the input capacitor as
close as possible to the PVIN and PGND pins of the device. A larger value or higher voltage rating may be used
to improve input filtering. Use X7R, X5R, or B types; do not use Y5V or F. DC bias characteristics of ceramic
capacitors must be considered when selecting case sizes like 0603(1608), 0805(2012), or smaller profile. The
input filter capacitor supplies current to the PMOS switch in the first half of each cycle and reduces the voltage
ripple imposed on the input power source. A ceramic capacitor's low ESR provides the best noise filtering of
input voltage spikes caused by this rapidly changing current.
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
15
LM3209-G3
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
www.ti.com
OUTPUT CAPACITOR SELECTION
Use a 4.7 µF, 6.3V, X7R, X5R, or B types; do not use Y5V or F. DC bias characteristics of ceramic capacitors
must be considered. DC bias characteristics vary from manufacturer to manufacturer, and DC bias curves should
be requested from them as part of the capacitor selection process.
The output filter capacitor smooths out current flow from the inductor to the load, helps maintain a steady output
voltage during transient load changes and reduces output voltage ripple. These capacitors must be selected with
sufficiency capacitance and low ESR to perform these functions. Note that the output voltage ripple is dependent
on the inductor current ripple and the Equivalent Series Resistance of the output capacitor (ESR). The ESR is
frequency dependent (as well as temperature dependent); make sure the value used for calculations is at the
switching frequency of the part.
Table 3. SUGGESTED CAPACITORS
Model
Vendor
C1608X5R0J106K
TDK
ECJ1VB0J475K
Panasonic
GRM188R60J475ME84D
Murata
GRM219R61A475KE19
Murata
10 µF for CIN
4.7 µF for COUT
RECOMMENDED EXTERNAL COMPONENT COMBINATIONS FOR VCON TRANSIENT
Achieving optimum Output Voltage (VCON) Transient is expected to require both an inductor with smaller
inductance degradation and an output capacitor with modest capacitance. FDK MIPSZ2520D2R2 and Panasonic
ECJ1VB0J475K are one sample of the external component combination.
An inductor with a large inductance drop at high DC bias causes slower charging current to the output capacitor.
An output capacitor with less capacitance drop at high voltage will cause a big overshoot. However, an output
capacitor with a large capacitance drop generates bigger output voltage ripple.
DSBGA PACKAGE ASSEMBLY AND USE
Use of the DSBGA package requires specialized board layout, precision mounting and careful re-flow
techniques, as detailed in Texas Instruments Application Note 1112. Refer to the section Surface Mount
Technology (SMD) Assembly Considerations. For best results in assembly, alignment ordinals on the PC board
should be used to facilitate placement of the device. The pad style used with DSBGA package must be the
NSMD (non-solder mask defined) type. This means that the solder-mask opening is larger than the pad size.
This prevents a lip that otherwise forms if the solder-mask and pad overlap, from holding the device off the
surface of the board and interfering with mounting. See Application Note 1112 for specific instructions how to do
this.
The 12-bump package used for LM3209-G3 has 300 micron solder balls and requires 10.82 mil pads for
mounting on the circuit board. The trace to each pad should enter the pad with a 90° entry angle to prevent
debris from being caught in deep corners. Initially, the trace to each pad should be 7 mil wide, for a section
approximately 7 mil long, as a thermal relief. Then each trance should neck up or down to its optimal width. The
important criterion is symmetry. This ensures the solder bumps on the LM3209-G3 re-flow evenly and that the
device solders level to the board. In particular, special attention must be paid to the pads for bumps A3, B3, and
D3. Because PVIN and PGND are typically connected to large copper planes, inadequate thermal relief can
result in late or inadequate re-flow of these bumps.
The DSBGA package is optimized for the smallest possible size in applications with red or infrared opaque
cases. Because the DSBGA package lacks the plastic encapsulation characteristic of larger devices, it is
vulnerable to light. Backside metallization and/or epoxy coating, along with front-side shading by the printed
circuit board, reduce this sensitivity. However, the package has exposed die edges. In particular, DSBGA
devices are sensitive to light (in the red and infrared range) shining on the package's exposed die edges.
16
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
LM3209-G3
www.ti.com
SNVS626B – NOVEMBER 2009 – REVISED MARCH 2013
BOARD LAYOUT CONSIDERATIONS
PC board layout is an important part of DC-DC converter design. Poor board layout can disrupt the performance
of a DC-DC converter and surrounding circuitry by contributing to EMI, ground bounce, and resistive voltage loss
in the traces.
1. Place the LM3209-G3, inductor and filter capacitors close together and make the traces short. The traces
between these components carry relatively high switching currents and act as antennas. Following this rule
reduces radiated noise. Special care must be given to place the input filter capacitor very close to the PVIN
and PGND pins.
2. Connect the ground pins of the LM3209-G3 and filter capacitors together using a generous component-side
copper fill as a pseudo-ground plane. Then connect this to the ground-plane (if one is used) with several
vias. This reduces ground-plane noise by preventing the switching currents from circulating through the
ground plane. It also reduces ground bounce at the LM3209-G3 by giving it a low-impedance ground
connection.
3. Use wide traces between the power components and for power connections to the DC-DC converter circuit.
This reduces voltage errors caused by resistive losses across the traces.
4. Route noise sensitive traces such as the voltage feedback path away from noisy traces between the power
components. The voltage feedback trace must remain close to the LM3209-G3 circuit and should be routed
directly from FB to VOUT at the output capacitor and should be routed opposite to noisy components. This
reduces EMI radiated onto the DC-DC converter's own voltage feedback trace.
5. Place noise sensitive circuitry, such as radio IF blocks, away from the DC-DC converter, CMOS digital
blocks, and other noisy circuitry. Interference with noise-sensitive circuitry in the system can be reduced
through distance.
In mobile phones, for example, a common practice is to place the DC-DC converter on one corner of the board,
arrange the CMOS digital circuitry around it (since this also generates noise), and then place sensitive
preamplifiers and IF stages on the diagonally opposing corner. Often, the sensitive circuitry is shielded with a
metal pan and power to it is post-regulated to reduce conducted noise, using low-dropout linear regulators.
Submit Documentation Feedback
Copyright © 2009–2013, Texas Instruments Incorporated
Product Folder Links: LM3209-G3
17
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM3209TLE-G3/NOPB
ACTIVE
DSBGA
YZR
12
250
RoHS & Green
SNAGCU
Level-1-260C-UNLIM
-30 to 85
09G3
LM3209TLX-G3/NOPB
ACTIVE
DSBGA
YZR
12
3000
RoHS & Green
SNAGCU
Level-1-260C-UNLIM
-30 to 85
09G3
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
工商网监
湘ICP备2023018690号
