LM3224
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SNVS277C – DECEMBER 2004 – REVISED MARCH 2013
LM3224 615kHz/1.25MHz Step-up PWM DC/DC Converter
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FEATURES
DESCRIPTION
•
•
The LM3224 is a step-up DC/DC converter with a
0.15Ω (typ.), 2.45A (typ.) internal switch and pin
selectable operating frequency. With the ability to
convert 3.3V to multiple outputs of 8V, -8V, and 23V,
the LM3224 is an ideal part for biasing TFT displays.
With the high current switch it is also ideal for driving
high current white LEDs for flash applications. The
LM3224 can be operated at switching frequencies of
615kHz and 1.25MHz allowing for easy filtering and
low noise. An external compensation pin gives the
user flexibility in setting frequency compensation,
which makes possible the use of small, low ESR
ceramic capacitors at the output. An external softstart pin allows the user to control the amount of
inrush current during start up. The LM3224 is
available in a low profile 8-lead VSSOP package.
1
2
•
•
•
Operating Voltage Range of 2.7V to 7V
615kHz/1.25MHz Pin Selectable Frequency
Operation
Over Temperature Protection
Optional Soft-Start Function
8-Lead VSSOP Package
APPLICATIONS
•
•
•
•
•
•
TFT Bias Supplies
Handheld Devices
Portable Applications
GSM/CDMA Phones
Digital Cameras
White LED Flash/Torch Applications
Typical Application Circuit
VIN
L
D
5
6
3
8
Battery or
Power Source
CIN
SW
VIN
FSLCT
FB
VC
2
VOUT
GND
4
1
RC
Optional
RFB1
LM3224
SHDN
SS
7
RFB2
COUT
CSS
CC
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM3224
SNVS277C – DECEMBER 2004 – REVISED MARCH 2013
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Connection Diagram
1
8
VC
SS
FB
FSLCT
7
2
3
6
SHDN
VIN
GND
SW
4
5
Figure 1. 8-Lead Plastic VSSOP
Top View
Package Number DGK0008A
PIN DESCRIPTIONS
Pin
2
Name
Function
1
VC
Compensation network connection. Connected to the output of the voltage error amplifier.
2
FB
Output voltage feedback input.
3
SHDN
4
GND
Analog and power ground.
5
SW
Power switch input. Switch connected between SW pin and GND pin.
6
VIN
Analog power input.
7
FSLCT
8
SS
Shutdown control input, active low. This pin has an internal pulldown resistor so the default condition
is off. The pin must be pulled high to turn on the device.
Switching frequency select input. VIN = 1.25MHz. Ground = 615kHz.
Soft-start Pin.
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Block Diagram
FSLCT
¦
SS
Duty
Cycle Limit
Oscillator
Load Current
Measurement
SW
+
PWM
COMP
-
-
FB
BG
Set
Reset
Drive
Reset
Driver
LOGIC
ERROR
AMP
UVP
OVP
+
BG
Thermal
SD
OVP
COMP
+
BG
+
Thermal
Shutdown
Bandgap Voltage
Reference
VC
Shutdown
Comparator
SHDN
UVP
COMP
VIN
GND
General Description
The LM3224 utilizes a PWM control scheme to regulate the output voltage over all load conditions. The operation
can best be understood referring to the block diagram and Figure 21 of the Operation section. At the start of
each cycle, the oscillator sets the driver logic and turns on the NMOS power device conducting current through
the inductor, cycle 1 of Figure 21 (a). During this cycle, the voltage at the VC pin controls the peak inductor
current. The VC voltage will increase with larger loads and decrease with smaller. This voltage is compared with
the summation of the SW voltage and the ramp compensation. The ramp compensation is used in PWM
architectures to eliminate the sub-harmonic oscillations that occur during duty cycles greater than 50%. Once the
summation of the ramp compensation and switch voltage equals the VC voltage, the PWM comparator resets the
driver logic turning off the NMOS power device. The inductor current then flows through the schottky diode to the
load and output capacitor, cycle 2 of Figure 21 (b). The NMOS power device is then set by the oscillator at the
end of the period and current flows through the NMOS power device once again.
The LM3224 has dedicated protection circuitry running during normal operation to protect the IC. The Thermal
Shutdown circuitry turns off the NMOS power device when the die temperature reaches excessive levels. The
UVP comparator protects the NMOS power device during supply power startup and shutdown to prevent
operation at voltages less than the minimum input voltage. The OVP comparator is used to prevent the output
voltage from rising at no loads allowing full PWM operation over all load conditions. The LM3224 also features a
shutdown mode decreasing the supply current to 0.1µA (typ.).
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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LM3224
SNVS277C – DECEMBER 2004 – REVISED MARCH 2013
Absolute Maximum Ratings (1)
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(2) (2)
VIN
7.5V
SW Voltage
21V
FB Voltage
(3)
7V
VC Voltage
(4)
1.26V ± 0.3V
SHDN Voltage
7.5V
FSLCT
7.5V
Maximum Junction Temperature
150°C
Power Dissipation (5)
Internally Limited
Lead Temperature
300°C
Vapor Phase (60 sec.)
215°C
Infrared (15 sec.)
ESD Susceptibility
(6)
220°C
Human Body Model
Machine Model
(1)
(2)
(3)
(4)
(5)
(6)
2kV
200V
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications
Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the
device is intended to be functional, but device parameter specifications may not be ensured. For ensured specifications and test
conditions, see the Electrical Characteristics().
The FB pin should never exceed VIN.
Under normal operation the VC pin may go to voltages above this value. This maximum rating is for the possibility of a voltage being
applied to the pin, however the VC pin should never have a voltage directly applied to it.
The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature is calculated
using: PD (MAX) = (TJ(MAX) − TA)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and
the regulator will go into thermal shutdown.
The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin. The machine model is a 200pF
capacitor discharged directly into each pin.
Operating Conditions
Operating Junction Temperature Range
−40°C to +125°C
Storage Temperature
−65°C to +150°C
Supply Voltage
2.7V to 7V
Maximum Output Voltage
4
20V
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Electrical Characteristics (1)
Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating
Temperature Range ( TJ = −40°C to +125°C). VIN = 2.7V, FSLCT = SHDN = VIN, and IL = 0A, unless otherwise specified.
Symbol
IQ
Typ
Max
(1)
Units
FB = 2V (Not Switching)
1.3
2.0
mA
VSHDN = 0V
0.1
2.0
µA
1.2285
1.26
1.2915
V
1.9
2.45
2.8
Parameter
Quiescent Current
VFB
Feedback Voltage
ICL (3)
Switch Current Limit
Min
Conditions
VIN = 2.7V
(1)
(4)
VIN = 3V, VOUT = 8V
2.1
VIN = 3V, VOUT = 5V
2.2
2.7V ≤ VIN ≤ 7V
%VFB/ΔVIN
Feedback Voltage Line
Regulation
IB
FB Pin Bias Current
ISS
SS Pin Current
7.5
VSS
SS Pin Voltage
1.2090
VIN
Input Voltage Range
gm
Error Amp Transconductance
AV
Error Amp Voltage Gain
DMAX
Maximum Duty Cycle
fS
Switching Frequency
ISHDN
(5) (6)
Shutdown Pin Current
(2)
0.085
0.15
35
250
nA
11
13
µA
1.2430
1.2622
7
V
87
135
µmho
2.7
ΔI = 5µA
40
A
78
%/V
V/V
85
92.5
FSLCT = Ground
450
615
750
kHz
%
FSLCT = VIN
0.9
1.25
1.5
MHz
VSHDN = 2.7V
2.4
5.0
µA
VSHDN = 0.3V
0.3
1.2
IL
Switch Leakage Current
VSW = 20V
0.2
8.0
µA
RDSON
Switch RDSON
VIN = 2.7V, ISW = 1A
0.15
0.4
Ω
ThSHDN
Shutdown Threshold
Output High
1.2
Output Low
UVP
On Threshold
(1)
(2)
(3)
(4)
(5)
(6)
0.8
2.3
Off Threshold
0.8
V
0.3
2.5
2.6
V
V
2.7
V
All limits ensured at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are
100% production tested. All limits at temperature extremes are ensured via correlation using standard Statistical Quality Control (SQC)
methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
Typical numbers are at 25°C and represent the most likely norm.
Duty cycle affects current limit due to ramp generator.
Current limit at 0% duty cycle. See Typical Performance Characteristics for Switch Current Limit vs. VIN
Bias current flows into FB pin.
The FB pin should never exceed VIN.
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Typical Performance Characteristics
SHDN Pin Current vs. SHDN Pin Voltage
SS Pin Current vs. Temperature
7
11.8
11.6
VIN = 7.0V
TJ = -40oC
SS PIN CURRENT (PA)
SHDN PIN CURRENT (PA)
6
5
TJ
4
=2
o
5C
3
2
o
TJ = 125 C
1
11.4
11.2
11.0
10.8
VIN = 2.7V
10.6
10.4
0
0.0
1.0
2.0
4.0
3.0
5.0
6.0
10.2
-40 -25 -10 5
7.0
SHDN PIN VOLTAGE (V)
TEMPERATURE (oC)
Figure 2.
Figure 3.
FSLCT Pin Current vs. FSLCT Pin Voltage
FB Pin Current vs. Temperature
70
8
FSLCT = VIN
FB = 1.265V
60
TJ = -40oC
5
FB PIN CURRENT (nA)
FSLCT PIN CURRENT (PA)
7
6
o
TJ =
25 C
4
3
TJ = 125oC
2
50
40
VIN = 2.7V
30
VIN = 7.0V
20
10
1
0
2.6 3.0 3.4 3.8 4.2 4.6 5.0 5.4 5.8 6.2 6.6 7.0
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
FSLCT VOLTAGE (V)
TEMPERATURE (oC)
Figure 4.
Figure 5.
NMOS RDSON vs. Input Voltage
0.23
615kHz Non-switching IQ vs. Input Voltage
1.8
ISW = 1.5A
0.21
NON-SWITCHING IQ (mA)
1.7
NMOS RDSON (:)
0.19
0.17
o
TA = 85 C
0.15
0.13
0.11
TA = 25oC
0.09
0.07
TA = -40oC
1.6
TJ = 125oC
1.5
1.4
TJ = 25oC
TJ = -40oC
1.3
1.2
1.1
0.05
2.6 3.0 3.4 3.8 4.2 4.6 5.0 5.4 5.8 6.2 6.6 7.0
INPUT VOLTAGE (V)
1.0
2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0
INPUT VOLTAGE (V)
Figure 6.
6
20 35 50 65 80 95 110 125
Figure 7.
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Typical Performance Characteristics (continued)
1.25MHz Non-switching IQ vs. Input Voltage
615kHz Switching IQ vs. Input Voltage
2.0
4.5
1.7
1.6
1.5
TJ = -40oC
4.0
1.8
SWITCHING IQ (mA)
NON-SWITCHING IQ (mA)
1.9
TJ = 125oC
TJ = 25oC
1.4
TJ = -40oC
1.3
3.5
TJ = 25oC
3.0
TJ = 125oC
2.5
2.0
1.2
1.5
1.1
1.0
2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0
1.0
2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 8.
Figure 9.
1.25MHz Switching IQ vs. Input Voltage
8.0
615kHz Switching IQ vs. Temperature
4.50
o
TJ = -40 C
4.25
7.0
VIN = 7.0V
5.0
4.0
SWITCHING IQ (mA)
SWITCHING IQ (mA)
4.00
6.0
TJ = 25oC
TJ = 125oC
3.0
2.0
3.75
3.50
3.25
3.00
2.75
2.50
2.25
0.0
2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0
2.00
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
INPUT VOLTAGE (V)
Figure 10.
Figure 11.
1.25MHz Switching IQ vs. Temperature
615kHz Switching Frequency vs. Temperature
7.5
640
SWITCHING FREQUENCY (kHz)
7.0
VIN = 7.0V
6.5
SWITCHING IQ (mA)
VIN = 2.7V
1.0
6.0
5.5
5.0
4.5
4.0
3.5
VIN = 2.7V
3.0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
630
620
VIN = 7.0V
610
600
VIN = 2.7V
590
580
570
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
Figure 12.
Figure 13.
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Typical Performance Characteristics (continued)
1.25MHz Switching Frequency vs. Temperature
615kHz Maximum Duty Cycle vs. Temperature
94.0
VIN = 7.0V
1.32
VIN = 7.0V
MAXIMUM DUTY CYCLE (%)
SWITCHING FREQUENCY (MHz)
1.34
1.30
1.28
1.26
1.24
VIN = 2.7V
1.22
1.20
93.5
93.0
92.5
VIN = 2.7V
92.0
1.18
1.16
-40 -25 -10 5 20 35 50 65 80 95 110 125
91.5
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 14.
Figure 15.
1.25MHz Maximum Duty Cycle vs. Temperature
Switch Current Limit vs. VIN
2.3
94.0
2.2
VIN = 7.0V
93.0
2.1
CURRENT LIMIT (A)
MAXIMUM DUTY CYCLE (%)
93.5
92.5
92.0
91.5
VIN = 2.7V
VOUT = 8V
2.0
1.9
VOUT = 12V
1.8
91.0
1.7
90.5
1.6
VOUT = 15V
90.0
-40 -25 -10 5
1.5
2.6 3.0 3.4 3.8 4.2 4.6 5.0 5.4 5.8
20 35 50 65 80 95 110 125
INPUT VOLTAGE (V)
TEMPERATURE (oC)
Figure 16.
Figure 17.
Switch Current Limit vs. Temperature
Switch Current Limit vs. Temperature
2.20
2.10
VOUT = 8V
VIN = 3.0V
2.00
2.10
VIN = 3.0V
CURRENT LIMIT (A)
CURRENT LIMIT (A)
VOUT = 15V
VIN = 5.5V
2.15
2.05
2.00
VIN = 4.2V
1.95
1.90
1.90
VIN = 4.2V
1.80
1.70
VIN = 5.5V
1.60
1.85
1.50
1.80
1.75
-40
-20
-30
0
-10
20
10
40
30
60
50
1.40
-40
80
70
90
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 18.
8
-20
0
20
40
60
80
-30 -10
10
30
50
70
90
Figure 19.
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Typical Performance Characteristics (continued)
1.25MHz Efficiency vs. Load Current
100
90
EFFICIENCY (%)
80
V OUT =8V
V IN = 2.7V
VIN =5.5V
70
V IN = 3.3V
60
VIN = 4. 2V
50
40
30
20
10
0
0. 1
1
10
100
1000
10000
LOAD CURRENT (mA)
Figure 20.
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OPERATION
L
D
COUT
VIN
RLOAD
PWM
L
X
+
+
L
COUT
VIN
R LOAD
V IN
COUT
R LOAD
V OUT
V OUT
-
-
Cycle 1
(a)
Cycle 2
(b)
Figure 21. Simplified Boost Converter Diagram
(a) First Cycle of Operation (b) Second Cycle Of Operation
CONTINUOUS CONDUCTION MODE
The LM3224 is a current-mode, PWM boost regulator. A boost regulator steps the input voltage up to a higher
output voltage. In continuous conduction mode (when the inductor current never reaches zero at steady state),
the boost regulator operates in two cycles.
In the first cycle of operation, shown in Figure 21 (a), the transistor is closed and the diode is reverse biased.
Energy is collected in the inductor and the load current is supplied by COUT.
The second cycle is shown in Figure 21 (b). During this cycle, the transistor is open and the diode is forward
biased. The energy stored in the inductor is transferred to the load and output capacitor.
The ratio of these two cycles determines the output voltage. The output voltage is defined approximately as:
VOUT =
VIN
1-D
, D' = (1-D) =
VIN
VOUT
where
•
D is the duty cycle of the switch
(1)
D and D′ will be required for design calculations.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the feedback pin and a resistor divider connected to the output as shown in the
typical operating circuit. The feedback pin voltage is 1.26V, so the ratio of the feedback resistors sets the output
voltage according to the following equation:
VOUT - 1.26
:
RFB1 = RFB2 x
1.26
(2)
10
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SOFT-START CAPACITOR
The LM3224 has a soft-start pin that can be used to limit the inductor inrush current on start-up. The external SS
pin is used to tailor the soft-start for a specific application but is not required for all applications and can be left
open when not needed. When used, a current source charges the external soft-start capacitor, Css. The softstart time can be estimated as:
Tss = Css*1.24V/Iss
(3)
THERMAL SHUTDOWN
The LM3224 includes thermal shutdown protection. If the die temperature exceeds 140°C the regulator will shut
off the power switch, significantly reducing power dissipation in the device. The switch will remain off until the die
temperature is reduced to approximately 120°C. If the cause of the excess heating is not removed (excessive
ambient temperature, excessive power dissipation, or both) the device will continue to cycle on and off in this
manner to protect from damage.
INTRODUCTION TO COMPENSATION
IL (A)
VIN VOUT
L
VIN
L
'i L
IL_AVG
t (s)
D*Ts
Ts
(a)
ID (A)
VIN VOUT
L
ID_AVG
=IOUT_AVG
t (s)
D*Ts
Ts
(b)
Figure 22. (a) Inductor current. (b) Diode current.
The LM3224 is a current mode PWM boost converter. The signal flow of this control scheme has two feedback
loops, one that senses switch current and one that senses output voltage.
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To keep a current programmed control converter stable above duty cycles of 50%, the inductor must meet
certain criteria. The inductor, along with input and output voltage, will determine the slope of the current through
the inductor (see Figure 22 (a)). If the slope of the inductor current is too great, the circuit will be unstable above
duty cycles of 50%. A 10µH to 15µH inductor is recommended for most 615 kHz applications, while a 4.7µH to
10µH inductor may be used for most 1.25 MHz applications. If the duty cycle is approaching the maximum of
85%, it may be necessary to increase the inductance by as much as 2X. See Inductor and Diode Selection for
more detailed inductor sizing.
The LM3224 provides a compensation pin (VC) to customize the voltage loop feedback. It is recommended that a
series combination of RC and CC be used for the compensation network, as shown in the typical application
circuit. For any given application, there exists a unique combination of RC and CC that will optimize the
performance of the LM3224 circuit in terms of its transient response. The series combination of RC and CC
introduces a pole-zero pair according to the following equations:
fZC =
1
Hz
2SRCCC
(4)
1
fPC =
Hz
2S(RC + RO)CC
where
•
RO is the output impedance of the error amplifier (approximately 900kΩ)
(5)
For most applications, performance can be optimized by choosing values within the range 5kΩ ≤ RC ≤ 100kΩ (RC
can be up to 200kΩ if CC2 is used, see High Output Capacitor ESR Compensation) and 680pF ≤ CC ≤ 10nF.
Refer to the Applications Information section for recommended values for specific circuits and conditions. Refer
to the Compensation section for other design requirement.
COMPENSATION
This section will present a general design procedure to help insure a stable and operational circuit. The designs
in this datasheet are optimized for particular requirements. If different conversions are required, some of the
components may need to be changed to ensure stability. Below is a set of general guidelines in designing a
stable circuit for continuous conduction operation, in most all cases this will provide for stability during
discontinuous operation as well. The power components and their effects will be determined first, then the
compensation components will be chosen to produce stability.
INDUCTOR AND DIODE SELECTION
Although the inductor sizes mentioned earlier are fine for most applications, a more exact value can be
calculated. To ensure stability at duty cycles above 50%, the inductor must have some minimum value
determined by the minimum input voltage and the maximum output voltage. This equation is:
L>
VINRDSON
0.144 fs
D -1
( D'
) (in H)
where
•
•
•
fs is the switching frequency
D is the duty cycl
RDSON is the ON resistance of the internal switch taken from the graph "NMOS RDSON vs. Input Voltage" in the
Typical Performance Characteristics section.
(6)
This equation is only good for duty cycles greater than 50% (D>0.5), for duty cycles less than 50% the
recommended values may be used. The corresponding inductor current ripple as shown in Figure 22 (a) is given
by:
VIND
(in Amps)
'iL =
2Lfs
(7)
12
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The inductor ripple current is important for a few reasons. One reason is because the peak switch current will be
the average inductor current (input current or ILOAD/D') plus ΔiL. As a side note, discontinuous operation occurs
when the inductor current falls to zero during a switching cycle, or ΔiL is greater than the average inductor
current. Therefore, continuous conduction mode occurs when ΔiL is less than the average inductor current. Care
must be taken to make sure that the switch will not reach its current limit during normal operation. The inductor
must also be sized accordingly. It should have a saturation current rating higher than the peak inductor current
expected. The output voltage ripple is also affected by the total ripple current.
The output diode for a boost regulator must be chosen correctly depending on the output voltage and the output
current. The typical current waveform for the diode in continuous conduction mode is shown in Figure 22 (b). The
diode must be rated for a reverse voltage equal to or greater than the output voltage used. The average current
rating must be greater than the maximum load current expected, and the peak current rating must be greater
than the peak inductor current. During short circuit testing, or if short circuit conditions are possible in the
application, the diode current rating must exceed the switch current limit. Using Schottky diodes with lower
forward voltage drop will decrease power dissipation and increase efficiency.
DC GAIN AND OPEN-LOOP GAIN
Since the control stage of the converter forms a complete feedback loop with the power components, it forms a
closed-loop system that must be stabilized to avoid positive feedback and instability. A value for open-loop DC
gain will be required, from which you can calculate, or place, poles and zeros to determine the crossover
frequency and the phase margin. A high phase margin (greater than 45°) is desired for the best stability and
transient response. For the purpose of stabilizing the LM3224, choosing a crossover point well below where the
right half plane zero is located will ensure sufficient phase margin.
To ensure a bandwidth of ½ or less of the frequency of the RHP zero, calculate the open-loop DC gain, ADC.
After this value is known, you can calculate the crossover visually by placing a −20dB/decade slope at each pole,
and a +20dB/decade slope for each zero. The point at which the gain plot crosses unity gain, or 0dB, is the
crossover frequency. If the crossover frequency is less than ½ the RHP zero, the phase margin should be high
enough for stability. The phase margin can also be improved by adding CC2 as discussed later in this section.
The equation for ADC is given below with additional equations required for the calculation:
gmROD'
RFB2
{[(ZcLeff)// RL]//RL} (in dB)
ADC(DB) = 20log10
RFB1 + RFB2 RDSON
)
(
(8)
2fs
Zc #
nD'
Leff =
n = 1+
(in rad/s)
(9)
L
(D')2
(10)
2mc
(no unit)
m1
(11)
(12)
mc ≊ 0.072fs (in V/s)
m1 #
VINRDSON
L
(in V/s)
where
•
•
•
•
RL is the minimum load resistance
VIN is the minimum input voltage
gm is the error amplifier transconductance found in the Electrical Characteristics table
RDSON is the value chosen from the graph "NMOS RDSON vs. Input Voltage" in the Typical Performance
Characteristics section
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INPUT AND OUTPUT CAPACITOR SELECTION
The switching action of a boost regulator causes a triangular voltage waveform at the input. A capacitor is
required to reduce the input ripple and noise for proper operation of the regulator. The size used is dependant on
the application and board layout. If the regulator will be loaded uniformly, with very little load changes, and at
lower current outputs, the input capacitor size can often be reduced. The size can also be reduced if the input of
the regulator is very close to the source output. The size will generally need to be larger for applications where
the regulator is supplying nearly the maximum rated output or if large load steps are expected. A minimum value
of 10µF should be used for the less stressful condtions while a 22µF to 47µF capacitor may be required for
higher power and dynamic loads. Larger values and/or lower ESR may be needed if the application requires very
low ripple on the input source voltage.
The choice of output capacitors is also somewhat arbitrary and depends on the design requirements for output
voltage ripple. It is recommended that low ESR (Equivalent Series Resistance, denoted RESR) capacitors be used
such as ceramic, polymer electrolytic, or low ESR tantalum. Higher ESR capacitors may be used but will require
more compensation which will be explained later on in the section. The ESR is also important because it
determines the peak to peak output voltage ripple according to the approximate equation:
ΔVOUT ≊ 2ΔiLRESR (in Volts)
(14)
A minimum value of 10µF is recommended and may be increased to a larger value. After choosing the output
capacitor you can determine a pole-zero pair introduced into the control loop by the following equations:
fP1 =
fZ1 =
1
(in Hz)
2S(RESR + RL)COUT
(15)
1
(in Hz)
2SRESRCOUT
where
•
RL is the minimum load resistance corresponding to the maximum load current
(16)
The zero created by the ESR of the output capacitor is generally very high frequency if the ESR is small. If low
ESR capacitors are used it can be neglected. If higher ESR capacitors are used see the High Output Capacitor
ESR Compensation section. Some suitable capacitor vendors include Vishay, Taiyo-Yuden, and TDK.
RIGHT HALF PLANE ZERO
A current mode control boost regulator has an inherent right half plane zero (RHP zero). This zero has the effect
of a zero in the gain plot, causing an imposed +20dB/decade on the rolloff, but has the effect of a pole in the
phase, subtracting another 90° in the phase plot. This can cause undesirable effects if the control loop is
influenced by this zero. To ensure the RHP zero does not cause instability issues, the control loop should be
designed to have a bandwidth of less than ½ the frequency of the RHP zero. This zero occurs at a frequency of:
VOUT(D')2
(in Hz)
RHPzero =
2S,LOADL
where
•
ILOAD is the maximum load current.
(17)
SELECTING THE COMPENSATION COMPONENTS
The first step in selecting the compensation components RC and CC is to set a dominant low frequency pole in
the control loop. Simply choose values for RC and CC within the ranges given in the Introduction to
Compensation section to set this pole in the area of 10Hz to 500Hz. The frequency of the pole created is
determined by the equation:
fPC =
1
(in Hz)
2S(RC + RO)CC
where
•
14
RO is the output impedance of the error amplifier, approximately 900kΩ
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Since RC is generally much less than RO, it does not have much effect on the above equation and can be
neglected until a value is chosen to set the zero fZC. fZC is created to cancel out the pole created by the output
capacitor, fP1. The output capacitor pole will shift with different load currents as shown by the equation, so setting
the zero is not exact. Determine the range of fP1 over the expected loads and then set the zero fZC to a point
approximately in the middle. The frequency of this zero is determined by:
fZC =
1
(in Hz)
2SCCRC
(19)
Now RC can be chosen with the selected value for CC. Check to make sure that the pole fPC is still in the 10Hz to
500Hz range, change each value slightly if needed to ensure both component values are in the recommended
range.
HIGH OUTPUT CAPACITOR ESR COMPENSATION
When using an output capacitor with a high ESR value, or just to improve the overall phase margin of the control
loop, another pole may be introduced to cancel the zero created by the ESR. This is accomplished by adding
another capacitor, CC2, directly from the compensation pin VC to ground, in parallel with the series combination of
RC and CC. The pole should be placed at the same frequency as fZ1, the ESR zero. The equation for this pole
follows:
1
(in Hz)
fPC2 =
2SCC2(RC //RO)
(20)
To ensure this equation is valid, and that CC2 can be used without negatively impacting the effects of RC and CC,
fPC2 must be greater than 10fZC.
CHECKING THE DESIGN
With all the poles and zeros calculated the crossover frequency can be checked as described in the section DC
Gain and Open-loop Gain. The compensation values can be changed a little more to optimize performance if
desired. This is best done in the lab on a bench, checking the load step response with different values until the
ringing and overshoot on the output voltage at the edge of the load steps is minimal. This should produce a
stable, high performance circuit. For improved transient response, higher values of RC should be chosen. This
will improve the overall bandwidth which makes the regulator respond more quickly to transients. If more detail is
required, or the most optimum performance is desired, refer to a more in depth discussion of compensating
current mode DC/DC switching regulators.
POWER DISSIPATION
The output power of the LM3224 is limited by its maximum power dissipation. The maximum power dissipation is
determined by the formula
PD = (Tjmax - TA)/θJA
where
•
•
•
Tjmax is the maximum specidfied junction temperature (125°C)
TA is the ambient temperature
θJA is the thermal resistance of the package
(21)
LAYOUT CONSIDERATIONS
The input bypass capacitor CIN, as shown in the typical operating circuit, must be placed close to the IC. This will
reduce copper trace resistance which effects input voltage ripple of the IC. For additional input voltage filtering, a
100nF bypass capacitor can be placed in parallel with CIN, close to the VIN pin, to shunt any high frequency noise
to ground. The output capacitor, COUT, should also be placed close to the IC. Any copper trace connections for
the COUT capacitor can increase the series resistance, which directly effects output voltage ripple. The feedback
network, resistors RFB1 and RFB2, should be kept close to the FB pin, and away from the inductor, to minimize
copper trace connections that can inject noise into the system. Trace connections made to the inductor and
schottky diode should be minimized to reduce power dissipation and increase overall efficiency. For more detail
on switching power supply layout considerations see Application Note Layout Guidelines for Switching Power
Supplies (SNVA021).
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APPLICATION INFORMATION
D5
D4
L
10 PH
23V
D7
C6
1 PF
C5
1 PF
C4
1 PF
VIN = 2.7V - 5.5V
D6
C7
1 PF
C1
4.7 PF
D1
D3
-8V
C2
D2 4.7 PF
5
SW
7
6
FSLCT
VIN
3
RFB1
LM3224
SHDN
8V
160k
8
SS
VC
GND
4
1
CIN
22 PF
RC
30k
CSS
RFB2
COUT1
COUT2
30k
10 PF
10 PF
CC2
68 pF
CC
1 nF
Figure 23. Triple Output TFT Bias (615 kHz operation)
TRIPLE OUTPUT TFT BIAS
The circuit in Figure 23 shows how the LM3224 can be configured to provide outputs of 8V, −8V, and 23V,
convenient for biasing TFT displays. The 8V output is regulated, while the −8V and 23V outputs are unregulated.
The 8V output is generated by a typical boost topology. The basic operation of the boost converter is described
in the OPERATION section. The output voltage is set with RFB1 and RFB2 by:
RFB1 = RFB2
VOUT - 1.26
1.26
:
(22)
The compensation network of RC and CC are chosen to optimally stabilize the converter. The inductor also
affects the stability. When operating at 615 kHz, a 10uH inductor is recommended to insure the converter is
stable at duty cycles greater than 50%. Refer to the COMPENSATION section for more information.
The -8V output is derived from a diode inverter. During the second cycle, when the transistor is open, D2
conducts and C1 charges to 8V minus a diode drop (≊0.4V if using a Schottky). When the transistor opens in the
first cycle, D3 conducts and C1's polarity is reversed with respect to the output at C2, producing -8V.
16
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The 23V output is realized with a series of capacitor charge pumps. It consists of four stages: the first stage
includes C4, D4, and the LM3224 switch; the second stage uses C5, D5, and D1; the third stage includes C6,
D6, and the LM3224 switch; the final stage is C7 and D7. In the first stage, C4 charges to 8V when the LM3224
switch is closed, which causes D5 to conduct when the switch is open. In the second stage, the voltage across
C5 is VC4 + VD1 - VD5 = VC4 ≊ 8V when the switch is open. However, because C5 is referenced to the 8V
output, the voltage at C5 is 16V when referenced to ground. In the third stage, the 16V at C5 appears across C6
when the switch is closed. When the switch opens, C6 is referenced to the 8V output minus a diode drop, which
raises the voltage at C6 with respect to ground to about 24V. Hence, in the fourth stage, C7 is charged to 24V
when the switch is open. From the first stage to the last, there are three diode drops that make the output voltage
closer to 24 - 3xVDIODE (about 22.8V if a 0.4V forward drop is assumed).
L
4.7 PH
2.9V - 4.2V
D
5
6
3
8
Battery or
Power Source
SW
FSLCT
VIN
SHDN
FB
VC
22 PF
High Current
White LED
LM3224
SS
CIN
7
2
GND
1
4
RSET
1.8:
(700 mA)
RC
COUT
10 PF
ceramic
2k
Pull high for FLASH or
constant full current,
PWM for TORCH or
partial current.
Optional
Disconnect
FET
CC
2.2 nF
Figure 24. PWM White LED Flash/Torch Driver
L
4.7 PH
2.9V - 4.2V
D
5
6
3
8
Battery or
Power Source
SW
FSLCT
VIN
SHDN
22 PF
FB
SS
2
GND
1
4
RC
2k
Pull high for TORCH.
Pull Flash Enable high
for FLASH.
High Current
White LED
LM3224
VC
CIN
7
CC
2.2 nF
RTORCH
RFLASH
6.2:
(200 mA)
2.55:
(500 mA)
Torch
Enable
Flash
Enable
COUT
10 PF
ceramic
Figure 25. Continuously Operating White LED Flash/Torch Driver
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LM3224
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The LM3224 can be configured to drive high current white LEDs for the flash and torch functions of a digital
camera, camera phone, or any other similar light source. The flash/torch can be set up with the circuit in
Figure 24 by using the resistor RSET to determine the amount of current that will flow through the LED using the
equation:
ILED = VFB/RSET
(23)
If the flash and torch modes will both be used the resistor RSET can be chosen for the higher current flash value.
To flash the circuit pull the SHDN high for the time duration needed for the flash. To enable a lower current torch
mode a PWM signal can be applied to the SHDN pin. The torch current would then be approximately the percent
ON time of the PWM signal multiplied by the flash (or maximum) current. The optional disconnect FET can be
used to eliminate leakage current through the LEDs when the part is off and also to disconnect the LED when
the input voltage exceeds the forward voltage drop of the LED. The maximum output current the LM3224 can
supply in this configuration is shown in Table 1.
Figure 25 is another method of driving a high current white LED. This circuit has a higher component count but
allows the switcher to remain on continuously for torch mode reducing stress on the supply. The two FETs also
double for a disconnect function as described above. In this circuit the device and the torch enable FET are
turned on setting a lower current through the LED. When flash is needed the flash enable FET is turned on to
increase the current for the amount of time desired. The minimum ensured maximum output current for this
circuit is the same as for Figure 24.
Table 1. Maximum LED Drive current
(FSW=1.25MHz, L=4.7µH, LED VFMAX=4V (VOUT=5.26V)
VIN
LED Drive Current (mA)
4.2
1077
4.1
1047
4.0
1017
3.9
987
3.8
958
3.7
929
3.6
900
3.5
871
3.4
842
3.3
814
3.2
785
3.1
757
3.0
729
2.9
701
2.8
673
2.7
646
Table 2. Some Recommended Inductors (Others May Be Used)
18
Manufacturer
Inductor
Contact Information
Coilcraft
DO3316 and DT3316 series
www.coilcraft.com
800-3222645
TDK
SLF10145 series
www.component.tdk.com
847-803-6100
Pulse
P0751 and P0762 series
www.pulseeng.com
Sumida
CDRH8D28 and CDRH8D43 series
www.sumida.com
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SNVS277C – DECEMBER 2004 – REVISED MARCH 2013
Table 3. Some Recommended Input And Output Capacitors (Others May Be
Used)
Manufacturer
Capacitor
Contact Information
Vishay Sprague
293D, 592D, and 595D series tantalum
www.vishay.com
407-324-4140
Taiyo Yuden
High capacitance MLCC ceramic
www.t-yuden.com
408-573-4150
Cornell Dubilier
ESRD seriec Polymer Aluminum Electrolytic
SPV and AFK series V-chip series
www.cde.com
MuRata
High capacitance MLCC ceramic
www.murata.com
L
4.7 PH
2.7V - 4.2V
5
6
3
8
Battery or
Power Source
5V, 650 mA
D
20.5k
SW
FSLCT
VIN
SHDN
LM3224
FB
SS
VC
CIN
22 PF
7
2
GND
1
4
COUT
RC
10 PF
ceramic
20k
6.98k
CC
1 nF
Figure 26. 1.25MHz, 5V Output
L
10 PH
2.9V - 4.2V
5
6
3
8
Battery or
Power Source
8V, 500mA
D
160k
SW
FSLCT
VIN
SHDN
SS
CIN
22 PF
7
LM3224
FB
VC
2
GND
1
4
COUT
RC
30k
30k
CC2
22 PF
ceramic
68 pF
CC
1 nF
Figure 27. 1.25MHz, 8V Output
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L
10 PH
2.9V - 4.2V
5
6
3
8
Battery or
Power Source
12V, 300mA
D
332k
SW
FSLCT
VIN
SHDN
SS
CIN
LM3224
FB
VC
22 PF
7
2
GND
1
4
COUT
RC
22 PF
ceramic
40.2k
39.2k
CC
1.5 nF
Figure 28. 1.25MHz, 12V Output
L
15 PH
2.9V - 4.2V
5
6
3
8
Battery or
Power Source
CIN
22 PF
15V, 220mA
D
301k
SW
FSLCT
VIN
SHDN
SS
7
LM3224
FB
VC
2
GND
1
4
COUT
RC
40.2k
27.4k
22 PF
ceramic
CC
1.5 nF
Figure 29. 1.25MHz, 15V Output
20
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SNVS277C – DECEMBER 2004 – REVISED MARCH 2013
REVISION HISTORY
Changes from Revision B (March 2013) to Revision C
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 20
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM3224MM-ADJ/NOPB
ACTIVE
VSSOP
DGK
8
1000
RoHS & Green
SN
Level-1-260C-UNLIM
LM3224MMX-ADJ/NOPB
ACTIVE
VSSOP
DGK
8
3500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SEKB
SEKB
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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