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LM3410, LM3410-Q1
SNVS541H – OCTOBER 2007 – REVISED AUGUST 2016
LM3410, LM3410-Q1 525-kHz and 1.6-MHz, Constant-Current Boost and SEPIC LED Driver
With Internal Compensation
1 Features
3 Description
•
•
The LM3410 and LM3410-Q1 constant current LED
driver are a monolithic, high frequency, PWM DC-DC
converter, available in 6-pin WSON, 8-pin MSOPPowerPad™, and 5-pin SOT-23 packages. With a
minimum of external components the LM3410 and
LM3410-Q1 are easy to use. It can drive 2.8-A
(typical) peak currents with an internal 170-mΩ
NMOS switch. Switching frequency is internally set to
either 525 kHz or 1.6 MHz, allowing the use of
extremely small surface mount inductors and chip
capacitors. Even though the operating frequency is
high, efficiencies up to 88% are easy to achieve.
External shutdown is included, featuring an ultra-low
standby current of 80 nA. The LM3410 and LM3410Q1 use current-mode control and internal
compensation to provide high-performance over a
wide range of operating conditions. Additional
features include PWM dimming, cycle-by-cycle
current limit, and thermal shutdown.
1
•
•
•
•
•
•
•
•
•
•
Qualified for Automotive Applications
AEC-Q100 Test Guidance With the Following:
– Device Temperature Grade 1: –40°C to 125°C
Ambient Operating Temperature Range
– Device HBM ESD Classification Level 2
– Device CDM ESD Classification Level C6
Space-Saving SOT-23 and WSON Packages
Input Voltage From 2.7 V to 5.5 V
Output Voltage From 3 V to 24 V
2.8-A (Typical) Switch Current Limit
High Switching Frequency
– 525 KHz (LM3410Y)
– 1.6 MHz (LM3410X)
170-mΩ NMOS Switch
190-mV Internal Voltage Reference
Internal Soft Start
Current-Mode, PWM Operation
Thermal Shutdown
PART NUMBER
LM3410,
LM3410Q
2 Applications
•
•
•
•
•
Device Information(1)
LED Backlight Current Sources
LiIon Backlight OLED and HB LED Drivers
Handheld Devices
LED Flash Drivers
Automotive Applications
BODY SIZE (NOM)
3.00 mm × 3.00 mm
MSOP-PowerPAD (8)
2.90 mm × 1.60 mm
SOT-23 (5)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Boost Application Circuit
L1
PACKAGE
WSON (6)
Typical Efficiency (LM3410X)
D1
DIMM
4
DIM
5
C1
VIN
L M3410
VIN
3
FB
2
GND
1
LEDs
C2
SW
R1
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM3410, LM3410-Q1
SNVS541H – OCTOBER 2007 – REVISED AUGUST 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
6
Detailed Description .............................................. 8
7.1
7.2
7.3
7.4
8
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Overview ................................................................... 8
Functional Block Diagram ....................................... 10
Feature Description................................................. 10
Device Functional Modes........................................ 10
Application and Implementation ........................ 11
8.1 Application Information............................................ 11
8.2 Typical Applications ................................................ 19
9 Power Supply Recommendations...................... 31
10 Layout................................................................... 32
10.1 Layout Guidelines ................................................. 32
10.2 Layout Examples................................................... 32
10.3 Thermal Considerations ........................................ 33
11 Device and Documentation Support ................. 40
11.1
11.2
11.3
11.4
11.5
11.6
11.7
11.8
Device Support......................................................
Documentation Support ........................................
Related Links ........................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
40
41
41
41
41
41
41
41
12 Mechanical, Packaging, and Orderable
Information ........................................................... 42
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision G (April 2013) to Revision H
Page
•
Added Device Information table, ESD Ratings table, Thermal Information table, Detailed Description section,
Feature Description section, Device Functional Modes section, Application and Implementation section, Typical
Application section, Power Supply Recommendations section, Layout section, Device and Documentation Support
section, and Mechanical, Packaging, and Orderable Information section.............................................................................. 1
•
Added AEC-Q100 Test Guidance bullets to Features............................................................................................................ 1
•
Changed RθJA value for NGG package from 80°C/W : to 55.3°C/W ...................................................................................... 4
•
Changed RθJA value for DGN package from 80°C/W : to 53.7°C/W ...................................................................................... 4
•
Changed RθJA value for DBV package from 118°C/W : to 164.2°C/W ................................................................................... 4
•
Changed RθJC(top) value for NGG package from 18°C/W : to 65.9°C/W ................................................................................. 4
•
Changed RθJC(top) value for DGN package from 18°C/W : to 61.4°C/W ................................................................................. 4
•
Changed RθJC(top) value for DBV package from 60°C/W : to 115.3°C/W ................................................................................ 4
Changes from Revision F (May 2013) to Revision G
•
2
Page
Changed layout of National Semiconductor Data Sheet to TI format .................................................................................... 1
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SNVS541H – OCTOBER 2007 – REVISED AUGUST 2016
5 Pin Configuration and Functions
NGG Package
6-Pin WSON
Top View
PGND
1
VIN
2
DIM
3
DAP
DGN Package
8-Pin MSOP-PowerPad
Top View
6
SW
5
AGND
4
FB
NC
1
PGND
2
8
NC
7
SW
DAP
VIN
3
6
AGND
DIM
4
5
FB
Not to scale
Not to scale
DBV Package
5-Pin SOT-23
Top View
SW
1
GND
2
FB
3
5
VIN
4
DIM
Not to scale
Pin Functions
PIN
I/O
DESCRIPTION
—
—
Signal ground pin. Place the bottom resistor of the feedback network as close
as possible to this pin and FB.
4
4
I
Dimming and shutdown control input. Logic high enables operation. Duty
Cycle from 0% to 100%. Do not allow this pin to float or be greater than VIN +
0.3 V.
4
5
3
I
Feedback pin. Connect FB to external resistor to set output current.
DAP
DAP
—
—
Die attach pad. Signal and Power ground. Connect to PGND and AGND on
top layer. Place 4 to 6 vias from DAP to bottom layer GND plane.
—
—
2
—
Signal and power ground pin. Place the bottom resistor of the feedback
network as close as possible to this pin.
NC
—
1, 8
—
—
No connection
PGND
1
2
—
—
Power ground pin. Place PGND and output capacitor GND close together.
SW
6
7
1
O
Output switch. Connect to the inductor, output diode.
VIN
2
3
5
I
Supply voltage pin for power stage, and input supply voltage.
NAME
WSON
MSOPPowerPAD
SOT-23
AGND
5
6
DIM
3
FB
GND
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SNVS541H – OCTOBER 2007 – REVISED AUGUST 2016
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2)
Input voltage
Operating juction temperature
MIN
MAX
VIN
–0.5
7
SW
–0.5
26.5
FB
–0.5
3
DIM
–0.5
7
(3)
, TJ
Storage temperature, Tstg
(1)
(2)
(3)
–65
UNIT
V
150
°C
150
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
Thermal shutdown occurs if the junction temperature exceeds the maximum junction temperature of the device.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
(1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±1000
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VIN
MAX
2.7
5.5
V
0
VIN
V
3
24
V
–40
125
°C
400
mW
(1)
VDIM
DIM control input
VSW
Switch output
TJ
Operating junction temperature
Power dissipation (Internal)
(1)
MIN
Input voltage
SOT-23
UNIT
Do not allow this pin to float or be greater than VIN + 0.3 V.
6.4 Thermal Information
LM3410, LM3410-Q1
NGG
(WSON)
DGN
(MSOPPowerPAD)
DBV
(SOT-23)
6 PINS
8 PINS
5 PINS
55.3
53.7
164.2
°C/W
RθJC(top) Junction-to-case (top) thermal resistance
65.9
61.4
115.3
°C/W
RθJB
Junction-to-board thermal resistance
29.6
37.3
27
°C/W
ψJT
Junction-to-top characterization parameter
1.1
7.1
12.8
°C/W
ψJB
Junction-to-board characterization parameter
29.7
37
26.5
°C/W
RθJC(bot) Junction-to-case (bottom) thermal resistance
9.3
6.8
—
°C/W
THERMAL METRIC (1)
RθJA
(1)
4
Junction-to-ambient thermal resistance
0 LFPM Air Flow
UNIT
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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SNVS541H – OCTOBER 2007 – REVISED AUGUST 2016
6.5 Electrical Characteristics
Typical values apply for TJ = 25°C; Minimum and maximum limits apply for TJ = –40°C to 125°C and VIN = 5 V (unless
otherwise noted). Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference
purposes only.
PARAMETER
TEST CONDITIONS
VFB
Feedback voltage
ΔVFB/VIN
Feedback voltage line regulation
IFB
Feedback input bias current
fSW
Switching frequency
DMAX
Maximum duty cycle
DMIN
Minimum duty cycle
RDS(ON)
Switch on resistance
ICL
Switch current limit
SU
Start-up time
VDIM_H
0.06
0.1
1
1200
1600
2000
LM3410Y
360
525
680
LM3410X
88%
92%
LM3410Y
90%
95%
LM3410X
5%
LM3410Y
2%
MSOP and SOT-23
170
330
WSON
190
350
2.8
Undervoltage lockout
11
LM3410Y, VFB = 0.25 V
3.4
7
80
2.3
VIN falling
1.7
0.4
Enable threshold voltage
1.8
Switch leakage
VSW = 24 V
Dimming pin current
Sink and source
Thermal shutdown temperature
(1)
kHz
mΩ
mA
nA
2.65
1.9
Shutdown threshold voltage
µA
µs
7
VIN rising
mV
A
LM3410X, VFB = 0.25 V
All versions, VDIM = 0 V
UNIT
%/V
LM3410X
IDIM
(1)
202
VIN = 2.7 V to 5.5 V
ISW
TSD
MAX
190
20
Quiescent current (shutdown)
UVLO
TYP
178
2.1
Quiescent current (switching)
IQ
MIN
V
V
1
µA
100
nA
165
°C
Thermal shutdown occurs if the junction temperature exceeds the maximum junction temperature of the device.
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SNVS541H – OCTOBER 2007 – REVISED AUGUST 2016
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6.6 Typical Characteristics
All curves taken at VIN = 5 V with the 50-mA boost configuration shown in Figure 18. TJ = 25°C, unless otherwise specified.
RSET = 4 Ω
Figure 1. LM3410X Efficiency vs VIN
500-Hz DIM Frequency
6
Figure 2. LM3410X Start-Up Signature
D = 50%
Figure 3. Four 3.3-V LEDs
Figure 4. DIM Frequency and Duty Cycle vs Average ILED
Figure 5. Current Limit vs Temperature
Figure 6. RDS(ON) vs Temperature
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Typical Characteristics (continued)
All curves taken at VIN = 5 V with the 50-mA boost configuration shown in Figure 18. TJ = 25°C, unless otherwise specified.
LM3410X
LM3410Y
Figure 7. Oscillator Frequency vs Temperature
Figure 8. Oscillator Frequency vs Temperature
Figure 9. VFB vs Temperature
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SNVS541H – OCTOBER 2007 – REVISED AUGUST 2016
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7 Detailed Description
7.1 Overview
The LM3410 and LM3410-Q1 are a constant frequency PWM, boost regulator IC. It delivers a minimum of 2.1-A
peak switch current. The device operates very similar to a voltage regulated boost converter except that the
device regulates the output current that passes through LEDs. The current magnitude is set with a series
resistor. The converter regulates to the feedback voltage (190 mV) created by the multiplication of the series
resistor and the LED current. The regulator has a preset switching frequency of either 525 kHz or 1.6 MHz. This
high frequency allows the LM3410 or LM3410-Q1 to operate with small surface mount capacitors and inductors,
resulting in a DC-DC converter that requires a minimum amount of board space. The LM3410 and LM3410-Q1
are internally compensated and requires few external components, making usage simple. The LM3410 and
LM3410-Q1 use current-mode control to regulate the LED current.
The LM3410 and LM3410-Q1 supply a regulated LED current by switching the internal NMOS control switch at
constant frequency and variable duty cycle. A switching cycle begins at the falling edge of the reset pulse
generated by the internal oscillator. When this pulse goes low, the output control logic turns on the internal
NMOS control switch. During this ON time, the SW pin voltage (VSW) decreases to approximately GND, and the
inductor current (IL) increases with a linear slope. IL is measured by the current sense amplifier, which generates
an output proportional to the switch current. The sensed signal is summed with the regulator’s corrective ramp
and compared to the error amplifier’s output, which is proportional to the difference between the feedback
voltage and reference voltage (VREF). When the PWM comparator output goes high, the output switch turns off
until the next switching cycle begins. During the switch OFF time, inductor current discharges through diode D1,
which forces the SW pin to swing to the output voltage plus the forward voltage (VD) of the diode. The regulator
loop adjusts the duty cycle (D) to maintain a regulated LED current.
IL
L1
Q1
VIN
Control
VO
D1
IC
+
VSW
C 1
I LED
Figure 10. Simplified Boost Topology Schematic
8
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Overview (continued)
V OUT + VD
Vsw ( t)
t
VIN
VL(t)
t
VIN - VOUT - VD
IL (t)
iL
t
I DIODE(t)
t
(
iL - - iOUT )
I Capacitor(t)
t
- i OUT
'v
VOUT(t)
DTS
TS
Figure 11. Typical Waveforms
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7.2 Functional Block Diagram
DIM
VIN
ThermalSHDN
Control Logic
+
Ramp
Artificial
UVLO=2.3V
Oscillator
+
-
cv
1. 6 MHz
+
S
R
SW
+
NMOS
+
R
Q
-
VFB
+
VREF = 190 mV
Internal
Compensation
ILIMIT
ISENSE
+
GND
Copyright © 2016, Texas Instruments Incorporated
7.3 Feature Description
7.3.1 Current Limit
The LM3410 and LM3410-Q1 use cycle-by-cycle current limiting to protect the internal NMOS switch. This
current limit does not protect the output from excessive current during an output short circuit. The input supply is
connected to the output by the series connection of an inductor and a diode. If a short circuit is placed on the
output, excessive current can damage both the inductor and diode.
7.3.2 DIM Pin and Shutdown Mode
The average LED current can be controlled using a PWM signal on the DIM pin. The duty cycle can be varied
from 0 to 100%, to either increase or decrease LED brightness. PWM frequencies from 1 Hz to 25 kHz can be
used. For controlling LED currents down to the µA levels, it is best to use a PWM signal frequency from 200 to
1 kHz. The maximum LED current would be achieved using a 100% duty cycle, that is the DIM pin always high.
7.4 Device Functional Modes
7.4.1 Thermal Shutdown
Thermal shutdown limits total power dissipation by turning off the output switch when the IC junction temperature
exceeds 165°C. After thermal shutdown occurs, the output switch does not turn on until the junction temperature
drops to approximately 150°C.
10
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 Boost Converter
8.1.1.1 Setting the LED Current
I LED
V FB
R SET
Figure 12. Setting ILED
The LED current is set using the following equation:
VFB
R SET
= ILED
where
•
RSET is connected between the FB pin and GND.
(1)
8.1.1.2 LED-Drive Capability
When using the LM3410 or LM3410-Q1 in the typical application configuration, with LEDs stacked in series
between the VOUT and FB pin, the maximum number of LEDs that can be placed in series is dependent on the
maximum LED forward voltage (VFMAX).
(VFMAX × NLEDs) + 190 mV < 24 V
(2)
When inserting a value for maximum VFMAX the LED forward voltage variation over the operating temperature
range must be considered.
8.1.1.3 Inductor Selection
The inductor value determines the input ripple current. Lower inductor values decrease the physical size of the
inductor, but increase the input ripple current. An increase in the inductor value decreases the input ripple
current.
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Application Information (continued)
'i L
I L (t)
iL
VIN
L
VIN - VOUT
L
DTS
TS
t
Figure 13. Inductor Current
2'iL § VIN ·
=¨
¸
DTS ¨© L ¹
§ VIN ·
¸¸ x DTS
'iL = ¨¨
© 2L ¹
(3)
The Duty Cycle (D) for a Boost converter can be approximated by using the ratio of output voltage (VOUT) to input
voltage (VIN).
§ 1 ·= 1
=¨
1 - D¸ Dc
VOUT
VIN
¹
©
(4)
Therefore:
VOUT - VIN
D=
VOUT
(5)
Power losses due to the diode (D1) forward voltage drop, the voltage drop across the internal NMOS switch, the
voltage drop across the inductor resistance (RDCR) and switching losses must be included to calculate a more
accurate duty cycle (see Calculating Efficiency and Junction Temperature for a detailed explanation). A more
accurate formula for calculating the conversion ratio is:
V OUT
V IN
=
K
'¶
where
•
η equals the efficiency of the device application.
(6)
Or:
K
12
VOUT u ILED
VIN u IIN
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(7)
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Application Information (continued)
Therefore:
VOUT KVIN
VOUT
D
(8)
Inductor ripple in a LED driver circuit can be greater than what would normally be allowed in a voltage regulator
Boost and Sepic design. A good design practice is to allow the inductor to produce 20% to 50% ripple of
maximum load. The increased ripple is unlikely to be a problem when illuminating LEDs.
From the previous equations, the inductor value is then obtained.
§ VIN ·
L = ¨¨
¸ x DTS
©2'iL¹
where
•
1 / TS = fSW
(9)
Ensure that the minimum current limit (2.1 A) is not exceeded, so the peak current in the inductor must be
calculated. The peak current (Lpk I) in the inductor is calculated by Equation 10:
ILpk = IIN + ΔIL or ILpk = IOUT /D' + ΔiL
(10)
When selecting an inductor, make sure that it is capable of supporting the peak input current without saturating.
Inductor saturation results in a sudden reduction in inductance and prevent the regulator from operating correctly.
Because of the speed of the internal current limit, the peak current of the inductor only needs to be specified for
the required maximum input current. For example, if the designed maximum input current is 1.5 A and the peak
current is 1.75 A, then the inductor must be specified with a saturation current limit of >1.75 A. There is no need
to specify the saturation or peak current of the inductor at the 2.8-A typical switch current limit.
Because of the operating frequency of the LM3410 and LM3410-Q1, ferrite based inductors are preferred to
minimize core losses. This presents little restriction because the variety of ferrite-based inductors is huge. Lastly,
inductors with lower series resistance (DCR) provides better operating efficiency. For recommended inductor
value examples, see Typical Applications.
8.1.1.4 Input Capacitor
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and ESL (Equivalent
Series Inductance). TI recommens an input capacitance from 2.2 µF to 22 µF depending on the application. The
capacitor manufacturer specifically states the input voltage rating. Make sure to check any recommended
deratings and also verify if there is any significant change in capacitance at the operating input voltage and the
operating temperature. The ESL of an input capacitor is usually determined by the effective cross sectional area
of the current path. At the operating frequencies of the LM3410 and LM3410-Q1, certain capacitors may have an
ESL so large that the resulting impedance (2πfL) is higher than that required to provide stable operation. As a
result, TI recommends surface mount capacitors. Multilayer ceramic capacitors (MLCC) are good choices for
both input and output capacitors and have very low ESL. For MLCCs TI recommends use of X7R or X5R
dielectrics. Consult the capacitor manufacturer's datasheet for rated capacitance variation over operating
conditions.
8.1.1.5 Output Capacitor
The LM3410 and LM3410-Q1 operate at frequencies allowing the use of ceramic output capacitors without
compromising transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing
output ripple. The output capacitor is selected based upon the desired output ripple and transient response. The
initial current of a load transient is provided mainly by the output capacitor. The output impedance therefore
determines the maximum voltage perturbation. The output ripple of the converter is a function of the capacitor’s
reactance and its equivalent series resistance (ESR) (see Equation 11).
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Application Information (continued)
§
'V OUT = 'iL x RESR + ¨
©
V OUT x D
·
2 x fSW x ROUT x COUT ¸
¹
(11)
When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the
output ripple is approximately sinusoidal and 90° phase shifted from the switching action.
Given the availability and quality of MLCCs and the expected output voltage of designs using the LM3410 or
LM3410-Q1, there no need to review any other capacitor technologies. Another benefit of ceramic capacitors is
their ability to bypass high frequency noise. A certain amount of switching edge noise couples through parasitic
capacitances in the inductor to the output. A ceramic capacitor bypasses this noise while a tantalum does not.
Because the output capacitor is one of the two external components that control the stability of the regulator
control loop, most applications requires a minimum at 0.47 µF of output capacitance. Like the input capacitor, TI
recommends X7R or X5R as multilayer ceramic capacitors. Again, verify actual capacitance at the desired
operating voltage and temperature.
8.1.1.6 Diode
The diode (D1) conducts during the switch off time. TI recommends Schottky diode for its fast switching times
and low forward voltage drop. The diode must be chosen so that its current rating is greater than:
ID1 ≥ IOUT
(12)
The reverse breakdown rating of the diode must be at least the maximum output voltage plus appropriate margin.
8.1.1.7 Output Overvoltage Protection
A simple circuit consisting of an external Zener diode can be implemented to protect the output and the LM3410
or LM3410-Q1 device from an overvoltage fault condition. If an LED fails open, or is connected backwards, an
output open circuit condition occurs. No current is conducted through the LEDs, and the feedback node equals
zero volts. The LM3410 or LM3410-Q1 reacts to this fault by increasing the duty cycle, thinking the LED current
has dropped. A simple circuit that protects the device is shown in Figure 14.
Zener diode D2 and resistor R3 is placed from VOUT in parallel with the string of LEDs. If the output voltage
exceeds the breakdown voltage of the Zener diode, current is drawn through the Zener diode, R3 and sense
resistor R1. Once the voltage across R1 and R3 equals the feedback voltage of 190 mV, the LM3410 and
LM3410-Q1 limits their duty cycle. No damage occurs to the device, the LEDs, or the Zener diode. Once the fault
is corrected, the application will work as intended.
14
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Application Information (continued)
V SW
D1
LEDs
O
V
P
D2
C2
R3
V FB
R1
Figure 14. Overvoltage Protection Circuit
8.1.2 SEPIC Converter
The LM3410 or LM3410-Q1 can easily be converted into a SEPIC converter. A SEPIC converter has the ability
to regulate an output voltage that is either larger or smaller in magnitude than the input voltage. Other converters
have this ability as well (CUK and Buck-Boost), but usually create an output voltage that is opposite in polarity to
the input voltage. This topology is a perfect fit for Lithium Ion battery applications where the input voltage for a
single cell Li-Ion battery varies from 2.7 V to 4.5 V and the output voltage is somewhere in between. Most of the
analysis of the LM3410 Boost Converter is applicable to the LM3410 SEPIC Converter.
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Application Information (continued)
V IN
L1
C1
LM 3410
1
6
2
5
3
4
VO
D1
C3
C2
L2
HB / OLED
R2
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 15. HB or OLED SEPIC Converter Schematic
8.1.2.1 SEPIC Equations
SEPIC Conversion ratio without loss elements:
VOUT
VIN
=
D
'¶
(13)
Therefore:
D=
VOUT
VOUT + VIN
(14)
Small ripple approximation:
In a well-designed SEPIC converter, the output voltage, and input voltage ripple, the inductor ripple IL1 and IL2 is
small in comparison to the DC magnitude. Therefore it is a safe approximation to assume a DC value for these
components. The main objective of the Steady State Analysis is to determine the steady state duty cycle, voltage
and current stresses on all components, and proper values for all components.
In a steady-state converter, the net volt-seconds across an inductor after one cycle equals zero. Also, the charge
into a capacitor equals the charge out of a capacitor in one cycle.
Therefore:
IL2
IL2
16
§ D' ·
¨¨ ¸¸ u IL1
©D¹
and
§D·
¨ ' ¸ u ILED
©D ¹
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Application Information (continued)
Substituting IL1 into IL2
IL2 = ILED
(16)
The average inductor current of L2 is the average output load.
VL(t )
AREA 1
t (s)
AREA 2
DTS
TS
Figure 16. Inductor Volt-Second Balance Waveform
Applying Charge balance on C1:
VC3 =
D'( VOUT)
D
(17)
Because there are no DC voltages across either inductor, and capacitor C3 is connected to Vin through L1 at
one end, or to ground through L2 on the other end, we can say that
VC3 = VIN
(18)
Therefore:
VIN =
D'( VOUT)
D
(19)
This verifies the original conversion ratio equation.
It is important to remember that the internal switch current is equal to IL1 and IL2 during the D interval. Design the
converter so that the minimum ensured peak switch current limit (2.1 A) is not exceeded.
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Application Information (continued)
8.1.2.2 Steady State Analysis with Loss Elements
- iL1(t)
v ()
C1 t
+
i
i sw
iD1(t)
vD1(t)
L 2 (t)
VIN
iC 2(t)
-
+
RL1
i C1( t )
+
L1(t)
vL2(t )
+
vC2(t)
-
Ron
vO(t)
-
v
+
RL 2
Copyright © 2016, Texas Instruments Incorporated
Figure 17. SEPIC Simplified Schematic
8.1.2.2.1 Details
Using inductor volt-second balance and capacitor charge balance, the following equations are derived:
IL2 = (ILED)
(20)
IL1 = (ILED) × (D/D')
(21)
and
VOUT
VIN
·
§
¸
¨
1
D
§ ·
¸
= ¨¨ ' ¸¸ ¨
¸
2·
·
§
§
© D ¹ ¨§
¨¨1+ VD + R L2 ·¸ + ¨ D ¸ §¨ RON ·¸ + ¨ D ¸ §¨ RL1·¸¸
¨¨© VOUT R ¸¹ ¨ '2 ¸ © R ¹ ¨ ' 2 ¸ © R ¹¸
©D ¹
©D ¹
¸
¨
¹
©
ROUT =
(22)
VOUT
ILED
(23)
Therefore:
·
§
¸
¨
1
¸
¨
K=
¸
¨§
2
·
·
§
§
·
¨ ¨1+ VD + R L2 ¸ + ¨ D ¸ §¨ R ON ·¸ + ¨ D ¸ §¨ R L1 ·¸ ¸
¸
¨
2
2
¨ © VOUT ROUT¹ ¨ D' ¸ ©ROUT ¹ ¨ D' ¸ ©R OUT¹ ¸
¹
¹
©
©
¸
¨
¹
©
(24)
All variables are known except for the duty cycle (D). A quadratic equation is needed to solve for D. A less
accurate method of determining the duty cycle is to assume efficiency, and calculate the duty cycle.
VOUT
VIN
18
=
§ D ·xK
¨1 - D¸
©
¹
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Application Information (continued)
VOUT
·
§
D=¨
(VIN x K) + VOUT ¸¹
©
(26)
Table 1. Efficiencies for Typical SEPIC Applications
EXAMPLE 1
EXAMPLE 2
EXAMPLE 3
VIN
2.7 V
VIN
3.3 V
VIN
5V
VOUT
3.1 V
VOUT
3.1 V
VOUT
3.1 V
IIN
770 mA
IIN
600 mA
IIN
375 mA
ILED
500 mA
ILED
500 mA
ILED
500 mA
η
75%
η
80%
η
83%
8.2 Typical Applications
8.2.1 Low Input Voltage, 1.6-MHz, 3 to 5 White LED Output at 50-mA Boost Converter
L1
D1
DIMM
4
DIM
L M3410
VIN
5
VIN
C1
3
FB
2
GND
1
LEDs
C2
SW
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 18. Boost Schematic
8.2.1.1 Design Requirements
For this design example, use the parameters listed in Table 2 as the input parameters.
Table 2. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
50 mA
VOUT
14.6 V (four 3.6-V LEDs in series plus 190 mV)
RD
8 Ω (dynamic resistance of 4 LEDs in series)
ΔILp–p
100 mA (maximum)
ΔVOUTp–p
250 mV (maximum)
8.2.1.2 Detailed Design Procedure
This design procedure uses the worst-case minimum input voltage and a nominal 4 LED series load for
calculations.
8.2.1.2.1 Set the LED Current (R1)
Rearranging the LED current equation the current sense resistor R1 can be found using Equation 27.
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R1 =
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VFB
190mV
=
ILED
50mA
(27)
3.8 Ω is not a standard value so a standard value of R1 = 3.83 Ω is chosen.
8.2.1.2.2 Calculate Maximum Duty Cycle (DMAX)
The maximum duty cycle is required for calculating the inductor value and the minimum output capacitance.
Assuming an approximate conversion efficiency (η) of 90% DMAX is calculated using Equation 28.
DMAX =
VOUT - × VIN(min) 14.6V - 0.9 × 2.7V
=
= 0.834
VOUT
14.6V
(28)
8.2.1.2.3 Calculate the Inductor Value (L1)
Using the maximum duty cycle, the minimum input voltage, and the maximum inductor ripple current (ΔiLp–p) the
minimum inductor value to achieve the maximum ripple current is calculated using Equation 29.
VIN(min) × DMAX × TS
2.7V × 0.834 × 625ns
L1 = F
p
G = l
2 × 100mA
2 × ¨iL-PP
H
(29)
To ensure the maximum inductor ripple current requirement is met with a 20% inductor tolerance an inductor
value of L1 = 10 µH is selected.
8.2.1.2.4 Calculate the Output Capacitor (C2)
To maintain a maximum of 250-mV output voltage ripple the dynamic resistance of the LED stack (RD) must be
used. Assuming a ceramic capacitor is used so the ESR can be neglected this minimum amount of capacitance
can be found using Equation 30.
C2 •
VOUT × DMAX
14.6V × 0.834
=
=
2 × fSW × RD × VOUT 2 × 1.6MHz ×
× 14.6V
F
(30)
1.9 µF is not a standard value so a value of C2 = 2.2 µF is selected.
8.2.1.2.5 Input Capacitor (C1) and Schottky Diode (D1)
TI recommends an input capacitor from 2.2 µF to 22 µF. This is a relatively low power design optimized for a
small footprint. For a good balance of input filtering and small size a 6.3-V capacitor with a value of C1 = 10 µF is
selected. The output voltage with a 5 LED load is over 18 V and the reverse voltage of the schottky diode must
be greater than this voltage. To give some headroom to avoid reverse breakdown and to maintain small size and
reliability the diode selected is D1 = 30 V, 500 mA.
20
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8.2.1.3 Application Curves
Figure 20. PWM Dimming
Figure 19. Efficiency versus Input Voltage
8.2.2 LM3410X SOT-23: 5 × 1206 Series LED String Application
D1
L1
LEDs
VIN
LM3410
DIMM
C1
4
3
2
R2
5
C2
1
R1
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Figure 21. LM3410X (1.6 MHz) 5 × 3.3-V LED String Application Diagram
8.2.2.1 Design Requirements
For this design example, use the parameters listed in Table 3 as the input parameters.
Table 3. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
≊50 mA
VOUT
≊16.5 V (five 3.3-V LEDs in series)
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Table 4. Part Values
PART
VALUE
U1
2.8-A ISW LED Driver
C1, Input capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
2.2 µF, 25 V, X5R
D1, Catch diode
0.4-Vf Schottky 500 mA, 30 VR
L1
10 µH, 1.2 A
R1
4.02 Ω, 1%
R2
100 kΩ, 1%
LEDs
SMD-1206, 50 mA, Vf ≊ 3.6 V
8.2.3 LM3410Y SOT-23: 5 × 1206 Series LED String Application
D1
L1
LEDs
VIN
LM3410
DIMM
C1
4
3
2
R2
5
C2
1
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 22. LM3410Y (525 kHz) 5 × 3.3-V LED String Application Diagram
8.2.3.1 Design Requirements
For this design example, use the parameters listed in Table 5 as the input parameters.
Table 5. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
≊50 mA
VOUT
≊16.5 V (five 3.3-V LEDs in series)
Table 6. Part Values
PART
22
VALUE
U1
2.8-A ISW LED Driver
C1, Input capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
2.2 µF, 25 V, X5R
D1, Catch diode
0.4-Vf Schottky 500 mA, 30 VR
L1
15 µH, 1.2 A
R1
4.02 Ω, 1%
R2
100 kΩ, 1%
LEDs
SMD-1206, 50 mA, Vf ≊ 3.6 V
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8.2.4 LM3410X WSON: 7 × 5 LED Strings Backlighting Application
L1
LEDs
D1
VIN
LM3410
C1
R2
DIMM
1
6
2
5
3
4
ILED
C2
ISET
R1
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Figure 23. LM3410X (1.6 MHz) 7 × 5 × 3.3-V LEDs Backlighting Application Diagram
8.2.4.1 Design Requirements
For this design example, use the parameters listed in Table 7 as the input parameters.
Table 7. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
≊25 mA
VOUT
≊16.7 V (seven strings of five 3.3-V LEDs in series)
Table 8. Part Values
PART
VALUE
U1
2.8-A ISW LED Driver
C1, Input capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
4.7 µF, 25 V, X5R
D1, Catch Diode
0.4-Vf Schottky 500 mA, 30 VR
L1
8.2 µH, 2 A
R1
1.15 Ω, 1%
R2
100 kΩ, 1%
LEDs
SMD-1206, 50 mA, Vf ≊ 3.6 V
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8.2.5 LM3410X WSON: 3 × HB LED String Application
L1
D1
VIN
LM3410
C1
R2
DIMM
1
6
2
5
3
4
HB - LEDs
C2
R3
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 24. LM3410X (1.6 MHz) 3 × 3.4-V LED String Application Diagram
8.2.5.1 Design Requirements
For this design example, use the parameters listed in Table 9 as the input parameters.
Table 9. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
≊340 mA
VOUT
≊11 V (three 3.4-V LEDs in series)
Table 10. Part Values
PART
24
VALUE
U1
2.8-A ISW LED Driver
C1, Input capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
2.2 µF, 25 V, X5R
D1, Catch diode
0.4-Vf Schottky 500 mA, 30 VR
L1
10 µH, 1.2 A
R1
1 Ω, 1%
R2
100 kΩ, 1%
R3
1.5 Ω, 1%
HB – LEDs
340 mA, Vf ≊ 3.6 V
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8.2.6 LM3410Y SOT-23: 5 × 1206 Series LED String Application With OVP
L1
LEDs
D1
VIN
DIMM
LM3410
C1
OVP
4
R2
3
C2
2
5
D2
1
R3
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 25. LM3410Y (525 kHz) 5 × 3.3-V LED String Application With OVP Diagram
8.2.6.1 Design Requirements
For this design example, use the parameters listed in Table 11 as the input parameters.
Table 11. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
≊50 mA
VOUT
≊16.5 V (five 3.3-V LEDs in series)
Table 12. Part Values
PART
VALUE
U1
2.8-A ISW LED Driver
C1, Input capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
2.2 µF, 25 V, X5R
D1, Catch diode
0.4-Vf Schottky 500 mA, 30 VR
D2
18 V Zener diode
L1
15 µH, 0.7 A
R1
4.02 Ω, 1%
R2
100 kΩ, 1%
R3
100 Ω, 1%
LEDs
SMD-1206, 50 mA, Vf ≊ 3.6 V
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8.2.7 LM3410X SEPIC WSON: HB or OLED Illumination Application
V IN
L1
C1
LM 3410
1
6
2
5
3
4
VO
D1
C3
C2
L2
HB / OLED
R2
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 26. LM3410X (1.6 MHz) HB or OLED Illumination Application Diagram
8.2.7.1 Design Requirements
For this design example, use the parameters listed in Table 13 as the input parameters.
Table 13. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
≊300 mA
VOUT
≊3.8 V
Table 14. Part Values
PART
26
VALUE
U1
2.8-A ISW LED Driver
C1, Input capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
10 µF, 6.3 V, X5R
C3
2.2 µF, 25 V, X5R
D1, Catch diode
0.4-Vf Schottky 1 A, 20 VR
L1 and L2
4.7 µH, 3 A
R1
665 mΩ, 1%
R2
100 kΩ, 1%
HB – LEDs
350 mA, Vf ≊ 3.6 V
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8.2.8 LM3410X WSON: Boost Flash Application
VIN
L1
C1
D1
VO
LM3410
1
6
2
5
3
4
C2
LEDs
FLASH CTRL
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 27. LM3410X (1.6 MHz) Boost Flash Application Diagram
8.2.8.1 Design Requirements
For this design example, use the parameters listed in Table 15 as the input parameters.
Table 15. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
≊1 A (pulse)
VOUT
≊8 V
Table 16. Part Values
PART
VALUE
U1
2.8-A ISW LED Driver
C1, Input capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
10 µF, 16 V, X5R
D1, Catch diode
0.4-Vf Schottky 500 mA, 30 VR
L1
4.7 µH, 3 A
R1
200 mΩ, 1%
LEDs
500 mA, Vf ≊ 3.6 V, IPULSE = 1 A
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8.2.9 LM3410X SOT-23: 5 × 1206 Series LED String Application With VIN > 5.5 V
D1
L1
LEDs
V PWR
LM3410
DIMM
C1
R3
4
2
R2
5
D2
3
C2
1
C3
R1
Copyright © 2016, Texas Instruments Incorporated
Figure 28. LM3410X (1.6 MHz) 5 × 1206 Series LED String Application With VIN > 5.5 V Diagram
8.2.9.1 Design Requirements
For this design example, use the parameters listed in Table 17 as the input parameters.
Table 17. Design Parameters
PARAMETER
EXAMPLE VALUE
VPWR
9 V to 14 V
ILED
≊50 mA
VOUT
≊16.5 V (five 3.3-V LEDs in series)
Table 18. Part Values
PART
28
VALUE
U1
2.8-A ISW LED Driver
C1, Input VPWRcapacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
2.2 µF, 25 V, X5R
C3, Input VIN capacitor
0.1 µF, 6.3 V, X5R
D1, Catch diode
0.43-Vf Schottky 500 mA, 30 VR
D2
3.3 V Zener, SOT-23
L1
10 µH, 1.2 A
R1
4.02 Ω, 1%
R2
100 kΩ, 1%
R3
576 Ω, 1%
LEDs
SMD-1206, 50 mA, Vf ≊ 3.6 V
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8.2.10 LM3410X WSON: Camera Flash or Strobe Circuit Application
VIN
L1
C1
VO
D1
C3
LM3410
1
6
2
5
3
4
L2
R2
C2
LED( s)
Q2
R3
R1
R4
Q1
FLASH CTRL
Copyright © 2016, Texas Instruments Incorporated
Figure 29. LM3410X (1.6 MHz) Camera Flash or Strobe Circuit Application Diagram
8.2.10.1 Design Requirements
For this design example, use the parameters listed in Table 19 as the input parameters.
Table 19. Design Parameters
PARAMETER
EXAMPLE VALUE
VIN
2.7 V to 5.5 V
ILED
≊1.5 A (flash)
VOUT
≊7.5 V
Table 20. Part Values
PART
VALUE
U1
2.8-A ISW LED Driver
C1, Input capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
220 µF, 10 V, tantalum
C3 capacitor
10 µF, 16 V, X5R
D1, Catch diode
0.43-Vf Schottky 1 A, 20 VR
L1
3.3 µH, 2.7 A
R1
1 Ω, 1%
R2
37.4 kΩ, 1%
R3
100 kΩ, 1%
R4
0.15 Ω, 1%
Q1 and Q2
30 V, ID = 3.9 A
LEDs
SMD-1206, 50 mA, Vf ≊ 3.6 V, IPULSE = 1.5 A
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8.2.11 LM3410X SOT-23: 5 × 1206 Series LED String Application With VIN and VPWR Rail > 5.5 V
L1
D1
LEDs
VPWR
LM3410
DIMM
C1
4
2
R2
VIN
3
5
C2
1
C3
R1
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Figure 30. LM3410X (1.6 MHz) 5 × 1206 Series LED String Application With VIN and VPWR Rail > 5.5 V
Diagram
8.2.11.1 Design Requirements
For this design example, use the parameters listed in Table 21 as the input parameters.
Table 21. Design Parameters
PARAMETER
EXAMPLE VALUE
VPWR
9 V to 14 V
VIN
2.7 V to 5.5 V
ILED
≊50 mA
VOUT
≊16.5 V (five 3.3-V LEDs in series)
Table 22. Part Values
PART
30
VALUE
U1
2.8-A ISW LED Driver
C1, Input VPWRcapacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
2.2 µF, 25 V, X5R
C3, Input VIN capacitor
0.1 µF, 6.3 V, X5R
D1, Catch diode
0.43-Vf Schottky 500 mA, 30 VR
L1
10 µH, 1.2 A
R1
4.02 Ω, 1%
R2
100 kΩ, 1%
LEDs
SMD-1206, 50 mA, Vf ≊ 3.6 V
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8.2.12 LM3410X WSON: Boot-Strap Circuit to Extend Battery Life
V IN
C4
L1
D2
C1
C3
VO
D1
LM3410
1
6
2
5
3
4
L2
C2
R3
D3
R1
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Figure 31. LM3410X (1.6 MHz) Boot-Strap Circuit to Extend Battery Life
8.2.12.1 Design Requirements
For this design example, use the parameters listed in Table 3 as the input parameters.
Table 23. Design Parameters
PARAMETER
EXAMPLE VALUE
1.9 V to 5.5 V
VIN
>2.3 V (typical) for start-up
≊300 mA
ILED
Table 24. Part Values
PART
VALUE
U1
2.8-A ISW LED Driver
C1, Input VPWR capacitor
10 µF, 6.3 V, X5R
C2, Output capacitor
10 µF, 6.3 V, X5R
C3, Input VIN capacitor
0.1 µF, 6.3 V, X5R
D1, Catch diode
0.43-Vf Schottky 1 A, 20 VR
D2 and D3
Dual small signal Schottky
L1 and L2
3.3 µH, 3 A
R1
665 mΩ, 1%
R3
100 kΩ, 1%
HB – LEDs
350 mA, Vf ≊ 3.4 V
9 Power Supply Recommendations
Any DC output power supply may be used provided it has a high enough voltage and current range for the
particular application required.
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10 Layout
10.1 Layout Guidelines
When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The
most important consideration when completing a boost converter layout is the close coupling of the GND
connections of the COUT capacitor and the PGND pin. The GND ends must be close to one another and be
connected to the GND plane with at least two vias. There must be a continuous ground plane on the bottom
layer of a two-layer board except under the switching node island. The FB pin is a high impedance node and the
FB trace must be kept short to avoid noise pickup and inaccurate regulation. The RSET feedback resistor must be
placed as close as possible to the IC, with the AGND of RSET (R1) placed as close as possible to the AGND of
the IC. Radiated noise can be decreased by choosing a shielded inductor. The remaining components must also
be placed as close as possible to the IC. See AN-1229 SIMPLE SWITCHER® PCB Layout Guidelins (SNVA054)
for further considerations and the LM3410 demo board as an example of a four-layer layout.
For certain high power applications, the PCB land may be modified to a dog bone shape (see Figure 33).
Increasing the size of ground plane and adding thermal vias can reduce the RθJA for the application.
10.2 Layout Examples
LEDs
COPPER
PCB
R1
PGND
DIM
FB
4
PGND 1
6
SW
VIN
2
5
AGND
DIM
3
4
FB
3
AGND
5
C2
VIN
VSW
VO
2
6
1
PGND
D1
C1
SW
L1
COPPER
Figure 32. Boost PCB Layout Guidelines
Figure 33. PCB Dog Bone Layout
LED1
VO
PGND
C2
R1
L2
D1
FB
DIM
4
3
AGND
5
2
VIN
C1
C3
6
1
PGND
SW
L1
VIN
The layout guidelines described for the LM3410 boost-converter are applicable to the SEPIC OLED Converter. This is
a proper PCB layout for a SEPIC Converter.
Figure 34. HB or OLED SEPIC PCB Layout
32
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10.3 Thermal Considerations
10.3.1 Design
When designing for thermal performance, many variables must be considered, such as ambient temperature,
airflow, external components, and PCB design.
The surrounding maximum air temperature is fairly explanatory. As the temperature increases, the junction
temperature increases. This may not be linear though. As the surrounding air temperature increases, resistances
of semiconductors, wires and traces increase. This decreases the efficiency of the application, and more power
is converted into heat, and increases the silicon junction temperatures further.
Forced air can drastically reduce the device junction temperature. Air flow reduces the hot spots within a design.
Warm airflow is often much better than a lower ambient temperature with no airflow.
Choose components that are efficient, and the mutual heating between devices can be reduced.
The PCB design is a very important step in the thermal design procedure. The LM3410 and LM3410-Q1 are
available in three package options (6-pin WSON, 8-pin MSOP, and 5-pin SOT-23). The options are electrically
the same, but there are differences between the package sizes and thermal performances. The WSON and
MSOP have thermal die attach pads (DAP) attached to the bottom of the packages, and are therefore capable of
dissipating more heat than the SOT-23 package. It is important that the customer choose the correct package for
the application. A detailed thermal design procedure has been included in this data sheet. This procedure helps
determine which package is correct, and common applications are analyzed.
There is one significant thermal PCB layout design consideration that contradicts a proper electrical PCB layout
design consideration. This contradiction is the placement of external components that dissipate heat. The
greatest external heat contributor is the external Schottky diode. Increasing the distance between the LM3410 or
LM3410-Q1 and the Schottky diode may reduce the mutual heating effect. This, however, creates electrical
performance issues. It is important to keep the device, the output capacitor, and Schottky diode physically close
to each other (see Layout Guidelines). The electrical design considerations outweigh the thermal considerations.
Other factors that influence thermal performance are thermal vias, copper weight, and number of board layers.
Heat energy is transferred from regions of high temperature to regions of low temperature via three basic
mechanisms: radiation, conduction and convection. Conduction and convection are the dominant heat transfer
mechanism in most applications.
The data sheet values for each packages thermal impedances are given to allow comparison of the thermal
performance of one package against another. To achieve a comparison between packages, all other variables
must be held constant in the comparison (PCB size, copper weight, thermal vias, power dissipation, VIN, VOUT,
load current, and others). This provides indication of package performance, but it would be a mistake to use
these values to calculate the actual junction temperature in an application.
10.3.2 LM3410 and LM3410-Q1 Thermal Models
Heat is dissipated from the LM3410, LM3410-Q1, and other devices. The external loss elements include the
Schottky diode, inductor, and loads. All loss elements mutually increase the heat on the PCB, and therefore
increase each other’s temperatures.
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Thermal Considerations (continued)
L1
D1
IL(t)
VOUT(t)
VIN
Q1
C1
Figure 35. Thermal Schematic
RTCASE-AMB
TCASE
CTCASE-AMB
RTJ-CASE
CTJ-CASE
INTERNAL
PDISS
SMALL
LARGE
PDISS-TOP
TAMBIENT
PDISS-PCB
TJUNCTION
RTJ-PCB
CTJ-PCB
DEVICE
EXTERNAL
PDISS
RTPCB-AMB
TPCB
CTPCB-AMB
PCB
Figure 36. Associated Thermal Model
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Thermal Considerations (continued)
10.3.3 Calculating Efficiency and Junction Temperature
Use Equation 31 to calculate RθJA.
R TJA =
T J - TA
PDissipation
(31)
A common error when calculating RθJA is to assume that the package is the only variable to consider.
Other variables are:
• Input voltage, output voltage, output current, RDS(ON)
• Ambient temperature and air flow
• Internal and external components' power dissipation
• Package thermal limitations
• PCB variables (copper weight, thermal vias, and component placement)
Another common error when calculating junction temperature is to assume that the top case temperature is the
proper temperature when calculating RθJC. RθJC represents the thermal impedance of all six sides of a package,
not just the top side. This document refers to a thermal impedance called RΨJC. RΨJC represents a thermal
impedance associated with just the top case temperature. This allows for the calculation of the junction
temperature with a thermal sensor connected to the top case.
The complete LM3410 and LM3410-Q1 boost converter efficiency can be calculated using Equation 32.
K
POUT
PIN
or
POUT
POUT PLOSS
K
where
•
PLOSS is the sum of two types of losses in the converter, switching and conduction
(32)
Conduction losses usually dominate at higher output loads, where as switching losses remain relatively fixed and
dominate at lower output loads.
To calculate losses in the LM3410 or LM3410-Q1 device, use Equation 33.
PLOSS = PCOND + PSW + PQ
where
•
PQ = quiescent operating power loss
(33)
Conversion ratio of the boost converter with conduction loss elements inserted is calculated with Equation 34.
VOUT
VIN
§
·
¨
¸
1
1 §¨1- Dc x VD·¸ ¨
¸
= ¨
¨
¸
R DCR + (D x R DSON)¸
VIN ¹ ¨
Dc ©
¸
2R
¨ 1+
¸
c
D
OUT
©
¹
where
•
RDCR is the Inductor series resistance
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Thermal Considerations (continued)
VOUT
ROUT =
ILED
(35)
If the loss elements are reduced to zero, the conversion ratio simplifies to Equation 36.
VOUT
VIN
V OUT
V IN
=
=
1
'¶
(36)
K
'¶
(37)
Therefore:
K = Dc
VOUT
VIN
D c x VD
§
·
1¸
¨
VIN
¨
¸
=¨
R
+ (D x R DSON) ¸
¸
¨ 1 + DCR
¸
¨
2
D c R OUT
¹
©
(38)
Only calculations for determining the most significant power losses are discussed. Other losses totaling less than
2% are not discussed.
A simple efficiency calculation that takes into account the conduction losses is Equation 39.
§
Dc x VD
·
1¨
¸
VIN
¨
¸
K|¨
R
+ (D x R DSON ) ¸
¸
¨ 1 + DCR
¸
¨
2R
c
D
©
OUT
¹
(39)
The diode, NMOS switch, and inductor (DCR) losses are included in this calculation. Setting any loss element to
zero simplifies the equation.
VD is the forward voltage drop across the Schottky diode. It can be obtained from Electrical Characteristics.
Conduction losses in the diode are calculated with Equation 40.
PDIODE = VD × ILED
(40)
Depending on the duty cycle, this can be the single most significant power loss in the circuit. Choose a diode that
has a low forward voltage drop. Another concern with diode selection is reverse leakage current. Depending on
the ambient temperature and the reverse voltage across the diode, the current being drawn from the output to
the NMOS switch during time (D) could be significant, this may increase losses internal to the LM3410 or
LM3410-Q1 and reduce the overall efficiency of the application. See the Schottky diode manufacturer’s data
sheets for reverse leakage specifications.
Another significant external power loss is the conduction loss in the input inductor. The power loss within the
inductor can be simplified to Equation 41,
PIND = IIN2RDCR
(41)
Or Equation 42.
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Thermal Considerations (continued)
·
§I 2 R
PIND = ¨ O DCR ¸¸
¨ D'
¹
©
(42)
The LM3410 and LM3410-Q1 conduction loss is mainly associated with the internal power switch.
PCOND-NFET = I2SW-rms × RDS(ON) × D
(43)
'i
IIN
ISW(t)
t
Figure 37. LM3410 and LM3410-Q1 Switch Current
2
Isw rms
IIND
1 § 'i ·
Du 1
¨
¸ | IIND D
3 © IIND ¹
(44)
(small ripple approximation)
PCOND-NFET = IIN2 × RDS(ON) × D
(45)
Or
2
PCOND - NFET
§I LED·
= ¨ ' ¸ x R DSON x D
©D ¹
(46)
The value for RDS(ON) must be equal to the resistance at the desired junction temperature for analyzation. As an
example, at 125°C and RDS(ON) = 250 mΩ (See Typical Characteristics for value).
Switching losses are also associated with the internal power switch. They occur during the switch ON and OFF
transition periods, where voltages and currents overlap resulting in power loss.
The simplest means to determine this loss is empirically measuring the rise and fall times (10% to 90%) of the
switch at the switch node.
PSWR = 1/2 (VOUT × IIN × fSW × tRISE)
PSWF = 1/2 (VOUT × IIN × fSW × tFALL)
PSW = PSWR + PSWF
(47)
(48)
(49)
Table 25. Typical Switch-Node Rise and Fall Times
VIN (V)
VOUT (V)
tRISE (ns)
tFALL (ns)
3
5
6
4
5
12
6
5
3
12
8
7
5
18
10
8
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10.3.3.1 Quiescent Power Losses
IQ is the quiescent operating current, and is typically around 1.5 mA.
PQ = IQ × VIN
(50)
10.3.3.2 RSET Power Losses
RSET power loss is calculated with Equation 51.
2
VFB
PRSET =
R SET
(51)
10.3.4 Example Efficiency Calculation
Operating Conditions:5 × 3.3-V LEDs + 190 mVREF ≊ 16.7 V
Table 26. Operating Conditions
PARAMETER
VALUE
VIN
3.3 V
VOUT
16.7 V
ILED
50 mA
VD
0.45 V
fSW
1.6 MHz
IQ
3 mA
tRISE
10 ns
tFALL
10 ns
RDS(ON)
225 mΩ
LDCR
75 mΩ
D
0.82
IIN
0.31 A
ΣPCOND + PSW + PDIODE + PIND + PQ = PLOSS
(52)
Quiescent Power Loss:
PQ = IQ × VIN = 10 mW
(53)
Switching Power Loss:
PSWR = 1/2(VOUT × IIN × fSW × tRISE) ≊ 40 mW
PSWF = 1/2(VOUT × IIN × fSW × tFALL) ≊ 40 mW
PSW = PSWR + PSWF = 80 mW
(54)
(55)
(56)
Internal NFET Power Loss:
RDS(ON) = 225 mΩ
PCONDUCTION = IIN2 × D × RDS(ON) = 17 mW
IIN = 310 mA
(57)
(58)
(59)
Diode Loss:
VD = 0.45 V
PDIODE = VD × ILED = 23 mW
(60)
(61)
Inductor Power Loss:
RDCR = 75 mΩ
PIND = IIN2 × RDCR = 7 mW
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(62)
(63)
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Table 27. Total Power Losses
PARAMETER
VALUE
LOSS PARAMETER
LOSS VALUE
—
—
VIN
3.3 V
VOUT
16.7 V
ILED
50 mA
POUT
VD
0.45 V
PDIODE
fSW
1.6 MHz
IQ
10 ns
PSWR
40 mW
tRISE
10 ns
PSWF
40 mW
IQ
3 mA
PQ
10 mW
RDS(ON)
225 mΩ
PCOND
17 mW
LDCR
75 mΩ
PIND
D
0.82
η
85%
—
—
825 W
23 mW
—
—
7 mW
—
PLOSS
—
137 mW
PINTERNAL = PCOND + PSW = 107 mW
(64)
10.3.5 Calculating RθJA and RΨJC
R TJA =
TJ - TA
PDissipatio n
: R
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