User's Guide
SNOA559B – October 2011 – Revised May 2013
AN-2127 LM3448 A19 Edison Retrofit Evaluation Board
1
Introduction
This demonstration board highlights the performance of a LM3448 non-isolated LED driver solution that
can be used to power a single LED string consisting of eight to twelve series connected LEDs from a 85
VRMS to 135 VRMS, 60 Hz input power supply.
This is a two-layer board using the bottom and top layer for component placement. The demonstration
board can be modified to adjust the LED forward current, the number of series connected LEDs that are
driven and the switching frequency. The topology used for this evaluation board eliminates the need for
passive power factor correction and results in high power factor with minimal component count which
results in a size that can fit in a standard A19 Edison socket. This board will also operate correctly and
dim smoothly using most standard TRIAC dimmers.
Refer to the LM3448 Phase Dimmable Offline LED Driver with Integrated FET (SNOSB51) data sheet for
detailed information regarding the LM3448 device. A schematic and layout have also been included along
with measured performance characteristics. A bill of materials is also included that describes the parts
used on this demonstration board.
2
Key Features
•
•
•
•
3
Applications
•
•
•
•
4
Drop-in compatibility with TRIAC dimmers
Line injection circuitry enables PFC values greater than 0.85
Adjustable LED current and switching frequency
Flicker free operation
Retrofit TRIAC Dimming
Solid State Lighting
Industrial and Commercial Lighting
Residential Lighting
Performance Specifications
Based on an LED Vf = 3V
Symbol
Parameter
Min
Typ
Max
VIN
Input voltage
85VRMS
120VRMS
135VRMS
VOUT
LED string voltage
-
36V
-
ILED
LED string average current
-
181mA
-
POUT
Output power
-
6.5W
-
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1
Performance Specifications
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Figure 1. Demo Board
LED CURRENT (mA)
200
150
100
50
0
20
40
60
80
100
INPUT VOLTAGE (VRMS)
120
Figure 2. LED Current vs. Line Voltage (using TRIAC Dimmer)
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Typical Performance Characteristics
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5
Typical Performance Characteristics
TJ=25°C and VCC=12V, unless otherwise specified.
84
0.90
12 LEDs
10 LEDs
8 LEDs
0.88
POWER FACTOR
EFFICIENCY (%)
82
80
78
12 LEDs
10 LEDs
8 LEDs
0.86
0.84
0.82
76
0.80
74
80
90
100 110 120 130
INPUT VOLTAGE (VRMS)
Figure 3. Efficiency vs. Line Voltage
350
90
100 110 120 130
INPUT VOLTAGE VRMS
140
Figure 4. Power Factor vs. Line Voltage
8
12 LEDs
10 LEDs
8 LEDs
12 LEDs
10 LEDs
8 LEDs
7
250
POUT(W)
LED CURRENT (mA)
300
0.78
80
140
200
6
150
5
100
50
4
80
90
100 110 120 130
INPUT VOLTAGE (VRMS)
140
80
90
100 110 120 130
INPUT VOLTAGE VRMS
140
Figure 5. LED Current vs. Line Voltage
Figure 6. Output Power vs. Line Voltage
Figure 7. SW FET Drain Voltage Waveform
(VIN=120VRMS, 12 LEDs, ILED=181mA)
Figure 8. COFF Voltage (CH1), Inductor Current (CH4)
(VIN=120VRMS, 12 LEDs, ILED=181mA)
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EMI Performance
6
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EMI Performance
120V, 6.5W Conducted EMI Scans
4
Figure 9. LINE – CISPR/FCC Class B Peak Scan
Figure 10. NEUTRAL – CISPR/FCC Class B Peak Scan
Figure 11. LINE – CISPR/FCC Class B Average Scan
Figure 12. NEUTRAL – CISPR/FCC Class B Average
Scan
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Circuit Operation With Forward Phase TRIAC Dimmer
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7
Circuit Operation With Forward Phase TRIAC Dimmer
The dimming operation of the circuit was verified using a forward phase TRIAC dimmer. Waveforms
captured at different dimmer settings are shown below:
Figure 13. Forward phase circuit at full brightness
Figure 14. Forward phase circuit at 90° firing angle
Figure 15. Forward phase circuit at 135° firing angle
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Circuit Operation With Reverse Phase Dimmer
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Circuit Operation With Reverse Phase Dimmer
The circuit operation was also verified using a reverse phase dimmer and waveforms captured at different
dimmer settings are shown below:
Figure 16. Reverse phase circuit at full brightness
Figure 17. Reverse phase circuit at 90° firing angle
Figure 18. Reverse phase circuit at 135° firing angle
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Thermal Performance
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9
Thermal Performance
The board temperature was measured using an IR camera (HIS-3000, Wahl) while running under the
following conditions: VIN = 120VRMS, ILED = 181mA, # of LEDs = 12, POUT = 6.5W.
NOTE: Thermal performance is highly dependent on the user's final end-application enclosure, heatsinking methods, ambient operating temperature, and PCB board layout in addition to the electrical
operating conditions. This LM3448 evaluation board is optimized to supply 6.5W of output power at room
temperature without exceeding the thermal limitations of the LM3448. However higher output power levels
can be achieved if precautions are taken not to exceed the power dissipation limits of the LM3448
package or die junction temperature. Please see the LM3448 datasheet for additional details regarding its
thermal specifications.
•
•
•
•
•
Cursor
Cursor
Cursor
Cursor
Cursor
1: 65.3°C
2: 60.1°C
3: 67.6°C
4: 64.9°C
5: 65.6°C
Figure 19. Top Side - Thermal Scan
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Thermal Performance
•
•
•
•
Cursor
Cursor
Cursor
Cursor
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1: 68.1°C
2: 64.7°C
3: 62.6°C
4: 61.7°C
Figure 20. Bottom Side - Thermal Scan
8
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LM3448 Device Pin-Out
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10
LM3448 Device Pin-Out
SW
1
16 SW
SW
2
15 SW
NC
3
14 NC
BLDR
4
13 ISNS
GND
5
12 GND
VCC
6
11 FLTR2
ASNS
7
10 COFF
FLTR1
8
9
DIM
Figure 21. Device Pin-Out
Table 1. Pin Description 16 Pin Narrow SOIC
Pin #
Name
1, 2, 15,
16
SW
Description
Drain connection of internal 600V MOSFET.
3, 14
NC
No connect. Provides clearance between high voltage and low voltage pins. Do not tie to GND.
4
BLDR
Bleeder pin. Provides the input signal to the angle detect circuitry. A 230Ω internal resistor ensures BLDR is
pulled down for proper angle sense detection.
5, 12
GND
Circuit ground connection.
6
VCC
7
ASNS
PWM output of the TRIAC dim decoder circuit. Outputs a 0 to 4V PWM signal with a duty cycle proportional
to the TRIAC dimmer on-time.
8
FLTR1
First filter input. The 120Hz PWM signal from ASNS is filtered to a DC signal and compared to a 1 to 3V,
5.85 kHz ramp to generate a higher frequency PWM signal with a duty cycle proportional to the TRIAC
dimmer firing angle. Pull above 4.9V (typical) to TRI-STATE® DIM.
9
DIM
Input/output dual function dim pin. This pin can be driven with an external PWM signal to dim the LEDs. It
may also be used as an output signal and connected to the DIM pin of other LM3448/LM3445 devices or
LED drivers to dim multiple LED circuits simultaneously.
10
COFF
OFF time setting pin. A user set current and capacitor connected from the output to this pin sets the
constant OFF time of the switching controller.
11
FLTR2
Second filter input. A capacitor tied to this pin filters the PWM dimming signal to supply a DC voltage to
control the LED current. Could also be used as an analog dimming input.
13
ISNS
Input voltage pin. This pin provides the power for the internal control circuitry and gate driver. Connect a
22µF (minimum) bypass capacitor to ground.
LED current sense pin (internally connected to MOSFET source). Connect a resistor from ISNS to GND to
set the maximum LED current.
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Demo Board Wiring Overview
Demo Board Wiring Overview
LED -
TP4
LED +
TP3
TP1 LINE
J5
J10
11
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TP2
NEUTRAL
Figure 22. Wiring Connection Diagram
Table 2. Test Points
12
Test
Point
Name
I/O
Description
TP3
LED +
Output
LED Constant Current Supply
Supplies voltage and constant-current to anode of LED string.
TP4
LED -
Output
LED Return Connection (not GND)
Connects to cathode of LED string. Do NOT connect to GND.
TP1
LINE
Input
AC Line Voltage
Connects directly to AC line or output of TRIAC dimmer of a 120VAC system.
TP2
NEUTRAL
Input
AC Neutral
Connects directly to AC neutral of a 120VAC system.
Demo Board Assembly
Figure 23. Top View
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Design Guide
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Figure 24. Bottom View
13
Design Guide
D1
V+
V+
R1
R2
C1
D2
C10
R3
R7
C6
Q1
C16
VLED+
R4
R22
C5
D8
VCC
D7
D4
C4
+
C3
C8
R8
C2
L1
VLED±
L2
L3
C13
R5
R6
LM3448
LINE
9 DIM
NEUTRAL
FLTR1 8
R9
COFF
LINE EMI FILTER
10 COFF
ASNS 7
VCC
C14
R15
C15
11 FLTR2
VCC 6
12 GND
GND 5
13 ISNS
BLDR 4
R16
COFF
R14
14 NC
NC 3
15 SW
SW 2
16 SW
SW 1
U1
C12
COFF Current Source
Figure 25. Evaluation Board Schematic
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Design Guide
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13.1 Buck Converter
The following section explains how to design a non-isolated buck converter using the LM3448. Refer to
the LM3448 datasheet for specific details regarding the function of the LM3448 device. All reference
designators refer to the Evaluation Board Schematic in Figure 25 unless otherwise noted. The circuit
operates in open-loop based on a constant off-time that is set by selecting appropriate circuit components.
Like an incandescent lamp, the driver is compatible with both forward and reverse phase dimmers.
AC-Coupled Line Injection
By injecting a voltage VINJECT which is proportional to the line voltage into the FLTR2 pin (see Figure 26),
input current shaping is obtained which improves power factor performance. By AC-coupling the VINJECT
signal through capacitor C14, improved line-regulation of the LED current is also achieved (see
Figure 27).
VINJECT
t
Figure 26. FLTR2 Waveform with No Dimmer
V+
R2
LM3448
R7
C14
VINJECT
R15
11 FLTR2
C15
Figure 27. AC-Coupled Line-Injection Circuit
Figure 28 shows how line shaping of the input current is implemented. Peak voltage at the FLTR2 pin
should be kept below 1.25V otherwise current limit will be tripped. A good starting point is to set up the
resistor divider consisting of resistors R2, R7 and R15 to provide a VINJECT peak input voltage of 1.0V at
the input of capacitor C14 at the nominal input voltage. Recommended values for the AC-coupling
capacitor C14 is 0.47µF and for the FLTR2 capacitor C15 is 0.1µF.
With a 1.0V VINJECT voltage, the voltage at the FLTR2 pin at the maximum and minimum input voltages can
be calculated using the following equations,
(1)
These VFLTR2 voltages will be used later to determine ripple and peak inductor currents.
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750 mV
50k
DIM DECODER
ASNS
As line voltage increases, the voltage across the
inductor increases, and the peak current increases.
370k
Tri-State
4.9V
RFLTR1
PWM
I-LIM
FLTR1
RAMP
LED Current
1.27V
CFLTR1
RAMP GEN.
5.9 kHz
3V
1V
1k
ISNS
1V
RSNS
DIM
LEADING EDGE BLANKING
FLTR2
The PWM reference increases
as the line voltage increases.
GND
125 ns
CFLTR2
Figure 28. Typical Operation of FLTR2 Pin
Off-time, On-time and Switching Frequency
The AC mains voltage at the line frequency fL is assumed to be perfectly sinusoidal and the diode bridge
ideal. This yields a perfect rectified sinusoid at the input to the buck converter. The maximum, nominal and
minimum peak input voltages are defined as follows,
(2)
The LM3448 will operate as a constant off-time regulator, and so tOFF will be constant throughout all
operating points. The on-time tON (and subsequently the switching frequency fSW) will vary depending on
input voltage and LED stack voltage values. For this buck converter operating in continuous conduction
mode (CCM), the minimum on-time tON(MIN) can be determined for a maximum desired switching frequency
fSW(MAX)at the maximum peak input voltage,
(3)
The off-time tOFF is now calculated where TS(MIN) is the minimum switching period,
(4)
It is important to note that there is a minimum on-time of 200ns that needs to be met in order for proper
LED driver operation.
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Design Guide
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Output Power and Current Sense Resistor
322
9.8
250
9.1
298
9.1
230
8.3
274
8.3
210
7.6
250
7.6
190
6.8
226
6.8
170
6.1
202
6.1
150
5.3
178
5.3
130
4.6
154
4.6
3.8
130
110
1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0
POUT (W)
9.8
ILED (mA)
270
POUT (W)
ILED (mA)
Due to the interaction of the AC-coupled line-injection voltage with the FLTR2 signal, the equations for
determining the correct sense resistor RSNS (shown as R14 in the evaluation board schematic) for a
desired output power POUT are complex and beyond the scope of this document. Instead, performance
graphs showing the relationship between LED current, POUT and RSNS are shown in Figure 29, Figure 30
and Figure 31 for common stack voltages of 8, 10 and 12 LEDs. By referring to these graphs, users can
choose R14 values that will meet their LED current and output power requirements.
3.8
1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 2.1 2.2 2.3 2.4 2.5 2.6
2.1 2.2 2.3 2.4 2.5 2.6
RSNS (W)
RSNS (W)
Figure 30. ILED vs. POUT vs. RSNS
for 10 LEDs (Vf=3.0V)
400
9.8
370
9.1
340
8.3
310
7.6
280
6.8
250
6.1
220
5.3
190
4.6
POUT (W)
ILED (mA)
Figure 29. ILED vs. POUT vs. RSNS
for 12 LEDs (Vf=3.0V)
160
3.8
1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 2.1 2.2 2.3 2.4 2.5 2.6
RSNS (W)
Figure 31. ILED vs. POUT vs. RSNS
for 8 LEDs (Vf=3.0V)
Inductor
Peak inductor currents will need to be calculated as shown below based on the VFLTR2 voltages and chosen
sense resistor R14 at the maximum and minimum peak input voltages,
(5)
14
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Inductor ripple current will need to be specified by the user based on desired EMI performance, inductor
size and other operating conditions. The following equations show how to calculate for maximum and
minimum inductor ripple currents respectively by basing the ripple (i.e.ΔiL(%) as a percentage of maximum
peak inductor currents,
(6)
It is recommended that this buck converter design operate in CCM over the full range of operating peak
input voltages, and so the minimum inductor peak current at VIN-PK(MIN) should not go below zero,
(7)
The inductor value can be calculated based on the minimum on-time, LED output voltage and the
specified inductor ripple current ΔiL-PK(VIN-PK-MAX) at the maximum peak input voltage as described below,
(8)
COFF Current Source
The current source used to establish the constant off-time is shown in Figure 32.
VCC
R16
COFF
C12
Figure 32. COFF Current Source Circuit
Capacitor C12 will be charged with current from the VCC supply through resistor R16. The COFF pin
threshold will therefore be tripped based on the following capacitor equation,
(9)
where,
(10)
Solving for off-time tOFF results in,
(11)
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Re-arranging the above equation results in R16 being calculated where C12 is typically chosen as value
around 470pF,
(12)
Additionally, the maximum on-time tON(MAX) and corresponding minimum switching frequency fSW(MIN) and
maximum switching period TS(MAX) occur at the minimum peak input voltage. Using the previously
calculated inductor value, these values can now be calculated as,
(13)
Maximum and minimum duty cycles, DMAX and DMIN, will occur at the minimum and maximum peak input
voltages respectively,
(14)
Switching MOSFET (SW FET)
Peak and RMS SW FET currents are calculated along with maximum SW FET power dissipation based on
the SW FET RDS-ON value using the following equations,
(15)
(16)
and,
(17)
Current Limit
The peak inductor current limit ILIM should be approximately 25% higher than the maximum operating peak
inductor current,
(18)
The sense resistor will need to be able to dissipate the maximum power,
(19)
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Re-circulating Diode
The main re-circulating diode (D4) should be sized to block the maximum reverse voltage VRD4(MAX),
operate at the maximum peak IDR-PK(MAX) and RMS currents ID4-RMS(MAX), and dissipate the maximum power
PD4(MAX) as determined by the following equations,
(20)
(21)
(22)
(23)
NOTE: For proper converter operation, the chosen diode should have a reverse recovery time that is less
than the LM3448's leading edge blanking time of 125ns.
13.2 Bias Supplies and Capacitances
The VCC bias supply circuit is shown in Figure 33. The passFET (Q1) is used in its linear region to standoff the line voltage from the LM3448 regulator. Both the VCC startup current and discharging of the EMI
filter capacitance for proper phase angle detection are handled by Q1. Therefore Q1 has to block the
maximum peak input voltage and have both sufficient surge and power handling capability with regards to
its safe operating area (SOA). The design equations are,
(24)
(25)
(26)
Note that if additional TRIAC holding current is to be sourced through Q1, then the transistor will need to
be sized appropriately to handle the additional current and power dissipation requirements.
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V+
R1
R3
Q1
VCC
R22
D8
D7
C8
R8
LM3448
VCC 6
Figure 33. Bias Supply Circuit
Input Capacitance
The input capacitors C1 and C10 have to be able to provide energy during the worst-case switching period
at the peak of the AC voltage input. They should be high frequency, high stability capacitors (usually
metallized film capacitors, either polypropylene or polyester) with an AC voltage rating equal to the
maximum input voltage. They should also have a DC voltage rating exceeding the maximum peak input
voltage plus half of the peak to peak input voltage ripple specification. The minimum required input
capacitance is calculated given the same ripple specification,
(27)
Output Capacitance
C3 should be a high quality electrolytic capacitor with a voltage rating greater than the specified LED stack
voltage. Given the desired voltage ripple, the minimum output capacitance is calculated,
(28)
13.3 Input Filter
Background
Since the LM3448 is used for AC to DC systems, electromagnetic interference (EMI) filtering is critical to
pass the necessary standards for both conducted and radiated EMI. This filter will vary depending on the
output power, the switching frequencies, and the layout of the PCB. There are two major components to
EMI: differential noise and common-mode noise. Differential noise is typically represented in the EMI
spectrum below approximately 500kHz while common-mode noise shows up at higher frequencies.
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LINE
R5
L1
C5
C16
C6
D2
C2
V+
R4
NEUTRAL
R6
L2
Figure 34. Input EMI Filter
Conducted
Figure 34 shows a typical filter used with this LM3448 flyback design. In order to conform to conducted
standards, a fourth order filter is implemented using inductors and "X" rated AC capacitors. If sized
properly, this filter design can provide ample attenuation of the switching frequency and lower order
harmonics contributing to differential noise. This combination of filter components along with any
necessary damping can easily provide a passing conducted EMI signature.
Radiated
Conforming to radiated EMI standards is much more difficult and is completely dependent on the entire
system including the enclosure. Reduction of dV/dt on switching edges and PCB layout iterations are
frequently necessary. Consult available literature and/or an EMI specialist for help with this. Several
iterations of component selection and layout changes may be necessary before passing a specific
radiated EMI standard.
Interaction with Dimmers
In general input filters and forward phase dimmers do not work well together. The TRIAC needs a
minimum amount of holding current to function. The converter itself is demanding a certain amount of
current from the input to provide to its output, and the input filter is providing or taking current depending
upon the dV/dt of the capacitors. The best way to deal with this problem is to minimize filter capacitance
and increase the regulated hold current until there is enough current to satisfy the dimmer and filter
simultaneously.
13.4 Inrush Limiting and Damping
Inrush
With a forward phase dimmer, a very steep rising edge causes a large inrush current every cycle as
shown in Figure 35. Series resistance (R5, R6) can be placed between the filter and the TRIAC to limit the
effect of this current on the converter and to provide some of the necessary holding current at the same
time. This will degrade efficiency but some inrush protection is always necessary in any AC system due to
startup. The size of R5 and R6 are best found experimentally as they provide attenuation for the whole
system.
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Design Calculations
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Triac Fires Æ Inrush Spike
Iin(t)
0
t
Potential Misfire
Figure 35. Inrush Current Spike
Damper
The inrush spike can also excite a resonance between the input filter of the TRIAC and the input filter of
the converter. The associated interaction can cause the current to ring negative, as shown in Figure 35,
thereby shutting off the TRIAC. A TRIAC damper can be placed between the dimmer and the EMI filter to
absorb some of the ringing energy and reduce the potential for misfires. The damper is also best sized
experimentally due to the large variance in TRIAC input filters. Resistors R5 and R6 can also be increased
to help dampen the ringing at the expense of some efficiency and power factor performance.
14
Design Calculations
The following is a step-by-step procedure with calculations for a 120V, 6.5W non-isolated buck converter
design.
14.1 Specifications
VIN(MAX) = 135VAC
VIN(NOM) = 120VAC
VIN(MIN) = 85VAC
POUT = 6.5W
VOUT = 36V
ILED = 181mA
Efficiency,η = 80%
fL = 60Hz
fSW(MAX) =75kHz
TS(MIN) =13.33µs
ΔvOUT = 1V
ΔvIN-PK = 35V
SW FET VDS(MAX) = 600V
SW FET RDS-ON = 3.5Ω
Vf(D4) = 0.8V
VCC = 12V
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VZ(D7)=12V
R8=49.9kΩ
VGS(Q1)=0.7V
14.2 Preliminary Calculations
Nominal peak input voltage:
(29)
Calculate minimum on-time and verify it's greater than 200ns:
(30)
Calculate off-time:
(31)
From Figure 29, choose R14=2.0Ω for 6.5W output power with 12 LEDs.
14.3 FLTR2 AC-LINE Injection
Choose VINJECT(NOM)=1.0V
Choose R2=R7=274kΩ
Calculate R15:
(32)
or,
(33)
Calculate maximum FLTR2 pin voltage and verify it is less than 1.25V:
(34)
Calculate minimum FLTR2 pin voltage:
(35)
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21
Design Calculations
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14.4 Inductor
Calculate peak inductor currents at the minimum and maximum peak input voltages:
(36)
Calculate inductor ripple currents at the minimum and maximum peak input voltages based on 80% of
maximum peak inductor currents:
(37)
Verify that converter is in CCM operation at the minimum peak input voltage:
(38)
Calculate inductor value:
(39)
14.5 COFF Current Source
Choose capacitor C12=470pF.
Calculate resistor R16:
(40)
Calculate maximum on-time, minimum switching frequency and maximum switching period:
(41)
Calculate maximum and minimum duty cycles:
(42)
22
AN-2127 LM3448 A19 Edison Retrofit Evaluation Board
SNOA559B – October 2011 – Revised May 2013
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Copyright © 2011–2013, Texas Instruments Incorporated
Design Calculations
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14.6 SW FET
Calculate maximum peak SW FET current:
(43)
Calculate maximum RMS SW FET current:
(44)
Calculate maximum power dissipation:
(45)
14.7 Current Limit
Calculate peak inductor current limit:
(46)
Power dissipation:
(47)
Resulting component choice:
(48)
14.8 Re-circulating Diode
Maximum reverse blocking voltage:
(49)
Maximum peak diode current:
(50)
Maximum RMS diode current:
(51)
Maximum power dissipation:
(52)
Resulting component choice:
(53)
SNOA559B – October 2011 – Revised May 2013
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23
Design Calculations
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14.9 PassFET
Calculate maximum peak voltage:
(54)
Calculate current:
(55)
Calculate maximum power dissipation:
(56)
Resulting component choice:
(57)
14.10 Input Capacitance
Minimum capacitance:
(58)
AC Voltage rating:
(59)
DC Voltage rating:
(60)
Resulting component choice:
(61)
14.11 Output Capacitance
Minimum capacitance:
(62)
Voltage rating:
(63)
Resulting component choice:
(64)
24
AN-2127 LM3448 A19 Edison Retrofit Evaluation Board
SNOA559B – October 2011 – Revised May 2013
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Copyright © 2011–2013, Texas Instruments Incorporated
Evaluation Board Schematic
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15
Evaluation Board Schematic
D1
V+
V+
R1
R2
C1
D2
C10
R3
R7
C6
Q1
C16
VLED+
R4
R22
C5
D8
VCC
D7
D4
C4
+
C3
C8
R8
C2
L1
VLED±
L2
L3
C13
R5
R6
LM3448
LINE
9 DIM
NEUTRAL
FLTR1 8
R9
COFF
LINE EMI FILTER
10 COFF
ASNS 7
VCC
C14
R15
C15
11 FLTR2
VCC 6
12 GND
GND 5
13 ISNS
BLDR 4
R16
COFF
R14
14 NC
NC 3
15 SW
SW 2
16 SW
SW 1
U1
C12
COFF Current Source
WARNING
The LM3448 evaluation board has exposed high voltage
components that present a shock hazard. Caution must be taken
when handling the evaluation board. Avoid touching the evaluation
board and removing any cables while the evaluation board is
operating. Isolating the evaluation board rather than the
oscilloscope is highly recommended.
SNOA559B – October 2011 – Revised May 2013
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Bill of Materials
16
26
www.ti.com
Bill of Materials
Part ID
Description
Manufacturer
Part Number
C1, C10
CAP CER 47000PF 500V X7R 1210
Johanson Dielectrics
501S41W473KV4E
C2, C6
CAP FILM MKP .015UF 310VAC X2
Vishay/BC Comp
BFC233820153
C3
CAP 470UF 50V ELECT PW RADIAL
Nichicon
UPW1H471MHD
C4
DNP
DNP
DNP
C5, C16
CAP CER .15UF 250V X7R 1210
TDK
C3225X7R2E154K
C8
Ceramic, X5R, 16V, 20%
MuRata
GRM32ER61C476ME15L
C12
Ceramic, X7R, 50V, 10%
MuRata
GRM188R71H471KA01D
C13, C15
Ceramic, X7R, 16V, 10%
MuRata
GRM188R71C104KA01D
C14
Ceramic, X7R, 16V, 10%
MuRata
GRM188R71C474KA88D
D1, D8
DIODE SCHOTTKY 1A 200V PWRDI 123
Diodes Inc.
DFLS1200-7
D2
RECT BRIDGE GP 400V 0.5A MINIDIP
Diodes Inc.
RH04DICT-ND
D4
DIODE FAST 1A 300V SMA
Fairchild
ES1F
D7
DIODE ZENER 15V 500MW SOD-123
Fairchild Semi
MMSZ5245B
J5, J10
CONN HEADER .312 VERT 2POS TIN
Tyco Electronics
1-1318301-2
L1, L2
INDUCTOR 4700UH .13A RADIAL
TDK Corp
TSL0808RA-472JR13-PF
L3
820uH, Shielded Drum Core,
Coilcraft Inc.
MSS1038-824KL
Q1
MOSFET N-CH 240V 260MA SOT-89
Infineon Technologies
BSS87 L6327
R1, R3
1%, 0.25W
Vishay-Dale
CRCW1206200kFKEA
R2, R7
1%, 0.25W
Vishay-Dale
CRCW1206274kFKEA
R4
RES 430 OHM 1/2W 5% 2010 SMD
Vishay\Dale
CRCW2010430RJNEF
R5, R6
RES 33 OHM 3W 5% AXIAL
TT Electronics/Welwyn
ULW3-33RJA1
R8
1%, 0.1W
Vishay-Dale
CRCW060349K9FKEA
R9
1%, 0.1W
Vishay-Dale
CRCW060348K7FKEA
R14
RES, 2.00 ohm, 1%, 0.25W, 1206
Vishay-Dale
CRCW12062R00FNEA
R15
RES, 3.16k ohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW06033K16FKEA
R16
RES, 226k ohm, 1%, 0.1W, 0603
Vishay-Dale
CRCW0603226KFKEA
R22
1%, 0.125W
Vishay-Dale
CRCW080540R2FKEA
TP1, TP2,
TP3, TP4
Terminal, Turret, TH, Double
Keystone Electronics
1502-2
U1
LM3448 LED Driver
Texas Instruments
LM3448
AN-2127 LM3448 A19 Edison Retrofit Evaluation Board
SNOA559B – October 2011 – Revised May 2013
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PCB Layout
www.ti.com
17
PCB Layout
NOTE: Spacing between traces and components of this evaluation board are based on high voltage
recommendations for designs that will be potted. Users are cautioned to satisfy themselves as to the
suitability of this design for the intended end application and take any necessary precautions where high
voltage layout and spacing rules must be followed.
Figure 36. Top Layer
Figure 37. Bottom Layer
SNOA559B – October 2011 – Revised May 2013
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Copyright © 2011–2013, Texas Instruments Incorporated
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