LM34914SDX

LM34914 www.ti.com SNVS453B – MAY 2006 – REVISED MARCH 2013 LM34914 Ultra Small 1.25A Step-Down Switching Regulator with Intelligent Current Limit Check for Samples: LM34914 FEATURES DESCRIPTION • • • The LM34914 Step-Down Switching Regulator features all the functions needed to implement a low cost, efficient, buck bias regulator capable of supplying at least 1.25A to the load. To reduce excessive switch current due to the possibility of a saturating inductor the valley current limit threshold changes with input and output voltages, and the ontime is reduced when current limit is detected. This buck regulator contains a 44V N-Channel Buck Switch, and is available in the thermally enhanced 3 mm x 3 mm WSON-10 package. The feedback regulation scheme requires no loop compensation, results in fast load transient response, and simplifies circuit implementation. The operating frequency remains constant with line and load variations due to the inverse relationship between the input voltage and the on-time. The valley current limit results in a smooth transition from constant voltage to constant current mode when current limit is detected, reducing the frequency and output voltage, without the use of foldback. Additional features include: VCC undervoltage lock-out, thermal shutdown, gate drive undervoltage lock-out, and maximum duty cycle limit. 1 2 • • • • • • • • • • Input Voltage Range: 8V to 40V Integrated N-Channel Buck Switch Valley Current Limit Varies with VIN and VOUT to Reduce Excessive Inductor Current On-time is Reduced when in Current Limit Integrated Start-Up Regulator No Loop Compensation Required Ultra-Fast Transient Response Maximum Switching Frequency: 1.3 MHz Operating Frequency Remains Nearly Constant with Load Current and Input Voltage Variations Programmable Soft-Start Precision Internal Reference Adjustable Output Voltage Thermal Shutdown TYPICAL APPLICATIONS • • • High Efficiency Point-Of-Load (POL) Regulator Non-Isolated Buck Regulator Secondary High Voltage Post Regulator Package • • WSON-10 (3 mm x 3mm) Exposed Thermal Pad Dissipation For Improved Heat Basic Step Down Regulator 8V - 40V Input VCC VIN C1 C3 LM34914 RON BST RON/SD L1 C4 SHUT DOWN VOUT SW D1 SS R1 R3 ISEN C5 FB RTN C2 R2 SGND 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2006–2013, Texas Instruments Incorporated LM34914 SNVS453B – MAY 2006 – REVISED MARCH 2013 www.ti.com Connection Diagram 1 2 3 4 5 10 SW VIN BST VCC ISEN RON/SD SGND SS RTN FB 9 8 7 6 10-Lead WSON PIN DESCRIPTIONS Pin Number Name Description Application Information 1 SW Switching Node Internally connected to the buck switch source. Connect to the inductor, diode, and bootstrap capacitor. 2 BST Boost pin for bootstrap capacitor Connect a 0.022 µF capacitor from SW to this pin. The capacitor is charged each off-time via an internal diode. 3 ISEN Current sense The re-circulating current flows out of this pin to the freewheeling diode. 4 SGND Sense Ground Re-circulating current flows into this pin to the current sense resistor. 5 RTN Circuit Ground Ground for all internal circuitry other than the current limit detection. 6 FB Feedback input from the regulated output Internally connected to the regulation and over-voltage comparators. The regulation level is 2.5V. 7 SS Softstart An internal current source charges an external capacitor to 2.5V, providing the softstart function. 8 RON/SD On-time control and shutdown An external resistor from VIN to this pin sets the buck switch on-time. Grounding this pin shuts down the regulator. 9 VCC Output from the startup regulator Nominally regulated at 7.0V. Connect a 0.1 µF capacitor from this pin to RTN. An external voltage (8V to 14V) can be applied to this pin to reduce internal dissipation. An internal diode connects VCC to VIN. 10 VIN Input supply voltage Operating input range is 8.0V to 40V. EP Exposed Pad Exposed metal pad on the underside of the device. It is recommended to connect this pad to the PC board ground plane to aid in heat dissipation. These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 2 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 LM34914 www.ti.com SNVS453B – MAY 2006 – REVISED MARCH 2013 Absolute Maximum Ratings (1) (2) VIN to RTN 44V BST to RTN 52V SW to RTN (Steady State) -1.5V BST to VCC 44V VIN to SW 44V BST to SW 14V VCC to RTN 14V SGND to RTN -0.3V to +0.3V Current out of ISEN See text SS to RTN -0.3V to 4V All Other Inputs to RTN -0.3 to 7V ESD Rating (3) Human Body Model 2kV Storage Temperature Range -65°C to +150°C JunctionTemperature 150°C (1) (2) (3) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For specifications and test conditions, see Electrical Characteristics. If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and specifications. The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin. Operating Ratings (1) VIN Voltage 8.0V to 40V −40°C to + 125°C Junction Temperature (1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For specifications and test conditions, see Electrical Characteristics. Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 12V, RON = 200kΩ (1) (2). Symbol Parameter Conditions Min Typ Max Units 6.6 7.0 7.4 V Start-Up Regulator, VCC VCCReg UVLOVCC (1) (2) (3) VCC regulated output Vin > 9V VIN-VCC dropout voltage ICC = 0 mA, VCC = UVLOVCC + 250 mV 1.3 V VCC output impedance (0 mA ≤ ICC ≤ 5 mA) VIN = 8V 155 VIN = 40V 0.16 VCC current limit (3) VCC = 0V 11 mA VCC under-voltage lockout threshold VCC increasing 5.7 V UVLOVCC hysteresis VCC decreasing 150 mV UVLOVCC filter delay 100 mV overdrive IIN operating current Non-switching, FB = 3V IIN shutdown current RON/SD = 0V Ω 3 µs 0.57 0.85 mA 80 160 µA For detailed information on soldering plastic WSON packages, visit www.ti.com/packaging. Typical specifications represent the most likely parametric norm at 25°C operation. VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 3 LM34914 SNVS453B – MAY 2006 – REVISED MARCH 2013 www.ti.com Electrical Characteristics (continued) Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 12V, RON = 200kΩ(1)(2). Symbol Parameter Conditions Min Typ Max Units 0.33 0.7 Ω 4.2 5.5 V Switch Characteristics Rds(on) Buck Switch Rds(on) ITEST = 200 mA UVLOGD Gate Drive UVLO VBST - VSW Increasing 3.0 UVLOGD hysteresis 470 mV VSS Pull-up voltage 2.5 V ISS Internal current source 12.5 µA Softstart Pin Current Limit ILIM Threshold VIN = 8V, VFB = 2.4V 1.0 1.2 1.4 VIN = 30V, VFB = 2.4V 0.9 1.1 1.3 VIN = 30V, VFB = 1.0V 0.85 1.05 1.25 Response time 150 A ns On Timer tON - 1 On-time (normal operation) VIN = 10V, RON = 200 kΩ tON - 2 On-time (normal operation) VIN = 40V, RON = 200 kΩ 2.1 655 ns tON - 3 On-time (current limit) VIN = 10V, RON = 200 kΩ 1.13 µs Shutdown threshold at RON/SD Voltage at RON/SD rising Shutdown Threshold hysteresis Voltage at RON/SD falling 0.4 2.8 0.8 3.4 1.2 µs V 32 mV 265 ns Off Timer tOFF Minimum Off-time Regulation and Over-Voltage Comparators (FB Pin) VREF FB regulation threshold SS pin = steady state 2.445 2.50 2.550 V FB over-voltage threshold 2.9 V FB bias current 15 nA 175 °C 20 °C Thermal Shutdown TSD Thermal shutdown temperature Junction temperature rising Thermal shutdown hysteresis Thermal Resistance (4) 4 θJA Junction to Ambient 0 LFPM Air Flow (4) 30 °C/W θJC Junction to Case (4) 8 °C/W Value shown assumes a 4-layer PC board and 4 vias to conduct heat from beneath the package. Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 LM34914 www.ti.com SNVS453B – MAY 2006 – REVISED MARCH 2013 Typical Performance Characteristics Unless otherwise specified the following conditions apply: TJ = 25°C Typical Efficiency Performance VCC vs VIN 100 7.5 Vin = 8V 95 90 24V VCC (V) EFFICIENCY (%) 7.0 12V 85 40V 80 75 6.5 6.0 5.5 VOUT = 5V FS = 275 kHz 70 0 200 400 600 800 5.0 1000 6.5 7.0 7.5 8.0 8.5 9.0 LOAD CURRENT (mA) VIN (V) Figure 1. Figure 2. VCC vs ICC ON-Time vs VIN and RON 8 10 VIN = 9V 6 ON-TIME (Ps) 4 3 2 600k 200k 3.0 VIN = 8V 5 VCC (V) 400k VIN t 10V 7 100k 1.0 0.3 VCC Externally Loaded 1 RON = 45k FS = 200 kHZ 0.1 0 2 0 4 8 6 ICC (mA) 10 0 12 10 Figure 3. 20 VIN (V) 40 30 Figure 4. Valley Current Limit Threshold vs. FB and VIN Voltage at the RON/SD Pin 3.0 1.3 1.2 VIN = 8V 15V 24V 1.1 34V 1.0 40V RON/SD PIN VOLTAGE (V) VALLEY CURRENT LIMIT THRESHOLD (A) RON = 45k 100k 2.0 500k 1.0 0 0.9 0 0.5 1.0 1.5 2.0 2.5 0 10 20 30 40 VIN (V) VFB (V) Figure 5. Figure 6. Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 5 LM34914 SNVS453B – MAY 2006 – REVISED MARCH 2013 www.ti.com Typical Performance Characteristics (continued) Unless otherwise specified the following conditions apply: TJ = 25°C Input Shutdown and Operating Current Into VIN INPUT CURRENT (PA) 800 600 Operating Current 400 200 Shutdown Current 0 0 10 20 30 40 VIN (V) Figure 7. 6 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 LM34914 www.ti.com SNVS453B – MAY 2006 – REVISED MARCH 2013 Typical Application Circuit and Block Diagram Input 8V - 40V 7V START-UP REGULATOR VIN C5 LM34914 VCC VCC THERMAL SHUTDOWN C3 UVLO C1 ON TIMER GND RON RON START ' Ton FINISH RON/SD MINIMUM OFF TIMER START FINISH 0.8V BST Gate Drive SD UVLO 2.5V SS 12.5 PA VIN C4 LOGIC LEVEL SHIFT Driver C6 L1 SW FB REGULATION COMPARATOR 2.9V VOUT D1 OVER-VOLTAGE COMPARATOR CURRENT LIMIT COMPARATOR + - RTN VIN FB CL Threshold Adjust RSENSE + ISEN R1 R3 R2 C2 41 m: SGND Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 7 LM34914 SNVS453B – MAY 2006 – REVISED MARCH 2013 www.ti.com VIN 7.0V UVLO VCC SW Pin Inductor Current 2.5V SS Pin VOUT t1 t2 Figure 8. Startup Sequence Functional Description The LM34914 Step Down Switching Regulator features all the functions needed to implement a low cost, efficient buck bias power converter capable of supplying at least 1.25A to the load. This high voltage regulator contains an N-Channel buck switch, is easy to implement, and is available in the thermally enhanced 3mm x 3mm WSON10 package. The regulator’s operation is based on a constant on-time control scheme where the on-time is determined by VIN. This feature results in the operating frequency remaining relatively constant with load and input voltage variations. The feedback control scheme requires no loop compensation resulting in very fast load transient response. The valley current limit scheme protects against excessively high currents if the output is short circuited when VIN is high. To aid in controlling excessive switch current due to a possible saturating inductor the valley current limit threshold changes with input and output voltages, and the on-time is reduced by approximately 50% when current limit is detected.The LM34914 can be applied in numerous applications to efficiently regulate down higher voltages. Additional features include: Thermal shutdown, VCC under-voltage lockout, gate drive under-voltage lock-out, and maximum duty cycle limit. Control Circuit Overview The LM34914 buck DC-DC regulator employs a control scheme based on a comparator and a one-shot on-timer, with the output voltage feedback (FB) compared to an internal reference (2.5V). If the FB voltage is below the reference the buck switch is switched on for a time period determined by the input voltage and a programming resistor (RON). Following the on-time the switch remains off until the FB voltage falls below the reference, but not less than the minimum off-time forced by the LM34914. The buck switch is then turned on for another on-time period. 8 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 LM34914 www.ti.com SNVS453B – MAY 2006 – REVISED MARCH 2013 When in regulation, the LM34914 operates in continuous conduction mode at heavy load currents and discontinuous conduction mode at light load currents. In continuous conduction mode the inductor’s current is always greater than zero, and the operating frequency remains relatively constant with load and line variations. The minimum load current for continuous conduction mode is one-half the inductor’s ripple current amplitude. The approximate operating frequency is calculated as follows: FS = VOUT x (VIN ± 1.5) 1.15 x 10-10 x (RON + 1.4k) x VIN (1) The buck switch duty cycle is equal to: DC = VOUT tON tON + tOFF = tON x FS = VIN (2) In discontinuous conduction mode, where the inductor’s current reaches zero during the off-time forcing a longerthan-normal off-time, the operating frequency is lower than in continuous conduction mode, and varies with load current. Conversion efficiency is maintained at light loads since the switching losses decrease with the reduction in load and frequency. The approximate discontinuous operating frequency can be calculated as follows: FS = VOUT2 x L1 x 1.5 x 1020 RL x RON2 (3) where RL = the load resistance, and L1 is the circuit’s inductor. The output voltage is set by the two feedback resistors (R1, R2 in the Block Diagram). The regulated output voltage is calculated as follows: VOUT = 2.5 x (R1 + R2) / R2 (4) Output voltage regulation is based on supplying ripple voltage to the feedback input (FB pin), normally obtained from the output voltage ripple through the feedback resistors. The LM34914 requires a minimum of 25 mVp-p of ripple voltage at the FB pin, requiring the ripple voltage at VOUT be higher by the gain factor of the feedback resistor ratio. The output ripple voltage is created by the inductor’s ripple current passing through R3 which is in series with the output capacitor. For applications where reduced ripple is required at VOUT, see Applications Information. If the voltage at FB rises above 2.9V, due to a transient at VOUT or excessive inductor current which creates higher than normal ripple at VOUT, the internal over-voltage comparator immediately shuts off the internal buck switch. The next on-time starts when the voltage FB falls below 2.5V and the inductor current falls below the current limit threshold. ON-Time Timer The on-time for the LM34914 is determined by the RON resistor and the input voltage (VIN), calculated from: 1.15 x 10 tON = -10 x (RON + 1.4k) (VIN - 1.5) + 50 ns (5) The inverse relationship with VIN results in a nearly constant frequency as VIN is varied. To set a specific continuous conduction mode switching frequency (FS), the RON resistor is determined from the following: VOUT x (VIN - 1.5) - 1.4k RON = -10 FS x 1.15 x 10 x VIN (6) Equation 1, Equation 5, and Equation 6 are valid only during normal operation - i.e., the circuit is not in current limit. When the LM34914 operates in current limit, the on-time is reduced by approximately 50%. This feature reduces the peak inductor current which may be excessively high if the load current and the input voltage are simultaneously high. This feature operates on a cycle-by-cycle basis until the load current is reduced and the output voltage resumes its normal regulated value. Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 9 LM34914 SNVS453B – MAY 2006 – REVISED MARCH 2013 www.ti.com Shutdown The LM34914 can be remotely shut down by taking the RON/SD pin below 0.8V. See Figure 9. In this mode the SS pin is internally grounded, the on-timer is disabled, and bias currents are reduced. Releasing the RON/SD pin allows the circuit to resume operation. The voltage at the RON/SD pin is normally between 1.5V and 3.0V, depending on VIN and the RON resistor. VIN Input Voltage RON LM34914 RON/SD STOP RUN Figure 9. Shutdown Implementation Current Limit Current limit detection occurs during the off-time by monitoring the recirculating current flowing out of the ISEN pin. Referring to the Typical Application Circuit and Block Diagram, during the off-time the inductor current flows through the load, into SGND, through the internal sense resistor, out of ISEN and through D1 to the inductor. If that current exceeds the current limit threshold the current limit comparator output delays the start of the next ontime period. The next on-time starts when the current out of ISEN is below the threshold and the voltage at FB falls below 2.5V. The operating frequency is typically lower due to longer-than-normal off-times. The valley current limit threshold is a function of the input voltage (VIN) and the output voltage sensed at FB, as shown in the graph “Valley Current Limit Threshold vs. VFB and VIN”. This feature reduces the inductor current’s peak value at high line and load. To further reduce the inductor’s peak current, the next cycle’s on-time is reduced by approximately 50% if the voltage at FB is below its threshold when the inductor current reduces to the current limit threshold (VOUT is low due to current limiting). Figure 10 illustrates the inductor current waveform during normal operation and in current limit. During the first “Normal Operation” the load current is IOUT1, the average of the ripple waveform. As the load resistance is reduced, the inductor current increases until it exceeds the current limit threshold. During the “Current Limited” portion of Figure 10, the current limit threshold lowers since the high load current causes VOUT (and the voltage at FB) to reduce. The on-time is reduced by approximately 50%, resulting in lower ripple amplitude for the inductor’s current. During this time the LM34914 is in a constant current mode, with an average load current equal to the current limit threshold + ΔI/2 (IOUT2). Normal operation resumes when the load current is reduced to IOUT3, allowing VOUT, the current limit threshold, and the on-time to return to their normal values. Note that in the second period of “Normal Operation”, even though the inductor’s peak current exceeds the current limit threshold during part of each cycle, the circuit is not in current limit since the current falls below the threshold before the feedback voltage reduces to its threshold. The peak current allowed through the buck switch, and the ISEN pin, is 2A, and the maximum allowed average current is 1.5A. 10 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 LM34914 www.ti.com SNVS453B – MAY 2006 – REVISED MARCH 2013 Inductor Current IOUT2 Current Limit Threshold IOUT3 TON 'I 2 IOUT1 Feedback Voltage @ FB Pin TON 2.5V Normal Operation Load Current Increases Current Limited Normal Operation Load Current Decreases Figure 10. Inductor Current - Normal and Current Limit Operation N - Channel Buck Switch and Driver The LM34914 integrates an N-Channel buck switch and associated floating high voltage gate driver. The gate driver circuit works in conjunction with an external bootstrap capacitor and an internal high voltage diode. A 0.022 µF capacitor (C4) connected between BST and SW provides the voltage to the driver during the on-time. During each off-time, the SW pin is at approximately -1V, and C4 is recharged for the next on-time from VCC through the internal diode. The minimum off-time ensures a minimum time each cycle to recharge the bootstrap capacitor. Softstart The softstart feature allows the converter to gradually reach a steady state operating point, thereby reducing start-up stresses and current surges. Upon turn-on, after VCC reaches the under-voltage threshold, an internal 12.5 µA current source charges up the external capacitor at the SS pin to 2.5V (t2 in Figure 8). The ramping voltage at SS (and the non-inverting input of the regulation comparator) ramps up the output voltage in a controlled manner. An internal switch grounds the SS pin if VCC is below the under-voltage lockout threshold, or if the RON/SD pin is grounded. Thermal Shutdown The LM34914 should be operated so the junction temperature does not exceed 125°C. If the junction temperature increases above that, an internal Thermal Shutdown circuit activates (typically) at 175°C, taking the controller to a low power reset state by disabling the buck switch and the on-timer. This feature helps prevent catastrophic failures from accidental device overheating. When the junction temperature reduces below 155°C (typical hysteresis = 20°C), normal operation resumes. Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 11 LM34914 SNVS453B – MAY 2006 – REVISED MARCH 2013 www.ti.com APPLICATIONS INFORMATION EXTERNAL COMPONENTS The following guidelines can be used to select the external components (see the Block Diagram). First determine the following operating parameters: - Output voltage (VOUT) - Minimum and maximum input voltage (VIN(min) and VIN(max)) - Minimum and maximum load current (IOUT(min) and IOUT(max)) - Switching Frequency (FS) R1 and R2: These resistors set the output voltage. The ratio of these resistors is calculated from: R1/R2 = (VOUT/2.5V) - 1 (7) R1 and R2 should be chosen from standard value resistors in the range of 1.0 kΩ - 10 kΩ which satisfy the above ratio. RON: The resistor sets the on-time, and consequently, the switching frequency. Its value can be determined using Equation 6 based on the frequency, or Equation 5 if a specific on-time is required. The minimum allowed value for RON is calculated from: RON t 100 ns x (VIN(MAX) ± 1.5V) 1.15 x 10-10 - 1.4 k: (8) L1: The main parameter affected by the inductor is the output current ripple amplitude (IOR). The minimum load current is used to determine the maximum allowable ripple. In order to maintain continuous conduction mode the valley should not reach 0 mA. This is not a requirement of the LM34914, but serves as a guideline for selecting L1. For this case, the maximum ripple current is: IOR(MAX) = 2 x IOUT(min) (9) If the minimum load current is zero, use 20% of IOUT(max) for IOUT(min) in Equation 9. The ripple calculated in Equation 6 is then used in the following equation: VOUT x (VIN (max) - VOUT) L1 = IOR (max) x FS x VIN (max) (10) where Fs is the switching frequency. This provides a minimum value for L1. The next larger standard value should be used, and L1 should be rated for the peak current level, equal to IOUT(max) + IOR(max)/2. C2 and R3: Since the LM34914 requires a minimum of 25 mVp-p of ripple at the FB pin for proper operation, the required ripple at VOUT is increased by R1 and R2. This necessary ripple is created by the inductor ripple current flowing through R3, and to a lesser extent by C2 and its ESR. The minimum inductor ripple current is calculated using Equation 10, rearranged to solve for IOR at minimum VIN. VOUT x (VIN (min) - VOUT) IOR (min) = L1 x FS x VIN (min) (11) The minimum value for R3 is then equal to: R3(min) = 25 mV x (R1 + R2) R2 x IOR (min) (12) Typically R3 is less than 5Ω. C2 should generally be no smaller than 3.3 µF, although that is dependent on the frequency and the desired output characteristics. C2 should be a low ESR good quality ceramic capacitor. Experimentation is usually necessary to determine the minimum value for C2, as the nature of the load may require a larger value. A load which creates significant transients requires a larger value for C2 than a nonvarying load. 12 Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 LM34914 www.ti.com SNVS453B – MAY 2006 – REVISED MARCH 2013 D1: A Schottky diode is recommended. Ultra-fast recovery diodes are not recommended as the high speed transitions at the SW pin may inadvertently affect the IC’s operation through external or internal EMI. The diode should be rated for the maximum input voltage (VIN(max)), the maximum load current (IOUT(max)), and the peak current which occurs when the current limit and maximum ripple current are reached simultaneously. The diode’s average power dissipation is calculated from: PD1 = VF x IOUT x (1-D) (13) where VF is the diode's forward voltage drop, and D is the duty cycle. C1 and C5: C1’s purpose is to supply most of the switch current during the on-time, and limit the voltage ripple at VIN, on the assumption that the voltage source feeding VIN has an output impedance greater than zero. If the source’s dynamic impedance is high (effectively a current source), it supplies the average input current, but not the ripple current. At maximum load current, when the buck switch turns on, the current into VIN suddenly increases to the lower peak of the inductor’s ripple current, ramps up to the upper peak, then drop to zero at turn-off. The average current during the on-time is the load current. For a worst case calculation, C1 must supply this average load current during the maximum on-time. C1 is calculated from: IOUT (max) x tON C1 = 'V (14) where tON is the maximum on-time, and ΔV is the allowable ripple voltage at VIN. C5’s purpose is to help avoid transients and ringing due to long lead inductance leading to the VIN pin. A low ESR, 0.1 µF ceramic chip capacitor is recommended, and must be located close to the VIN and RTN pins. C3: The capacitor at the VCC output provides not only noise filtering and stability, but also prevents false triggering of the VCC UVLO at the buck switch on/off transitions. C3 should be no smaller than 0.1 µF, and should be a good quality, low ESR, ceramic capacitor. C3’s value, and the VCC current limit, determine a portion of the turn-on-time (t1 in Figure 8). C4: The recommended value for C4 is 0.022 µF. A high quality ceramic capacitor with low ESR is recommended as C4 supplies a surge current to charge the buck switch gate at turn-on. A low ESR also helps ensure a complete recharge during each off-time. C6: The capacitor at the SS pin determines the softstart time, i.e. the time for the output voltage, to reach its final value (t2 in Figure 8). The capacitor value is determined from the following: C6 = t2 x 12.5 PA 2.5V (15) PC BOARD LAYOUT The LM34914 regulation, over-voltage, and current limit comparators are very fast, and respond to short duration noise pulses. Layout considerations are therefore critical for optimum performance. The layout must be as neat and compact as possible, and all of the components must be as close as possible to their associated pins. The current loop formed by D1, L1, C2 and the SGND and ISEN pins should be as small as possible. The ground connection from SGND and RTN to C1 should be as short and direct as possible. If it is expected that the internal dissipation of the LM34914 will produce excessive junction temperatures during normal operation, good use of the PC board’s ground plane can help to dissipate heat. The exposed pad on the bottom of the IC package can be soldered to a ground plane, and that plane should extend out from beneath the IC, and be connected to ground plane on the board’s other side with several vias, to help dissipate the heat. The exposed pad is internally connected to the IC substrate. Additionally the use of wide PC board traces, where possible, can help conduct heat away from the IC. Judicious positioning of the PC board within the end product, along with the use of any available air flow (forced or natural convection) can help reduce the junction temperatures. Submit Documentation Feedback Copyright © 2006–2013, Texas Instruments Incorporated Product Folder Links: LM34914 13 LM34914 SNVS453B – MAY 2006 – REVISED MARCH 2013 www.ti.com LOW OUTPUT RIPPLE CONFIGURATIONS For applications where low output ripple is required, the following options can be used to reduce or nearly eliminate the ripple. a) Reduced ripple configuration: In Figure 11, Cff is added across R1 to AC-couple the ripple at VOUT directly to the FB pin. This allows the ripple at VOUT to be reduced to a minimum of 25 mVp-p by reducing R3, since the ripple at VOUT is not attenuated by the feedback resistors. The minimum value for Cff is determined from: Cff = tON (max) (R1//R2) (16) where tON(max) is the maximum on-time, which occurs at VIN(min). The next larger standard value capacitor should be used for Cff. R1 and R2 should each be towards the upper end of the 1kΩ to 10kΩ range. L1 SW VOUT LM34914 Cff R1 R3 FB R2 C2 Figure 11. Reduced Ripple Configuration b) Minimum ripple configuration: If the application requires a lower value of ripple (