Order
Now
Product
Folder
Support &
Community
Tools &
Software
Technical
Documents
Reference
Design
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
LM34925 用于隔离式 DC-DC 转换器的集成二次侧偏置稳压器
1 特性
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
1
3 说明
宽输入电压范围：7.5V 至 100V
集成了 100mA 高侧和低侧开关
无需肖特基二极管
恒定导通时间控制
无需环路补偿
超快瞬态响应
接近恒定的运行频率
智能峰值电流限制
可调节输出电压（以 1.225V 为基准电压）
2% 的反馈基准电压精度
频率可调至 1MHz
可调低压闭锁 (UVLO)
远程关断
热关断
封装：
– 8 引脚晶圆级小外形无引线 (WSON)
– 8 引脚小外形尺寸 (SO) PowerPAD™
LM34925 稳压器具有 实现低成本高效率的隔离式偏置
稳压器所需的全部功能。此高压稳压器包含两个 100V
N 通道 MOSFET 开关：一个高侧降压开关和一个低侧
同步开关。LM34625 所采用的恒定导通时间 (COT) 控
制方案无需环路补偿，可提供出色的瞬态响应。此稳压
器的运行接通时间与输入电压成反比。此特性使得工作
频率能够保持相对恒定。使用集成感测电路来执行智能
峰值电流限制。其他 特性 包括一个用于抑制低压条件
下运行的可编程输入欠压比较器。保护 特性 包括热关
断和 VCC 欠压锁定 (UVLO)。LM34925 器件采用 8 引
脚 WSON 和
8 引脚 SO PowerPAD 塑料封装。
器件信息(1)
器件型号
LM34925
封装尺寸（标称值）
4.89mm × 3.90mm
WSON (8)
4.00mm x 4.00mm
(1) 如需了解所有可用封装，请参阅数据表末尾的可订购产品附
录。
2 应用
•
•
封装
SO PowerPAD (8)
隔离式电信偏压电源
隔离式汽车和工业用电子元件
典型应用原理图
VOUT2
D1
+
LM34925
VIN
7.5V-100V
CIN
2
+
4
RUV2
BST
VIN
SW
RON
VCC
UVLO
FB
RUV1
COUT2
7
+
8
X1
CBST
VOUT1
NP
RON
3
NS
RTN
1
6
D2
5
+
CVCC
Rr
RFB2
+
RFB1
COUT1
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
English Data Sheet: SNVS846
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
目录
1
2
3
4
5
6
7
特性 ..........................................................................
应用 ..........................................................................
说明 ..........................................................................
修订历史记录 ...........................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
4
5
6.1
6.2
6.3
6.4
6.5
6.6
6.7
5
5
5
5
6
7
7
Absolute Maximum Ratings .....................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Detailed Description .............................................. 9
7.1 Overview ................................................................... 9
7.2 Functional Block Diagram ......................................... 9
7.3 Feature Description................................................. 10
7.4 Device Functional Modes........................................ 14
8
Application and Implementation ........................ 15
8.1 Application Information............................................ 15
8.2 Typical Application .................................................. 15
9 Power Supply Recommendations...................... 20
10 Layout................................................................... 20
10.1 Layout Guidelines ................................................. 20
10.2 Layout Example .................................................... 20
11 器件和文档支持 ..................................................... 21
11.1
11.2
11.3
11.4
11.5
11.6
文档支持................................................................
接收文档更新通知 .................................................
社区资源................................................................
商标 .......................................................................
静电放电警告.........................................................
Glossary ................................................................
21
21
21
21
21
21
12 机械、封装和可订购信息 ....................................... 21
4 修订历史记录
Changes from Revision F (November 2014) to Revision G
Page
•
Deleted lead temperature and related footnote from Abs Max table...................................................................................... 5
•
Changed /moved Storage temp from Handling ratings to Absolute Maximum Ratings, changed Handling ratings title
to ESD ratings ........................................................................................................................................................................ 5
•
Changed 14 V to 13 V in the VCC Regulator section ........................................................................................................... 11
Changes from Revision E (December 2013) to Revision F
Page
•
已添加 引脚配置和功能 部分、处理额定值 表、开关特性 表、功能 说明 部分、器件功能模式、应用和实施 部分、电
源建议 部分、布局 部分、器件和文档支持 部分，以及机械、封装和可订购信息 部分 .......................................................... 1
•
Changed Thermal Information table ....................................................................................................................................... 5
•
Changed TON vs VIN and RON in Typical Characteristics ....................................................................................................... 7
•
Changed Control Overview section, Ripple Configuration table .......................................................................................... 10
•
Changed Soft-Start Circuit, Isolated Fly-Buck Converter graphics. ..................................................................................... 14
Changes from Revision D (December 2013) to Revision E
Page
•
Added Thermal Parameters ................................................................................................................................................... 5
2
版权 © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
Changes from Revision C (September 2013) to Revision D
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
Page
•
已更改 按照 TI 标准，对文档格式进行了通篇更改 ................................................................................................................. 1
•
已更改 将特性、引脚说明 和建议运行条件 中的最低工作输入电压由 9V 更改为 7.5V .......................................................... 1
•
Added Absolute Maximum Junction Temperature.................................................................................................................. 5
Changes from Revision B (September, 2013) to Revision C
•
Added SW to RTN (100-ns transient) in Absolute Maximum Ratings ................................................................................... 5
Changes from Revision A (February 2013) to Revision B
•
Page
Page
Changed layout of National Data Sheet to TI format ........................................................................................................... 20
Copyright © 2012–2017, Texas Instruments Incorporated
3
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
5 Pin Configuration and Functions
DDA Package
8-Pin SO PowerPAD
Top View
RTN
1
VIN
2
UVLO
3
RON
4
SO
PowePAD-8
Exp Pad
8
SW
7
BST
6
VCC
5
FB
NGU Package
8-Pin WSON With Exposed Thermal Pad
Top View
RTN
1
VIN
2
UVLO
3
RON
4
8 SW
WSON-8
Exp Pad
7 BST
6 VCC
5 FB
Pin Functions
PIN
NO.
NAME
I/O
1
RTN
—
2
VIN
3
DESCRIPTION
APPLICATION INFORMATION
Ground
Ground connection of the integrated circuit.
I
Input Voltage
Operating input range is 7.5 V to 100 V.
UVLO
I
Input Pin of Undervoltage
Comparator
Resistor divider from VIN to UVLO to GND programs the undervoltage
detection threshold. An internal current source is enabled when UVLO is
above 1.225 V to provide hysteresis. When UVLO pin is pulled below
0.66 V externally, the parts goes in shutdown mode.
4
RON
I
On-Time Control
A resistor between this pin and VIN sets the switch on-time as a function
of VIN. Minimum recommended on-time is 100ns at max input voltage.
5
FB
I
Feedback
This pin is connected to the inverting input of the internal regulation
comparator. The regulation level is 1.225 V.
6
VCC
O
Output from the Internal
High Voltage Series Pass
Regulator. Regulated at
7.6 V.
The internal VCC regulator provides bias supply for the gate drivers and
other internal circuitry. A 1-μF decoupling capacitor is recommended.
7
BST
I
Bootstrap Capacitor
An external capacitor is required between the BST and SW pins (0.01 μF
ceramic). The BST pin capacitor is charged by the VCC regulator through
an internal diode when the SW pin is low.
8
SW
O
Switching Node
Power switching node. Connect to the output inductor and bootstrap
capacitor.
—
EP
—
Exposed Pad
Exposed pad must be connected to RTN pin. Connect to system ground
plane on application board for reduced thermal resistance.
4
Copyright © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
6 Specifications
6.1 Absolute Maximum Ratings (1)
MIN
MAX
UNIT
VIN, UVLO to RTN
–0.3
100
V
SW to RTN
–1.5
VIN + 0.3
V
–5
VIN + 0.3
V
BST to VCC
100
V
BST to SW
13
V
SW to RTN (100-ns transient)
RON to RTN
–0.3
100
V
VCC to RTN
–0.3
13
V
FB to RTN
–0.3
5
V
150
°C
150
°C
Maximum junction temperature (2)
Storage temperature Tstg
(1)
(2)
–55
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Recommended Operating Conditions are
conditions under which operation of the device is intended to be functional. For verified specifications and test conditions, see the
Electrical Characteristics. The RTN pin is the GND reference electrically connected to the substrate.
High junction temperatures degrade operating lifetimes. Operating lifetime is de-rated for junction temperatures greater than 125°C.
6.2 ESD Ratings
MIN
V(ESD)
(1)
(2)
Electrostatic discharge
MAX
UNIT
2
kV
750
V
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all
pins (1)
Charged device model (CDM), per JEDEC specification
JESD22-C101, all pins (2)
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted) (1)
MIN
MAX
VIN voltage
7.5
100
V
Operating junction temperature (2)
–40
125
°C
(1)
(2)
UNIT
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Recommended Operating Conditions are
conditions under which operation of the device is intended to be functional. For verified specifications and test conditions, see the
Electrical Characteristics. The RTN pin is the GND reference electrically connected to the substrate.
High junction temperatures degrade operating lifetimes. Operating lifetime is de-rated for junction temperatures greater than 125°C.
6.4 Thermal Information
LM34925
THERMAL METRICS (1)
NGU
DDA
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
41.3
41.1
°C/W
RθJCbot
Junction-to-case (bottom) thermal resistance
3.2
2.4
°C/W
ΨJB
Junction-to-board thermal characteristic parameter
19.2
24.4
°C/W
RθJB
Junction-to-board thermal resistance
19.1
30.6
°C/W
RθJCtop
Junction-to-case (top) thermal resistance
34.7
37.3
°C/W
ΨJT
Junction-to-top thermal characteristic parameter
0.3
6.7
°C/W
(1)
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report (SPRA953).
Copyright © 2012–2017, Texas Instruments Incorporated
5
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
6.5 Electrical Characteristics
Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over –40°C to 125°C junction temperature
range, unless otherwise stated. VIN = 48 V unless otherwise stated. (see (1)).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
6.25
7.6
8.55
UNIT
VCC SUPPLY
VCC Reg
VCC Regulator Output
VIN = 48 V, ICC = 20 mA
VCC Current Limit
VIN = 48 V (2)
VCC UVLO Threshold
(VCC Increasing)
–40°C ≤ TJ ≤ 125°C
26
4.15
VCC UVLO Hysteresis
V
mA
4.5
4.9
300
V
mV
VCC Drop Out Voltage
VIN = 8 V, ICC = 20 mA
2.3
IIN Operating Current
Non-switching, FB = 3 V
V
1.75
mA
IIN Shutdown Current
UVLO = 0 V
50
225
µA
Buck Switch RDS(ON)
ITEST = 100 mA, BST-SW = 7 V
0.8
1.8
Ω
Synchronous RDS(ON)
ITEST = 200 mA
0.45
1
Ω
Gate Drive UVLO
VBST − VSW Rising
3
3.6
V
SWITCH CHARACTERISTICS
2.4
Gate Drive
UVLO Hysteresis
260
mV
UNDERVOLTAGE SENSING FUNCTION
UV Threshold
UV Rising
1.19
1.225
1.26
V
UV Hysteresis Input Current
UV = 2.5 V
–10
–20
–29
µA
Remote Shutdown Threshold
Voltage at UVLO falling
0.32
0.66
V
110
mV
Remote Shutdown Hysteresis
REGULATION AND OVERVOLTAGE COMPARATORS
FB Regulation Level
Internal reference trip point for switch ON
FB Overvoltage Threshold
Trip point for switch OFF
1.2
FB Bias Current
1.225
1.25
V
1.62
V
60
nA
CURRENT LIMIT
Current Limit Threshold
150
Current Limit Response Time
Time to switch off
OFF-Time Generator (Test 1)
OFF-Time Generator (Test 2)
270
370
mA
150
ns
FB = 0.1 V, VIN = 48 V
12
µs
FB = 1 V, VIN = 48 V
2.5
µs
165
°C
20
°C
THERMAL SHUTDOWN
Tsd
Thermal Shutdown Temperature
Thermal Shutdown Hysteresis
(1)
(2)
6
All limits are verified by design. All electrical characteristics having room temperature limits are tested during production at TA = 25°C. All
hot and cold limits are verified by correlating the electrical characteristics to process and temperature variations and applying statistical
process control.
VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.
Copyright © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
6.6 Switching Characteristics
Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over –40°C to 125°C junction temperature range
unless otherwise stated. VIN = 48 V unless otherwise stated.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ON-TIME GENERATOR
TON Test 1
VIN = 32 V, RON = 100 kΩ
270
350
460
ns
TON Test 2
VIN = 48 V, RON = 100 kΩ
188
250
336
ns
TON Test 3
VIN = 75 V, RON = 250 kΩ
250
370
500
ns
TON Test 4
VIN = 10 V, RON = 250 kΩ
1880
3200
4425
ns
MINIMUM OFF-TIME
Minimum Off-Timer
FB = 0 V
144
ns
6.7 Typical Characteristics
100
EFFICIENCY (%)
90
80
VIN=24V
VIN=36V
70
60
VIN=48V
50
40
30
VOUT2=10V, IOUT1=0
20
30
40
50 60 70 80 90 100 110
LOAD CURRENT (mA)
Figure 1. Efficiency at 750 kHz, VOUT1 = 10 V
Figure 2. VCC vs VIN
Figure 3. VCC vs ICC
Copyright © 2012–2017, Texas Instruments Incorporated
Figure 4. ICC vs External VCC
7
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
Typical Characteristics (continued)
8
Figure 5. TON vs VIN and RON
Figure 6. TOFF (ILIM) vs VFB and VIN
Figure 7. IIN vs VIN (Operating, Non-Switching)
Figure 8. IIN vs VIN (Shutdown)
Copyright © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
7 Detailed Description
7.1 Overview
The LM34925 step-down switching regulator features all the functions needed to implement a low cost, efficient,
isolated bias supply. This high-voltage regulator contains 100-V, N-channel buck, and synchronous switches, is
easy to implement, and is provided in thermally enhanced SO PowerPAD-8 and WSON-8 packages. The
regulator operation is based on a constant on-time control scheme using an on-time inversely proportional to VIN.
This control scheme does not require loop compensation. Current limit is implemented with forced off-time
inversely proportional to VOUT. This scheme ensures short circuit protection while providing minimum foldback.
The simplified block diagram of the LM34925 device is shown in the Functional Block Diagram section.
The LM34925 device can be applied in numerous applications to efficiently regulate down higher voltages. This
regulator is well suited for 48-V telecom and automotive power bus ranges. Protection features include: thermal
shutdown, undervoltage lockout, minimum forced off-time, and an intelligent current limit.
7.2 Functional Block Diagram
LM34925
START-UP
REGULATOR
VIN
VCC
V UVLO
20 µA
4.5V
UVLO
THERMAL
SHUTDOWN
UVLO
1.225V
SD
VDD REG
BST
0.66V
SHUTDOWN
BG REF
VIN
DISABLE
ON/OFF
TIMERS
RON
1.225V
SW
COT CONTROL
LOGIC
FEEDBACK
FB
OVER-VOLTAGE
1.62V
RTN
Copyright © 2012–2017, Texas Instruments Incorporated
CURRENT
LIMIT
ONE-SHOT
ILIM
COMPARATOR
+
VILIM
9
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
7.3 Feature Description
7.3.1 Control Overview
The LM34925 regulator employs a control principle based on a comparator and a one-shot on-timer, with the
output voltage feedback (FB) compared to an internal reference (1.225 V). If the FB voltage is below the
reference the internal buck switch is switched on for the one-shot timer period, which is a function of the input
voltage and the programming resistor (RT). Following the on-time the switch remains off until the FB voltage falls
below the reference, and the forced minimum off-time has expired. When the FB pin voltage falls below the
reference and the off-time one-shot period expires, the buck switch is then turned on for another on-time oneshot period. This will continue until regulation is achieved and the FB voltage is approximately equal to 1.225 V
(typ).
In a synchronous buck converter, the low-side (sync) FET is ‘on’ when the high-side (buck) FET is ‘off’. The
inductor current ramps up when the high-side switch is ‘on’ and ramps down when the high-side switch is ‘off’.
There is no diode emulation feature in this IC, and therefore, the inductor current may ramp in the negative
direction at light load. This causes the converter to operate in continuous conduction mode (CCM) regardless of
the output loading. The operating frequency remains relatively constant with load and line variations.
The operating frequency can be determined from Equation 1.
VOUT1
f SW =
. x RON
where
K = 9 × 10–11
(1)
The output voltage (VOUT) is set by two external resistors (RFB1, RFB2). The regulated output voltage is
determined from Equation 2.
VOUT = 1.225V x
RFB2 + RFB1
RFB1
(2)
This regulator regulates the output voltage based on ripple voltage at the feedback input, requiring a minimum
amount of ESR for the output capacitor (COUT). A minimum of 250 mV of ripple voltage at the feedback pin (FB)
is required for the LM34925 device. In cases where the capacitor ESR is too small, additional series resistance
may be required (RC in Figure 9).
For applications where lower output voltage ripple is required, the output can be taken directly from a low ESR
output capacitor, as shown in Figure 9. However, RC slightly degrades the load regulation.
L1
VOUT
SW
LM34925
RFB2
FB
RC
+
RFB1
COUT
VOUT
(low ripple)
Figure 9. Low Ripple Output Configuration
7.3.2 VCC Regulator
The LM34925 device contains an internal high voltage linear regulator with a nominal output of 7.6 V. The input
pin (VIN) can be connected directly to the line voltages up to 100 V. The VCC regulator is internally current limited
to 30 mA. The regulator sources current into the external capacitor at VCC. This regulator supplies current to
internal circuit blocks including the synchronous MOSFET driver and the logic circuits. When the voltage on the
VCC pin reaches the Undervoltage Lockout threshold of 4.5 V, the IC is enabled.
The VCC regulator contains an internal diode connection to the BST pin to replenish the charge in the gate drive
boot capacitor when SW pin is low.
10
Copyright © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
Feature Description (continued)
At high-input voltages, the power dissipated in the high voltage regulator is significant and can limit the overall
achievable output power. As an example, with the input at 48 V and switching at high frequency, the VCC
regulator may supply up to 7 mA of current resulting in 48 V × 7 mA = 336 mW of power dissipation. If the VCC
voltage is driven externally by an alternate voltage source, between 8.55 V and 13 V, the internal regulator is
disabled. This reduces the power dissipation in the IC.
7.3.3 Regulation Comparator
The feedback voltage at FB is compared to an internal 1.225-V reference. In normal operation, when the output
voltage is in regulation, an on-time period is initiated when the voltage at FB falls below 1.225 V. The high-side
switch will stay on for the on-time, causing the FB voltage to rise above 1.225 V. After the on-time period, the
high-side switch will stay off until the FB voltage again falls below 1.225 V. During start-up, the FB voltage will be
below 1.225 V at the end of each on-time causing the high-side switch to turn on immediately after the minimum
forced off-time of 144 ns. The high-side switch can be turned off before the on-time is over, if peak current in the
inductor reaches the current limit threshold.
7.3.4 Overvoltage Comparator
The feedback voltage at FB is compared to an internal 1.62-V reference. If the voltage at FB rises above 1.62 V
the on-time pulse is immediately terminated. This condition can occur if the input voltage and/or the output load
changes suddenly. The high-side switch will not turn on again until the voltage at FB falls below 1.225 V.
7.3.5 On-Time Generator
The on-time for the LM34925 device is determined by the RON resistor, and is inversely proportional to the input
voltage (VIN), resulting in a nearly constant frequency as VIN is varied over its range. The on-time equation for the
LM34925 can is determined by Equation 3.
TON =
10-10 x RON
VIN
(3)
See Figure 5. RON should be selected for a minimum on-time (at maximum VIN) greater than 100 ns, for proper
operation. This requirement limits the maximum frequency for high VIN.
7.3.6 Current Limit
The LM34925 contains an intelligent current limit off-timer. If the current in the buck switch exceeds 270 mA, the
present cycle is immediately terminated, and a non-resetable off-timer is initiated. The length of off-time is
controlled by the FB voltage and the input voltage VIN. As an example, when FB = 0 V and VIN = 48 V, a
maximum off-time is set to 16 μs. This condition occurs when the output is shorted, and during the initial part of
start-up. This amount of time ensures safe short circuit operation up to the maximum input voltage of
100 V.
In cases of overload where the FB voltage is above zero volts (not a short circuit) the current limit off-time is
reduced. Reducing the off-time during less severe overloads reduces the amount of foldback, recovery time, and
start-up time. The off-time is calculated from Equation 4.
TOFF(ILIM)
0.07 u VIN
Ps
VFB 0.2 V
(4)
The current limit protection feature is peak limited, the maximum average output will be less than the peak.
7.3.7 N-Channel Buck Switch and Driver
The LM34925 device integrates an N-Channel Buck switch and associated floating high voltage gate driver. The
gate driver circuit works in conjunction with an external bootstrap capacitor and an internal high voltage diode. A
0.01-uF ceramic capacitor connected between the BST pin and SW pin provides the voltage to the driver during
the on-time. During each off-time, the SW pin is at approximately 0 V, and the bootstrap capacitor charges from
VCC through the internal diode. The minimum off-timer, set to 144 ns, ensures a minimum time each cycle to
recharge the bootstrap capacitor.
Copyright © 2012–2017, Texas Instruments Incorporated
11
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
Feature Description (continued)
7.3.8 Synchronous Rectifier
The LM34925 device provides an internal synchronous N-Channel MOSFET rectifier. This MOSFET provides a
path for the inductor current to flow when the high-side MOSFET is turned off.
The synchronous rectifier has no diode emulation mode, and is designed to keep the regulator in continuous
conduction mode even during light loads which would otherwise result in discontinuous operation. This feature
specifically allows the user to design a secondary regulator using a transformer winding off the main inductor to
generate the alternate regulated output voltage.
7.3.9 Undervoltage Detector
The LM34925 device contains a dual level Undervoltage Lockout (UVLO) circuit. A summary of threshold
voltages and operational states is provided in the Device Functional Modes section. When the UVLO pin voltage
is below 0.66 V, the controller is in a low current shutdown mode. When the UVLO pin voltage is greater than
0.66 V but less than 1.225 V, the controller is in standby mode. In standby mode the VCC bias regulator is active
while the regulator output is disabled. When the VCC pin exceeds the VCC undervoltage thresholds and the UVLO
pin voltage is greater than 1.225 V, normal operation begins. An external set-point voltage divider from VIN to
GND can be used to set the minimum operating voltage of the regulator.
UVLO hysteresis is accomplished with an internal 20-μA current source that is switched on or off into the
impedance of the set-point divider. When the UVLO threshold is exceeded, the current source is activated to
quickly raise the voltage at the UVLO pin. The hysteresis is equal to the value of this current times the resistance
RUV2.
If the UVLO pin is wired directly to the VIN pin, the regulator will begin operation once the VCC undervoltage is
satisfied.
VIN
CIN
2
VIN
+
RUV2
LM34925
3
UVLO
RUV1
Figure 10. UVLO Resistor Setting
7.3.10 Thermal Protection
The LM34925 device should be operated so the junction temperature does not exceed 150°C during normal
operation. An internal Thermal Shutdown circuit is provided to protect the LM34925 device in the event of a
higher than normal junction temperature. When activated, typically at 165°C, the controller is forced into a low
power reset state, disabling the buck switch and the VCC regulator. This feature prevents catastrophic failures
from accidental device overheating. When the junction temperature reduces below 145°C (typical hysteresis =
20°C), the VCC regulator is enabled, and normal operation is resumed.
7.3.11 Ripple Configuration
LM34925 uses Constant-On-Time (COT) control scheme, in which the on-time is terminated by an on-timer, and
the off-time is terminated by the feedback voltage (VFB) falling below the reference voltage (VREF). Therefore, for
stable operation, the feedback voltage must decrease monotonically, in phase with the inductor current during
the off-time. Furthermore, this change in feedback voltage (VFB) during off-time must be large enough to
suppress any noise component present at the feedback node.
12
Copyright © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
Feature Description (continued)
Table 1 shows three different methods for generating appropriate voltage ripple at the feedback node. Type 1
and Type 2 ripple circuits couple the ripple at the output of the converter to the feedback node (FB). The output
voltage ripple has two components:
1. Capacitive ripple caused by the inductor current ripple charging/discharging the output capacitor.
2. Resistive ripple caused by the inductor current ripple flowing through the ESR of the output capacitor.
The capacitive ripple is not in phase with the inductor current. As a result, the capacitive ripple does not
decrease monotonically during the off-time. The resistive ripple is in phase with the inductor current and
decreases monotonically during the off-time. The resistive ripple must exceed the capacitive ripple at the output
node (VOUT) for stable operation. If this condition is not satisfied unstable switching behavior is observed in COT
converters, with multiple on-time bursts in close succession followed by a long off-time.
Type 3 ripple method uses Rr and Cr and the switch node (SW) voltage to generate a triangular ramp. This
triangular ramp is ac coupled using Cac to the feedback node (FB). Since this circuit does not use the output
voltage ripple, it is ideally suited for applications where low output voltage ripple is required. See AN-1481
Controlling Output Ripple and Achieving ESR Independence in Constant On-Time (COT) Regulator Designs
(SNVA166) for more details for each ripple generation method.
Table 1. Ripple Configuration
TYPE 1
LOWEST COST CONFIGURATION
TYPE 2
REDUCED RIPPLE CONFIGURATION
VOUT
TYPE 3
MINIMUM RIPPLE CONFIGURATION
VOUT
L1
VOUT
L1
L1
R FB2
Cac
R FB2
RC
To FB
C OUT
COUT
R FB2
GND
R FB1
GND
25 mV VOUT
x
ûIL(MIN) VREF
Cr
Cac
To FB
R FB1
RC >
Rr
RC
C OUT
To FB
R FB1
GND
C>
5
gsw (RFB2||RFB1)
25 mV
RC >
ûIL(MIN)
Cr
1000 pF
Cac
100 nF
Rr Cr d
(VIN(MIN)
VOUT ) u TON
25 mV
7.3.12 Soft Start
A soft-start feature can be implemented with the LM34925 device using an external circuit. As shown in
Figure 11, the soft-start circuit consists of one capacitor, C1, two resistors, R1 and R2, and a diode, D. During the
initial start-up, the VCC voltage is established prior to the VOUT voltage. Capacitor C1 is discharged and diode D
is thereby forward biased to pull up the FB pin voltage. The FB voltage exceeds the reference voltage (1.225 V)
and switching is therefore disabled. As capacitor C1 charges, the voltage at node B gradually decreases and
switching commences. VOUT will gradually rise to maintain the FB voltage at the reference voltage. Once the
voltage at node B is less than a diode drop above the FB voltage, the soft-start sequence is finished and D is
reverse biased.
During the initial part of the start-up, the FB voltage can be approximated as shown in Equation 5. Please note
that the effect of R1 has been ignored to simplify the calculation.
RFB1 x RFB2
VFB = (VCC - VD) x
R2 x (RFB1 + RFB2) + RFB1 x RFB2
(5)
Copyright © 2012–2017, Texas Instruments Incorporated
13
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
C1 is charged after the first start up. Diode D1 is added to discharge C1 when the input voltage experiences a
momentary drop to initialize the soft-start sequence.
To achieve the desired soft start, the following design guidance is recommended:
(1) R2 is selected so that VFB is higher than 1.225 V for a VCC of 4.5 V, but is lower than 5 V when VCC is 8.55 V.
If an external VCC is used, VFB should not exceed 5 V at maximum VCC.
(2) C1 is selected to achieve the desired start-up time which can be determined from Equation 6.
RFB1 x RFB2
)
tS = C1 x (R2 +
RFB1 + RFB2
(6)
(3) R1 is used to maintain the node B voltage at zero after the soft start is finished. A value larger than the
feedback resistor divider is preferred. Note that the effect of resistor R1 is ignored in Equation 5.
Based on the schematic shown in Figure 11, selecting C1 = 1 uF, R2 = 1 kΩ, R1 = 30 kΩ results in a soft-start
time of about 2 ms.
VOUT
VCC
C1
RFB2
R2
To FB
D
D1
B
RFB1
R1
Figure 11. Soft-Start Circuit
7.4 Device Functional Modes
The UVLO pin controls the operating mode of the LM34925 device (see Table 2 for the detailed functional
states).
Table 2. UVLO Mode
UVLO
VCC
MODE
< 0.66 V
Disabled
Shutdown
VCC regulator disabled.
Switching disabled.
0.66 V — 1.225 V
Enabled
Standby
VCC regulator enabled
Switching disabled.
VCC < 4.5 V
Standby
VCC regulator enabled.
Switching disabled.
VCC > 4.5 V
Operating
> 1.225 V
14
DESCRIPTION
VCC enabled.
Switching enabled.
Copyright © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM34925 device is step-down dc-to-dc converter. The device is typically used to convert a higher dc voltage
to a lower dc voltage with a maximum available output current of 100 mA. Use the following design procedure to
select component values for the LM34925 device. Alternately, use the WEBENCH® software to generate a
complete design. The WEBENCH software uses an iterative design procedure and accesses a comprehensive
database of components when generating a design. This section presents a simplified discussion of the design
process.
8.2 Typical Application
8.2.1 Application Circuit: 20-V to 95-V Input and 10-V, 100-mA Output Isolated Fly-Buck™ Converter
VOUT2
D1
+
N2
COUT2
1 µF
X1
LM34925
BST
VIN
20V-95V
0.01 µF
+
CBST
+
150 µH
VOUT1
SW
46.4 NŸ 1 nF
Rr
Cr
VIN
CBYP
0.1 µF
CIN
1 µF
N1
+
RON
RUV2
127 NŸ
RON
130 NŸ
RUV1
8.25 NŸ
0.1 µF
+
Cac
RFB2
VCC
UVLO
RTN
COUT1
1 µF
FB
+
D2
7.32 NŸ
CVCC
1 µF
RFB1
1 NŸ
Figure 12. Isolated Fly-Buck Converter Using LM34925
8.2.1.1 Design Requirements
Selection of external components is illustrated through a design example. The design example specifications are
shown in Table 3.
Table 3. Buck Converter Design Specifications
DESIGN PARAMETERS
Input Voltage Range
VALUE
20 V to 95 V
Primary Output Voltage
10 V
Secondary (Isolated) Output Voltage
9.5 V
Maximum Output Current (Primary + Secondary)
Maximum Power Output
Nominal Switching Frequency
Copyright © 2012–2017, Texas Instruments Incorporated
100 mA
1W
750 kHz
15
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Transformer Turns Ratio
The transformer turns ratio is selected based on the ratio of the primary output voltage to the secondary
(isolated) output voltage. In this design example, the two outputs are nearly equal and a 1:1 turns ratio
transformer is selected. Therefore, N2 / N1 = 1.
If the secondary (isolated) output voltage is significantly higher or lower than the primary output voltage, a turns
ratio less than or greater than 1 is recommended. The primary output voltage is normally selected based on the
input voltage range such that the duty cycle of the converter does not exceed 50% at the minimum input voltage.
This condition is satisfied if VOUT1 < VIN_MIN / 2.
8.2.1.2.2 Total IOUT
The total primary referred load current is calculated by multiplying the isolated output load(s) by the turns ratio of
the transformer as shown in Equation 7.
IOUT(MAX)
IOUT1 IOUT2 u
N2
N1
0.1 A
(7)
8.2.1.2.3 RFB1, RFB2
The feedback resistors are selected to set the primary output voltage. The selected value for RFB1 is 1 kΩ. RFB2
can be calculated using the following equations to set VOUT1 to the specified value of 10 V. A standard resistor
value of 7.32 kΩ is selected for RFB2.
RFB2
VOUT1 = 1.225V x (1 +
)
RFB1
(8)
VOUT1
- 1) x RFB1 = 7.16 k:
: RFB2 = (
1.225
(9)
8.2.1.2.4 Frequency Selection
Equation 1 is used to calculate the value of RON required to achieve the desired switching frequency.
VOUT1
f SW =
. x RON
(10)
–11
Where K = 9 × 10
For VOUT1 of 10 V and fSW of 750 kHz, the calculated value of RON is 148 kΩ. A lower value of 130 kΩ is selected
for this design to allow for second order effects at high switching frequency that are not included in Equation 1.
8.2.1.2.5 Transformer Selection
A coupled inductor or a flyback-type transformer is required for this topology. Energy is transferred from primary
to secondary when the low-side synchronous switch of the buck converter is conducting.
The maximum inductor primary ripple current that can be tolerated without exceeding the buck switch peak
current limit threshold (0.15 A minimum) is given by Equation 11.
'IL1
N2 ·
§
¨ 0.15 IOUT1 IOUT2 u N1 ¸ u 2
©
¹
0.1A
(11)
Using the maximum peak-to-peak inductor ripple current ΔIL1 from Equation 11, the minimum inductor value is
given by Equation 12.
L1
VIN(MAX)
VOUT
'IL1 u ¦SW
u
VOUT
VIN(MAX)
119.3 PH
(12)
A higher value of 150 µH is selected to insure the high-side switch current does not exceed the minimum peak
current limit threshold.
16
Copyright © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
8.2.1.2.6 Primary Output Capacitor
f
In a conventional buck converter the output ripple voltage is calculated as shown in Equation 13.
'IL1
'VOUT =
x f x COUT1
(13)
To limit the primary output ripple voltage ΔVOUT1 to approximately 50 mV, an output capacitor COUT1 of 0.33 µF is
required.
Figure 13 shows the primary winding current waveform (IL1) of a Fly-Buck converter. The reflected secondary
winding current adds to the primary winding current during the buck switch off-time. Because of this increased
current, the output voltage ripple is not the same as in conventional buck converter. The output capacitor value
calculated in Equation 13 should be used as the starting point. Optimization of output capacitance over the entire
line and load range must be done experimentally. If the majority of the load current is drawn from the secondary
isolated output, a better approximation of the primary output voltage ripple is given by Equation 14.
'VOUT1
N2 ·
§
¨ IOUT2 u N1 ¸ u TON(MAX)
©
¹
| 67 mV
COUT1
(14)
TON(MAX) x IOUT2 x N2/N1
IL1
IOUT2
IL2
TON(MAX) x IOUT2
Figure 13. Current Waveforms for COUT1 Ripple Calculation
A standard 1-µF, 25-V capacitor is selected for this design. If lower output voltage ripple is required, a higher
value should be selected for COUT1 and/or COUT2.
8.2.1.2.7 Secondary Output Capacitor
A simplified waveform for secondary output current (IOUT2) is shown in Figure 14.
IOUT2
IL2
TON(MAX) x IOUT2
Figure 14. Secondary Current Waveforms for COUT2 Ripple Calculation
The secondary output current (IOUT2) is sourced by COUT2 during on-time of the buck switch, TON. Ignoring the
current transition times in the secondary winding, the secondary output capacitor ripple voltage can be calculated
using Equation 15.
IOUT2 x TON (MAX)
'VOUT2 =
COUT2
(15)
For a 1:1 transformer turns ratio, the primary and secondary voltage ripple equations are identical. Therefore,
COUT2 is chosen to be equal to COUT1 (1 µF) to achieve comparable ripple voltages on primary and secondary
outputs.
If lower output voltage ripple is required, a higher value should be selected for COUT1 and/or COUT2.
Copyright © 2012–2017, Texas Instruments Incorporated
17
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
8.2.1.2.8 Type III Feedback Ripple Circuit
Type III ripple circuit as described in Ripple Configuration is required for the Fly-Buck topology. Type I and Type
II ripple circuits use series resistance and the triangular inductor ripple current to generate ripple at VOUT and the
FB pin. The primary ripple current of a Fly-Buck is the combination or primary and reflected secondary currents
as illustrated in Figure 13. In the Fly-Buck topology, Type I and Type II ripple circuits suffer from large jitter as the
reflected load current affects the feedback ripple.
VOUT
L1
Rr
Cac
C OUT
Cr
R FB2
GND
To FB
R FB1
Figure 15. Type III Ripple Circuit
Selecting the Type III ripple components using the equations from Ripple Configuration will guarantee that the FB
pin ripple is be greater than the capacitive ripple from the primary output capacitor COUT1. The feedback ripple
component values are chosen as shown in Equation 16.
Cr = 1000 pF
Cac = 0.1 PF
(VIN (MIN) - VOUT) x TON
RrCr d
50 mV
(16)
The calculated value for Rr is 66 kΩ. This value provides the minimum ripple for stable operation. A smaller
resistance should be selected to allow for variations in TON, COUT1 and other components. For this design, Rr
value of 46.4 kΩ is selected.
8.2.1.2.9 Secondary Diode
The reverse voltage across secondary-rectifier diode D1 when the high-side buck switch is off can be calculated
using Equation 17.
N2
VD1 =
VIN
N1
(17)
For a VIN_MAX of 95 V and the 1:1 turns ratio of this design, a 100-V Schottky is selected.
8.2.1.2.10 VCC and Bootstrap Capacitor
A 1-µF capacitor of 16-V or higher rating is recommended for the VCC regulator bypass capacitor.
A good value for the BST pin bootstrap capacitor is 0.01-µF with a 16-V or higher rating.
8.2.1.2.11 Input Capacitor
The input capacitor is typically a combination of a smaller bypass capacitor located near the regulator IC and a
larger bulk capacitor. The total input capacitance should be large enough to limit the input voltage ripple to a
desired amplitude. For input ripple voltage ΔVIN, CIN can be calculated using Equation 18.
CIN t
18
IOUT(MAX)
4 u ¦ u '9IN
(18)
Copyright © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
Choosing a ΔVIN of 0.5 V gives a minimum CIN of 0.067 μF. A standard value of 0.1 μF is selected for CBYP in this
design. A bulk capacitor of higher value reduces voltage spikes due to parasitic inductance between the power
source to the converter. A standard value of 1 μF is selected for CIN in this design. The voltage ratings of the two
input capacitors should be greater than the maximum input voltage under all conditions.
8.2.1.2.12 UVLO Resistors
UVLO resistors RUV1 and RUV2 set the undervoltage lockout threshold and hysteresis according to Equation 19
and Equation 20.
VIN (HYS) = IHYS x RUV2
(19)
VIN (UVLO, rising) = 1.225V x
R
( RUV2
+ 1)
(20)
UV1
Where IHYS = 20 μA, typical.
For a UVLO hysteresis of 2.5 V and UVLO rising threshold of 20 V, Equation 19 and Equation 20 require RUV1 of
8.25 kΩ and RUV2 of 127 kΩ and these values are selected for this design example.
8.2.1.2.13 VCC Diode
Diode D2 is an optional diode connected between VOUT1 and the VCC regulator output pin. When VOUT1 is more
than one diode drop greater than the VCC voltage, the VCC bias current is supplied from VOUT1. This results in
reduced power losses in the internal VCC regulator which improves converter efficiency. VOUT1 must be set to a
voltage at least one diode drop higher than 8.55 V (the maximum VCC voltage) if D2 is used to supply bias
current.
8.2.2 Application Curves
Figure 16. Efficiency at 750 kHz, VOUT1 = 10 V
Copyright © 2012–2017, Texas Instruments Incorporated
Figure 17. Steady State Waveform
(VIN = 48 V, IOUT1 = 0 mA, IOUT2 = 100 mA)
19
LM34925
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
www.ti.com.cn
9 Power Supply Recommendations
LM34925 is a power management device. The power supply for the device is any dc voltage source within the
specified input range.
10 Layout
10.1 Layout Guidelines
A proper layout is essential for optimum performance of the circuit. In particular, the following guidelines should
be observed:
• CIN: The loop consisting of input capacitor (CIN), VIN pin, and RTN pin carries switching currents. Therefore
the input capacitor should be placed close to the IC, directly across VIN and RTN pins and the connections to
these two pins should be direct to minimize the loop area. In general it is not possible to accommodate all of
input capacitance near the IC. A good practice is to use a 0.1-μF or 0.47-μF capacitor directly across the VIN
and RTN pins close to the IC, and the remaining bulk capacitor as close as possible (see Figure 18).
• CVCC and CBST: The VCC and bootstrap (BST) bypass capacitors supply switching currents to the high and
low-side gate drivers. These two capacitors should also be placed as close to the IC as possible, and the
connecting trace lengths and loop area should be minimized (see Figure 18).
• The Feedback trace carries the output voltage information and a small ripple component that is necessary for
proper operation of LM34925. Therefore care should be taken while routing the feedback trace so avoid
coupling any noise to this pin. In particular, feedback trace should not run close to magnetic components, or
parallel to any other switching trace.
• SW trace: SW node switches rapidly between VIN and GND every cycle and is therefore a possible source of
noise. SW node area should be minimized. In particular SW node should not be inadvertently connected to a
copper plane or pour.
10.2 Layout Example
RTN
1
8
SW
VIN
2
7
BST
UVLO
3
6
VCC
RON
4
5
FB
CIN
CVCC
Figure 18. Placement of Bypass Capacitors
20
版权 © 2012–2017, Texas Instruments Incorporated
LM34925
www.ti.com.cn
ZHCSD58G – JUNE 2012 – REVISED NOVEMBER 2017
11 器件和文档支持
11.1 文档支持
11.1.1 相关文档
请参阅如下相关文档：
• AN-1481《在恒定导通时间
(COT)
稳压器设计中控制输出纹波并获得
号：SNVA166）
• AN-2285《LM34925 隔离评估板》（文献编号：SNVA676）
• AN-2292《设计隔离式降压 (Fly-buck) 转换器》（文献编号：SNVA674）
ESR
非相关性》（文献编
11.2 接收文档更新通知
要接收文档更新通知，请转至 TI.com 上的器件产品文件夹。单击右上角的通知我 进行注册，即可每周接收产品信
息更改摘要。有关更改的详细信息，请查看任何已修订文档中包含的修订历史记录。
11.3 社区资源
下列链接提供到 TI 社区资源的连接。链接的内容由各个分销商“按照原样”提供。这些内容并不构成 TI 技术规范，
并且不一定反映 TI 的观点；请参阅 TI 的 《使用条款》。
TI E2E™ 在线社区 TI 的工程师对工程师 (E2E) 社区。此社区的创建目的在于促进工程师之间的协作。在
e2e.ti.com 中，您可以咨询问题、分享知识、拓展思路并与同行工程师一道帮助解决问题。
设计支持
TI 参考设计支持 可帮助您快速查找有帮助的 E2E 论坛、设计支持工具以及技术支持的联系信息。
11.4 商标
PowerPAD, Fly-Buck, E2E are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 静电放电警告
这些装置包含有限的内置 ESD 保护。 存储或装卸时，应将导线一起截短或将装置放置于导电泡棉中，以防止 MOS 门极遭受静电损
伤。
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 机械、封装和可订购信息
以下页面包含机械、封装和可订购信息。这些信息是指定器件的最新可用数据。这些数据如有变更，恕不另行通知
和修订此文档。如欲获取此数据表的浏览器版本，请参阅左侧的导航。
版权 © 2012–2017, Texas Instruments Incorporated
21
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM34925MR/NOPB
ACTIVE SO PowerPAD
DDA
8
95
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
S000YB
LM34925MRX/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
S000YB
LM34925SD/NOPB
ACTIVE
WSON
NGU
8
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L34925
LM34925SDX/NOPB
ACTIVE
WSON
NGU
8
4500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L34925
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
