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LM34927
SNVS799H – APRIL 2012 – REVISED NOVEMBER 2017
LM34927 Integrated Secondary-Side Bias Regulator for Isolated DC-DC Converters
1 Features
3 Description
•
•
The LM34927 regulator features all of the functions
needed to implement a low-cost, efficient, isolated
bias regulator. This high-voltage regulator contains
two 100-V N-channel MOSFET switches — a highside buck switch and a low-side synchronous switch.
The Constant on-time (COT) control scheme
employed in the LM34927 device requires no loop
compensation and provides excellent transient
response. The regulator operates with an on-time that
is inversely proportional to the input voltage. This
feature allows the operating frequency to remain
relatively constant. An intelligent peak current limit is
implemented with integrated sense circuit. Other
features include a programmable input undervoltage
comparator to inhibit operation during low-voltage
conditions. Protection features include thermal
shutdown and VCC undervoltage lockout (UVLO). The
LM34927 device is offered in WSON-8 and SO
PowerPAD-8 plastic packages.
1
•
•
•
•
•
•
•
•
•
•
•
•
•
Wide 7.5-V to 100-V Input Range
Integrated 600-mA High-Side
and Low-Side Switches
No Schottky Required
Constant On-Time Control
No Loop Compensation Required
Ultra-Fast Transient Response
Nearly Constant Operating Frequency
Intelligent Peak Current Limit
Adjustable Output Voltage From 1.225 V
Precision 2% Feedback Reference
Frequency Adjustable to 1 MHz
Adjustable Undervoltage Lockout (UVLO)
Remote Shutdown
Thermal Shutdown
Packages:
– 8-pin WSON
– 8-pin SO PowerPAD™
Device Information(1)
PART NUMBER
PACKAGE
LM34927
2 Applications
•
•
BODY SIZE (NOM)
SO PowerPAD (8)
4.89 mm × 3.90 mm
WSON (8)
4.00 mm × 4.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Isolated Telecom Bias Supply
Isolated Automotive and Industrial Electronics
space
space
Typical Application
D1
VOUT2
+
LM34927
VIN
9V-100V
CIN
2
+
4
RUV2
BST
VIN
RON
VCC
UVLO
FB
RUV1
RTN
1
COUT2
7
+
SW 8
RON
3
NS
X1
CBST
VOUT1
NP
6
D2
5
+
CVCC
Rr
RFB2
+
RFB1
COUT1
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM34927
SNVS799H – APRIL 2012 – REVISED NOVEMBER 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
4
5
6.1
6.2
6.3
6.4
6.5
6.6
6.7
5
5
5
5
6
7
7
Absolute Maximum Ratings .....................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 10
7.1 Overview ................................................................. 10
7.2 Functional Block Diagram ....................................... 10
7.3 Feature Description................................................. 11
7.4 Device Functional Modes........................................ 15
8
Application and Implementation ........................ 16
8.1 Application Information............................................ 16
8.2 Typical Applications ................................................ 16
9 Power Supply Recommendations...................... 21
10 Layout................................................................... 21
10.1 Layout Guidelines ................................................. 21
10.2 Layout Example .................................................... 21
10.3 Thermal Curves..................................................... 21
11 Device and Documentation Support ................. 22
11.1
11.2
11.3
11.4
11.5
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
22
22
22
22
22
12 Mechanical, Packaging, and Orderable
Information ........................................................... 22
4 Revision History
Changes from Revision G (December 2014) to Revision H
Page
•
Deleted lead temperature from Abs Max table ...................................................................................................................... 5
•
Changed 14 V to 13 V in VCC Regulator section ................................................................................................................. 12
Changes from Revision F (December 2013) to Revision G
Page
•
Added Pin Configuration and Functions section, ESD Rating table, Feature Description, Device Functional Modes,
Application and Implementation, Power Supply Recommendations, Layout, Device and Documentation Support, and
Mechanical, Packaging, and Orderable Information sections ............................................................................................... 1
•
Changed Thermal Information table ....................................................................................................................................... 5
•
Changed TON vs VIN and RON in Typical Characteristics ....................................................................................................... 7
•
Deleted 3-W Isolated Bias Application Schematic, Lowest Part Count Isolated Application Schematic, and Typical
Buck Configuration graphics................................................................................................................................................... 9
•
Changed Control Overview section ...................................................................................................................................... 11
•
Changed Soft-Start Circuit, Isolated Fly-Buck Converter graphics. ..................................................................................... 15
Changes from Revision E (December 2013) to Revision F
Page
•
Added Thermal Parameters. .................................................................................................................................................. 5
•
Added Thermal Derating Curve............................................................................................................................................ 21
2
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Changes from Revision D (September 2013) to Revision E
Page
•
Changed min. operating input voltge from 9 V to 7.5 V in Features and Typical Application; format to TI standard ........... 1
•
Added abs max junction temp; changed min op. input V from 9 to 7.5 V in ROC table ........................................................ 5
Changes from Revision C (July 2013) to Revision D
•
Page
Added SW to RTN (100-ns transient) to Absolute Maximum Ratings .................................................................................. 5
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SNVS799H – APRIL 2012 – REVISED NOVEMBER 2017
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5 Pin Configuration and Functions
DDA Package
8-Pin SO PowerPAD
Top View
RTN
1
VIN
2
UVLO
3
RON
4
SO
PowePAD-8
Exp Pad
8
SW
7
BST
6
VCC
5
FB
Connect Exposed Pad to RTN
8-Pin WSON With Exposed Thermal Pad
NGU Package
Top View
RTN
1
VIN
2
UVLO
3
RON
4
8 SW
WSON-8
Exp Pad
7 BST
6 VCC
5 FB
Pin Functions
PIN
NO.
NAME
I/O
1
RTN
—
2
VIN
DESCRIPTION
APPLICATION INFORMATION
Ground
Ground connection of the integrated circuit.
I
Input voltage
Operating input range is 7.5 V to 100 V.
Resistor divider from VIN to UVLO to GND programs the
undervoltage detection threshold. An internal current source is
enabled when UVLO is above 1.225 V to provide hysteresis.
When UVLO pin is pulled below 0.66 V externally, the parts goes
in shutdown mode.
3
UVLO
I
Input pin of undervoltage
comparator
4
RON
I
On-time control
A resistor between this pin and VIN sets the switch on-time as a
function of VIN. Minimum recommended on-time is 100 ns at max
input voltage.
5
FB
I
Feedback
This pin is connected to the inverting input of the internal
regulation comparator. The regulation level is 1.225 V.
6
VCC
O
Output from the internal high
voltage series pass regulator;
regulated at
7.6 V.
7
BST
I
Bootstrap capacitor
An external capacitor is required between the BST and SW pins
(0.01-μF ceramic). The BST pin capacitor is charged by the VCC
regulator through an internal diode when the SW pin is low.
8
SW
O
Switching node
Power switching node. Connect to the output inductor and
bootstrap capacitor.
—
EP
—
Exposed pad
Exposed pad must be connected to RTN pin. Connect to system
ground plane on application board for reduced thermal resistance.
4
The internal VCC regulator provides bias supply for the gate
drivers and other internal circuitry. A 1.0-μF decoupling capacitor
is recommended.
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6 Specifications
6.1 Absolute Maximum Ratings
(1)
See
MIN
MAX
UNIT
VIN, UVLO to RTN
–0.3
100
V
SW to RTN
–1.5
VIN + 0.3
V
–5
VIN + 0.3
V
100
V
SW to RTN (100-ns transient)
BST to VCC
BST to SW
13
V
RON to RTN
–0.3
100
V
VCC to RTN
–0.3
13
V
FB to RTN
–0.3
Maximum junction temperature (2)
Storage temperature range
(1)
(2)
–55
5
V
150
°C
150
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
High junction temperatures degrade operating lifetimes. Operating lifetime is de-rated for junction temperatures greater than 125°C.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±750
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
VIN voltage
7.5
100
V
Operating junction temperature (1)
–40
125
°C
(1)
UNIT
High junction temperatures degrade operating lifetimes. Operating lifetime is de-rated for junction temperatures greater than 125°C.
6.4 Thermal Information
LM34927
THERMAL METRICS
(1)
NGU (WSON)
DDA (SO
PowerPAD)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
41.3
41.1
°C/W
RθJCbot
Junction-to-case (bottom) thermal resistance
3.2
2.4
°C/W
ΨJB
Junction-to-board thermal characteristic parameter
19.2
24.4
°C/W
RθJB
Junction-to-board thermal resistance
19.1
30.6
°C/W
RθJCtop
Junction-to-case (top) thermal resistance
34.7
37.3
°C/W
ΨJT
Junction-to-top thermal characteristic parameter
0.3
6.7
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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SNVS799H – APRIL 2012 – REVISED NOVEMBER 2017
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6.5 Electrical Characteristics
Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over –40°C to 125°C junction temperature
range, unless otherwise stated. VIN = 48 V unless otherwise stated. See (1).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
6.25
7.6
8.55
UNIT
VCC SUPPLY
VCC Reg
VCC Regulator Output
VIN = 48 V, ICC = 20 mA
VCC Current Limit
VIN = 48 V (2)
26
VCC UVLO Threshold (VCC increasing)
4.15
VCC UVLO Hysteresis
4.5
4.9
300
VCC Dropout Voltage
VIN = 8 V, ICC = 20 mA
IIN Operating Current
Non-switching, FB = 3 V
IIN Shutdown Current
UVLO = 0 V
V
mA
V
mV
2.3
V
1.75
mA
50
225
µA
UNDERVOLTAGE SENSING FUNCTION
UV Threshold
UV Rising
1.19
1.225
1.26
V
UV Hysteresis Input Current
UV = 2.5 V
–10
–20
–29
µA
Remote Shutdown Threshold
Voltage at UVLO Falling
0.32
0.66
V
110
mV
Remote Shutdown Hysteresis
REGULATION AND OVERVOLTAGE COMPARATORS
FB Regulation Level
Internal Reference Trip Point for Switch ON
FB Overvoltage Threshold
Trip Point for Switch OFF
1.2
FB Bias Current
1.225
1.25
V
1.62
V
60
nA
SWITCH CHARACTERISTICS
Buck Switch RDS(ON)
ITEST = 200 mA, BST-SW = 7 V
Synchronous RDS(ON)
ITEST = 200 mA
Gate Drive UVLO
VBST − VSW Rising
2.4
Gate Drive UVLO Hysteresis
0.8
1.8
Ω
0.45
1
Ω
3
3.6
V
260
mV
CURRENT LIMIT
Current Limit Threshold
0.7
Current Limit Response Time
Time to switch off
OFF-Time Generator (Test 1)
OFF-Time Generator (Test 2)
1.02
1.3
A
150
ns
FB = 0.1 V, VIN = 48 V
12
µs
FB = 1.0 V, VIN = 48 V
2.5
µs
165
°C
20
°C
THERMAL SHUTDOWN
Tsd
Thermal Shutdown Temperature
Thermal Shutdown Hysteresis
(1)
(2)
6
All limits are specified by design. All electrical characteristics having room temperature limits are tested during production at TA = 25°C.
All hot and cold limits are specified by correlating the electrical characteristics to process and temperature variations and applying
statistical process control.
VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.
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6.6 Switching Characteristics
Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over –40°C to 125°C junction temperature range
unless otherwise stated. VIN = 48 V unless otherwise stated.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ON-TIME GENERATOR
TON Test 1
VIN = 32 V, RON = 100 kΩ
270
350
460
ns
TON Test 2
VIN = 48 V, RON = 100 kΩ
188
250
336
ns
TON Test 3
VIN = 75 V, RON = 250 kΩ
250
370
500
ns
TON Test 4
VIN = 10 V, RON = 250 kΩ
1880
3200
4425
ns
MINIMUM OFF-TIME
Minimum Off-Timer
FB = 0 V
144
ns
6.7 Typical Characteristics
Figure 1. Efficiency at 750 kHz, VOUT1 = 10 V
Figure 2. VCC vs VIN
Figure 3. VCC vs ICC
Figure 4. ICC vs External VCC
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Typical Characteristics (continued)
8
Figure 5. TON vs VIN and RON
Figure 6. TOFF (ILIM) vs VFB and VIN
Figure 7. IIN vs VIN (Operating, Non-switching)
Figure 8. IIN vs VIN (Shutdown)
Figure 9. Efficiency
Figure 10. Buck Transient Response 36 VIN
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Typical Characteristics (continued)
Figure 11. Buck Transient Response 36 VIN
Figure 12. Buck Transient Response 48 VIN
Figure 13. Buck Transient Response 48 VIN
Figure 14. Buck Transient Response 72 VIN
Figure 15. Buck Transient Response 72 VIN
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7 Detailed Description
7.1 Overview
The LM34927 step-down switching regulator features all the functions needed to implement a low-cost, efficient,
isolated-bias supply. This high-voltage regulator contains 100 V, N-channel buck and synchronous switches, is
easy to implement, and is provided in thermally enhanced SO PowerPAD-8 and WSON-8 packages. The
regulator operation is based on a constant on-time control scheme using an on-time inversely proportional to VIN.
This control scheme does not require loop compensation. Current limit is implemented with forced off-time
inversely proportional to VOUT. This scheme ensures short-circuit protection while providing minimum foldback.
The simplified block diagram of the LM34927 device is shown in Functional Block Diagram.
The LM34927 device can be applied in numerous applications to efficiently regulate down higher voltages. This
regulator is well suited for 48-V telecom and automotive power bus ranges. Protection features include: thermal
shutdown, UVLO, minimum forced off-time, and an intelligent current limit.
7.2 Functional Block Diagram
LM34927
START-UP
REGULATOR
VIN
VCC
V UVLO
20 µA
4.5V
UVLO
THERMAL
SHUTDOWN
UVLO
1.225V
SD
VDD REG
BST
0.66V
SHUTDOWN
BG REF
VIN
DISABLE
ON/OFF
TIMERS
RON
1.225V
SW
COT CONTROL
LOGIC
FEEDBACK
FB
OVER-VOLTAGE
1.62V
RTN
10
CURRENT
LIMIT
ONE-SHOT
ILIM
COMPARATOR
+
VILIM
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7.3 Feature Description
7.3.1 Control Overview
The LM34927 regulator employs a control principle based on a comparator and a one-shot on-timer, with the
output voltage feedback (FB) compared to an internal reference (1.225 V). If the FB voltage is below the
reference the internal buck switch is switched on for the one-shot timer period, which is a function of the input
voltage and the programming resistor (RT). Following the on-time the switch remains off until the FB voltage falls
below the reference, and the forced minimum off-time has expired. When the FB pin voltage falls below the
reference and the off-time one-shot period expires, the buck switch is then turned on for another on-time oneshot period. This continues until regulation is achieved and the FB voltage is approximately equal to 1.225 V
(typ).
In a synchronous buck converter, the low-side (sync) FET is on when the high side (buck) FET is off. The
inductor current ramps up when the high-side switch is on and ramps down when the high side switch is off.
There is no diode emulation feature in this IC, and therefore, the inductor current may ramp in the negative
direction at light load. This causes the converter to operate in continuous conduction mode (CCM) regardless of
the output loading. The operating frequency remains relatively constant with load and line variations. The
operating frequency can be calculated as shown in Equation 1.
VOUT1
f SW =
. x RON
(1)
Where K = 9 × 10–11
The output voltage (VOUT) is set by two external resistors ®FB1, RFB2). The regulated output voltage is calculated
as shown in Equation 2.
VOUT = 1.225V x
RFB2 + RFB1
RFB1
(2)
This regulator regulates the output voltage based on ripple voltage at the feedback input, requiring a minimum
amount of ESR for the output capacitor ©OUT). A minimum of 25 mV of ripple voltage at the feedback pin (FB) is
required for the LM34927 device. In cases where the capacitor ESR is too small, additional series resistance
may be required ®C in Figure 16).
For applications where lower output voltage ripple is required, the output can be taken directly from a low ESR
output capacitor, as shown in Figure 16. However, RC slightly degrades the load regulation.
L1
VOUT
SW
LM34927
RFB2
FB
RC
+
RFB1
COUT
VOUT
(low ripple)
Figure 16. Low Ripple Output Configuration
7.3.2 VCC Regulator
The LM34927 device contains an internal high voltage linear regulator with a nominal output of 7.6 V. The input
pin (VIN) can be connected directly to the line voltages up to 100 V. The VCC regulator is internally current limited
to 30 mA. The regulator sources current into the external capacitor at VCC. This regulator supplies current to
internal circuit blocks including the synchronous MOSFET driver and the logic circuits. When the voltage on the
VCC pin reaches the UVLO threshold of 4.5 V, the IC is enabled.
The VCC regulator contains an internal diode connection to the BST pin to replenish the charge in the gate drive
boot capacitor when SW pin is low.
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Feature Description (continued)
At high input voltages, the power dissipated in the high-voltage regulator is significant and can limit the overall
achievable output power. As an example, with the input at 48 V and switching at high frequency, the VCC
regulator may supply up to 7 mA of current resulting in 48 V × 7 mA = 336 mW of power dissipation. If the VCC
voltage is driven externally by an alternate voltage source, from 8.55 V to 13 V, the internal regulator is disabled.
This reduces the power dissipation in the IC.
7.3.3 Regulation Comparator
The feedback voltage at FB is compared to an internal 1.225 V reference. In normal operation, when the output
voltage is in regulation, an on-time period is initiated when the voltage at FB falls below 1.225 V. The high-side
switch will stay on for the on-time, causing the FB voltage to rise above 1.225 V. After the on-time period, the
high-side switch will stay off until the FB voltage again falls below 1.225 V. During start-up, the FB voltage will be
below 1.225 V at the end of each on-time causing the high-side switch to turn on immediately after the minimum
forced off-time of 144 ns. The high-side switch can be turned off before the on-time is over if peak current in the
inductor reaches the current limit threshold.
7.3.4 Overvoltage Comparator
The feedback voltage at FB is compared to an internal 1.62-V reference. If the voltage at FB rises above 1.62 V
the on-time pulse is immediately terminated. This condition can occur if the input voltage and/or the output load
changes suddenly. The high-side switch will not turn on again until the voltage at FB falls below 1.225 V.
7.3.5 On-Time Generator
The on-time for the LM34927 device is determined by the RON resistor, and is inversely proportional to the input
voltage (VIN), resulting in a nearly constant frequency as VIN is varied over its range. The on-time equation for the
LM34927 device is shown in Equation 3.
TON =
10-10 x RON
VIN
(3)
See Figure 5. RON should be selected for a minimum on-time (at maximum VIN) greater than 100 ns for proper
operation. This requirement limits the maximum frequency for each application.
7.3.6 Current Limit
The LM34927 device contains an intelligent current limit off-timer. If the current in the buck switch exceeds
1.02 A, the present cycle is immediately terminated, and a non-resetable off-timer is initiated. The length of offtime is controlled by the FB voltage and the input voltage VIN. As an example, when FB = 0 V and VIN = 48 V, a
maximum off-time is set to 16 μs. This condition occurs when the output is shorted, and during the initial part of
start up. This amount of time ensures safe short circuit operation up to the maximum input voltage of
100 V.
In cases of overload where the FB voltage is above zero volts (not a short circuit) the current limit off-time is
reduced. Reducing the off-time during less severe overloads reduces the amount of foldback, recovery time, and
start-up time. The off-time is calculated from Equation 4.
TOFF(ILIM)
0.07 u VIN
Ps
VFB 0.2 V
(4)
The current limit protection feature is peak limited, the maximum average output will be less than the peak.
7.3.7 N-Channel Buck Switch and Driver
The LM34927 device integrates an N-Channel Buck switch and associated floating high-voltage gate driver. The
gate driver circuit works in conjunction with an external bootstrap capacitor and an internal high-voltage diode. A
0.01-µF ceramic capacitor connected between the BST pin and SW pin provides the voltage to the driver during
the on-time. During each off-time, the SW pin is at approximately 0 V and the bootstrap capacitor charges from
VCC through the internal diode. The minimum off-timer, set to 144 ns, ensures a minimum time each cycle to
recharge the bootstrap capacitor.
12
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Feature Description (continued)
7.3.8 Synchronous Rectifier
The LM34927 device provides an internal synchronous N-channel MOSFET rectifier. This MOSFET provides a
path for the inductor current to flow when the high-side MOSFET is turned off.
The synchronous rectifier has no diode emulation mode, and is designed to keep the regulator in continuous
conduction mode even during light loads which would otherwise result in discontinuous operation. This feature
specifically lets the user design a secondary regulator using a transformer winding off the main inductor to
generate the alternate regulated output voltage.
7.3.9 Undervoltage Detector
The LM34927 device contains a dual-level UVLO circuit. When the UVLO pin voltage is below 0.66 V, the
controller is in a low current shutdown mode. When the UVLO pin voltage is greater than 0.66 V but less than
1.225 V, the controller is in standby mode. In standby mode the VCC bias regulator is active while the regulator
output is disabled. When the VCC pin exceeds the VCC undervoltage thresholds and the UVLO pin voltage is
greater than 1.225 V, normal operation begins. An external set-point voltage divider from VIN to GND can be
used to set the minimum operating voltage of the regulator.
UVLO hysteresis is accomplished with an internal 20-μA current source that is switched on or off into the
impedance of the set-point divider. When the UVLO threshold is exceeded, the current source is activated to
quickly raise the voltage at the UVLO pin. The hysteresis is equal to the value of this current times the resistance
RUV2.
If the UVLO pin is wired directly to the VIN pin, the regulator will begin operation once the VCC undervoltage is
satisfied.
VIN
2
CIN
VIN
+
RUV2
LM34927
3
UVLO
RUV1
Figure 17. UVLO Resistor Setting
7.3.10 Thermal Protection
The LM34927 device should be operated so the junction temperature does not exceed 150°C during normal
operation. An internal Thermal Shutdown circuit is provided to protect the LM34927 in the event of a higher than
normal junction temperature. When activated, typically at 165°C, the controller is forced into a low-power reset
state, disabling the buck switch and the VCC regulator. This feature prevents catastrophic failures from accidental
device overheating. When the junction temperature reduces below 145°C (typical hysteresis = 20°C), the VCC
regulator is enabled, and normal operation is resumed.
7.3.11 Ripple Configuration
LM34927 uses COT control scheme, in which the on-time is terminated by an on-timer, and the off-time is
terminated by the feedback voltage (VFB) falling below the reference voltage (VREF). Therefore, for stable
operation, the feedback voltage must decrease monotonically, in phase with the inductor current during the offtime. Furthermore this change in feedback voltage (ΔVFB) during off-time must be large enough to suppress any
noise component present at the feedback node.
Table 1 shows three different methods for generating appropriate voltage ripple at the feedback node. Type 1
and Type 2 ripple circuits couple the ripple at the output of the converter to the feedback node (FB). The output
voltage ripple has two components:
• Capacitive ripple caused by the inductor current ripple charging/discharging the output capacitor.
• Resistive ripple caused by the inductor current ripple flowing through the ESR of the output capacitor.
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Feature Description (continued)
The capacitive ripple is not in phase with the inductor current. As a result, the capacitive ripple does not
decrease monotonically during the off-time. The resistive ripple is in phase with the inductor current and
decreases monotonically during off-time. The resistive ripple must exceed the capacitive ripple at the output node
(VOUT) for stable operation. If this condition is not satisfied, unstable switching behavior is observed in COT
converters, with multiple on-time bursts in close succession followed by a long off-time.
Type 3 ripple method uses Rr and Cr and the switch node (SW) voltage to generate a triangular ramp. This
triangular ramp is ac coupled using Cac to the feedback node (FB). Because this circuit does not use the output
voltage ripple, it is ideally suited for applications where low output voltage ripple is required. See AN-1481
Controlling Output Ripple and Achieving ESR Independence in Constant On-Time (COT) Regulator Designs
(SNVA166) for more details for each ripple generation method.
Table 1. Ripple Configuration
TYPE 1
LOWEST COST CONFIGURATION
TYPE 2
REDUCED RIPPLE CONFIGURATION
TYPE 3
MINIMUM RIPPLE CONFIGURATION
VOUT
VOUT
L1
VOUT
L1
L1
Cac
R FB2
R FB2
RC
To FB
R FB1
R FB1
GND
25 mV VOUT
x
ûIL(MIN) VREF
Cr
R FB2
C OUT
Cac
To FB
C OUT
RC >
Rr
RC
GND
C OUT
To FB
R FB1
GND
C>
(5)
Cr = 3300 pF
Cac = 100 nF
(VIN(MIN) - VOUT) x TON
R rC r <
25 mV
5
gsw (RFB2||RFB1)
25 mV
RC >
ûIL(MIN)
(6)
(7)
7.3.12 Soft Start
A soft-start feature can be implemented with the LM34927 device using an external circuit. As shown in
Figure 18, the soft-start circuit consists of one capacitor C1, two resistors R1 and R2, and a diode D. During the
initial start-up, the VCC voltage is established prior to the VOUT voltage. Capacitor C1 is discharged and D is
thereby forward biased to pull up the FB voltage. The FB voltage exceeds the reference voltage (1.225 V) and
switching is therefore disabled. As capacitor C1 charges, the voltage at node B gradually decreases and
switching commences. VOUT will gradually rise to maintain the FB voltage at the reference voltage. Once the
voltage at node B is less than a diode drop above the FB voltage, the soft-start sequence is finished and D is
reverse biased.
During the initial part of the start up, the FB voltage can be approximated as shown in Equation 8. The effect of
R1 has been ignored to simplify the calculation.
VFB = (VCC - VD) x
RFB1 x RFB2
R2 x (RFB1 + RFB2) + RFB1 x RFB2
(8)
C1 is charged after the first start up. Diode D1 is optional and can be added to discharge C1 and initialize the
soft-start sequence when the input voltage experiences a momentary drop.
14
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To achieve the desired soft start, the following design guidance is recommended:
• R2 is selected so that VFB is higher than 1.225 V for a VCC of 4.5 V, but is lower than 5 V when VCC is 8.55 V.
If an external VCC is used, VFB should not exceed 5 V at maximum VCC.
• C1 is selected to achieve the desired start-up time which can be determined from Equation 9.
RFB1 x RFB2
)
tS = C1 x (R2 +
RFB1 + RFB2
(9)
• R1 is used to maintain the node B voltage at zero after the soft start is finished. A value larger than the
feedback resistor divider is preferred. The effect of resistor R1 is ignored.
Using component values shown in Figure 19, selecting C1 = 1 µF, R2 = 1 kΩ, R1 = 30 kΩ results in a soft-start
time of about 2 ms.
VOUT
VCC
C1
RFB2
R2
To FB
D
D1
B
RFB1
R1
Figure 18. Soft-Start Circuit
7.4 Device Functional Modes
Table 2. Undervoltage Detector
UVLO
VCC
MODE
DESCRIPTION
< 0.66 V
Disabled
Shutdown
VCC regulator disabled.
Switching disabled.
0.66 V to 1.225 V
Enabled
Standby
VCC regulator enabled.
Switching disabled.
VCC < 4.5 V
Standby
VCC regulator enabled.
Switching disabled.
VCC > 4.5 V
Operating
> 1.225 V
VCC enabled.
Switching enabled.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM34927 device is step-down DC-DC converter. The device is typically used to convert a higher DC voltage
to a lower DC voltage with a maximum available output current of 600 mA. Use the following design procedure to
select component values for the LM34927 device. Alternately, use the WEBENCH® software to generate a
complete design. The WEBENCH software uses an iterative design procedure and accesses a comprehensive
database of components when generating a design. This section presents a simplified discussion of the design
process.
8.2 Typical Applications
8.2.1 Application Circuit: 20-V to 95-V Input and 10-V, 300-mA Output Isolated Fly-Buck™ Converter
VOUT2
D1
+
N2
X1
1:1
LM34927
BST
VIN
20V-95V
0.01 µF
+
CBST
+
N1
33 µH
VOUT1
SW
46.4 NŸ 1 nF
Rr
Cr
VIN
CIN
2.2 µF
COUT2
1 µF
CBYP
0.47 µF
+
RON
RUV2
127 NŸ
RON
130 NŸ
RUV1
8.25 NŸ
0.1 µF
RTN
COUT1
1 µF
RFB2
VCC
UVLO
+
Cac
FB
+
D2
7.32 NŸ
CVCC
1 µF
RFB1
1 NŸ
Figure 19. Isolated Fly-Buck Converter Using LM34927
8.2.1.1 Design Requirements
Selection of external components is illustrated through a design example. The design example specifications are
shown in Table 3.
Table 3. Buck Converter Design Specifications
DESIGN PARAMETERS
VALUE
Input voltage range
20 V to 95 V
Primary output voltage
10 V
Secondary (isolated) output voltage
9.5 V
Maximum output current (primary + secondary)
300 mA
Maximum power output
3W
Nominal switching frequency
16
750 kHz
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8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Transformer Turns Ratio
The transformer turns ratio is selected based on the ratio of the primary output voltage to the secondary
(isolated) output voltage. In this design example, the two outputs are nearly equal and a 1:1 turns ratio
transformer is selected. Therefore, N2 / N1 = 1.
If the secondary (isolated) output voltage is significantly higher or lower than the primary output voltage, a turns
ratio less than or greater than 1 is recommended. The primary output voltage is normally selected based on the
input voltage range such that the duty cycle of the converter does not exceed 50% at the minimum input voltage.
This condition is satisfied if VOUT1 < VIN_MIN / 2.
8.2.1.2.2 Total IOUT
The total primary referred load current is calculated by multiplying the isolated output loads by the turns ratio of
the transformer as shown in Equation 10.
IOUT(MAX)
IOUT1 IOUT2 u
N2
N1
0.3 A
(10)
8.2.1.2.3 RFB1, RFB2
The feedback resistors are selected to set the primary output voltage. The selected value for RFB1 is 1 kΩ. RFB2
can be calculated using the following equations to set VOUT1 to the specified value of 10 V. A standard resistor
value of 7.32 kΩ is selected for RFB2.
VOUT1 = 1.225V x (1 +
RFB2
)
RFB1
(11)
VOUT1
- 1) x RFB1 = 7.16 k:
: RFB2 = (
1.225
(12)
8.2.1.2.4 Frequency Selection
Equation 13 is used to calculate the value of RON required to achieve the desired switching frequency.
VOUT1
f SW =
. x RON
where
K = 9 × 10–11
(13)
VOUT1 of 10 V and fSW of 750 kHz, the calculated value of RON is 148 kΩ. A lower value of 130 kΩ is selected for
this design to allow for second-order effects at high switching frequency that are not included in Equation 13.
8.2.1.2.5 Transformer Selection
A coupled inductor or a flyback-type transformer is required for this topology. Energy is transferred from primary
to secondary when the low-side synchronous switch of the buck converter is conducting.
The maximum inductor primary ripple current that can be tolerated without exceeding the buck switch peak
current limit threshold (0.7 A minimum) is given by Equation 14.
'IL1
N2 ·
§
¨ 0.7 IOUT1 IOUT2 u N1 ¸ u 2
©
¹
0.8 A
(14)
Using the maximum peak-to-peak inductor ripple current ΔIL1 from Equation 14, the minimum inductor value is
given by Equation 15.
L1
VIN(MAX)
VOUT
'IL1 u fSW
u
VOUT
VIN(MAX)
14.9 PH
(15)
A higher value of 33 µH is selected to insure the high-side switch current does not exceed the minimum peak
current limit threshold. With this inductance, the inductor current ripple is ΔIL1= 0.36 A at the maximum VIN.
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8.2.1.2.6 Primary Output Capacitor
In a conventional buck converter the output ripple voltage is calculated as shown in Equation 16.
f
'VOUT =
'IL1
x f x COUT1
(16)
To limit the primary output ripple voltage ΔVOUT1 to approximately 50 mV, an output capacitor COUT1 of 1.2 µF
would be required for a conventional buck.
Figure 20 shows the primary winding current waveform (IL1) of a Fly-Buck converter. The reflected secondary
winding current adds to the primary winding current during the buck switch off-time. Because of this increased
current, the output voltage ripple is not the same as in conventional buck converter. The output capacitor value
calculated in Equation 16 should be used as the starting point. Optimization of output capacitance over the entire
line and load range must be done experimentally. If the majority of the load current is drawn from the secondary
isolated output, a better approximation of the primary output voltage ripple is given by Equation 17.
'VOUT1
N2 ·
§
¨ IOUT2 u N1 ¸ u TON(MAX)
©
¹
| 67 mV
COUT1
(17)
TON(MAX) x IOUT2 x N2/N1
IL1
IOUT2
IL2
TON(MAX) x IOUT2
Figure 20. Current Waveforms for COUT1 Ripple Calculation
A standard 1-µF, 25-V capacitor is selected for this design. If lower output voltage ripple is required, a higher
value should be selected for COUT1 and/or COUT2.
8.2.1.2.7 Secondary Output Capacitor
A simplified waveform for secondary output current (IOUT2) is shown in Figure 21.
IOUT2
IL2
TON(MAX) x IOUT2
Figure 21. Secondary Current Waveforms for COUT2 Ripple Calculation
The secondary output current (IOUT2) is sourced by COUT2 during on-time of the buck switch, TON. Ignoring the
current transition times in the secondary winding, the secondary output capacitor ripple voltage can be calculated
using Equation 18.
IOUT2 x TON (MAX)
'VOUT2 =
COUT2
(18)
For a 1:1 transformer turns ratio, the primary and secondary voltage ripple equations are identical. Therefore,
COUT2 is chosen to be equal to COUT1 (1 µF) to achieve comparable ripple voltages on primary and secondary
outputs.
If lower output voltage ripple is required, a higher value should be selected for COUT1 and/or COUT2.
18
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8.2.1.2.8 Type III Feedback Ripple Circuit
Type III ripple circuit as described in Ripple Configuration is required for the Fly-Buck topology. Type I and Type
II ripple circuits use series resistance and the triangular inductor ripple current to generate ripple at VOUT and the
FB pin. The primary ripple current of a Fly-Buck is the combination or primary and reflected secondary currents
as illustrated in Figure 20. In the Fly-Buck topology, Type I and Type II ripple circuits suffer from large jitter as the
reflected load current affects the feedback ripple.
VOUT
L1
Rr
Cac
C OUT
Cr
R FB2
GND
To FB
R FB1
Figure 22. Type III Ripple Circuit
Selecting the Type III ripple components using the equations from Ripple Configuration will ensure that the FB
pin ripple is be greater than the capacitive ripple from the primary output capacitor COUT1. The feedback ripple
component values are chosen as shown in Equation 19.
Cr = 1000 pF
Cac = 0.1 PF
RrCr d
(VIN (MIN) - VOUT) x TON
50 mV
(19)
The calculated value for Rr is 66 kΩ. This value provides the minimum ripple for stable operation. A smaller
resistance should be selected to allow for variations in TON, COUT1 and other components. For this design, Rr
value of 46.4 kΩ is selected.
8.2.1.2.9 Secondary Diode
The reverse voltage across secondary-rectifier diode D1 when the high-side buck switch is off can be calculated
using Equation 20.
VD1 =
N2
VIN
N1
(20)
For a VIN_MAX of 95 V and the 1:1 turns ratio of this design, a 100-V Schottky is selected.
8.2.1.2.10 VCC and Bootstrap Capacitor
A 1-µF capacitor of 16 V or higher rating is recommended for the VCC regulator bypass capacitor.
A good value for the BST pin bootstrap capacitor is 0.01-µF with a voltage rating of 16 or higher.
8.2.1.2.11 Input Capacitor
The input capacitor is typically a combination of a smaller bypass capacitor located near the regulator IC and a
larger bulk capacitor. The total input capacitance should be large enough to limit the input voltage ripple to a
desired amplitude. For input ripple voltage ΔVIN, CIN can be calculated using Equation 21.
CIN t
IOUT(MAX)
4 u ¦ u '9IN
(21)
Choosing a ΔVIN of 0.5 V gives a minimum CIN of 0.2 μF. A standard value of 0.47 μF is selected for CBYP in this
design. A bulk capacitor of higher value reduces voltage spikes due to parasitic inductance between the power
source to the converter. A standard value of 2.2 μF is selected for CIN in this design. The voltage ratings of the
two input capacitors should be greater than the maximum input voltage under all conditions.
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8.2.1.2.12 UVLO Resistors
UVLO resistors RUV1 and RUV2 set the undervoltage lockout threshold and hysteresis according to Equation 22
and Equation 23.
VIN (HYS) = IHYS x RUV2
(22)
VIN (UVLO, rising) = 1.225V x
R
( RUV2
+ 1)
(23)
UV1
Where IHYS = 20 μA, typical.
For a UVLO hysteresis of 2.5 V and UVLO rising threshold of 20 V, Equation 22 and Equation 23 require RUV1 of
8.25 kΩ and RUV2 of 127 kΩ and these values are selected for this design example.
8.2.1.2.13 VCC Diode
Diode D2 is an optional diode connected between VOUT1 and the VCC regulator output pin. When VOUT1 is more
than one diode drop greater than the VCC voltage, the VCC bias current is supplied from VOUT1. This results in
reduced power losses in the internal VCC regulator which improves converter efficiency. VOUT1 must be set to a
voltage at least one diode drop higher than 8.55 V (the maximum VCC voltage) if D2 is used to supply bias
current.
8.2.1.3 Application Curves
Figure 23. Efficiency at 750 kHz, VOUT1 = 10 V
Figure 24. Steady State Waveform (VIN = 48 V, IOUT1 = 100
mA, IOUT2 = 200 mA)
Figure 25. Step Load Response (VIN = 48 V, IOUT1 = 0, Step Load on IOUT2 = 100 mA to 200 mA)
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9 Power Supply Recommendations
LM34927 is a power-management device. The power supply for the device is any dc voltage source within the
specified input range.
10 Layout
10.1 Layout Guidelines
A proper layout is essential for optimum performance of the circuit. In particular, the following guidelines should
be observed:
1. CIN: The loop consisting of input capacitor ©IN), VIN pin, and RTN pin carries switching currents. Therefore,
place the input capacitor close to the IC, directly across VIN and RTN pins and the connections to these two
pins should be direct to minimize the loop area. In general it is not possible to accommodate all of input
capacitance near the IC. A good practice is to use a 0.1-µF or 0.47-μF capacitor directly across the VIN and
RTN pins close to the IC, and the remaining bulk capacitor as close as possible (see Placement of Bypass
Capacitors ).
2. CVCC and CBST: The VCC and bootstrap (BST) bypass capacitors supply switching currents to the high- and
low-side gate drivers. These two capacitors must also be placed as close to the IC as possible, and the
connecting trace lengths and loop area should be minimized (See Figure 26).
3. The Feedback trace carries the output voltage information and a small ripple component that is necessary for
proper operation of LM34927 device. Therefore take care while routing the feedback trace so avoid coupling
any noise to this pin. In particular, feedback trace should not run close to magnetic components, or parallel to
any other switching trace.
4. SW trace: SW node switches rapidly between VIN and GND every cycle and is therefore a possible source of
noise. SW node area should be minimized. In particular SW node should not be inadvertently connected to a
copper plane or pour.
10.2 Layout Example
RTN
1
VIN
2
8
SW
7
BST
CIN
SO
PowerPAD-8
UVLO
3
6
VCC
RON
4
5
FB
CVCC
Figure 26. Placement of Bypass Capacitors
10.3 Thermal Curves
0.7
Output Current (A)
0.6
0.5
VIN=54V
fsw=125kHz
VOUT=3.3V
L=220uH
0.4
0.3
0.2
500LFM
200LFM
100LFM
No Flow
0.1
0.0
0
20
40
60
80
100
120
Ambient Temperature (ƒC)
140
C003
Figure 27. Thermal Derating Curve
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11 Device and Documentation Support
11.1 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.2 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.3 Trademarks
PowerPAD, Fly-Buck, E2E are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.5 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
22
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PACKAGE OPTION ADDENDUM
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2-Jun-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
LM34927MR/NOPB
ACTIVE SO PowerPAD
DDA
8
95
RoHS & Green
NIPDAU | SN
Level-3-260C-168 HR
-40 to 125
L34927
Samples
LM34927MRX/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAU | SN
Level-3-260C-168 HR
-40 to 125
L34927
Samples
LM34927SD/NOPB
ACTIVE
WSON
NGU
8
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L34927
Samples
LM34927SDX/NOPB
ACTIVE
WSON
NGU
8
4500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
L34927
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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