LM2524D, LM3524D
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LM2524D/LM3524D Regulating Pulse Width Modulator
Check for Samples: LM2524D, LM3524D
FEATURES
1
•
2
•
•
•
•
•
•
•
•
Fully Interchangeable With Standard LM3524
Family
±1% Precision 5V Reference With Thermal
Shut-Down
Output Current to 200 mA DC
60V Output Capability
Wide Common Mode Input Range for ErrorAmp
One Pulse per Period (Noise Suppression)
Improved Max. Duty Cycle at High Frequencies
Double Pulse Suppression
Synchronize Through Pin 3
DESCRIPTION
The LM3524D family is an improved version of the
industry standard LM3524. It has improved
specifications and additional features yet is pin for pin
compatible with existing 3524 families. New features
reduce the need for additional external circuitry often
required in the original version.
The LM3524D has a ±1% precision 5V reference.
The current carrying capability of the output drive
transistors has been raised to 200 mA while reducing
VCEsat and increasing VCE breakdown to 60V. The
common mode voltage range of the error-amp has
been raised to 5.5V to eliminate the need for a
resistive divider from the 5V reference.
In the LM3524D the circuit bias line has been isolated
from the shut-down pin. This prevents the oscillator
pulse amplitude and frequency from being disturbed
by shut-down. Also at high frequencies (≃300 kHz)
the max. duty cycle per output has been improved to
44% compared to 35% max. duty cycle in other
3524s.
In addition, the LM3524D can now be synchronized
externally, through pin 3. Also a latch has been
added to insure one pulse per period even in noisy
environments. The LM3524D includes double pulse
suppression logic that insures when a shut-down
condition is removed the state of the T-flip-flop will
change only after the first clock pulse has arrived.
This feature prevents the same output from being
pulsed twice in a row, thus reducing the possibility of
core saturation in push-pull designs.
Connection Diagram
Figure 1. Top View
See Package Number NFG
See Package Number D
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM2524D, LM3524D
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Block Diagram
2
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1) (2)
Supply Voltage
40V
Collector Supply Voltage
LM2524D
55V
LM3524D
40V
Output Current DC (each)
200 mA
Oscillator Charging Current (Pin 7)
5 mA
Internal Power Dissipation
Operating Junction Temperature Range
1W
(3)
LM2524D
−40°C to +125°C
LM3524D
0°C to +125°C
Maximum Junction Temperature
150°
−65°C to +150°C
Storage Temperature Range
Lead Temperature (Soldering 4 sec.)
(1)
NFG, D Pkg.
260°C
Absolute maximum ratings indicate limits beyond which damage to the device may occur. DC and AC electrical specifications do not
apply when operating the device beyond its rated operating conditions.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
For operation at elevated temperatures, devices in the NFG package must be derated based on a thermal resistance of 86°C/W,
junction to ambient. Devices in the D package must be derated at 125°C/W, junction to ambient.
(2)
(3)
Electrical Characteristics (1)
Symbol
Parameter
Conditions
LM2524D
LM3524D
Typ
Tested
Limit (2)
Design
Limit (3)
Typ
Tested
Limit (2)
5
4.85
4.80
5
4.75
Units
Design
Limit (3)
REFERENCE SECTION
VREF
Output Voltage
5.15
5.20
VRLine
Line Regulation
VIN = 8V to 40V
10
15
30
10
25
50
mVMax
VRLoad
Load Regulation
IL = 0 mA to 20 mA
10
15
25
10
25
50
mVMax
ΔVIN/ΔVREF
Ripple Rejection
f = 120 Hz
66
IOS
Short Circuit Current
VREF = 0
5.25
10 Hz ≤ f ≤ 10 kHz
40
Long Term Stability
TA = 125°C
20
550
dB
25
mA Min
50
180
Output Noise
VMax
66
25
50
NO
VMin
200
100
40
mA Max
μVrms
100
Max
20
mV/kHr
OSCILLATOR SECTION
fOSC
Max. Freq.
RT = 1k, CT = 0.001 μF (4)
fOSC
Initial Accuracy
RT = 5.6k, CT = 0.01 μF (4)
500
RT = 2.7k, CT = 0.01 μF (4)
kHzMin
22.5
22.5
kHzMax
34
30
kHzMin
46
kHzMax
20
38
38
42
(2)
(3)
(4)
kHzMin
17.5
20
(1)
350
17.5
Unless otherwise stated, these specifications apply for TA = TJ = 25°C. Boldface numbers apply over the rated temperature range:
LM2524D is −40° to 85°C and LM3524D is 0°C to 70°C. VIN = 20V and fOSC = 20 kHz.
Tested limits are ensured and 100% tested in production.
Design limits are ensured (but not 100% production tested) over the indicated temperature and supply voltage range. These limits are
not used to calculate outgoing quality level.
The value of a Ct capacitor can vary with frequency. Careful selection of this capacitor must be made for high frequency operation.
Polystyrene was used in this test. NPO ceramic or polypropylene can also be used.
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Electrical Characteristics(1) (continued)
Symbol
Parameter
Conditions
ΔfOSC
Freq. Change with VIN
VIN = 8 to 40V
ΔfOSC
Freq. Change with Temp.
TA = −55°C to +125°C
at 20 kHz RT = 5.6k,
CT = 0.01 μF
VOSC
Output Amplitude (Pin 3)
RT = 5.6k, CT = 0.01 μF
tPW
LM2524D
Typ
Tested
Limit (2)
0.5
1
LM3524D
Design
Limit (3)
5
Tested
Limit (2)
0.5
1.0
Design
Limit (3)
2.4
Output Pulse Width (Pin 3) RT = 5.6k, CT = 0.01 μF
0.5
1.5
Sawtooth Peak Voltage
RT = 5.6k, CT = 0.01 μF
3.4
3.6
3.8
Sawtooth Valley Voltage
RT = 5.6k, CT = 0.01 μF
1.1
0.8
0.6
2
8
10
Units
%Max
5
3
(5)
Typ
%
3
2.4
VMin
0.5
1.5
μsMax
3.8
VMax
0.6
VMin
2
10
mVMax
ERROR-AMP SECTION
VIO
Input Offset Voltage
VCM = 2.5V
IIB
Input Bias Current
VCM = 2.5V
1
8
10
1
10
μAMax
IIO
Input Offset Current
VCM = 2.5V
0.5
1.0
1
0.5
1
μAMax
ICOSI
Compensation Current
(Sink)
VIN(I) − VIN(NI) = 150 mV
65
μAMin
125
125
μAMax
Compensation Current
(Source)
VIN(NI) − VIN(I) = 150 mV
−125
−125
μAMin
AVOL
Open Loop Gain
RL = ∞, VCM = 2.5 V
VCMR
Common Mode Input
Voltage Range
CMRR
Common Mode Rejection
Ratio
GBW
Unity Gain Bandwidth
AVOL = 0 dB, VCM = 2.5V
VO
Output Voltage Swing
RL = ∞
ICOSO
65
95
95
−95
−95
−65
PSRR
Power Supply Rejection
Ratio
80
90
−65
60
1.5
1.4
1.5
VMin
5.5
5.4
5.5
VMax
80
dBMin
80
80
90
3
70
μAMax
74
2
60
dBMin
MHz
0.5
0.5
VMin
5.5
5.5
VMax
80
65
dbMin
VIN = 8 to 40V
80
Minimum Duty Cycle
Pin 9 = 0.8V,
[RT = 5.6k, CT = 0.01 μF]
0
0
0
0
%Max
Maximum Duty Cycle
Pin 9 = 3.9V,
[RT = 5.6k, CT = 0.01 μF]
49
45
49
45
%Min
Maximum Duty Cycle
Pin 9 = 3.9V,
[RT = 1k, CT = 0.001 μF]
44
35
44
35
%Min
Input Threshold
Zero Duty Cycle
1
1
V
3.5
3.5
V
−1
−1
μA
70
COMPARATOR SECTION
tON/tOSC
tON/tOSC
tON/tOSC
VCOMPZ
(Pin 9)
VCOMPM
Input Threshold (Pin 9)
IIB
Input Bias Current
Maximum Duty Cycle
CURRENT LIMIT SECTION
VSEN
Sense Voltage
V(Pin 2) − V(Pin 1) ≥ 150 mV
180
200
Sense Voltage T.C.
Common Mode Voltage
Range
(5)
4
V5 − V4 = 300 mV
mVMin
220
mVMax
200
220
TC-Vsense
180
0.2
0.2
mV/°C
−0.7
−0.7
VMin
1
1
VMax
OSC amplitude is measured open circuit. Available current is limited to 1 mA so care must be exercised to limit capacitive loading of fast
pulses.
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Electrical Characteristics(1) (continued)
Symbol
Parameter
Conditions
LM2524D
Typ
Tested
Limit (2)
1
0.5
LM3524D
Design
Limit (3)
Design
Limit (3)
Units
Typ
Tested
Limit (2)
1
0.5
VMin
1.5
VMax
SHUT DOWN SECTION
VSD
High Input Voltage
V(Pin 2) − V(Pin 1) ≥ 150 mV
1.5
ISD
High Input Current
I(pin 10)
1
1
mA
OUTPUT SECTION (EACH OUTPUT)
VCES
Collector Emitter Voltage
Breakdown
IC ≤ 100 μA
ICES
Collector Leakage Current
VCE = 60V
VCE = 55V
55
0.1
Saturation Voltage
VMin
μAMax
50
VCE = 40V
VCESAT
40
0.1
50
IE = 20 mA
0.2
0.5
0.2
0.7
IE = 200 mA
1.5
2.2
1.5
2.5
17
18
17
VMax
VEO
Emitter Output Voltage
IE = 50 mA
18
tR
Rise Time
VIN = 20V,
IE = −250 μA
RC = 2k
200
200
VMin
ns
tF
Fall Time
RC = 2k
100
100
ns
SUPPLY CHARACTERISTICS SECTION
VIN
T
IIN
(6)
(7)
Input Voltage Range
Thermal Shutdown Temp.
Stand By Current
After Turn-on
(6)
8
8
VMin
40
40
VMax
160
VIN = 40V
(7)
5
160
10
5
°C
10
mA
For operation at elevated temperatures, devices in the NFG package must be derated based on a thermal resistance of 86°C/W,
junction to ambient. Devices in the D package must be derated at 125°C/W, junction to ambient.
Pins 1, 4, 7, 8, 11, and 14 are grounded; Pin 2 = 2V. All other inputs and outputs open.
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Typical Performance Characteristics
6
Switching Transistor Peak Output Current
vs Temperature
Maximum Average Power Dissipation (NFG, D Packages)
Figure 2.
Figure 3.
Maximum & Minimum
Duty Cycle Threshold Voltage
Output Transistor
Saturation Voltage
Figure 4.
Figure 5.
Output Transistor Emitter
Voltage
Reference Transistor
Peak Output Current
Figure 6.
Figure 7.
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Typical Performance Characteristics (continued)
Standby Current
vs Voltage
Standby Current
vs Temperature
Figure 8.
Figure 9.
Current Limit Sense Voltage
Figure 10.
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TEST CIRCUIT
8
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Functional Description
Internal Voltage Regulator
The LM3524D has an on-chip 5V, 50 mA, short circuit protected voltage regulator. This voltage regulator
provides a supply for all internal circuitry of the device and can be used as an external reference.
For input voltages of less than 8V the 5V output should be shorted to pin 15, VIN, which disables the 5V
regulator. With these pins shorted the input voltage must be limited to a maximum of 6V. If input voltages of
6V–8V are to be used, a pre-regulator, as shown in Figure 11, must be added.
*Minimum CO of 10 μF required for stability.
Figure 11.
Oscillator
The LM3524D provides a stable on-board oscillator. Its frequency is set by an external resistor, RT and capacitor,
CT. A graph of RT, CT vs oscillator frequency is shown is Figure 12. The oscillator's output provides the signals
for triggering an internal flip-flop, which directs the PWM information to the outputs, and a blanking pulse to turn
off both outputs during transitions to ensure that cross conduction does not occur. The width of the blanking
pulse, or dead time, is controlled by the value of CT, as shown in Figure 13. The recommended values of RT are
1.8 kΩ to 100 kΩ, and for CT, 0.001 μF to 0.1 μF.
If two or more LM3524D's must be synchronized together, the easiest method is to interconnect all pin 3
terminals, tie all pin 7's (together) to a single CT, and leave all pin 6's open except one which is connected to a
single RT. This method works well unless the LM3524D's are more than 6″ apart.
A second synchronization method is appropriate for any circuit layout. One LM3524D, designated as master,
must have its RTCT set for the correct period. The other slave LM3524D(s) should each have an RTCT set for a
10% longer period. All pin 3's must then be interconnected to allow the master to properly reset the slave units.
The oscillator may be synchronized to an external clock source by setting the internal free-running oscillator
frequency 10% slower than the external clock and driving pin 3 with a pulse train (approx. 3V) from the clock.
Pulse width should be greater than 50 ns to insure full synchronization.
Figure 12.
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Figure 13.
Error Amplifier
The error amplifier is a differential input, transconductance amplifier. Its gain, nominally 86 dB, is set by either
feedback or output loading. This output loading can be done with either purely resistive or a combination of
resistive and reactive components. A graph of the amplifier's gain vs output load resistance is shown in
Figure 14.
Figure 14.
The output of the amplifier, or input to the pulse width modulator, can be overridden easily as its output
impedance is very high (ZO ≃ 5 MΩ). For this reason a DC voltage can be applied to pin 9 which will override the
error amplifier and force a particular duty cycle to the outputs. An example of this could be a non-regulating
motor speed control where a variable voltage was applied to pin 9 to control motor speed. A graph of the output
duty cycle vs the voltage on pin 9 is shown in Figure 15.
The duty cycle is calculated as the percentage ratio of each output's ON-time to the oscillator period. Paralleling
the outputs doubles the observed duty cycle.
10
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Figure 15.
The amplifier's inputs have a common-mode input range of 1.5V–5.5V. The on board regulator is useful for
biasing the inputs to within this range.
Current Limiting
The function of the current limit amplifier is to override the error amplifier's output and take control of the pulse
width. The output duty cycle drops to about 25% when a current limit sense voltage of 200 mV is applied
between the +CL and −CLsense terminals. Increasing the sense voltage approximately 5% results in a 0% output
duty cycle. Care should be taken to ensure the −0.7V to +1.0V input common-mode range is not exceeded.
In most applications, the current limit sense voltage is produced by a current through a sense resistor. The
accuracy of this measurement is limited by the accuracy of the sense resistor, and by a small offset current,
typically 100 μA, flowing from +CL to −CL.
Output Stages
The outputs of the LM3524D are NPN transistors, capable of a maximum current of 200 mA. These transistors
are driven 180° out of phase and have non-committed open collectors and emitters as shown in Figure 16.
Figure 16.
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Typical Applications
Figure 17. Positive Regulator, Step-Up Basic Configuration (IIN(MAX) = 80 mA)
(1)
Figure 18. Positive Regulator, Step-Up Boosted Current Configuration
12
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Figure 19. Positive Regulator, Step-Down Basic Configuration (IIN(MAX) = 80 mA)
(2)
Figure 20. Positive Regulator, Step-Down Boosted Current Configuration
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Figure 21. Boosted Current Polarity Inverter
(3)
Basic Switching Regulator Theory and Applications
The basic circuit of a step-down switching regulator circuit is shown in Figure 22, along with a practical circuit
design using the LM3524D in Figure 25.
Figure 22. Basic Step-Down Switching Regulator
The circuit works as follows: Q1 is used as a switch, which has ON and OFF times controlled by the pulse width
modulator. When Q1 is ON, power is drawn from VIN and supplied to the load through L1; VA is at approximately
VIN, D1 is reverse biased, and Co is charging. When Q1 turns OFF the inductor L1 will force VA negative to keep
the current flowing in it, D1 will start conducting and the load current will flow through D1 and L1. The voltage at
VAis smoothed by the L1, Co filter giving a clean DC output. The current flowing through L1 is equal to the
nominal DC load current plus some ΔIL which is due to the changing voltage across it. A good rule of thumb is to
set ΔILP-P ≃ 40% × Io.
14
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Figure 23. Relation of Switch Timing to Inductor Current in Step-Down Regulator
(4)
Neglecting VSAT, VD, and settling ΔIL+ = ΔIL−;
(5)
where T = Total Period
The above shows the relation between VIN, Vo and duty cycle.
(6)
as Q1 only conducts during tON.
(7)
The efficiency, η, of the circuit is:
(8)
ηMAX will be further decreased due to switching losses in Q1. For this reason Q1 should be selected to have the
maximum possible fT, which implies very fast rise and fall times.
Calculating Inductor L1
(9)
Since ΔIL+ =
ΔIL−
= 0.4Io
Solving the above for L1
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(10)
where: L1 is in Henrys
f is switching frequency in Hz
Also, see LM1578 data sheet for graphical methods of inductor selection.
Calculating Output Filter Capacitor Co
Figure 23 shows L1's current with respect to Q1's tON and tOFF times (VA is at the collector of Q1). This curent
must flow to the load and Co. Co's current will then be the difference between IL, and Io.
Ico = IL − Io
(11)
From Figure 23 it can be seen that current will be flowing into Co for the second half of tON through the first half of
tOFF, or a time, tON/2 + tOFF/2. The current flowing for this time is ΔIL/4. The resulting ΔVc or ΔVo is described by:
(12)
For best regulation, the inductor's current cannot be allowed to fall to zero. Some minimum load current Io, and
thus inductor current, is required as shown below:
(13)
Figure 24. Inductor Current Slope in Step-Down Regulator
A complete step-down switching regulator schematic, using the LM3524D, is illustrated in Figure 25. Transistors
Q1 and Q2 have been added to boost the output to 1A. The 5V regulator of the LM3524D has been divided in
half to bias the error amplifier's non-inverting input to within its common-mode range. Since each output
transistor is on for half the period, actually 45%, they have been paralleled to allow longer possible duty cycle, up
to 90%. This makes a lower possible input voltage. The output voltage is set by:
(14)
where VNI is the voltage at the error amplifier's non-inverting input.
16
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Resistor R3 sets the current limit to:
(15)
Figure 26 and Figure 27 show a PC board layout and stuffing diagram for the 5V, 1A regulator of Figure 25. The
regulator's performance is listed in Table 1.
*Mounted to Staver Heatsink No. V5-1.
Q1 = BD344
Q2 = 2N5023
L1 = >40 turns No. 22 wire on Ferroxcube No. K300502 Torroid core.
Figure 25. 5V, 1 Amp Step-Down Switching Regulator
Table 1.
Parameter
Conditions
Typical Characteristics
Output Voltage
VIN = 10V, Io = 1A
5V
Switching Frequency
VIN = 10V, Io = 1A
20 kHz
Short Circuit Current Limit
VIN = 10V
1.3A
Load Regulation
VIN = 10V
Io = 0.2 − 1A
3 mV
Line Regulation
ΔVIN = 10 − 20V,
Io = 1A
6 mV
Efficiency
VIN = 10V, Io = 1A
80%
Output Ripple
VIN = 10V, Io = 1A
10 mVp-p
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Figure 26. 5V, 1 Amp Switching Regulator, Foil Side
Figure 27. Stuffing Diagram, Component Side
The Step-Up Switching Regulator
Figure 28 shows the basic circuit for a step-up switching regulator. In this circuit Q1 is used as a switch to
alternately apply VIN across inductor L1. During the time, tON, Q1 is ON and energy is drawn from VIN and stored
in L1; D1 is reverse biased and Io is supplied from the charge stored in Co. When Q1 opens, tOFF, voltage V1 will
rise positively to the point where D1 turns ON. The output current is now supplied through L1, D1 to the load and
any charge lost from Co during tON is replenished. Here also, as in the step-down regulator, the current through
L1 has a DC component plus some ΔIL. ΔIL is again selected to be approximately 40% of IL. Figure 29 shows the
inductor's current in relation to Q1's ON and OFF times.
18
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Figure 28. Basic Step-Up Switching Regulator
Figure 29. Relation of Switch Timing to Inductor Current in Step-Up Regulator
(16)
Since ΔIL+ = ΔIL−, VINtON = VotOFF − VINtOFF,
and neglecting VSAT and VD1
(17)
The above equation shows the relationship between VIN, Vo and duty cycle.
In calculating input current IIN(DC), which equals the inductor's DC current, assume first 100% efficiency:
(18)
for η = 100%, POUT = PIN
(19)
This equation shows that the input, or inductor, current is larger than the output current by the factor (1 +
tON/tOFF). Since this factor is the same as the relation between Vo and VIN, IIN(DC) can also be expressed as:
(20)
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So far it is assumed η = 100%, where the actual efficiency or ηMAX will be somewhat less due to the saturation
voltage of Q1 and forward on voltage of D1. The internal power loss due to these voltages is the average IL
current flowing, or IIN, through either VSAT or VD1. For VSAT = VD1 = 1V this power loss becomes IIN(DC) (1V). ηMAX
is then:
(21)
(22)
This equation assumes only DC losses, however ηMAX is further decreased because of the switching time of Q1
and D1.
In calculating the output capacitor Co it can be seen that Co supplies Io during tON. The voltage change on Co
during this time will be some ΔVc = ΔVo or the output ripple of the regulator. Calculation of Co is:
(23)
where: Co is in farads, f is the switching frequency,
ΔVo is the p-p output ripple
Calculation of inductor L1 is as follows:
(24)
VIN is applied across L1
(25)
where: L1 is in henrys, f is the switching frequency in Hz
To apply the above theory, a complete step-up switching regulator is shown in Figure 30. Since VIN is 5V, VREF is
tied to VIN. The input voltage is divided by 2 to bias the error amplifier's inverting input. The output voltage is:
(26)
The network D1, C1 forms a slow start circuit.
20
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LM2524D, LM3524D
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SNVS766E – JUNE 2009 – REVISED MAY 2013
This holds the output of the error amplifier initially low thus reducing the duty-cycle to a minimum. Without the
slow start circuit the inductor may saturate at turn-on because it has to supply high peak currents to charge the
output capacitor from 0V. It should also be noted that this circuit has no supply rejection. By adding a reference
voltage at the non-inverting input to the error amplifier, see Figure 31, the input voltage variations are rejected.
The LM3524D can also be used in inductorless switching regulators. Figure 32 shows a polarity inverter which if
connected to Figure 30 provides a −15V unregulated output.
L1 = > 25 turns No. 24 wire on Ferroxcube No. K300502 Toroid core.
Figure 30. 15V, 0.5A Step-Up Switching Regulator
Figure 31. Replacing R3/R4 Divider in Figure 30 with Reference Circuit Improves Line Regulation
Figure 32. Polarity Inverter Provides Auxiliary −15V Unregulated Output from Circuit of Figure 30
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Product Folder Links: LM2524D LM3524D
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21
LM2524D, LM3524D
SNVS766E – JUNE 2009 – REVISED MAY 2013
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REVISION HISTORY
Changes from Revision D (May 2013) to Revision E
•
22
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 21
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PACKAGE OPTION ADDENDUM
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7-Jun-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
LM3524DM/NOPB
ACTIVE
SOIC
D
16
48
RoHS & Green
SN
Level-1-260C-UNLIM
0 to 125
LM3524DM
Samples
LM3524DMX/NOPB
ACTIVE
SOIC
D
16
2500
RoHS & Green
SN
Level-1-260C-UNLIM
0 to 125
LM3524DM
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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