LM4766
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LM4766 Overture™ Audio Power Amplifier Series
Dual 40W Audio Power Amplifier with Mute
Check for Samples: LM4766
FEATURES
DESCRIPTION
•
•
The LM4766 is a stereo audio amplifier capable of
delivering typically 40W per channel with the nonisolated "NDL" package and 30W per channel with
the isolated "NDB" package of continuous average
output power into an 8Ω load with less than 0.1%
(THD+N).
1
23
•
•
•
SPiKe Protection
Minimal Amount of External Components
Necessary
Quiet Fade-In/Out Mute Mode
Non-Isolated 15-Lead TO-220 Package
Wide Supply Range 20V - 78V
APPLICATIONS
•
•
•
High-End Stereo TVs
Component Stereo
Compact Stereo
KEY SPECIFICATIONS
•
•
THD+N at 1kHz at 2 x 30W Continuous
Average Output Power Into 8Ω 0.1% (Max)
THD+N at 1kHz at Continuous Average Output
Power of 2 x 30W Into 8Ω 0.009% (Typ)
The performance of the LM4766, utilizing its Self
Peak Instantaneous Temperature (°Ke) (SPiKe)
Protection Circuitry, places it in a class above
discrete and hybrid amplifiers by providing an
inherently, dynamically protected Safe Operating
Area (SOA). SPiKe Protection means that these parts
are safeguarded at the output against overvoltage,
undervoltage, overloads, including thermal runaway
and instantaneous temperature peaks.
Each amplifier within the LM4766 has an independent
smooth transition fade-in/out mute that minimizes
output pops. The IC's extremely low noise floor at
2µV and its extremely low THD+N value of 0.06% at
the rated power make the LM4766 optimum for highend stereo TVs or minicomponent systems.
Connection Diagram
Figure 1. Plastic Package
Top View
Non-Isolated TO-220 Package
See Package Number NDL0015A
Isolated PFM Package
See Package Number NDB0015B
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Overture is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM4766
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Typical Application
Numbers in parentheses represent pinout for amplifier B.
*Optional component dependent upon specific design requirements.
Figure 2. Typical Audio Amplifier Application Circuit
2
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS
(1) (2) (3)
(No Input)
Supply Voltage |VCC| + |VEE|
78V
(with Input)
74V
Common Mode Input Voltage
(VCC or VEE) and
|VCC| + |VEE| ≤ 60V
Differential Input Voltage
60V
Output Current
Internally Limited
Power Dissipation
(4)
ESD Susceptibility
(5)
Junction Temperature
62.5W
3000V
(6)
150°C
Thermal Resistance
Non-Isolated NDL-Package
θJC
1°C/W
Isolated NDB-Package
θJC
2°C/W
Soldering Information
NDL and NDB Packages
(1)
(2)
(3)
(4)
(5)
(6)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication
of device performance.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
All voltages are measured with respect to the GND pins (5, 10), unless otherwise specified.
For operating at case temperatures above 25°C, the device must be derated based on a 150°C maximum junction temperature and a
thermal resistance of θJC = 1°C/W (junction to case) for the NDL package. Refer to the section DETERMINING THE CORRECT HEAT
SINK in the APPLICATION INFORMATION section.
Human body model, 100pF discharged through a 1.5kΩ resistor.
The operating junction temperature maximum is 150°C, however, the instantaneous Safe Operating Area temperature is 250°C.
OPERATING RATINGS
(1) (2)
TMIN ≤ TA ≤ TMAX
Temperature Range
Supply Voltage |VCC| + |VEE|
(1)
(2)
(3)
260°C
−40°C to +150°C
Storage Temperature
(3)
−20°C ≤ TA ≤ +85°C
20V to 60V
All voltages are measured with respect to the GND pins (5, 10), unless otherwise specified.
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication
of device performance.
Operation is ensured up to 60V, however, distortion may be introduced from SPiKe Protection Circuitry if proper thermal considerations
are not taken into account. Refer to the APPLICATION INFORMATION section for a complete explanation.
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ELECTRICAL CHARACTERISTICS
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(1) (2)
The following specifications apply for VCC = +30V, VEE = −30V, IMUTE = −0.5mA with RL = 8Ω unless otherwise specified.
Limits apply for TA = 25°C.
Symbol
Parameter
Conditions
LM4766
Typical (3)
|VCC| + |VEE|
PO
(6) (7)
Power Supply Voltage
(5)
Output Power (Continuous
Average)
Limit
(4)
Units
(Limits)
20
V (min)
60
V (max)
40
30
W/ch (min)
30
25
W/ch (min)
GND − VEE ≥ 9V
18
NDL Package, VCC = ±30V,THD+N = 0.1%
(max),
f = 1kHz, f = 20kHz
NDB Package, VCC = ±26V (7),
THD+N = 0.1% (max), f = 1kHz, f = 20kHz
NDL Package, 30W/ch, RL = 8Ω,
20Hz ≤ f ≤ 20kHz, AV = 26dB
0.06
%
NDB Package, 25W/ch, RL = 8Ω,
20Hz ≤ f ≤ 20kHz, AV = 26dB
0.06
%
THD+N
Total Harmonic Distortion
Plus Noise
Xtalk
Channel Separation
f = 1kHz, VO = 10.9Vrms
60
SR (6)
Slew Rate
VIN = 1.2Vrms, trise = 2ns
9
5
V/μs (min)
Total Quiescent Power Supply
Current
Both Amplifiers VCM = 0V, VO = 0V, IO = 0mA
48
100
mA (max)
VOS (8)
Input Offset Voltage
VCM = 0V, IO = 0mA
1
10
mV (max)
IB
Input Bias Current
VCM = 0V, IO = 0mA
0.2
1
μA (max)
IOS
Input Offset Current
VCM = 0V, IO = 0mA
0.01
0.2
μA (max)
IO
Output Current Limit
|VCC| = |VEE| = 10V, tON = 10ms, VO = 0V
(8)
Itotal
(8)
VOD
Output Dropout Voltage
PSRR
(8)
CMRR
(8)
AVOL
(8)
GBWP
eIN
Power Supply Rejection Ratio
4
V (max)
|VO–VEE|, VEE = −20V, IO = −100mA
2.5
4
V (max)
VCC = 30V to 10V, VEE = −30V, VCM = 0V, IO
= 0mA
125
85
dB (min)
VCC = 30V, VEE = −30V to −10V VCM = 0V, IO
= 0mA
110
85
dB (min)
75
dB (min)
Open Loop Voltage Gain
RL = 2kΩ, ΔVO = 40V
115
80
dB (min)
Gain Bandwidth Product
fO = 100kHz, VIN = 50mVrms
8
2
MHz (min)
Input Noise
IHF–A Weighting Filter, RIN = 600Ω (Input
Referred)
2.0
8
μV (max)
(6)
Mute Attenuation
(8)
(9)
Apk (min)
4
110
AM
(6)
(7)
3
1.5
VCC = 50V to 10V, VEE = −10V to −50V, VCM
= 20V to −20V, IO = 0mA
Signal-to-Noise Ratio
(3)
(4)
(5)
4
|VCC–VO|, VCC = 20V, IO = +100mA
Common Mode Rejection Ratio
SNR
(1)
(2)
(9)
dB
PO = 1W, A–Weighted, Measured at 1kHz,
RS = 25Ω
98
dB
PO = 25W, A–Weighted Measured at 1kHz,
RS = 25Ω
112
dB
Pin 6,11 at 2.5V
115
80
dB (min)
All voltages are measured with respect to the GND pins (5, 10), unless otherwise specified.
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication
of device performance.
Typicals are measured at 25°C and represent the parametric norm.
Limits are specifications that all parts are tested in production to meet the stated values.
VEE must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled. In
addition, the voltage differential between VCC and VEE must be greater than 14V.
AC Electrical Test; Refer to Test Circuit #2 .
When using the isolated package (NDB), the θJC is 2°C/W verses 1°C/W for the non-isolated package (NDL). This increased thermal
resistance from junction to case requires a lower supply voltage for decreased power dissipation within the package. Voltages higher
than ±26V maybe used but will require a heat sink with less than 1°C/W thermal resistance to avoid activating thermal shutdown during
normal operation.
DC Electrical Test; Refer to Test Circuit #1 .
The output dropout voltage, VOD, is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltage
graph in the TYPICAL PERFORMANCE CHARACTERISTICS section.
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Test Circuit #1
(DC Electrical Test Circuit)
Figure 3.
Test Circuit #2
(AC Electrical Test Circuit)
Figure 4.
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BRIDGED AMPLIFIER APPLICATION CIRCUIT
Figure 5. Bridged Amplifier Application Circuit
Single Supply Application Circuit
*Optional components dependent upon specific design requirements.
Figure 6. Single Supply Amplifier Application Circuit
6
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Auxiliary Amplifier Application Circuit
Figure 7. Special Audio Amplifier Application Circuit
Equivalent Schematic
(excluding active protection circuitry)
Figure 8. LM4766 (One Channel Only)
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External Components Description
Components
RB
Prevents currents from entering the amplifier's non-inverting input which may be passed through to the load upon
power down of the system due to the low input impedance of the circuitry when the undervoltage circuitry is off.
This phenomenon occurs when the supply voltages are below 1.5V.
2
Ri
Inverting input resistance to provide AC gain in conjunction with Rf.
3
Rf
Feedback resistance to provide AC gain in conjunction with Ri.
4
Ci (1)
Feedback capacitor which ensures unity gain at DC. Also creates a highpass filter with Ri at fC = 1/(2πRiCi).
5
CS
Provides power supply filtering and bypassing. Refer to the SUPPLY BYPASSING section for proper placement
and selection of bypass capacitors.
6
RV (1)
(1)
Acts as a volume control by setting the input voltage level.
7
RIN
8
CIN (1)
Input capacitor which blocks the input signal's DC offsets from being passed onto the amplifier's inputs.
9
RSN (1)
Works with CSN to stabilize the output stage by creating a pole that reduces high frequency instabilities.
(1)
10
CSN
11
L
(1)
12
R
(1)
13
RA
14
CA
Sets the amplifier's input terminals DC bias point when CIN is present in the circuit. Also works with CIN to create a
highpass filter at fC = 1/(2πRINCIN). Refer to Figure 7.
Works with RSN to stabilize the output stage by creating a pole that reduces high frequency instabilities. The pole is
set at fC = 1/(2πRSNCSN). Refer to Figure 7.
Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce the Q of
the series resonant circuit. Also provides a low impedance at low frequencies to short out R and pass audio signals
to the load. Refer to Figure 7.
Provides DC voltage biasing for the transistor Q1 in single supply operation.
Provides bias filtering for single supply operation.
(1)
15
RINP
16
RBI
Provides input bias current for single supply operation. Refer to the CLICKS AND POPS application section for a
more detailed explanation of the function of RBI.
17
RE
Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the halfsupply point along with CA.
18
RM
Mute resistance set up to allow 0.5mA to be drawn from pin 6 or 11 to turn the muting function off.
→ RM is calculated using: RM ≤ (|VEE| − 2.6V)/l where l ≥ 0.5mA. Refer to the Mute Attenuation vs Mute Current
curves in the TYPICAL PERFORMANCE CHARACTERISTICS section.
19
CM
Mute capacitance set up to create a large time constant for turn-on and turn-off muting.
20
S1
Mute switch that mutes the music going into the amplifier when opened.
(1)
8
Functional Description
1
Limits the voltage difference between the amplifier's inputs for single supply operation. Refer to the CLICKS AND
POPS application section for a more detailed explanation of the function of RINP.
Optional components dependent upon specific design requirements.
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TYPICAL PERFORMANCE CHARACTERISTICS
THD+N vs Frequency
THD+N vs Frequency
Figure 9.
Figure 10.
THD+N vs Output Power
THD+N vs Output Power
Figure 11.
Figure 12.
THD+N vs Distribution
THD+N vs Distribution
Figure 13.
Figure 14.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
10
Channel Separation vs Frequency
Clipping Voltage vs Supply Voltage
Figure 15.
Figure 16.
Output Power vs Load Resistance
Output Power vs Supply Voltage
Figure 17.
Figure 18.
Power Dissipation vs Output Power
Power Dissipation vs Output Power
Figure 19.
Figure 20.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Max Heatsink Thermal Resistance (°C/W)
at the Specified Ambient Temperature (°C)
Note: The maximum heatsink thermal resistance values, θSA, in the table above were calculated using a θCS = 0.2°C/W due to thermal
compound.
Figure 21.
Safe Area
SPiKe Protection Response
Figure 22.
Figure 23.
Pulse Power Limit
Pulse Power Limit
Figure 24.
Figure 25.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
12
Pulse Response
Large Signal Response
Figure 26.
Figure 27.
Power Supply Rejection Ratio
Common-Mode Rejection Ratio
Figure 28.
Figure 29.
Open Loop Frequency Response
Supply Current vs Case Temperature
Figure 30.
Figure 31.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Input Bias Current vs Case Temperature
Mute Attenuation vs Mute Current (per Amplifier)
Figure 32.
Figure 33.
Mute Attenuation vs Mute Current (per Amplifier)
Output Power/Channel vs Supply Voltage
f = 1kHz, RL = 4Ω, 80kHz BW
Figure 34.
Figure 35.
Output Power/Channel vs Supply Voltage
f = 1kHz, RL = 6Ω, 80kHz BW
Output Power/Channel vs Supply Voltage
f = 1kHz, RL = 8Ω, 80kHz BW
Figure 36.
Figure 37.
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APPLICATION INFORMATION
MUTE MODE
The muting function of the LM4766 allows the user to mute the music going into the amplifier by drawing more
than 0.5mA out of each mute pin on the device. This is accomplished as shown in the Typical Application Circuit
where the resistor RM is chosen with reference to your negative supply voltage and is used in conjunction with a
switch. The switch when opened cuts off the current flow from pin 6 or 11 to −VEE, thus placing the LM4766 into
mute mode. Refer to the Mute Attenuation vs Mute Current curves in the TYPICAL PERFORMANCE
CHARACTERISTICS section for values of attenuation per current out of pins 6 or 11. The resistance RM is
calculated by the following equation:
RM ≤ (|−VEE| − 2.6V)/Ipin6
where
•
Ipin6 = Ipin11 ≥ 0.5mA.
(1)
Both pins 6 and 11 can be tied together so that only one resistor and capacitor are required for the mute
function. The mute resistance must be chosen such that greater than 1mA is pulled through the resistor RM so
that each amplifier is fully pulled out of mute mode. Taking into account supply line fluctuations, it is a good idea
to pull out 1mA per mute pin or 2 mA total if both pins are tied together.
UNDER-VOLTAGE PROTECTION
Upon system power-up, the under-voltage protection circuitry allows the power supplies and their corresponding
capacitors to come up close to their full values before turning on the LM4766 such that no DC output spikes
occur. Upon turn-off, the output of the LM4766 is brought to ground before the power supplies such that no
transients occur at power-down.
OVER-VOLTAGE PROTECTION
The LM4766 contains over-voltage protection circuitry that limits the output current to approximately 4.0APK while
also providing voltage clamping, though not through internal clamping diodes. The clamping effect is quite the
same, however, the output transistors are designed to work alternately by sinking large current spikes.
SPiKe PROTECTION
The LM4766 is protected from instantaneous peak-temperature stressing of the power transistor array. The Safe
Operating graph in the TYPICAL PERFORMANCE CHARACTERISTICS section shows the area of device
operation where SPiKe Protection Circuitry is not enabled. The waveform to the right of the SOA graph
exemplifies how the dynamic protection will cause waveform distortion when enabled. Please refer to AN-898 for
more detailed information.
THERMAL PROTECTION
The LM4766 has a sophisticated thermal protection scheme to prevent long-term thermal stress of the device.
When the temperature on the die reaches 165°C, the LM4766 shuts down. It starts operating again when the die
temperature drops to about 155°C, but if the temperature again begins to rise, shutdown will occur again at
165°C. Therefore, the device is allowed to heat up to a relatively high temperature if the fault condition is
temporary, but a sustained fault will cause the device to cycle in a Schmitt Trigger fashion between the thermal
shutdown temperature limits of 165°C and 155°C. This greatly reduces the stress imposed on the IC by thermal
cycling, which in turn improves its reliability under sustained fault conditions.
Since the die temperature is directly dependent upon the heat sink used, the heat sink should be chosen such
that thermal shutdown will not be reached during normal operation. Using the best heat sink possible within the
cost and space constraints of the system will improve the long-term reliability of any power semiconductor
device, as discussed in the DETERMINING THE CORRECT HEAT SINK section.
DETERMlNlNG MAXIMUM POWER DISSIPATION
Power dissipation within the integrated circuit package is a very important parameter requiring a thorough
understanding if optimum power output is to be obtained. An incorrect maximum power dissipation calculation
may result in inadequate heat sinking causing thermal shutdown and thus limiting the output power.
14
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Equation 2 exemplifies the theoretical maximum power dissipation point of each amplifier where VCC is the total
supply voltage.
PDMAX = VCC2/2π2RL
(2)
Thus by knowing the total supply voltage and rated output load, the maximum power dissipation point can be
calculated. The package dissipation is twice the number which results from Equation 2 since there are two
amplifiers in each LM4766. Refer to the graphs of Power Dissipation versus Output Power in the TYPICAL
PERFORMANCE CHARACTERISTICS section which show the actual full range of power dissipation not just the
maximum theoretical point that results from Equation 2.
DETERMINING THE CORRECT HEAT SINK
The choice of a heat sink for a high-power audio amplifier is made entirely to keep the die temperature at a level
such that the thermal protection circuitry does not operate under normal circumstances.
The thermal resistance from the die (junction) to the outside air (ambient) is a combination of three thermal
resistances, θJC, θCS, and θSA. In addition, the thermal resistance, θJC (junction to case), of the LM4766T is
1°C/W. Using Thermalloy Thermacote thermal compound, the thermal resistance, θCS (case to sink), is about
0.2°C/W. Since convection heat flow (power dissipation) is analogous to current flow, thermal resistance is
analogous to electrical resistance, and temperature drops are analogous to voltage drops, the power dissipation
out of the LM4766 is equal to the following:
PDMAX = (TJMAX−TAMB)/θJA
where
•
•
TJMAX = 150°C, TAMB is the system ambient temperature
θJA = θJC + θCS + θSA
(3)
Once the maximum package power dissipation has been calculated using Equation 2, the maximum thermal
resistance, θSA, (heat sink to ambient) in °C/W for a heat sink can be calculated. This calculation is made using
Equation 4 which is derived by solving for θSA in Equation 3.
θSA = [(TJMAX−TAMB)−PDMAX(θJC +θCS)]/PDMAX
(4)
Again it must be noted that the value of θSA is dependent upon the system designer's amplifier requirements. If
the ambient temperature that the audio amplifier is to be working under is higher than 25°C, then the thermal
resistance for the heat sink, given all other things are equal, will need to be smaller.
SUPPLY BYPASSING
The LM4766 has excellent power supply rejection and does not require a regulated supply. However, to improve
system performance as well as eliminate possible oscillations, the LM4766 should have its supply leads
bypassed with low-inductance capacitors having short leads that are located close to the package terminals.
Inadequate power supply bypassing will manifest itself by a low frequency oscillation known as “motorboating” or
by high frequency instabilities. These instabilities can be eliminated through multiple bypassing utilizing a large
tantalum or electrolytic capacitor (10μF or larger) which is used to absorb low frequency variations and a small
ceramic capacitor (0.1μF) to prevent any high frequency feedback through the power supply lines.
If adequate bypassing is not provided, the current in the supply leads which is a rectified component of the load
current may be fed back into internal circuitry. This signal causes distortion at high frequencies requiring that the
supplies be bypassed at the package terminals with an electrolytic capacitor of 470μF or more.
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BRIDGED AMPLIFIER APPLICATION
The LM4766 has two operational amplifiers internally, allowing for a few different amplifier configurations. One of
these configurations is referred to as “bridged mode” and involves driving the load differentially through the
LM4766's outputs. This configuration is shown in Figure 5. Bridged mode operation is different from the classical
single-ended amplifier configuration where one side of its load is connected to ground.
A bridge amplifier design has a distinct advantage over the single-ended configuration, as it provides differential
drive to the load, thus doubling output swing for a specified supply voltage. Consequently, theoretically four times
the output power is possible as compared to a single-ended amplifier under the same conditions. This increase in
attainable output power assumes that the amplifier is not current limited or clipped.
A direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal
power dissipation. For each operational amplifier in a bridge configuration, the internal power dissipation will
increase by a factor of two over the single ended dissipation. Thus, for an audio power amplifier such as the
LM4766, which has two operational amplifiers in one package, the package dissipation will increase by a factor
of four. To calculate the LM4766's maximum power dissipation point for a bridged load, multiply Equation 2 by a
factor of four.
This value of PDMAX can be used to calculate the correct size heat sink for a bridged amplifier application. Since
the internal dissipation for a given power supply and load is increased by using bridged-mode, the heatsink's θSA
will have to decrease accordingly as shown by Equation 4. Refer to the section, DETERMINING THE CORRECT
HEAT SINK for a more detailed discussion of proper heat sinking for a given application.
SINGLE-SUPPLY AMPLIFIER APPLICATION
The typical application of the LM4766 is a split supply amplifier. But as shown in Figure 6, the LM4766 can also
be used in a single power supply configuration. This involves using some external components to create a halfsupply bias which is used as the reference for the inputs and outputs. Thus, the signal will swing around halfsupply much like it swings around ground in a split-supply application. Along with proper circuit biasing, a few
other considerations must be accounted for to take advantage of all of the LM4766 functions, like the mute
function.
CLICKS AND POPS
In the typical application of the LM4766 as a split-supply audio power amplifier, the IC exhibits excellent “click”
and “pop” performance when utilizing the mute mode. In addition, the device employs Under-Voltage Protection,
which eliminates unwanted power-up and power-down transients. The basis for these functions are a stable and
constant half-supply potential. In a split-supply application, ground is the stable half-supply potential. But in a
single-supply application, the half-supply needs to charge up just like the supply rail, VCC. This makes the task of
attaining a clickless and popless turn-on more challenging. Any uneven charging of the amplifier inputs will result
in output clicks and pops due to the differential input topology of the LM4766.
To achieve a transient free power-up and power-down, the voltage seen at the input terminals should be ideally
the same. Such a signal will be common-mode in nature, and will be rejected by the LM4766. In Figure 6, the
resistor RINP serves to keep the inputs at the same potential by limiting the voltage difference possible between
the two nodes. This should significantly reduce any type of turn-on pop, due to an uneven charging of the
amplifier inputs. This charging is based on a specific application loading and thus, the system designer may need
to adjust these values for optimal performance.
As shown in Figure 6, the resistors labeled RBI help bias up the LM4766 off the half-supply node at the emitter of
the 2N3904. But due to the input and output coupling capacitors in the circuit, along with the negative feedback,
there are two different values of RBI, namely 10kΩ and 200kΩ. These resistors bring up the inputs at the same
rate resulting in a popless turn-on. Adjusting these resistors values slightly may reduce pops resulting from
power supplies that ramp extremely quick or exhibit overshoot during system turn-on.
16
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Product Folder Links: LM4766
LM4766
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SNAS031F – SEPTEMBER 1998 – REVISED MARCH 2013
AUDIO POWER AMPLlFIER DESIGN
Design a 30W/8Ω Audio Amplifier
Given:
Power Output
30Wrms
Load Impedance
8Ω
Input Level
1Vrms(max)
Input Impedance
47kΩ
Bandwidth
20Hz−20kHz ±0.25dB
A designer must first determine the power supply requirements in terms of both voltage and current needed to
obtain the specified output power. VOPEAK can be determined from Equation 5 and IOPEAK from Equation 6.
(5)
(6)
To determine the maximum supply voltage the following conditions must be considered. Add the dropout voltage
to the peak output swing VOPEAK, to get the supply rail at a current of IOPEAK. The regulation of the supply
determines the unloaded voltage which is usually about 15% higher. The supply voltage will also rise 10% during
high line conditions. Therefore the maximum supply voltage is obtained from the following equation.
Max supplies ≈ ± (VOPEAK + VOD) (1 + regulation) (1.1)
(7)
For 30W of output power into an 8Ω load, the required VOPEAK is 21.91V. A minimum supply rail of 25.4V results
from adding VOPEAK and VOD. With regulation, the maximum supplies are ±32V and the required IOPEAK is 2.74A
from Equation 6. It should be noted that for a dual 30W amplifier into an 8Ω load the IOPEAK drawn from the
supplies is twice 2.74APK or 5.48APK. At this point it is a good idea to check the Power Output vs Supply Voltage
to ensure that the required output power is obtainable from the device while maintaining low THD+N. In addition,
the designer should verify that with the required power supply voltage and load impedance, that the required
heatsink value θSA is feasible given system cost and size constraints. Once the heatsink issues have been
addressed, the required gain can be determined from Equation 8.
(8)
From Equation 8, the minimum AV is:
AV ≥ 15.5.
By selecting a gain of 21, and with a feedback resistor, Rf = 20kΩ, the value of Ri follows from Equation 9.
Ri = Rf (AV − 1)
(9)
Thus with Ri = 1kΩ a non-inverting gain of 21 will result. Since the desired input impedance was 47kΩ, a value of
47kΩ was selected for RIN. The final design step is to address the bandwidth requirements which must be stated
as a pair of −3dB frequency points. Five times away from a −3dB point is 0.17dB down from passband response
which is better than the required ±0.25dB specified. This fact results in a low and high frequency pole of 4Hz and
100kHz respectively. As stated in the External Components Description section, Ri in conjunction with Ci create a
high-pass filter.
Ci ≥ 1/(2π * 1kΩ * 4Hz) = 39.8μF;
use 39μF.
(10)
The high frequency pole is determined by the product of the desired high frequency pole, fH, and the gain, AV.
With a AV = 21 and fH = 100kHz, the resulting GBWP is 2.1MHz, which is less than the ensured minimum GBWP
of the LM4766 of 8MHz. This will ensure that the high frequency response of the amplifier will be no worse than
0.17dB down at 20kHz which is well within the bandwidth requirements of the design.
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Product Folder Links: LM4766
17
LM4766
SNAS031F – SEPTEMBER 1998 – REVISED MARCH 2013
www.ti.com
REVISION HISTORY
Changes from Revision E (March 2013) to Revision #IMPLIED
18
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Copyright © 1998–2013, Texas Instruments Incorporated
Product Folder Links: LM4766
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM4766T/NOPB
ACTIVE
TO-220
NDL
15
20
RoHS & Green
SN
Level-1-NA-UNLIM
0 to 70
LM4766T
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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