LM5000
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SNVS176D – MAY 2004 – REVISED MARCH 2007
LM5000 High Voltage Switch Mode Regulator
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FEATURES
DESCRIPTION
•
•
•
The LM5000 is a monolithic integrated circuit
specifically designed and optimized for flyback, boost
or forward power converter applications. The internal
power switch is rated for a maximum of 80V, with a
current limit set to 2A. Protecting the power switch
are current limit and thermal shutdown circuits. The
current mode control scheme provides excellent
rejection of line transients and cycle-by-cycle current
limiting. An external compensation pin and the built-in
slope compensation allow the user to optimize the
frequency compensation. Other distinctive features
include softstart to reduce stresses during start-up
and an external shutdown pin for remote ON/OFF
control. There are two operating frequency ranges
available. The LM5000-3 is pin selectable for either
300kHz (FS Grounded) or 700kHz (FS Open). The
LM5000-6 is pin selectable for either 600kHz (FS
Grounded) or 1.3MHz (FS Open). The device is
available in a low profile 16-lead TSSOP package or
a thermally enhanced 16-lead WSON package.
1
2
•
•
•
•
•
•
•
•
80V Internal Switch
Operating Input Voltage Range of 3.1V to 40V
Pin Selectable Operating Frequency
– 300kHz/700kHz (-3)
– 600kHz/1.3MHz (-6)
Adjustable Output Voltage
External Compensation
Input Undervoltage Lockout
Softstart
Current Limit
Over Temperature Protection
External Shutdown
Small 16-Lead TSSOP or 16-Lead WSON
Package
APPLICATIONS
•
•
•
•
•
Flyback Regulator
Forward Regulator
Boost Regulator
DSL Modems
Distributed Power Converters
Typical Application Circuit
Figure 1. LM5000 Flyback Converter
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM5000
SNVS176D – MAY 2004 – REVISED MARCH 2007
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Connection Diagram
Figure 2. Top View
PIN DESCRIPTIONS
2
Pin
Name
1
COMP
Function
Compensation network connection. Connected to the output of the voltage error amplifier. The RC
compenstion network should be connected from this pin to AGND. An additional 100pF high frequency
capacitor to AGND is recommended.
2
FB
3
SHDN
Output voltage feedback input.
Shutdown control input, Open = enable, Ground = disable.
4
AGND
Analog ground, connect directly to PGND.
5
PGND
Power ground.
6
PGND
Power ground.
7
PGND
Power ground.
8
PGND
Power ground.
9
SW
Power switch input. Switch connected between SW pins and PGND pins
10
SW
Power switch input. Switch connected between SW pins and PGND pins
11
SW
Power switch input. Switch connected between SW pins and PGND pins
12
BYP
Bypass-Decouple Capacitor Connection, 0.1µF ceramic capacitor recommended.
13
VIN
Analog power input. A small RC filter is recommended, to suppress line glitches. Typical values of 10Ω
and ≥ 0.1µF are recommended.
14
SS
Softstart Input. External capacitor and internal current source sets the softstart time.
15
FS
Switching frequency select input. Open = Fhigh. Ground = Flow
16
TEST
-
Exposed Pad
underside of WSON
package
Factory test pin, connect to ground.
Connect to system ground plane for reduced thermal resistance.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1) (2)
VIN
-0.3V to 40V
SW Voltage
-0.3V to 80V
FB Voltage
-0.3V to 5V
COMP Voltage
-0.3V to 3V
All Other Pins
-0.3V to 7V
Maximum Junction Temperature
Power Dissipation
150°C
(3)
Internally Limited
Lead Temperature
216°C
Infrared (15 sec.)
ESD Susceptibility
235°C
(4)
Human Body Model
2kV
Machine Model
200V
−65°C to +150°C
Storage Temperature
(1)
Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the
device is intended to be functional, but device parameter specifications may not be ensured. For ensured specifications and test
conditions, see the Electrical Characteristics.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. See the Electrical Characteristics table for the thermal resistance of various layouts.
The maximum allowable power dissipation at any ambient temperature is calculated using: PD (MAX) = (TJ(MAX) − TA)/θJA. Exceeding
the maximum allowable power dissipation will cause excessive die temperature, and the regulator will go into thermal shutdown.
The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin. The machine model is a 200pF
capacitor discharged directly into each pin.
(2)
(3)
(4)
Operating Conditions
Operating Junction Temperature Range (1)
−40°C to +125°C
Supply Voltage (1)
(1)
3.1V to 40V
Supply voltage, bias current product will result in aditional device power dissipation. This power may be significant. The thermal
dissipation design should take this into account.
Electrical Characteristics
Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating
Temperature Range (TJ = −40°C to +125°C) Unless otherwise specified. VIN = 12V and IL = 0A, unless otherwise specified.
Symbol
IQ
Quiescent Current
VFB
Feedback Voltage
ICL
Switch Current Limit
%VFB/ΔVIN
Feedback Voltage Line
Regulation
IB
FB Pin Bias Current (3)
(1)
(2)
(3)
Typ (2)
Max (1)
Units
FB = 2V (Not Switching)
FS = 0V
2.0
2.5
mA
FB = 2V (Not Switching)
FS = Open
2.1
2.5
mA
VSHDN = 0V
18
30
µA
1.2330
1.259
1.2840
V
1.35
2.0
2.7
A
0.001
0.04
%/V
55
200
nA
Parameter
Conditions
Min (1)
3.1V ≤ VIN ≤ 40V
All limits specified at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are
100% production tested. All limits at temperature extremes are specified via correlation using standard Statistical Quality Control (SQC)
methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
Typical numbers are at 25°C and represent the most likely norm.
Bias current flows into FB pin.
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Electrical Characteristics (continued)
Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating
Temperature Range (TJ = −40°C to +125°C) Unless otherwise specified. VIN = 12V and IL = 0A, unless otherwise specified.
Symbol
BV
Parameter
Output Switch Breakdown
Voltage
Conditions
Min (1)
TJ = 25°C, ISW = 0.1µA
80
TJ = -40°C to + 125°C, ISW =
0.5µA
76
ΔI = 5µA
150
Typ (2)
Max (1)
Units
V
VIN
Input Voltage Range
gm
Error Amp Transconductance
3.1
AV
Error Amp Voltage Gain
280
V/V
DMAX
Maximum Duty Cycle
LM5000-3
FS = 0V
85
90
%
Maximum Duty Cycle
LM5000-6
FS = 0V
85
90
%
410
40
V
750
µmho
TMIN
Minimum On Time
fS
Switching Frequency LM50003
FS = 0V
240
165
300
360
FS = Open
550
700
840
Switching Frequency LM50006
FS = 0V
485
600
715
1.055
1.3
1.545
MHz
ISHDN
Shutdown Pin Current
VSHDN = 0V
−1
-2
µA
IL
Switch Leakage Current
VSW = 80V
0.008
5
µA
RDSON
Switch RDSON
ISW = 1A
160
445
mΩ
ThSHDN
SHDN Threshold
Output High
FS = Open
0.9
Output Low
UVLO
0.6
kHz
V
0.6
0.3
V
On Threshold
2.74
2.92
3.10
V
Off Threshold
2.60
2. 77
2.96
V
14
µA
OVP
VCOMP Trip
ISS
Softstart Current
θJA
Thermal Resistance
4
ns
0.67
8
11
TSSOP, Package only
150
WSON, Package only
45
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V
°C/W
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Typical Performance Characteristics
Iq (non-switching) vs VIN @ fSW = 300kHz
Iq (non-switching) vs VIN @ fSW = 700kHz
3.000
3.000
2.800
2.800
2.600
2.600
2.400
-40oC
2.200
2.000
Iq (mA)
Iq (mA)
2.400
25oC
1.800
125oC
2.000
25oC
1.800
125oC
1.600
1.600
1.400
1.400
1.200
1.200
1.000
1.000
0
10
5
15
20
25
35
30
40
-40oC
2.200
0
5
10
15
VIN (V)
Figure 3.
Iq (switching) vs VIN @ fSW = 300kHz
10
9
9
8
8
7
7
6
o
-40 C
5
4
25
30
35
40
Figure 4.
Iq (mA)
Iq (mA)
10
20
VIN (V)
125oC
Iq (switching) vs VIN @ fSW = 700kHz
-40oC
25oC
125oC
6
5
4
25oC
3
3
2
2
0
1.2800
5
10
15
20
25
30
35
40
0
5
10
15
20
25
VIN (V)
VIN (V)
Figure 5.
Figure 6.
Vfb vs Temperature
30
35
40
RDS(ON) vs VIN @ ISW =1A
400
300
RDS(ON) (m:)
FEEDBACK VOLTAGE (V)
350
1.2700
1.2600
1.2500
125oC
250
200
25oC
150
100
-40oC
1.2400
50
1.2300
-40 -20
0
20
40
80 100 120
60
0
0
5
10
15
20
25
30
35
40
o
TEMPERATURE ( C)
VIN (V)
Figure 7.
Figure 8.
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Typical Performance Characteristics (continued)
Current Limit vs Temperature
Current Limit vs VIN
2
2
1.95
1.95
1.9
1.85
CURRENT LIMIT (A)
CURRENT LIMIT (A)
1.9
1.8
1.75
1.7
1.65
1.85
1.8
1.75
1.7
1.65
1.6
1.6
1.55
1.55
1.5
-40 -20
1.5
0
20
40
60
80 100 120
0
5
10
15
TEMPERATURE (oC)
Figure 9.
25
30
35
40
Figure 10.
fSW vs. VIN @ FS = Low (-3)
fSW vs. VIN @ FS = OPEN (-3)
770
315
750
FREQUENCY (KHZ)
310
305
fsw (kHz)
20
VIN (V)
300
295
290
730
710
690
670
650
285
630
0
5
10
15
20
25
30
35
40
0
5
10
15
20
25
VIN (V)
VIN (V)
Figure 11.
Figure 12.
fSW vs. Temperature @ FS = Low (-3)
30
35
40
fSW vs. Temperature @ FS = OPEN (-3)
330
770
320
750
730
fsw (kHz)
fSW (kHz)
310
300
710
690
290
670
280
650
270
-40
630
-20
0
20
40
60
80
100 120
-40 -20
0
20
40
60
80 100 120
TEMPERATURE (oC)
TEMPERATURE (oC)
Figure 13.
6
Figure 14.
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Typical Performance Characteristics (continued)
fSW vs. Temperature @ FS = OPEN (-6)
640
1.38
620
1.34
600
1.30
fSW (kHz)
fSW (kHz)
fSW vs. Temperature @ FS = Low (-6)
580
1.26
560
1.22
540
1.18
520
-40
-20
0
20
40
60
80
1.14
-40
100 120
-20
TEMPERATURE (oC)
0
40
60
80
100 120
TEMPERATURE (oC)
Figure 15.
Figure 16.
Error Amp. Transconductance vs Temp.
BYP Pin Voltage vs VIN
600
8
550
7
500
6
BYP PIN VOLTAGE (V)
Gm [Pmho]
20
450
400
350
300
125oC
-40oC
25oC
5
4
3
2
1
250
-40 -20
0
20
40
60
80 100 120
TEMPERATURE (oC)
0
0
5
10
15
20
25
30
35
40
VIN (V)
Figure 17.
Figure 18.
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Typical Application Diagrams
Figure 19. 300 kHz operation, 48V output
Figure 20. 700 kHz operation, 48V output
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Block Diagram
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BOOST REGULATOR OPERATION
The LM5000 utilizes a PWM control scheme to regulate the output voltage over all load conditions. The operation
can best be understood referring to the block diagram and Figure 21. At the start of each cycle, the oscillator
sets the driver logic and turns on the NMOS power device conducting current through the inductor, cycle 1 of
Figure 21 (a). During this cycle, the voltage at the COMP pin controls the peak inductor current. The COMP
voltage will increase with larger loads and decrease with smaller. This voltage is compared with the summation
of the SW volatge and the ramp compensation.The ramp compensation is used in PWM architectures to
eliminate the sub-harmonic oscillations that occur during duty cycles greater than 50%. Once the summation of
the ramp compensation and switch voltage equals the COMP voltage, the PWM comparator resets the driver
logic turning off the NMOS power device. The inductor current then flows through the output diode to the load
and output capacitor, cycle 2 of Figure 21 (b). The NMOS power device is then set by the oscillator at the end of
the period and current flows through the inductor once again.
The LM5000 has dedicated protection circuitry running during the normal operation to protect the IC. The
Thermal Shutdown circuitry turns off the NMOS power device when the die temperature reaches excessive
levels. The UVP comparator protects the NMOS power device during supply power startup and shutdown to
prevent operation at voltages less than the minimum input voltage. The OVP comparator is used to prevent the
output voltage from rising at no loads allowing full PWM operation over all load conditions. The LM5000 also
features a shutdown mode. An external capacitor sets the softstart time by limiting the error amp output range,
as the capacitor charges up via an internal 10µA current source.
The LM5000 is available in two operating frequency ranges. The LM5000-3 is pin selectable for either 300kHz
(FS Grounded) or 700kHz (FS Open). The LM5000-6 is pin selectable for either 600kHz (FS Grounded) or
1.3MHz (FS Open)
Operation
Figure 21. Simplified Boost Converter Diagram
(a) First Cycle of Operation (b) Second Cycle Of Operation
CONTINUOUS CONDUCTION MODE
The LM5000 is a current-mode, PWM regulator. When used as a boost regulator the input voltage is stepped up
to a higher output voltage. In continuous conduction mode (when the inductor current never reaches zero at
steady state), the boost regulator operates in two cycles.
In the first cycle of operation, shown in Figure 21 (a), the transistor is closed and the diode is reverse biased.
Energy is collected in the inductor and the load current is supplied by COUT.
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The second cycle is shown in Figure 21 (b). During this cycle, the transistor is open and the diode is forward
biased. The energy stored in the inductor is transferred to the load and output capacitor.
The ratio of these two cycles determines the output voltage. The output voltage is defined approximately as:
VIN
VOUT =
1-D
, D' = (1-D) =
VIN
VOUT
where
•
•
D is the duty cycle of the switch
D and D′ will be required for design calculations
(1)
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the feedback pin and a resistor divider connected to the output as shown in
Figure 19. The feedback pin is always at 1.259V, so the ratio of the feedback resistors sets the output voltage.
VOUT - 1.259
:
RFB1 = RFB2 x
1.259
(2)
INTRODUCTION TO COMPENSATION
Figure 22. (a) Inductor current. (b) Diode current.
The LM5000 is a current mode PWM regulator. The signal flow of this control scheme has two feedback loops,
one that senses switch current and one that senses output voltage.
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To keep a current programmed control converter stable above duty cycles of 50%, the inductor must meet
certain criteria. The inductor, along with input and output voltage, will determine the slope of the current through
the inductor (see Figure 22 (a)). If the slope of the inductor current is too great, the circuit will be unstable above
duty cycles of 50%.
The LM5000 provides a compensation pin (COMP) to customize the voltage loop feedback. It is recommended
that a series combination of RC and CC be used for the compensation network, as shown in Figure 19. The
series combination of RC and CC introduces pole-zero pair according to the following equations:
fZC =
1
Hz
2SRCCC
(3)
1
fPC =
Hz
2S(RC + RO)CC
where
•
RO is the output impedance of the error amplifier, 850kΩ
(4)
For most applications, performance can be optimized by choosing values within the range 5kΩ ≤ RC ≤ 20kΩ and
680pF ≤ CC ≤ 4.7nF.
COMPENSATION
This section will present a general design procedure to help insure a stable and operational circuit. The designs
in this datasheet are optimized for particular requirements. If different conversions are required, some of the
components may need to be changed to ensure stability. Below is a set of general guidelines in designing a
stable circuit for continuous conduction operation (loads greater than 100mA), in most all cases this will provide
for stability during discontinuous operation as well. The power components and their effects will be determined
first, then the compensation components will be chosen to produce stability.
INDUCTOR SELECTION
To ensure stability at duty cycles above 50%, the inductor must have some minimum value determined by the
minimum input voltage and the maximum output voltage. This equation is:
2
L>
VINRDSON
0.144 fs
( DD') -1
( DD') +1
(in H)
where
•
•
•
fs is the switching frequency
D is the duty cycle
RDSON is the ON resistance of the internal switch
This equation is only good for duty cycles greater than 50% (D>0.5).
VIND
(in Amps)
'iL =
2Lfs
(5)
(6)
The inductor ripple current is important for a few reasons. One reason is because the peak switch current will be
the average inductor current (input current) plus ΔiL. Care must be taken to make sure that the switch will not
reach its current limit during normal operation. The inductor must also be sized accordingly. It should have a
saturation current rating higher than the peak inductor current expected. The output voltage ripple is also affected
by the total ripple current.
DC GAIN AND OPEN-LOOP GAIN
Since the control stage of the converter forms a complete feedback loop with the power components, it forms a
closed-loop system that must be stabilized to avoid positive feedback and instability. A value for open-loop DC
gain will be required, from which you can calculate, or place, poles and zeros to determine the crossover
frequency and the phase margin. A high phase margin (greater than 45°) is desired for the best stability and
transient response. For the purpose of stabilizing the LM5000, choosing a crossover point well below where the
right half plane zero is located will ensure sufficient phase margin. A discussion of the right half plane zero and
checking the crossover using the DC gain will follow.
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OUTPUT CAPACITOR SELECTION
The choice of output capacitors is somewhat more arbitrary. It is recommended that low ESR (Equivalent Series
Resistance, denoted RESR) capacitors be used such as ceramic, polymer electrolytic, or low ESR tantalum.
Higher ESR capacitors may be used but will require more compensation which will be explained later on in the
section. The ESR is also important because it determines the output voltage ripple according to the approximate
equation:
ΔVOUT ≊ 2ΔiLRESR (in Volts)
(7)
After choosing the output capacitor you can determine a pole-zero pair introduced into the control loop by the
following equations:
fP1 =
fZ1 =
1
(in Hz)
2S(RESR + RL)COUT
1
2SRESRCOUT
(8)
(in Hz)
where
•
RL is the minimum load resistance corresponding to the maximum load current
(9)
The zero created by the ESR of the output capacitor is generally very high frequency if the ESR is small. If low
ESR capacitors are used it can be neglected. If higher ESR capacitors are used see the HIGH OUTPUT
CAPACITOR ESR COMPENSATION section.
RIGHT HALF PLANE ZERO
A current mode control boost regulator has an inherent right half plane zero (RHP zero). This zero has the effect
of a zero in the gain plot, causing an imposed +20dB/decade on the rolloff, but has the effect of a pole in the
phase, subtracting another 90° in the phase plot. This can cause undesirable effects if the control loop is
influenced by this zero. To ensure the RHP zero does not cause instability issues, the control loop should be
designed to have a bandwidth of ½ the frequency of the RHP zero or less. This zero occurs at a frequency of:
RHPzero =
VOUT(D')2
(in Hz)
2S,LOADL
where
•
ILOAD is the maximum load current
(10)
SELECTING THE COMPENSATION COMPONENTS
The first step in selecting the compensation components RC and CC is to set a dominant low frequency pole in
the control loop. Simply choose values for RC and CC within the ranges given in the INTRODUCTION TO
COMPENSATION section to set this pole in the area of 10Hz to 100Hz. The frequency of the pole created is
determined by the equation:
fPC =
1
(in Hz)
2S(RC + RO)CC
where
•
RO is the output impedance of the error amplifier, 850kΩ
(11)
Since RC is generally much less than RO, it does not have much effect on the above equation and can be
neglected until a value is chosen to set the zero fZC. fZC is created to cancel out the pole created by the output
capacitor, fP1. The output capacitor pole will shift with different load currents as shown by the equation, so setting
the zero is not exact. Determine the range of fP1 over the expected loads and then set the zero fZC to a point
approximately in the middle. The frequency of this zero is determined by:
fZC =
1
(in Hz)
2SCCRC
(12)
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Now RC can be chosen with the selected value for CC. Check to make sure that the pole fPC is still in the 10Hz to
100Hz range, change each value slightly if needed to ensure both component values are in the recommended
range. After checking the design at the end of this section, these values can be changed a little more to optimize
performance if desired. This is best done in the lab on a bench, checking the load step response with different
values until the ringing and overshoot on the output voltage at the edge of the load steps is minimal. This should
produce a stable, high performance circuit. For improved transient response, higher values of RC (within the
range of values) should be chosen. This will improve the overall bandwidth which makes the regulator respond
more quickly to transients. If more detail is required, or the most optimal performance is desired, refer to a more
in depth discussion of compensating current mode DC/DC switching regulators.
HIGH OUTPUT CAPACITOR ESR COMPENSATION
When using an output capacitor with a high ESR value, or just to improve the overall phase margin of the control
loop, another pole may be introduced to cancel the zero created by the ESR. This is accomplished by adding
another capacitor, CC2, directly from the compensation pin VC to ground, in parallel with the series combination of
RC and CC. The pole should be placed at the same frequency as fZ1, the ESR zero. The equation for this pole
follows:
fPC2 =
1
(in Hz)
2SCC2(RC //RO)
(13)
To ensure this equation is valid, and that CC2 can be used without negatively impacting the effects of RC and CC,
fPC2 must be greater than 10fPC.
CHECKING THE DESIGN
The final step is to check the design. This is to ensure a bandwidth of ½ or less of the frequency of the RHP
zero. This is done by calculating the open-loop DC gain, ADC. After this value is known, you can calculate the
crossover visually by placing a −20dB/decade slope at each pole, and a +20dB/decade slope for each zero. The
point at which the gain plot crosses unity gain, or 0dB, is the crossover frequency. If the crossover frequency is
at less than ½ the RHP zero, the phase margin should be high enough for stability. The phase margin can also
be improved some by adding CC2 as discussed earlier in the section. The equation for ADC is given below with
additional equations required for the calculation:
gmROD'
RFB2
{[(ZcLeff)// RL]//RL} (in dB)
ADC(DB) = 20log10
RFB1 + RFB2 RDSON
(14)
(
)
2fs (in rad/s)
Zc #
nD'
L
Leff =
(D')2
(15)
(16)
2mc
(no unit)
n = 1+
m1
(17)
(18)
mc ≊ 0.072fs (in A/s)
m1 #
VINRDSON
L
(in V/s)
where
•
•
•
RL is the minimum load resistance
VIN is the maximum input voltage
RDSON is the value chosen from the graph "RDSON vs. VIN " in the Typical Performance Characteristics
section
(19)
SWITCH VOLTAGE LIMITS
In a flyback regulator, the maximum steady-state voltage appearing at the switch, when it is off, is set by the
transformer turns ratio, N, the output voltage, VOUT, and the maximum input voltage, VIN (Max):
VSW(OFF) = VIN (Max) + (VOUT +VF)/N
where
•
14
VF is the forward biased voltage of the output diode, and is typically 0.5V for Schottky diodes and 0.8V for
ultra-fast recovery diodes
(20)
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SNVS176D – MAY 2004 – REVISED MARCH 2007
In certain circuits, there exists a voltage spike, VLL, superimposed on top of the steady-state voltage . Usually,
this voltage spike is caused by the transformer leakage inductance and/or the output rectifier recovery time. To
“clamp” the voltage at the switch from exceeding its maximum value, a transient suppressor in series with a
diode is inserted across the transformer primary.
If poor circuit layout techniques are used, negative voltage transients may appear on the Switch pin. Applying a
negative voltage (with respect to the IC's ground) to any monolithic IC pin causes erratic and unpredictable
operation of that IC. This holds true for the LM5000 IC as well. When used in a flyback regulator, the voltage at
the Switch pin can go negative when the switch turns on. The “ringing” voltage at the switch pin is caused by the
output diode capacitance and the transformer leakage inductance forming a resonant circuit at the
secondary(ies). The resonant circuit generates the “ringing” voltage, which gets reflected back through the
transformer to the switch pin. There are two common methods to avoid this problem. One is to add an RC
snubber around the output rectifier(s). The values of the resistor and the capacitor must be chosen so that the
voltage at the Switch pin does not drop below −0.4V. The resistor may range in value between 10Ω and 1 kΩ,
and the capacitor will vary from 0.001 μF to 0.1 μF. Adding a snubber will (slightly) reduce the efficiency of the
overall circuit.
The other method to reduce or eliminate the “ringing” is to insert a Schottky diode clamp between the SW pin
and the PGND pin. The reverse voltage rating of the diode must be greater than the switch off voltage.
OUTPUT VOLTAGE LIMITATIONS
The maximum output voltage of a boost regulator is the maximum switch voltage minus a diode drop. In a
flyback regulator, the maximum output voltage is determined by the turns ratio, N, and the duty cycle, D, by the
equation:
VOUT ≈ N × VIN × D/(1 − D)
(21)
The duty cycle of a flyback regulator is determined by the following equation:
(22)
Theoretically, the maximum output voltage can be as large as desired—just keep increasing the turns ratio of the
transformer. However, there exists some physical limitations that prevent the turns ratio, and thus the output
voltage, from increasing to infinity. The physical limitations are capacitances and inductances in the LM5000
switch, the output diode(s), and the transformer—such as reverse recovery time of the output diode (mentioned
above).
INPUT LINE CONDITIONING
A small, low-pass RC filter should be used at the input pin of the LM5000 if the input voltage has an unusually
large amount of transient noise. Additionally, the RC filter can reduce the dissipation within the device when the
input voltage is high.
Flyback Regulator Operation
The LM5000 is ideally suited for use in the flyback regulator topology. The flyback regulator can produce a single
output voltage, or multiple output voltages.
The operation of a flyback regulator is as follows: When the switch is on, current flows through the primary
winding of the transformer, T1, storing energy in the magnetic field of the transformer. Note that the primary and
secondary windings are out of phase, so no current flows through the secondary when current flows through the
primary. When the switch turns off, the magnetic field collapses, reversing the voltage polarity of the primary and
secondary windings. Now rectifier D5 is forward biased and current flows through it, releasing the energy stored
in the transformer. This produces voltage at the output.
The output voltage is controlled by modulating the peak switch current. This is done by feeding back a portion of
the output voltage to the error amp, which amplifies the difference between the feedback voltage and a 1.259V
reference. The error amp output voltage is compared to a ramp voltage proportional to the switch current (i.e.,
inductor current during the switch on time). The comparator terminates the switch on time when the two voltages
are equal, thereby controlling the peak switch current to maintain a constant output voltage.
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15
LM5000
SNVS176D – MAY 2004 – REVISED MARCH 2007
www.ti.com
Figure 23. LM5000 Flyback Converter
16
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LM5000
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SNVS176D – MAY 2004 – REVISED MARCH 2007
ITEM
PART NUMBER
DESCRIPTION
VALUE
C
1
C4532X7R2A105MT
Capacitor, CER, TDK
1µ, 100V
C
2
C4532X7R2A105MT
Capacitor, CER, TDK
1µ, 100V
C
3
C1206C224K5RAC
Capacitor, CER, KEMET
0.22µ, 50V
C
4
C1206C104K5RAC
Capacitor, CER, KEMET
0.1µ, 50V
C
5
C1206C104K5RAC
Capacitor, CER, KEMET
0.1µ, 50V
C
6
C1206C101K1GAC
Capacitor, CER, KEMET
100p, 100V
C
7
C1206C104K5RAC
Capacitor, CER, KEMET
0.1µ, 50V
C
8
C4532X7S0G686M
Capacitor, CER, TDK
68µ, 4V
C
9
C4532X7S0G686M
Capacitor, CER, TDK
68µ, 4V
C
10
C1206C221K1GAC
Capacitor, CER, KEMET
220p, 100V
C
11
C1206C102K5RAC
Capacitor, CER, KEMET
1000p, 500V
D
1
BZX84C10-NSA
Central, 10V Zener, SOT-23
D
2
CMZ5930B-NSA
Central, 16V Zener, SMA
D
3
CMPD914-NSA
Central, Switching, SOT-23
D
4
CMPD914-NSA
Central, Switching, SOT-23
D
5
CMSH3-40L-NSA
Central, Schottky, SMC
T
1
A0009-A
Coilcraft, Transformer
R
1
CRCW12064992F
Resistor
49.9K
R
2
CRCW12061001F
Resistor
1K
R
3
CRCW12061002F
Resistor
10K
R
4
CRCW12066191F
Resistor
6.19K
R
5
CRCW120610R0F
Resistor
10
R
6
CRCW12062003F
Resistor
200K
10K
R
7
CRCW12061002F
Resistor
Q
1
CXT5551-NSA
Central, NPN, 180V
U
1
LM5000-3
Regulator, TI
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Copyright © 2004–2007, Texas Instruments Incorporated
Product Folder Links: LM5000
17
PACKAGE OPTION ADDENDUM
www.ti.com
30-Sep-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM5000-3MTC
NRND
TSSOP
PW
16
92
Non-RoHS
& Green
Call TI
Level-1-260C-UNLIM
-40 to 125
LM5000
3MTC
LM5000-3MTC/NOPB
ACTIVE
TSSOP
PW
16
92
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
LM5000
3MTC
LM5000-3MTCX
NRND
TSSOP
PW
16
2500
Non-RoHS
& Green
Call TI
Level-1-260C-UNLIM
-40 to 125
LM5000
3MTC
LM5000-3MTCX/NOPB
ACTIVE
TSSOP
PW
16
2500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
LM5000
3MTC
LM5000SD-3/NOPB
ACTIVE
WSON
NHQ
16
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
5000-3
LM5000SD-6/NOPB
ACTIVE
WSON
NHQ
16
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
5000-6
LM5000SDX-3/NOPB
ACTIVE
WSON
NHQ
16
4500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
5000-3
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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