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LM5022
SNVS480J – JANUARY 2007 – REVISED JULY 2020
LM5022 60-V Low-Side Controller for Boost and SEPIC
1 Features
3 Description
•
•
•
The LM5022 device is a high-voltage, low-side, Nchannel MOSFET controller ideal for use in boost and
SEPIC regulators. It contains all of the features
required to implement single-ended primary
topologies. Output voltage regulation is based on
current-mode control, which eases the design of loop
compensation while providing inherent input voltage
feedforward. The LM5022 includes a start-up
regulator that operates over a wide input range of 6 V
to 60 V. The PWM controller is designed for highspeed capability including an oscillator frequency
range up to 2.2 MHz and total propagation delays
less than 100 ns. Additional features include an error
amplifier, precision reference, line undervoltage
lockout,
cycle-by-cycle
current
limit,
slope
compensation, soft start, external synchronization
capability, and thermal shutdown. The LM5022 is
available in the 10-pin VSSOP package.
1
•
•
•
•
•
•
•
•
Internal 60-V start-up regulator
1-A Peak MOSFET gate driver
VIN Range: 6 V to 60 V (operates down to 3 V
after start-up)
Duty cycle limit of 90%
Programmable UVLO with hysteresis
Cycle-by-cycle current limit
External synchronizable (AC-coupled)
Single resistor oscillator frequency set
Slope compensation
Adjustable soft start
10-Pin VSSOP package
2 Applications
•
•
Boost converters
SEPIC converters
Device Information(1)
PART NUMBER
LM5022
PACKAGE
VSSOP (10)
BODY SIZE (NOM)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application
VIN
L1
CIN
RS1
VIN
VO
D1
CO
Q1
OUT
RT
RUV2
RT
RUV1
LM5022
UVLO
CSS
RSNS
CS
SS
CCS
GND
CF
VCC
COMP
FB
R1
C2
RFB2
RFB1
C1
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM5022
SNVS480J – JANUARY 2007 – REVISED JULY 2020
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Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
7
Absolute Maximum Ratings ......................................
ESD Ratings ............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description .............................................. 9
7.1
7.2
7.3
7.4
Overview ................................................................... 9
Functional Block Diagram ......................................... 9
Feature Description................................................. 10
Device Functional Modes........................................ 12
8
Application and Implementation ........................ 14
8.1 Application Information............................................ 14
8.2 Typical Application .................................................. 14
9 Power Supply Recommendations...................... 29
10 Layout................................................................... 30
10.1 Layout Guidelines ................................................. 30
10.2 Layout Examples................................................... 31
11 Device and Documentation Support ................. 33
11.1
11.2
11.3
11.4
11.5
11.6
11.7
Device Support......................................................
Documentation Support ........................................
Receiving Notification of Documentation Updates
Support Resources ...............................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
33
33
33
33
33
33
33
12 Mechanical, Packaging, and Orderable
Information ........................................................... 33
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision I (December 2017) to Revision J
Page
•
Changed Figure 14 to correct Gnd trace between device-Gnd pin and Rsns........................................................................ 14
•
Corrected Equation 3 ........................................................................................................................................................... 16
Changes from Revision H (December 2016) to Revision I
•
Page
Changed numerator From: 1 To: C2 in Equation 55 ............................................................................................................ 24
Changes from Revision G (December 2013) to Revision H
Page
•
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section .................................................................................................. 1
•
Deleted soldering temperature (215°C Vapor phase maximum and 220°C Infrared maximum) ........................................... 4
•
Changed Junction to Ambient Thermal Resistance, RθJA, value From: 200 To: 161.5 .......................................................... 4
•
Changed slope compensation amplitude, VSLOPE, values From: 80 To: 83 (Minimum), From: 105 To: 110 (Typical),
and From: 130 To: 137 (Maximum)........................................................................................................................................ 5
Changes from Revision F (March 2013) to Revision G
•
Changed timing resistor equation. Incorrect change when converting to TI format ............................................................. 12
Changes from Revision E (March 2013) to Revision F
•
2
Page
Page
Changed layout of National Semiconductor Data Sheet to TI format .................................................................................... 1
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SNVS480J – JANUARY 2007 – REVISED JULY 2020
5 Pin Configuration and Functions
DGS Package
10-Pin VSSOP
Top View
VIN
1
10
SS
FB
2
9
RT/SYNC
COMP
3
8
CS
VCC
4
7
UVLO
OUT
5
6
GND
Not to scale
Pin Functions
PIN
NO.
NAME
I/O
DESCRIPTION
1
VIN
I
Source input voltage: Input to the start-up regulator. Operates from 6 V to 60 V.
2
FB
I
Feedback pin: Inverting input to the internal voltage error amplifier. The non-inverting input of the error
amplifier connects to a 1.25-V reference.
3
COMP
I/O
Error amplifier output and PWM comparator input: The control loop compensation components connect
between this pin and the FB pin.
4
VCC
O
Output of the internal, high-voltage linear regulator: This pin must be bypassed to the GND pin with a
ceramic capacitor.
5
OUT
O
Output of MOSFET gate driver: Connect this pin to the gate of the external MOSFET. The gate driver has
a 1-A peak current capability.
6
GND
—
System ground
7
UVLO
I
Input undervoltage lockout: Set the start-up and shutdown levels by connecting this pin to the input voltage
through a resistor divider. A 20-µA current source provides hysteresis.
8
CS
I
Current sense input: Input for the switch current used for current mode control and for current limiting.
9
RT/SYNC
I
Oscillator frequency adjust pin and synchronization input: An external resistor connected from this pin to
GND sets the oscillator frequency. This pin can also accept an AC-coupled input for synchronization from
an external clock.
10
SS
I
Soft-start pin: An external capacitor placed from this pin to ground is charged by a 10-µA current source,
creating a ramp voltage to control the regulator start-up.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2)
MIN
MAX
UNIT
VIN to GND
–0.3
65
V
VCC to GND
–0.3
16
V
RT/SYNC to GND
–0.3
5.5
V
–1.5 for < 100 ns
V
OUT to GND
All other pins to GND
–0.3
Power dissipation
7
Junction temperature, TJ (3)
Storage temperature, Tstg
(1)
(2)
(3)
V
Internally limited
–65
150
°C
150
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
High junction temperatures degrade operating lifetimes. Operating lifetime is derated for junction temperatures greater than 125°C.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
(3)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1) (2)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (3)
±750
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
The human-body model is a 100-pF capacitor discharged through a 1.5-kΩ resistor into each pin.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted) (1)
Supply voltage
MIN
MAX
6
60
UNIT
V
External voltage at VCC
7.5
14
V
Junction temperature
–40
125
°C
(1)
Device thermal limitations may limit usable range
6.4 Thermal Information
LM5022
THERMAL METRIC (1)
DGS (VSSOP)
UNIT
10 PINS
RθJA
Junction-to-ambient thermal resistance
RθJC(top)
Junction-to-case (top) thermal resistance
161.5
°C/W
56
°C/W
RθJB
ψJT
Junction-to-board thermal resistance
81.3
°C/W
Junction-to-top characterization parameter
5.7
°C/W
ψJB
Junction-to-board characterization parameter
80
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
—
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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6.5 Electrical Characteristics
Typical limits apply for TJ=25°C and are provided for reference purposes only; minimum and maximum limits apply over the
junction temperature (TJ) range of –40°C to 125°C. VIN = 24 V and RT = 27.4 kΩ (unless otherwise noted). (1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
1.225
1.25
1.275
V
6.6
7
7.4
SYSTEM PARAMETERS
VFB
FB pin voltage
STARTUP REGULATOR
10 V ≤ VIN ≤ 60 V, ICC = 1 mA
VCC
VCC regulation (2)
ICC
Supply current
OUT Pin Capacitance = 0,
VCC = 10 V
ICC-LIM
VCC current limit
VCC = 0 V (2) (3)
35
mA
VIN - VCC
Dropout voltage across bypass switch
ICC = 0 mA, ƒSW < 200 kHz,
6 V ≤ VIN ≤ 8.5 V
200
mV
VBYP-HI
Bypass switch turnoff threshold
VIN increasing
8.7
V
VBYP-HYS
Bypass switch threshold hysteresis
VIN Decreasing
260
mV
6 V ≤ VIN < 10V,
VCC Pin Open Circuit
V
5
3.5
15
VIN = 6 V
58
VIN = 8 V
53
VIN = 24 V
1.6
4
mA
ZVCC
VCC pin output impedance
0 mA ≤ ICC ≤ 5 mA
VCC-HI
VCC pin UVLO rising threshold
VCC-HYS
VCC pin UVLO falling hysteresis
IVIN
Startup regulator leakage
VIN = 60 V
150
500
µA
IIN-SD
Shutdown current
VUVLO = 0 V, VCC = Open Circuit
350
450
µA
Ω
5
V
300
mV
ERROR AMPLIFIER
GBW
Gain bandwidth
ADC
DC gain
ICOMP
COMP pin current sink capability
4
VFB = 1.5 V, VCOMP = 1 V
MHz
75
dB
5
17
mA
1.22
1.25
1.28
V
16
20
24
µA
UVLO
VSD
Shutdown threshold
ISD-HYS
Shutdown hysteresis current source
CURRENT LIMIT
tLIM-DLY
Delay from ILIM to output
VCS
Current limit threshold voltage
tBLK
Leading edge blanking time
RCS
CS pin sink impedance
CS steps from 0 V to 0.6 V, OUT
transitions to 90% of VCC
30
0.45
0.5
ns
0.55
65
Blanking active
V
ns
40
75
Ω
7
10
13
µA
0.35
0.55
0.75
V
SOFT START
ISS
Soft-start current source
VSS-OFF
Soft start to COMP offset
(1)
(2)
(3)
All Minimum and Maximum limits are specified by correlating the electrical characteristics to process and temperature variations and
applying statistical process control. The junction temperature (TJ in °C) is calculated from the ambient temperature (TA in °C) and power
dissipation (PD in Watts) as follows: TJ = TA + (PD × RθJA) where RθJA (in °C/W) is the package thermal impedance provided in Thermal
Information.
VCC provides bias for the internal gate drive and control circuits.
Device thermal limitations may limit usable range.
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Electrical Characteristics (continued)
Typical limits apply for TJ=25°C and are provided for reference purposes only; minimum and maximum limits apply over the
junction temperature (TJ) range of –40°C to 125°C. VIN = 24 V and RT = 27.4 kΩ (unless otherwise noted).(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OSCILLATOR
170 (4)
200
230
kHz
RT to GND = 27.4 kΩ
See
(4)
525
600
675
kHz
RT to GND = 16.2 kΩ
See (4)
865
990
1115
kHz
RT to GND = 84.5 kΩ
fSW
VSYNC-HI
Synchronization rising threshold
3.8
V
PWM COMPARATOR
tCOMP-DLY
Delay from COMP to OUT transition
VCOMP = 2 V, CS stepped
from 0 V to 0.4 V
DMIN
Minimum duty cycle
VCOMP = 0 V
DMAX
Maximum duty cycle
APWM
COMP to PWM comparator gain
VCOMP-OC
COMP pin open circuit voltage
VFB = 0 V
4.3
5.2
6.1
V
ICOMP-SC
COMP pin short circuit current
VCOMP = 0 V, VFB = 1.5 V
0.6
1.1
1.5
mA
83
110
137
mV
25
ns
0%
90%
95%
0.33
V/V
SLOPE COMPENSATION
VSLOPE
Slope compensation amplitude
MOSFET DRIVER
VSAT-HI
Output high saturation voltage (VCC – VOUT)
IOUT = 50 mA
0.25
0.75
V
VSAT-LO
Output low saturation voltage (VOUT)
IOUT = 100 mA
0.25
0.75
V
tRISE
OUT pin rise time
OUT Pin load = 1 nF
18
ns
tFALL
OUT pin fall time
OUT Pin load = 1 nF
15
ns
THERMAL CHARACTERISTICS
TSD
Thermal shutdown threshold
165
°C
TSD-HYS
Thermal shutdown hysteresis
25
°C
(4)
6
Specification applies to the oscillator frequency.
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6.6 Typical Characteristics
VO = 40 V
VIN = 24 V
Figure 1. Efficiency, Example Circuit BOM
TA = 25°C
Figure 2. VFB vs Temperature
TA = 25°C
Figure 3. VFB vs VIN
Figure 4. VCC vs VIN
TA = 25°C
RT = 16.2 KΩ
Figure 5. Maximum Duty Cycle vs ƒSW
Figure 6. ƒSW vs Temperature
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Typical Characteristics (continued)
TA = 25°C
8
Figure 7. RT vs ƒSW
Figure 8. SS vs Temperature
Figure 9. OUT Pin TRISE vs Gate Capacitance
Figure 10. OUT Pin TFALL vs Gate Capacitance
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7 Detailed Description
7.1 Overview
The LM5022 is a low-side, N-channel MOSFET controller that contains all of the features required to implement
single-ended power converter topologies. The LM5022 includes a high-voltage start-up regulator that operates
over a wide input range of 6 V to 60 V. The PWM controller is designed for high-speed capability including an
oscillator frequency range up to 2.2 MHz and total propagation delays less than 100 ns. Additional features
include an error amplifier, precision reference, input undervoltage lockout, cycle-by-cycle current limit, slope
compensation, soft start, oscillator sync capability, and thermal shutdown.
The LM5022 is designed for current-mode control power converters that require a single drive output, such as
boost and SEPIC topologies. The LM5022 provides all of the advantages of current-mode control including input
voltage feedforward, cycle-by-cycle current limiting, and simplified loop compensation.
7.2 Functional Block Diagram
BYPASS
SWITCH
(6 V to 8.7 V)
VCC
VIN
7-V SERIES
REGULATOR
REFERENCE
ENABLE
UVLO
1.25 V
+
-
5V
1.25 V
LOGIC
UVLO
HYSTERESIS
(20 µA)
RT/SYNC
CLK
OSC
DRIVER
45 µA
Max
Duty
Limit
0
S
Q
R
Q
OUT
5V
COMP
GND
5k
1.25 V
PWM
+
-
100 NŸ
FB
1.4 V
50 NŸ
SS
CS
0.5 V
2 NŸ
LOGIC
SS
10 µA
SS
+
-
CLK + LEB
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7.3 Feature Description
7.3.1 High-Voltage Start-Up Regulator
The LM5022 contains an internal high-voltage start-up regulator that allows the VIN pin to be connected directly
to line voltages as high as 60 V. The regulator output is internally current limited to 35 mA (typical). When power
is applied, the regulator is enabled and sources current into an external capacitor, CF, connected to the VCC pin.
The recommended capacitance range for CF is 0.1 µF to 100 µF. When the voltage on the VCC pin reaches the
rising threshold of 5 V, the controller output is enabled. The controller remains enabled until VCC falls below 4.7
V. In applications using a transformer, an auxiliary winding can be connected through a diode to the VCC pin.
This winding must raise the VCC pin voltage to above 7.5 V to shut off the internal start-up regulator. Powering
VCC from an auxiliary winding improves conversion efficiency while reducing the power dissipated in the
controller. The capacitance of CF must be high enough that it maintains the VCC voltage greater than the VCC
UVLO falling threshold (4.7 V) during the initial start-up. During a fault condition when the converter auxiliary
winding is inactive, external current draw on the VCC line must be limited such that the power dissipated in the
start-up regulator does not exceed the maximum power dissipation capability of the controller.
An external start-up or other bias rail can be used instead of the internal start-up regulator by connecting the
VCC and the VIN pins together and feeding the external bias voltage (7.5 V to 14 V) to the two pins.
7.3.2 Input Undervoltage Detector
The LM5022 contains an input undervoltage lockout (UVLO) circuit. UVLO is programmed by connecting the
UVLO pin to the center point of an external voltage divider from VIN to GND. The resistor divider must be
designed such that the voltage at the UVLO pin is greater than 1.25 V when VIN is in the desired operating
range. If the undervoltage threshold is not met, all functions of the controller are disabled and the controller
remains in a low power standby state. UVLO hysteresis is accomplished with an internal 20-µA current source
that is switched on or off into the impedance of the setpoint divider. When the UVLO threshold is exceeded, the
current source is activated to instantly raise the voltage at the UVLO pin. When the UVLO pin voltage falls below
the 1.25-V threshold the current source is turned off, causing the voltage at the UVLO pin to fall. The UVLO pin
can also be used to implement a remote enable or disable function. If an external transistor pulls the UVLO pin
below the 1.25-V threshold, the converter is disabled. This external shutdown method is shown in Figure 11.
VIN
VIN
RUV2
LM5022
UVLO
RUV1
ON/OFF
2N7000 or
Equivalent
GND
Figure 11. Enable or Disable Using UVLO
10
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Feature Description (continued)
7.3.3 Error Amplifier
An internal high gain error amplifier is provided within the LM5022. The noninverting input of the amplifier is
internally set to a fixed reference voltage of 1.25 V. The inverting input is connected to the FB pin. In nonisolated applications such as the boost converter the output voltage, VO, is connected to the FB pin through a
resistor divider. The control loop compensation components are connected between the COMP and FB pins. For
most isolated applications, the error amplifier function is implemented on the secondary side of the converter and
the internal error amplifier is not used. The internal error amplifier is configured as an open-drain output and can
be disabled by connecting the FB pin to ground. An internal 5-kΩ pullup resistor between a 5-V reference and
COMP can be used as the pullup for an opto-coupler in isolated applications.
7.3.4 Current Sensing and Current Limiting
The LM5022 provides a cycle-by-cycle over current protection function. Current limit is accomplished by an
internal current sense comparator. If the voltage at the current sense comparator input exceeds 0.5 V, the
MOSFET gate drive is immediately terminated. A small RC filter, placed near the controller, is recommended to
filter noise from the current sense signal. The CS input has an internal MOSFET which discharges the CS pin
capacitance at the conclusion of every cycle. The discharge device remains on an additional 65 ns after the
beginning of the new cycle to attenuate leading edge ringing on the current sense signal.
The LM5022 current sense and PWM comparators are very fast, and may respond to short duration noise
pulses. Layout considerations are critical for the current sense filter and sense resistor. The capacitor associated
with the CS filter must be placed very close to the device and connected directly to the pins of the controller (CS
and GND). If a current sense transformer is used, both leads of the transformer secondary must be routed to the
sense resistor and the current sense filter network. The current sense resistor can be placed between the source
of the primary power MOSFET and power ground, but it must be a low inductance type. When designing with a
current sense resistor all of the noise sensitive low-power ground connections must be connected together
locally to the controller and a single connection must be made to the high current power ground (sense resistor
ground point).
7.3.5 PWM Comparator and Slope Compensation
The PWM comparator compares the current ramp signal with the error voltage derived from the error amplifier
output. The error amplifier output voltage at the COMP pin is offset by 1.4 V and then further attenuated by a 3:1
resistor divider. The PWM comparator polarity is such that 0 V on the COMP pin results in a zero duty cycle at
the controller output. For duty cycles greater than 50%, current mode control circuits can experience
subharmonic oscillation. By adding an additional fixed-slope voltage ramp signal (slope compensation), this
oscillation can be avoided. Proper slope compensation damps the double pole associated with current mode
control (see the Control Loop Compensation section) and eases the design of the control loop compensator. The
LM5022 generates the slope compensation with a sawtooth-waveform current source with a slope of 45 µA ×
ƒSW, generated by the clock (see Figure 12). This current flows through an internal 2-kΩ resistor to create a
minimum compensation ramp with a slope of 100 mV × ƒSW (typical). The slope of the compensation ramp
increases when external resistance is added for filtering the current sense (RS1) or in the position RS2. As shown
in Figure 12 and the Functional Block Diagram, the sensed current slope and the compensation slope add
together to create the signal used for current limiting and for the control loop itself.
ISW
LM5022
45 µA
0
RS1
RS2
CS
2 NŸ
0.5V
RSNS
CSNS
+
Current
Limit
VCL
Figure 12. Slope Compensation
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Feature Description (continued)
In peak current mode control, the optimal slope compensation is proportional to the slope of the inductor current
during the power switch off-time. For boost converters, the inductor current slope while the MOSFET is off is
(VO – VIN) / L. This relationship is combined with the requirements to set the peak current limit and is used to
select RSNS and RS2 in the Application and Implementation section.
7.3.6 Soft Start
The soft start feature allows the power converter output to gradually reach the initial steady-state output voltage,
thereby reducing start-up stresses and current surges. At power on, after the VCC and input undervoltage
lockout thresholds are satisfied, an internal 10-µA current source charges an external capacitor connected to the
SS pin. The capacitor voltage ramps up slowly and limits the COMP pin voltage and the switch current.
7.3.7 MOSFET Gate Driver
The LM5022 provides an internal gate driver through the OUT pin that can source and sink a peak current of 1 A
to control external, ground-referenced N-channel MOSFETs.
7.3.8 Thermal Shutdown
Internal thermal shutdown circuitry is provided to protect the LM5022 in the event that the maximum junction
temperature is exceeded. When activated, typically at 165°C, the controller is forced into a low power standby
state, disabling the output driver and the VCC regulator. After the temperature is reduced (typical hysteresis is
25°C) the VCC regulator is re-enabled and the LM5022 performs a soft start.
7.4 Device Functional Modes
7.4.1 Oscillator, Shutdown, and SYNC
A single external resistor, RT, connected between the RT/SYNC and GND pins sets the LM5022 oscillator
frequency. To set the switching frequency (ƒSW), RT can be calculated with Equation 1.
RT
(1 - 8 ´ 10
=
-8
´ fSW
fSW ´ 5.77 ´ 10
)
-11
where
•
•
fSW is in Hz
RT is in Ω
(1)
The LM5022 can also be synchronized to an external clock. The external clock must have a higher frequency
than the free-running oscillator frequency set by the RT resistor. The clock signal must be capacitively coupled
into the RT/SYNC pin with a 100-pF capacitor as shown in Figure 13. A peak voltage level greater than 3.8 V at
the RT/SYNC pin is required for detection of the sync pulse. The sync pulse width must be set between 15 ns to
150 ns by the external components. The RT resistor is always required, whether the oscillator is free-running or
externally synchronized. The voltage at the RT/SYNC pin is internally regulated to 2 V, and the typical delay from
a logic high at the RT/SYNC pin to the rise of the OUT pin voltage is 120 ns. RT must be placed very close to the
device and connected directly to the pins of the controller (RT/SYNC and GND).
12
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Device Functional Modes (continued)
EXTERNAL
CLOCK
LM5022
CSS
RT/SYNC
100 pF
RT
15 ns to 150 ns
EXTERNAL
CLOCK
120 ns
(Typical)
OUT PIN
Figure 13. SYNC Operation
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The most common circuit controlled by the LM5022 is a non-isolated boost regulator. The boost regulator steps
up the input voltage and has a duty ratio D in Equation 2.
D
V O - V IN + V D
V O + VD
where
•
VD is the forward voltage drop of the output diode
(2)
The following is a design procedure for selecting all the components for the boost converter circuit shown in
Figure 14. The application is in-cabin automotive, meaning that the operating ambient temperature ranges from
–20°C to 85°C. This circuit operates in continuous conduction mode (CCM), where inductor current stays above
0 A at all times, and delivers an output voltage of 40 V ±2% at a maximum output current of 0.5 A. Additionally,
the regulator must be able to handle a load transient of up to 0.5 A while keeping VO within ±4%. The voltage
input comes from the battery or alternator system of an automobile, where the standard range of 9 V to 16 V and
transients of up to 32 V must not cause any malfunction.
8.2 Typical Application
L1
VIN = 9 V to 16 V
CIN1,2
D1
VO = 40 V
CINX
Q1
RS1
VIN
RS2
UVLO
CSS
LM5022
RT
RUV1
COX
OUT
RT
RUV2
CO1,2
SS
RSNS
CS
CCS
GND
CF
VCC
COMP
RFB2
FB
R1
C2
RFB1
C1
Figure 14. LM5022 Typical Application
14
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Typical Application (continued)
8.2.1 Design Requirements
For typical low-side controller applications, use the parameters listed in Table 1.
Table 1. Design Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage range
9 V to 16 V
Output voltage
40 V
Maximum output current
500 mA
Switching frequency
500 kHz
8.2.2 Detailed Design Procedure
Table 2 lists the bill of materials for this design example.
Table 2. BOM for Example Circuit
ID
PART NUMBER
TYPE
SIZE
PARAMETERS
QTY
VENDOR
U1
LM5022
Low-Side Controller
10-pin VSSOP
60 V
1
TI
Q1
Si4850EY
MOSFET
SO-8
60 V, 31 mΩ, 27 nC
1
Vishay
D1
CMSH2-60M
Schottky Diode
SMA
60 V, 2 A
1
Central Semi
L1
SLF12575T-M3R2
Inductor
12.5 × 12.5 × 7.5 mm
33 µH, 3.2 A, 40 mΩ
1
TDK
Cin1, Cin2
C4532X7R1H475M
Capacitor
1812
4.7 µF, 50 V, 3 mΩ
2
TDK
Co1, Co2
C5750X7R2A475M
Capacitor
2220
4.7 µF,100 V, 3 mΩ
2
TDK
Cf
C2012X7R1E105K
Capacitor
0805
1 µF, 25 V
1
TDK
Cinx
Cox
C2012X7R2A104M
Capacitor
0805
100 nF, 100 V
2
TDK
C1
VJ0805A561KXXAT
Capacitor
0805
560 pF 10%
1
Vishay
C2
VJ0805Y124KXXAT
Capacitor
0805
120 nF 10%
1
Vishay
Css
VJ0805Y103KXXAT
Capacitor
0805
10 nF 10%
1
Vishay
Ccs
VJ0805Y102KXXAT
Capacitor
0805
1 nF 10%
1
Vishay
R1
CRCW08053011F
Resistor
0805
3.01 kΩ 1%
1
Vishay
Rfb1
CRCW08056490F
Resistor
0805
649 Ω 1%
1
Vishay
Rfb2
CRCW08052002F
Resistor
0805
20 kΩ 1%
1
Vishay
Rs1
CRCW0805101J
Resistor
0805
100 Ω 5%
1
Vishay
Rs2
CRCW08053571F
Resistor
0805
3.57 kΩ 1%
1
Vishay
Rsns
ERJL14KF10C
Resistor
1210
100 mΩ, 1%, 0.5 W
1
Panasonic
Rt
CRCW08053322F
Resistor
0805
33.2 kΩ 1%
1
Vishay
Ruv1
CRCW08052611F
Resistor
0805
2.61 kΩ 1%
1
Vishay
Ruv2
CRCW08051002F
Resistor
0805
10 kΩ 1%
1
Vishay
8.2.2.1 Switching Frequency
The selection of switching frequency is based on the tradeoffs between size, cost, and efficiency. In general, a
lower frequency means larger, more expensive inductors and capacitors is required. A higher switching
frequency generally results in a smaller but less efficient solution, as the power MOSFET gate capacitances must
be charged and discharged more often in a given amount of time. For this application, a frequency of 500 kHz
was selected as a good compromise between the size of the inductor and efficiency. PCB area and component
height are restricted in this application. Following Equation 1, a 33.2-kΩ 1% resistor must be used to switch at
500 kHz.
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8.2.2.2 MOSFET
Selection of the power MOSFET is governed by tradeoffs between cost, size, and efficiency. Breaking down the
losses in the MOSFET is one way to determine relative efficiencies between different devices. For this example,
the SO 8-pin package provides a balance of a small footprint with good efficiency (see Q1 in Table 2).
Losses in the MOSFET can be broken down into conduction loss, gate charging loss, and switching loss.
Conduction, or I2R loss (PC) is approximately Equation 3.
2
ª§ I
º
O ·
«
Pc D u ¨¨
¸¸ u R DSON u 1.3 »
« 1 D¹
»
©
¼
(3)
The factor 1.3 accounts for the increase in MOSFET on-resistance due to heating. Alternatively, the factor of 1.3
can be ignored and the maximum on-resistance of the MOSFET can be used.
Gate charging loss, PG, results from the current required to charge and discharge the gate capacitance of the
power MOSFET and is approximated with Equation 4.
PG = VCC × QG × fSW
(4)
QG is the total gate charge of the MOSFET. Gate charge loss differs from conduction and switching losses
because the actual dissipation occurs in the LM5022 and not in the MOSFET itself. If no external bias is applied
to the VCC pin, additional loss in the LM5022 IC occurs as the MOSFET driving current flows through the VCC
regulator. This loss (PVCC) is estimated with Equation 5.
PVCC = (VIN – VCC) × QG × fSW
(5)
Switching loss (PSW) occurs during the brief transition period as the MOSFET turns on and off. During the
transition period both current and voltage are present in the channel of the MOSFET. The loss can be
approximated with Equation 6.
PSW = 0.5 × VIN × [IO / (1 – D)] × (tR + tF) × ƒSW
where
•
•
tR is the rise time of the MOSFET
tF is the fall time of the MOSFET
(6)
For this example, the maximum drain-to-source voltage applied across the MOSFET is VO plus the ringing due to
parasitic inductance and capacitance. The maximum drive voltage at the gate of the high-side MOSFET is VCC,
or 7 V typical. The MOSFET selected must be able to withstand 40 V plus any ringing from drain to source, and
be able to handle at least 7 V plus ringing from gate to source. A minimum voltage rating of 50-VD-S and 10-VG-S
MOSFET is used. Comparing the losses in a spreadsheet leads to a 60 VD-S rated MOSFET in SO-8 with an
RDSON of 22 mΩ (the maximum value is 31 mΩ), a gate charge of 27 nC, and rise and falls times of 10 ns and 12
ns, respectively.
8.2.2.3 Output Diode
The boost regulator requires an output diode D1 (see Figure 14) to carrying the inductor current during the
MOSFET off-time. The most efficient choice for D1 is a Schottky diode due to low forward drop and near-zero
reverse recovery time. D1 must be rated to handle the maximum output voltage plus any switching node ringing
when the MOSFET is on. In practice, all switching converters have some ringing at the switching node due to the
diode parasitic capacitance and the lead inductance. D1 must also be rated to handle the average output current,
IO.
The overall converter efficiency becomes more dependent on the selection of D1 at low duty cycles, where the
boost diode carries the load current for an increasing percentage of the time. This power dissipation can be
calculating by checking the typical diode forward voltage, VD, from the I-V curve on the data sheet of the diode
and then multiplying it by IO. Diode data sheets also provides a typical junction-to-ambient thermal resistance,
RθJA, which can be used to estimate the operating die temperature of the Schottky. Multiplying the power
dissipation (PD = IO × VD) by RθJA gives the temperature rise. The diode case size can then be selected to
maintain the Schottky diode temperature below the operational maximum.
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In this example, a Schottky diode rated to 60 V and 1 A is suitable, as the maximum diode current is 0.5 A. A
small case such as SOD-123 can be used if a small footprint is critical. Larger case sizes generally have lower
RθJA and lower forward voltage drop, so for better efficiency the larger SMA case size is used.
8.2.2.4 Boost Inductor
The first criterion for selecting an inductor is the inductance itself. In fixed-frequency boost converters, this value
is based on the desired peak-to-peak ripple current, ΔiL, which flows in the inductor along with the average
inductor current, IL. For a boost converter in CCM, IL is greater than the average output current, IO. The two
currents are related by Equation 7.
IL = IO / (1 – D)
(7)
As with switching frequency, the inductance used is a tradeoff between size and cost. Larger inductance means
lower input ripple current, however because the inductor is connected to the output during the off-time only, there
is a limit to the reduction in output ripple voltage. Lower inductance results in smaller, less expensive magnetics.
An inductance that gives a ripple current of 30% to 50% of IL is a good starting point for a CCM boost converter.
Minimum inductance must be calculated with Equation 8 at the extremes of input voltage to find the operating
condition with the highest requirement.
L1
V IN u D
f SW u 'i L
(8)
By calculating in terms of amperes, volts, and megahertz, the inductance value comes out in micro henries.
To ensure that the boost regulator operates in CCM, a second equation is required, and must also be evaluated
with Equation 9 at the corners of input voltage to find the minimum inductance required.
L2
D(1 D) u V IN
I O u f SW
(9)
By calculating in terms of volts, amps, and megahertz, the inductance value comes out in µH.
For this design, ΔiL is set to 40% of the maximum IL. Duty cycle is evaluated first at VIN(MIN) and at VIN(MAX).
Second, the average inductor current is evaluated at the two input voltages. Third, the inductor ripple current is
determined. Finally, the inductance can be calculated, and a standard inductor value selected that meets all the
criteria.
1. Inductance for Minimum Input Voltage (Equation 10, Equation 11, and Equation 12)
DVIN(MIN) = (40 – 9 + 0.5) / (40 + 0.5) = 78% IL-VIN(MIN) = 0.5 / (1 – 0.78) = 2.3 A ΔiL = 0.4 × 2.3 A = 0.92 A
L1
L2
VIN (MIN)
VIN (MIN)
9 u 0.78
0.5 u 0.92
(10)
15.3 PH
(11)
0.78 u 0.22 u 9
0.5 u 0.5
6.2 PH
(12)
2. Inductance for Maximum Input Voltage (Equation 13, Equation 14, and Equation 15)
DVIN(MAX) = (40 – 16 + 0.5) / (40 + 0.5) = 60% IL-VIN(MIAX) = 0.5 / (1 – 0.6) = 1.25 A ΔiL = 0.4 × 1.25 A = 0.5 A
L1
L2
VIN ( MIN )
VIN (MIN)
16 u 0.6
0.5 u 0.5
(13)
38.4 PH
0.6 u 0.4 u 16
0.5 u 0.5
(14)
15.4 PH
(15)
Maximum average inductor current occurs at VIN(MIN), and the corresponding inductor ripple current is 0.92 AP-P.
Selecting an inductance that exceeds the ripple current requirement at VIN(MIN) and the requirement to stay in
CCM for VIN(MAX) provides a tradeoff that allows smaller magnetics at the cost of higher ripple current at
maximum input voltage. For this example, a 33-µH inductor satisfies these requirements.
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The second criterion for selecting an inductor is the peak current carrying capability. This is the level above
which the inductor saturates. In saturation, the inductance can drop off severely, resulting in higher peak current
that can overheat the inductor or push the converter into current limit. In a boost converter, peak current, IPK, is
equal to the maximum average inductor current plus one half of the ripple current. First, the current ripple must
be determined under the conditions that give maximum average inductor current with Equation 16.
'i L
V IN x D
f SW u L
(16)
Maximum average inductor current occurs at VIN(MIN). Using the selected inductance of 33 µH yields Equation 17.
ΔiL = (9 × 0.78) / (0.5 × 33) = 425 mAP-P
(17)
The highest peak inductor current over all operating conditions is therefore Equation 18.
IPK = IL + 0.5 × ΔiL = 2.3 + 0.213 = 2.51 A
(18)
Hence, an inductor must be selected that has a peak current rating greater than 2.5 A and an average current
rating greater than 2.3 A. One possibility is an off-the-shelf 33 µH ±20% inductor that can handle a peak current
of 3.2 A and an average current of 3.4 A. Finally, the inductor current ripple is recalculated with Equation 19 at
the maximum input voltage.
ΔiL-VIN(MAX) = (16 × 0.6) / (0.5 × 33) = 0.58 AP-P
(19)
8.2.2.5 Output Capacitor
The output capacitor in a boost regulator supplies current to the load during the MOSFET on-time and also filters
the AC portion of the load current during the off-time. This capacitor determines the steady-state output voltage
ripple, ΔVO, a critical parameter for all voltage regulators. Output capacitors are selected based on their
capacitance, CO, their equivalent series resistance (ESR), and their RMS or AC current rating.
The magnitude of ΔVO is comprised of three parts, and in steady-state, the ripple voltage during the on-time is
equal to the ripple voltage during the off-time. For simplicity, the analysis is performed for the MOSFET turning
off (off-time) only. The first part of the ripple voltage is the surge created as the output diode D1 turns on. At this
point, inductor and diode current are at peak value, and the ripple voltage increase can be calculated with
Equation 20.
ΔVO1 = IPK × ESR
(20)
The second portion of the ripple voltage is the increase due to the charging of CO through the output diode. This
portion can be approximated with Equation 21.
ΔVO2 = (IO / CO) × (D / ƒSW)
(21)
The final portion of the ripple voltage is a decrease due to the flow of the diode and inductor current through the
ESR of the output capacitor. This decrease can be calculated with Equation 22.
ΔVO3 = ΔiL × ESR
(22)
The total change in output voltage is Equation 23.
ΔVO = ΔVO1 + ΔVO2 – ΔVO3
(23)
The combination of two positive terms and one negative term may yield an output voltage ripple with a net rise or
a net fall during the converter off-time. The ESR of the output capacitor or capacitors has a strong influence on
the slope and direction of ΔVO. Capacitors with high ESR such as tantalum and aluminum electrolytic create an
output voltage ripple that is dominated by ΔVO1 and ΔVO3, with a shape shown in Figure 15. Ceramic capacitors,
in contrast, have very low ESR and lower capacitance. The shape of the output ripple voltage is dominated by
ΔVO2, with a shape shown in Figure 16.
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ÂvO
VO
ÂvO
VO
ID
ID
Figure 15. ΔVO Using High-ESR Capacitors
Figure 16. ΔVO Using Low-ESR Capacitors
For this example, the small size and high temperature rating of ceramic capacitors make them a good choice.
The output ripple voltage waveform of Figure 16 is assumed, and the capacitance is selected first. The desired
ΔVO is ±2% of 40 V, or 0.8 VP-P. Beginning with the calculation for ΔVO2, the required minimum capacitance is in
Equation 24.
CO-MIN = (IO / ΔVO) × (DMAX / fSW) CO-MIN = (0.5 / 0.8) × (0.77 / 5 x 105) = 0.96 µF
(24)
The next higher standard 20% capacitor value is 1 µF, however, to provide margin for component tolerance and
load transients, two capacitors rated 4.7 µF each (CO= 9.4 µF) are used. Ceramic capacitors rated 4.7 µF ±20%
are available from many manufacturers. The minimum quality dielectric that is suitable for switching power supply
output capacitors is X5R, while X7R (or better) is preferred. Pay careful attention to the DC voltage rating and
case size, as ceramic capacitors can lose 60% or more of their rated capacitance at the maximum DC voltage.
This is the reason that ceramic capacitors are often de-rated to 50% of their capacitance at their working voltage.
The output capacitors for this example has a 100-V rating in a 2220 case size.
The typical ESR of the selected capacitors is 3 mΩ each, and in parallel is approximately 1.5 mΩ. The worstcase value for ΔVO1 occurs during the peak current at minimum input voltage in Equation 25.
ΔVO1 = 2.5 × 0.0015 = 4 mV
(25)
The worst-case capacitor charging ripple occurs at maximum duty cycle in Equation 26.
ΔVO2 = (0.5 / 9.4 × 10–6) × (0.77 / 5 × 105) = 82 mV
(26)
Finally, the worst-case value for ΔVO3 occurs when inductor ripple current is highest, at maximum input voltage in
Equation 27.
ΔVO3 = 0.58 × 0.0015 = 1 mV (negligible)
(27)
The output voltage ripple can be estimated by summing the three terms in Equation 28.
ΔVO = 4 mV + 82 mV - 1 mV = 85 mV
(28)
The RMS current through the output capacitor or capacitors can be estimated using the following, worst-case
equation in Equation 29.
IO
RMS
1.13 u I L u D u (1 D )
(29)
The highest RMS current occurs at minimum input voltage. For this example, the maximum output capacitor
RMS current is calculated with Equation 30.
IO-RMS(MAX) = 1.13 × 2.3 × (0.78 x 0.22)0.5 = 1.08 ARMS
(30)
These 2220 case size devices are capable of sustaining RMS currents of over 3 A each, making them more than
adequate for this application.
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8.2.2.6 VCC Decoupling Capacitor
The VCC pin must be decoupled with a ceramic capacitor placed as close as possible to the VCC and GND pins
of the LM5022. The decoupling capacitor must have a minimum X5R or X7R type dielectric to ensure that the
capacitance remains stable over voltage and temperature, and be rated to a minimum of 470 nF. One good
choice is a 1-µF device with X7R dielectric and 1206 case size rated to 25 V.
8.2.2.7 Input Capacitor
The input capacitors to a boost regulator control the input voltage ripple (ΔVIN) hold up the input voltage during
load transients and prevent impedance mismatch (also called power supply interaction) between the LM5022 and
the inductance of the input leads. Selection of input capacitors is based on their capacitance, ESR, and RMS
current rating. The minimum value of ESR can be selected based on the maximum output current transient,
ISTEP, using Equation 31.
ESR MIN
1 D u 'V IN
2 x I STEP
(31)
For this example, the maximum load step is equal to the load current or 0.5 A. The maximum permissable ΔVIN
during load transients is 4%P-P. ΔVIN and duty cycle are taken at minimum input voltage to give the worst-case
value in Equation 32.
ESRMIN = [(1 – 0.77) × 0.36] / (2 × 0.5) = 83 mΩ
(32)
The minimum input capacitance can be selected based on ΔVIN, based on the drop in VIN during a load transient,
or based on prevention of power supply interaction. In general, the requirement for greatest capacitance comes
from the power supply interaction. The inductance and resistance of the input source must be estimated, and if
this information is not available, they can be assumed to be 1 µH and 0.1 Ω, respectively. Minimum capacitance
is then estimated with Equation 33.
C MIN
2 u L S u VO u I O
VIN 2 u RS
(33)
As with ESR, the worst-case, highest minimum capacitance calculation comes at the minimum input voltage.
Using the default estimates for LS and RS, minimum capacitance is calculated with Equation 34.
C MIN
2 u 1Pu 40 u 0.5
9 2 u 0.1
4.9 PF
(34)
The next highest standard 20% capacitor value is 6.8 µF, but because the actual input source impedance and
resistance are not known, two 4.7-µF capacitors is used. In general, doubling the calculated value of input
capacitance provides a good safety margin. The final calculation is for the RMS current. For boost converters
operating in CCM, this can be estimated with Equation 35.
IRMS = 0.29 × ΔiL(MAX)
(35)
From the inductor section, maximum inductor ripple current is 0.58 A, hence the input capacitor or capacitors
must be rated to handle 0.29 × 0.58 = 170 mARMS.
The input capacitors can be ceramic, tantalum, aluminum, or almost any type, however, the low capacitance
requirement makes ceramic capacitors particularly attractive. As with the output capacitors, the minimum quality
dielectric used must be X5R, with X7R or better preferred. The voltage rating for input capacitors requirement
does not need to be as conservative as the output capacitors, as the requirement for capacitance decreases as
input voltage increases. For this example, the capacitor selected is 4.7 µF ±20%, rated to 50 V in the 1812 case
size. The RMS current rating of these capacitors is over 2 A each, more than enough for this application.
8.2.2.8 Current Sense Filter
Parasitic circuit capacitance, inductance, and gate drive current create a spike in the current sense voltage at the
point where Q1 turns on. To prevent this spike from terminating the on-time prematurely, every circuit must have
a low-pass filter that consists of CCS and RS1, shown in Figure 14. The time constant of this filter must be long
enough to reduce the parasitic spike without significantly affecting the shape of the actual current sense voltage.
The recommended range for RS1 is between 10 Ω and 500 Ω, and the recommended range for CCS is between
100 pF and 2.2 nF. For this example, the values of RS1 and CCS is 100 Ω and 1 nF, respectively.
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8.2.2.9 RSNS, RS2, and Current Limit
The current sensing resistor, RSNS, is used for steady-state regulation of the inductor current and to sense
overcurrent conditions. The slope compensation resistor is used to ensure control loop stability, and both
resistors affect the current limit threshold. The RSNS value selected must be low enough to keep the power
dissipation to a minimum, yet high enough to provide good signal-to-noise ratio for the current sensing circuitry.
RSNS and RS2 must be set so that the current limit comparator, with a threshold of 0.5 V, trips before the sensed
current exceeds the peak current rating of the inductor, without limiting the output power in steady state.
For this example, the peak current at VIN(MIN) is 2.5 A, while the inductor itself is rated to 3.2 A. The threshold for
current limit, ILIM, is set slightly between these two values to account for tolerance of the circuit components, at a
level of 3 A. The required resistor calculation must take both the switch current through RSNS and the
compensation ramp current flowing through the internal 2-kΩ RS1 and RS2 resistors into account. RSNS must be
selected first because it is a power resistor with more limited selection. Equation 36 and Equation 37 must be
evaluated at VIN(MIN) when duty cycle is highest.
R SNS
L u f SW u V CL
VO
V IN u 3 u D L u f SW uI LIM
(36)
R SNS
40
33 u 0.5 u 0.5
9 u 3 u 0.78 33 u 0.5 u 3
where
•
•
L is in µH
fSW in MHz
(37)
The closest 5% value is 100 mΩ. Power dissipation in RSNS can be estimated by calculating the average current.
The worst-case average current through RSNS occurs at minimum input voltage/maximum duty cycle and can be
calculated with Equation 38 and Equation 39.
PCS
2
ª§ I
º
·
«¨ O ¸ u R SNS » u D
«¨© 1 D ¸¹
»
¬
¼
(38)
(39)
PCS = [(0.5 / 0.22)2 × 0.1] × 0.78 = 0.4 W
For this example, a 0.1 Ω ±1%, thick-film chip resistor in a 1210 case size rated to 0.5 W is used.
With RSNS selected, RS2 can be determined using Equation 40 and Equation 41.
R S2
R S2
V CL
I ILIM u R SNS
2000
45P u D
0.5 3 u 0.1
45P u 0.78
R S1
(40)
2000
100
3598:
(41)
The closest 1% tolerance value is 3.57 kΩ.
8.2.2.10 Control Loop Compensation
The LM5022 uses peak current-mode PWM control to correct changes in output voltage due to line and load
transients. Peak current-mode provides inherent cycle-by-cycle current limiting, improved line transient response,
and easier control loop compensation.
The control loop is comprised of two parts. The first is the power stage, which consists of the pulse width
modulator, output filter, and the load. The second part is the error amplifier, which is an op-amp configured as an
inverting amplifier. Figure 17 shows the regulator control loop components.
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L
+
D
VIN
CO
RO
+
-
RSNS
RC
+
RFB2
R1
C2
C1
+
VREF
RFB1
+
-
Figure 17. Power Stage and Error Amplifier
One popular method for selecting the compensation components is to create Bode plots of gain and phase for
the power stage and error amplifier. Combined, they make the overall bandwidth and phase margin of the
regulator easy to determine. Software tools such as Excel, MathCAD, and Matlab are useful for observing how
changes in compensation or the power stage affect system gain and phase.
The power stage in a CCM peak current mode boost converter consists of the DC gain, APS, a single lowfrequency pole, ƒLFP, the ESR zero, ƒZESR, a right-half plane zero, ƒRHP, and a double pole resulting from the
sampling of the peak current. The power stage transfer function (also called the control-to-output transfer
function) can be written with Equation 42, Equation 43, and Equation 44.
æ
s öæ
s ö
ç1 +
÷ ç1 ÷
wZESR ø è
wRHP ø
è
GPS = APS ´
æ
s öæ
s
s2 ö
÷
+
ç1 +
÷ çç 1 +
Qn ´ wn w2n ø÷
wLEP ø è
è
where
•
A PS
22
the DC gain is defined as:
(42)
(1 D) u R O
2 u R SNS
where
(43)
RO = VO / IO
(44)
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The system ESR zero is calculated with Equation 45.
ZZESR
1
RCu CO
(45)
The low-frequency pole is calculated with Equation 46.
1
Z LEP
0.5 u R O
ESR u C O
(46)
The right-half plane zero is calculated with Equation 47.
§ V IN ·
ROu ¨
¸
¨ VO ¸
©
¹
L
Z RHP
2
(47)
The sampling double-pole quality factor is calculated with Equation 48.
Qn
1
ª
S« D
«¬
0.5
(1 D)
Se º
»
S n »¼
(48)
The sampling double corner frequency is calculated with Equation 49.
ωn = π × fSW
(49)
The natural inductor current slope is calculated with Equation 50.
Sn = RSNS × VIN / L
(50)
The external ramp slope is calculated with Equation 51.
Se = 45 µA × (2000 + RS1 + RS2)] × ƒSW
(51)
In Equation 43, DC gain is highest when input voltage and output current are at the maximum. In this example,
those conditions are VIN = 16 V and IO = 500 mA.
DC gain is 44 dB. The low-frequency pole, fP = ωP / 2π, is at 423 Hz, the ESR zero, fZ = ωZ / 2π, is at 5.6 MHz,
and the right-half plane zero, ƒRHP = ωRHP / 2π, is at 61 kHz. The sampling double-pole occurs at one-half of the
switching frequency. Proper selection of slope compensation (through RS2) is most evident the sampling double
pole. A well-selected RS2 value eliminates peaking in the gain and reduces the rate of change of the phase lag.
Gain and phase plots for the power stage are shown in Figure 18 and Figure 19.
60
180
45
120
POWER STAGE PHASE (°)
POWER STAGE GAIN (dB)
SPACE
30
15
0
-15
-30
100
1k
10k
100k
60
0
-60
-120
-180
100
1M
FREQUENCY (Hz)
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 18. Power Stage Gain and Phase
Figure 19. Power Stage Gain and Phase
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The single pole causes a rolloff in the gain of –20 dB/decade at lower frequency. The combination of the RHP
zero and sampling double pole maintain the slope out to beyond the switching frequency. The phase tends
towards –90° at lower frequency but then increases to –180° and beyond from the RHP zero and the sampling
double pole. The effect of the ESR zero is not seen because its frequency is several decades above the
switching frequency. The combination of increasing gain and decreasing phase makes converters with RHP
zeroes difficult to compensate. Setting the overall control loop bandwidth to 1/3 to 1/10 of the RHP zero
frequency minimizes these negative effects, but requires a compromise in the control loop bandwidth. If this loop
were left uncompensated, the bandwidth would be 89 kHz and the phase margin –54°. The converter would
oscillate, and therefore is compensated using the error amplifier and a few passive components.
The transfer function of the compensation block (GEA) can be derived by treating the error amplifier as an
inverting op amp with input impedance ZI and feedback impedance ZF. The majority of applications require a
Type II, or two-pole one-zero amplifier, shown in Figure 17. The LaPlace domain transfer function for this Type II
network is given by Equation 52.
G EA
ZF
ZI
1
R FB2 (C1
C2)
u
s u R1 u C2 1
§ s u R1 u C1 u C2
·
s¨
1¸
C1 C2
©
¹
(52)
Many techniques exist for selecting the compensation component values. The following method is based upon
setting the mid-band gain of the error amplifier transfer function first and then positioning the compensation zero
and pole:
1. Determine the desired control loop bandwidth: The control loop bandwidth (ƒ0dB) is the point at which the
total control loop gain (H = GPS × GEA) is equal to 0 dB. For this example, a low bandwidth of 10 kHz, or
approximately 1/6th of the RHP zero frequency, is chosen because of the wide variation in input voltage.
2. Determine the gain of the power stage at ƒ0dB: This value, A, can be read graphically from the gain plot of
GPS or calculated by replacing the ‘s’ terms in GPS with ‘2 πf0dB’. For this example, the gain at 10 kHz is
approximately 16 dB.
3. Calculate the negative of A and convert it to a linear gain: By setting the mid-band gain of the error amplifier
to the negative of the power stage gain at f0dB, the control loop gain equals 0 dB at that frequency. For this
example, –16 dB = 0.15 V/V.
4. Select the resistance of the top feedback divider resistor RFB2: This value is arbitrary, however, selecting a
resistance between 10 kΩ and 100 kΩ leads to practical values of R1, C1, and C2. For this example, RFB2 =
20 kΩ 1%.
5. Set Equation 55:
R1 = A × RFB2
(53)
For this example: R1 = 0.15 × 20000 = 3 kΩ
6. Select a frequency for the compensation zero, ƒZ1: The suggested placement for this zero is at the lowfrequency pole of the power stage, ƒLFP = ωLFP / 2π. For this example, ƒZ1 = ƒLFP = 423 Hz.
7. Set Equation 54.
C2
1
:
2S u R1 u fZ1
(54)
For this example, C2 = 125 nF
8. Select a frequency for the compensation pole, ƒP1: The suggested placement for this pole is at one-fifth of
the switching frequency. For this example, ƒP1 = 100 kHz
9. Set Equation 55.
C1 =
C2
2Œ×C2×R1×f P1 -1
(55)
For this example, C1 = 530 pF
24
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10. Plug the closest 1% tolerance values for RFB2 and R1, then the closest 10% values for C1 and C2 into GEA
and model the error amp: The open-loop gain and bandwidth of the internal error amplifier of the LM5022 are
75 dB and 4 MHz, respectively. Their effect on GEA can be modeled using Equation 56:
OPG
2S u GBW
2S u GBW
s
A DC
(56)
ADC is a linear gain, the linear equivalent of 75 dB is approximately 5600 V/V. C1 = 560 pF 10%, C2 = 120
nF 10%, R1 = 3.01 kΩ 1%
11. Plot or evaluate the actual error amplifier transfer function:
G EA
G EA u OPG
ACTUAL
1
G EA u OPG
(57)
60
OVERALL LOOP GAIN (dB)
40
20
0
-20
-40
-60
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 20. Overall Loop Gain and Phase
OVERALL LOOP PHASE (°)
180
120
60
0
-60
-120
-180
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 21. Overall Loop Gain and Phase
12. Plot or evaluate the complete control loop transfer function: The complete control loop transfer function is
obtained by multiplying the power stage and error amplifier functions together. The bandwidth and phase
margin can then be read graphically or evaluated numerically. The bandwidth of this example circuit at VIN =
16 V is 10.5 kHz with a phase margin of 66°.
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13. Re-evaluate at the corners of input voltage and output current: Boost converters exhibit significant change in
their loop response when VIN and IO change. With the compensation fixed, the total control loop gain and
phase must be checked to ensure a minimum phase margin of 45° over both line and load.
8.2.2.11 Efficiency Calculations
A reasonable estimation for the efficiency of a boost regulator controlled by the LM5022 can be obtained by
adding together the loss is each current carrying element and using Equation 58.
K
PO
PO
P total
loss
(58)
The following shows an efficiency calculation to complement the circuit design. Output power for this circuit is 40
V × 0.5 A = 20 W. Input voltage is assumed to be 13.8 V, and the calculations used assume that the converter
runs in CCM. Duty cycle for VIN = 13.8 V is 66%, and the average inductor current is 1.5 A.
8.2.2.11.1 Chip Operating Loss
This term accounts for the current drawn at the VIN pin. This current, IIN, drives the logic circuitry and the power
MOSFETs. The gate driving loss term from MOSFET is included in the chip operating loss. For the LM5022, IIN is
equal to the steady-state operating current, ICC, plus the MOSFET driving current, IGC. Power is lost as this
current passes through the internal linear regulator of the LM5022 in Equation 59.
IGC = QG × ƒSW IGC = 27 nC × 500 kHz = 13.5 mA
(59)
ICC is typically 3.5 mA (taken from the Electrical Characteristics). Chip operating loss is then calculated with
Equation 60.
PQ = VIN × (IQ + IGC) PQ = 13.8 × (3.5 m + 13.5m) = 235 mW
(60)
8.2.2.11.2 MOSFET Switching Loss
PSW = 0.5 × VIN × IL × (tR + tF) × fSW PSW = 0.5 × 13.8 × 1.5 × (10 ns + 12 ns) × 5 × 105 = 114 mW
(61)
8.2.2.11.3 MOSFET and RSNS Conduction Loss
PC = D × (IL2 × (RDSON × 1.3 + RSNS)) PC = 0.66 × (1.52 × (0.029 + 0.1)) = 192 mW
(62)
8.2.2.11.4 Output Diode Loss
The average output diode current is equal to IO or 0.5 A. The estimated forward drop (VD) is 0.5 V. The output
diode loss is Equation 63.
PD1 = IO × VD PD1 = 0.5 × 0.5 = 0.25 W
(63)
8.2.2.11.5 Input Capacitor Loss
This term represents the loss as input ripple current passes through the ESR of the input capacitor bank. In this
equation, ‘n’ is the number of capacitors in parallel. The 4.7-µF input capacitors selected have a combined ESR
of approximately 1.5 mΩ, and ΔiL for a 13.8-V input is 0.55 A in Equation 64 and Equation 65.
PCIN
I IN
2
RMS
u ESR
n
IIN-RMS = 0.29 × ΔiL = 0.29 × 0.55 = 0.16 A PCIN = [0.162 × 0.0015] / 2 = 0.02 mW (negligible)
(64)
(65)
8.2.2.11.6 Output Capacitor Loss
This term is calculated using the same method as the input capacitor loss, substituting the output capacitor RMS
current for VIN = 13.8 V. The combined ESR of the output capacitors is also approximately 1.5 mΩ in
Equation 66.
IO-RMS = 1.13 × 1.5 × (0.66 x 0.34)0.5 = 0.8 A PCO = [0.8 × 0.0015] / 2 = 0.6 mW
26
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8.2.2.11.7 Boost Inductor Loss
The typical DCR of the selected inductor is 40 mΩ in Equation 67.
PDCR = IL2 × DCR PDCR = 1.52 × 0.04 = 90 mW
(67)
Core loss in the inductor is estimated to be equal to the DCR loss, adding an additional 90 mW to the total
inductor loss.
8.2.2.11.8 Total Loss
PLOSS = Sum of All Loss Terms = 972 mW
(68)
8.2.2.11.9 Efficiency
η = 20 / (20 + 0.972) = 95%
(69)
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8.2.3 Application Curves
10V/DIV
VO
SW
10V/DIV
1 és/DIV
VIN = 9 V, IO = 0.5 A
Figure 23. Switch Node Voltage
Figure 22. Efficiency
10V/DIV
VO
VO
50 mV/DIV
SW
10V/DIV
1 és/DIV
1 és/DIV
VIN = 16 V, IO = 0.5 A
VIN = 9 V, IO = 0.5 A
Figure 24. Switch Node Voltage
Figure 25. Output Voltage Ripple AC Coupled
200 mA/DIV
IO
VO
VO
2V/DIV
50 mV/DIV
400 és/DIV
1 és/DIV
VIN = 9 V, IO = 50 mA to 0.5
A
VIN = 16 V, IO = 0.5 A
Figure 26. Output Voltage Ripple AC Coupled
28
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200 mA/DIV
IO
VO
1V/DIV
1 ms/DIV
VIN = 16 V, IO = 50 mA to 0.5 A
Figure 28. Load Transient Response
9 Power Supply Recommendations
The LM5022 is a power management device. The power supply for the device can be any DC voltage source
within the specified input range.
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10 Layout
10.1 Layout Guidelines
To produce an optimal power solution with the LM5022, good layout and design of the PCB are as critical as
component selection. The following are the several guidelines to create a good layout of the PCB, as based on
Figure 14:
1. Using a low-ESR ceramic capacitor, place CINX as close as possible to the VIN and GND pins of the
LM5022.
2. Using a low-ESR ceramic capacitor, place COX close to the load as possible of the LM5022.
3. Using a low-ESR ceramic capacitor, place CF close to the VCC and GND pins of the LM5022.
4. Minimize the loop area formed by the output capacitor connections (Co1, Co2) by D1 and Rsns. Make sure the
cathode of D1 and Rsns are positioned next to each other, and place Co1(+) and Co1(–) close to D1 cathode
and Rsns(–) respectively.
5. Rsns(+) must be connected to the CS pin with a separate trace made as short as possible. This trace must be
routed away from the inductor and the switch node (where D1, Q1, and L1 connect).
6. Minimize the trace length to the FB pin by positioning RFB1 and RFB2 close to the LM5022.
7. Route the VOUT sense path away from noisy node and connect it as close as possible to the positive side of
COX.
10.1.1 Filter Capacitors
The low-value ceramic filter capacitors are most effective when the inductance of the current loops that they filter
is minimized. Place CINX as close as possible to the VIN and GND pins of the LM5022. Place COX close to the
load, and CF next to the VCC and GND pins of the LM5022.
10.1.2 Sense Lines
The top of RSNS must be connected to the CS pin with a separate trace made as short as possible. Route this
trace away from the inductor and the switch node (where D1, Q1, and L1 connect). For the voltage loop, keep
RFB1/2 close to the LM5022 and run a trace from as close as possible to the positive side of COX to RFB2. As with
the CS line, the FB line must be routed away from the inductor and the switch node. These measures minimize
the length of high impedance lines and reduce noise pickup.
10.1.3 Compact Layout
Parasitic inductance can be reduced by keeping the power path components close together. As described in the
Layout Guidelines, keep the high slew-rate current loops as tight as possible. Short, thick traces or copper pours
(shapes) are best.
The switch node must be just large enough to connect all the components together without excessive heating
from the current it carries. The LM5022 (boost converter) operates in two distinct cycles whose high current
paths are shown in Figure 29.
+
-
Figure 29. Boost Converter Current Loops
30
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Layout Guidelines (continued)
The dark grey, inner loops represents the high current paths during the MOSFET on-time. The light grey, outer
loop represents the high current path during the off-time.
10.1.4 Ground Plane and Shape Routing
The diagram of Figure 29 is also useful for analyzing the flow of continuous current versus the flow of pulsating
currents. The circuit paths with current flow during both the on-time and off-time are considered to be continuous
current, while those that carry current during the on-time or off-time only are pulsating currents. Preference in
routing must be given to the pulsating current paths, as these are the portions of the circuit most likely to emit
EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as
any other circuit path. The continuous current paths on the ground net can be routed on the system ground plane
with less risk of injecting noise into other circuits. The path between the input source, input capacitor and the
MOSFET and the path between the output capacitor and the load are examples of continuous current paths. In
contrast, the path between the grounded side of the power switch and the negative output capacitor terminal
carries a large pulsating current. This path must be routed with a short, thick shape, preferably on the component
side of the PCB. Multiple vias in parallel must be used right at the negative pads of the input and output
capacitors to connect the component side shapes to the ground plane. Vias must not be placed directly at the
grounded side of the MOSFET (or RSNS) as they tend to inject noise into the ground plane. A second pulsating
current loop that is often ignored but must be kept small is the gate drive loop formed by the OUT and VCC pins,
Q1, RSNS, and capacitor CF.
10.2 Layout Examples
Figure 30. Top Layer and Top Overlay
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Layout Examples (continued)
Figure 31. Bottom Layer
32
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
WEBENCH® software uses an iterative design procedure and accesses comprehensive databases of
components. For more details, go to www.ti.com/webench.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
AN-1557 LM5022 Evaluation Board (SNVA203)
11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.4 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
11.5 Trademarks
E2E is a trademark of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.6 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
www.ti.com
5-Jul-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
LM5022MM
NRND
VSSOP
DGS
10
1000
Non-RoHS
& Green
Call TI
Level-1-260C-UNLIM
-40 to 125
5022
LM5022MM/NOPB
ACTIVE
VSSOP
DGS
10
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
5022
Samples
LM5022MME/NOPB
ACTIVE
VSSOP
DGS
10
250
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
5022
Samples
LM5022MMX/NOPB
ACTIVE
VSSOP
DGS
10
3500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
5022
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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