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LM5022-Q1
SNVSAG9 – MARCH 2016
LM5022-Q1 2.2MHz, 60 V Low-Side Controller For Boost and SEPIC
1 Features
3 Description
•
The LM5022-Q1 is a high voltage low-side N-channel
MOSFET controller ideal for use in boost and SEPIC
regulators. It contains all of the features needed to
implement single-ended primary topologies. Output
voltage regulation is based on current-mode control,
which eases the design of loop compensation while
providing inherent input voltage feed-forward. The
LM5022-Q1 includes a start-up regulator that
operates over a wide input range of 6 V to 60 V. The
PWM controller is designed for high-speed capability
including an oscillator frequency range up to 2.2 MHz
and total propagation delays less than 100 ns.
Additional features include an error amplifier,
precision reference, line undervoltage lockout, cycleby-cycle current limit, slope compensation, soft-start,
external synchronization capability, and thermal
shutdown. The LM5022-Q1 is available in the 10-pin
VSSOP package.
1
•
•
•
•
•
•
•
•
•
•
•
•
AEC-Q100 Grade 1 Qualified with the following
results:
– Device Temperature Grade 1: -40°C to 125°C
Ambient Operating Temperature Range
– Device HBM ESD Classification Level 2
– Device CDM ESD Classification Level C5
Internal 60-V Start-Up Regulator
1-A Peak MOSFET Gate Driver
VIN Range: 6 V to 60 V (operate down to 3 V after
startup)
Duty Cycle Limit of 90%
Programmable UVLO with Hysteresis
Cycle-by-Cycle Current Limit
Single Resistor Oscillator Frequency Set
Adjustable Switching Frequency to 2.2MHz
External Clock Synchronization
Slope Compensation
Adjustable Soft Start
10-Pin VSSOP Package
Device Information(1)
PART NUMBER
LM5022-Q1
PACKAGE
BODY SIZE (NOM)
VSSOP (10)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
2 Applications
•
•
Boost Converter
SEPIC Converter
Typical Application
VIN
L1
VO
D1
Q1
CIN
CO
RS1
VIN
OUT
RUV2
RT
UVLO
CSS
RUV1
SS
LM5022
RT
COMP
RSNS
CS
CCS
GND
CF
VCC
RFB2
FB
R1
C2
RFB1
C1
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM5022-Q1
SNVSAG9 – MARCH 2016
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Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
7
Absolute Maximum Ratings ......................................
ESD Ratings: LM5022-Q1 ........................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics ..........................................
Typical Characteristics ..............................................
7.4 Device Functional Modes........................................ 12
8
Application and Implementation ........................ 14
8.1 Application Information............................................ 14
8.2 Typical Application ................................................. 14
9 Power Supply Recommendations...................... 28
10 Layout................................................................... 28
10.1 Layout Guidelines ................................................. 28
10.2 Layout Example .................................................... 30
11 Device and Documentation Support ................. 31
11.1
11.2
11.3
11.4
11.5
Detailed Description .............................................. 9
7.1 Overview ................................................................... 9
7.2 Functional Block Diagram ......................................... 9
7.3 Feature Description................................................. 10
Device Support......................................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
31
31
31
31
31
12 Mechanical, Packaging, and Orderable
Information ........................................................... 31
4 Revision History
2
DATE
REVISION
NOTES
March 2016
*
Initial release.
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5 Pin Configuration and Functions
DGS Package
10-Pin VSSOP
Top View
1
2
3
4
5
VIN
SS
FB
RT
COMP
CS
VCC
UVLO
OUT
GND
10
9
8
7
6
Pin Functions
PIN
NO.
NAME
1
VIN
2
TYPE
DESCRIPTION
APPLICATION INFORMATION
I
Source input voltage
Input to the start-up regulator. Operates from 6 V to 60
V.
FB
I
Feedback pin
Inverting input to the internal voltage error amplifier.
The non-inverting input of the error amplifier connects
to a 1.25-V reference.
3
COMP
I/O
Error amplifier output and PWM
comparator input
The control loop compensation components connect
between this pin and the FB pin.
4
VCC
O
Output of the internal, high voltage linear
regulator.
This pin should be bypassed to the GND pin with a
ceramic capacitor.
5
OUT
O
Output of MOSFET gate driver
Connect this pin to the gate of the external MOSFET.
The gate driver has a 1-A peak current capability.
6
GND
-
System ground
7
UVLO
I
Input undervoltage lockout
8
CS
I
Current sense input
An external resistor connected from this pin to GND
sets the oscillator frequency. This pin can also accept
an AC-coupled input for synchronization from an
external clock.
An external capacitor placed from this pin to ground
will be charged by a 10-µA current source, creating a
ramp voltage to control the regulator start-up.
9
RT/SYNC
I
Oscillator frequency adjust pin and
synchronization input
10
SS
I
Soft-start pin
Set the start-up and shutdown levels by connecting
this pin to the input voltage through a resistor divider.
A 20-µA current source provides hysteresis.
Input for the switch current used for current mode
control and for current limiting.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2)
MIN
MAX
UNIT
VIN to GND
–0.3
65
V
VCC to GND
–0.3
16
V
RT/SYNC to GND
–0.3
5.5
V
OUT to GND
–1.5V for < 100 ns
All other pins to GND
–0.3
Power dissipation
7
Junction temperature (3)
Soldering information
150
°C
Vapor phase (60 sec.)
215
°C
Infrared (15 sec.)
220
°C
150
°C
Storage temperature, Tstg
(1)
(2)
(3)
V
Internally limited
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
High junction temperatures degrade operating lifetimes. Operating lifetime is de-rated for junction temperatures greater than 125°C.
6.2 ESD Ratings: LM5022-Q1
V(ESD)
(1)
(2)
VALUE
UNIT
Human body model (HBM), per AEC Q100-002 (1)
±2000
V
Charged device model (CDM), per AEC Q100-011 (2)
±750
V
AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification. This is the passing
level per ANSI/ESDA/JEDEC JS-001. JEDEC document JEP155 states that 500 V HBM allows safe manufacturing with a standard ESD
control process
Level listed above is the passing level per EIA-JEDEC JESD22-C101. JEDEC document JEP157 states that 250 V CDM allows safe
manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted) (1)
MIN
Supply voltage
NOM
MAX
6
60
UNIT
V
External voltage at VCC
7.5
14
V
Junction temperature
–40
125
°C
(1)
Operating Ratings are conditions under the device is intended to be functional. For specifications and test conditions, see Electrical
Characteristics
6.4 Thermal Information
LM5022-Q1
THERMAL METRIC
(1)
DGS (VSSOP)
UNIT
10 PINS
RθJA
Junction-to-ambient thermal resistance
RθJC(top)
Junction-to-case (top) thermal resistance
RθJB
Junction-to-board thermal resistance
ψJT
Junction-to-top characterization parameter
5.7
°C/W
ψJB
Junction-to-board characterization parameter
80
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
N/A
°C/W
(1)
4
161.5
°C/W
56
°C/W
81.3
°C/W
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
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6.5 Electrical Characteristics
Typical limits apply for TJ = 25°C and are provided for reference purposes only; minimum and maximum limits apply over the
junction temperature (TJ) range of –40°C to +125°C. VIN = 24 V and RT = 27.4 kΩ, unless otherwise indicated. (1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
1.225
1.250
1.275
V
6.6
7
7.4
3.5
4
SYSTEM PARAMETERS
VFB
FB Pin Voltage
START-UP REGULATOR
VCC Regulation
10 V ≤ VIN ≤ 60 V, ICC = 1 mA
VCC Regulation
6 V ≤ VIN < 10 V, VCC Pin Open Circuit
ICC
Supply Current
OUT Pin Capacitance = 0
VCC = 10 V
ICC-LIM
VCC Current Limit
VCC = 0 V, ( (3),
VIN - VCC
Dropout Voltage Across Bypass
Switch
ICC = 0 mA, ƒSW < 200 kHz
6 V ≤ VIN ≤ 8.5 V
VBYP-HI
Bypass Switch Turn-off Threshold
VIN increasing
8.7
V
VBYP-HYS
Bypass Switch Threshold Hysteresis
VIN Decreasing
260
mV
ZVCC
VCC Pin Output Impedance
0 mA ≤ ICC ≤ 5 mA
VCC (2)
(2)
)
5
15
35
mA
mA
200
VIN = 6 V
V
mV
58
VIN = 8 V
53
VIN = 24 V
1.6
Ω
VCC-HI
VCC Pin UVLO Rising Threshold
VCC-HYS
VCC Pin UVLO Falling Hysteresis
5
V
IVIN
Start-up Regulator Leakage
VIN = 60 V
150
500
µA
IIN-SD
Shutdown Current
VUVLO = 0 V, VCC = Open Circuit
350
450
µA
300
mV
ERROR AMPLIFIER
GBW
Gain Bandwidth
ADC
DC Gain
ICOMP
COMP Pin Current Sink Capability
VFB = 1.5 V
VCOMP = 1 V
4
MHz
75
dB
5
17
1.22
1.25
1.28
16
20
24
mA
UVLO
VSD
Shutdown Threshold
ISD-HYS
Shutdown
Hysteresis Current Source
V
µA
CURRENT LIMIT
tLIM-DLY
Delay from ILIM to Output
VCS
Current Limit Threshold Voltage
tBLK
Leading Edge Blanking Time
RCS
CS Pin Sink Impedance
CS steps from 0 V to 0.6 V
OUT transitions to 90% of VCC
30
0.434
0.5
ns
0.55
65
Blanking active
40
V
ns
75
Ω
SOFT-START
ISS
Soft-start Current Source
7
10
13
µA
VSS-OFF
Soft-start to COMP Offset
0.344
0.55
0.75
V
(1)
(2)
(3)
All minimum and maximum limits are specified by correlating the electrical characteristics to process and temperature variations and
applying statistical process control. The junction temperature (TJ in °C) is calculated from the ambient temperature (TA in °C) and power
dissipation (PD in Watts) as follows: TJ = TA + (PD • RθJA) where RθJA (in °C/W) is the package thermal impedance provided in the
Thermal Information section.
VCC provides bias for the internal gate drive and control circuits.
Device thermal limitations may limit usable range.
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Electrical Characteristics (continued)
Typical limits apply for TJ = 25°C and are provided for reference purposes only; minimum and maximum limits apply over the
junction temperature (TJ) range of –40°C to +125°C. VIN = 24 V and RT = 27.4 kΩ, unless otherwise indicated.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OSCILLATOR
fSW
VSYNC-HI
RT to GND = 84.5 kΩ
See (4)
170
200
230
kHz
RT to GND = 27.4 kΩ
See (4)
525
600
675
kHz
RT to GND = 16.2 kΩ
See (4)
865
990
1115
kHz
RT to GND = 6.65 kΩ
(4)
1910
2240
2570
kHz
See
Synchronization Rising Threshold
3.8
V
PWM COMPARATOR
tCOMP-DLY
Delay from COMP to OUT Transition
VCOMP = 2 V
CS stepped from 0 V to 0.4 V
25
DMIN
Minimum Duty Cycle
VCOMP = 0 V
DMAX
Maximum Duty Cycle
APWM
COMP to PWM Comparator Gain
VCOMP-OC
COMP Pin Open Circuit Voltage
VFB = 0 V
4.3
5.2
6.1
V
ICOMP-SC
COMP Pin Short Circuit Current
VCOMP = 0 V, VFB = 0V
0.6
1.1
1.5
mA
83
110
137
mV
ns
0%
90%
95%
0.33
V/V
SLOPE COMPENSATION
VSLOPE
Slope Compensation Amplitude
MOSFET DRIVER
VSAT-HI
Output High Saturation Voltage
(VCC – VOUT)
IOUT = 50 mA
0.25
0.75
V
VSAT-LO
Output Low Saturation Voltage
(VOUT)
IOUT = 100 mA
0.25
0.75
V
tRISE
OUT Pin Rise Time
OUT Pin load = 1 nF
18
ns
tFALL
OUT Pin Fall Time
OUT Pin load = 1 nF
15
ns
THERMAL CHARACTERISTICS
TSD
Thermal Shutdown Threshold
165
°C
TSD-HYS
Thermal Shutdown Hysteresis
25
°C
(4)
6
Specification applies to the oscillator frequency.
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6.6 Typical Characteristics
VO = 40 V
VIN = 24 V
Figure 1. Efficiency, Example Circuit BOM
TA = 25°C
Figure 2. VFB vs. Temperature
TA = 25°C
Figure 3. VFB vs. VIN
Figure 4. VCC vs. VIN
RT = 16.2 KΩ
TA = 25°C
Figure 5. Maximum Duty Cycle vs. ƒSW
Figure 6. ƒSW vs. Temperature
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Typical Characteristics (continued)
SWITCHING FREQUENCY (kHz)
2320
2310
2300
2290
2280
2270
2260
2250
2240
2230
2220
2210
-60
-40
-20
0
20
40
60
80
TEMPERATURE (oC)
100
120
140
RT = 6.65 KΩ
Figure 7. ƒSW vs. Temperature
Figure 8. SS vs. Temperature
Figure 9. OUT Pin TRISE vs. Gate Capacitance
Figure 10. OUT Pin TFALL vs. Gate Capacitance
85
75
RT (k:)
65
55
45
35
25
15
5
200
400
600 800 1000 1200 1400 1600 1800 2000 2200
SWITCHING FREQUNECY (kHz)
TA = 25°C
Figure 11. RT vs. ƒSW
8
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7 Detailed Description
7.1 Overview
The LM5022-Q1 is a low-side N-channel MOSFET controller that contains all of the features needed to
implement single ended power converter topologies. The LM5022-Q1 includes a high-voltage startup regulator
that operates over a wide input range of 6 V to 60 V. The PWM controller is designed for high speed capability
including an oscillator frequency range up to 2.2 MHz and total propagation delays less than 100 ns. Additional
features include an error amplifier, precision reference, input under-voltage lockout, cycle-by-cycle current limit,
slope compensation, soft-start, oscillator sync capability and thermal shutdown.
The LM5022-Q1 is designed for current-mode control power converters that require a single drive output, such
as boost and SEPIC topologies. The LM5022-Q1 provides all of the advantages of current-mode control
including input voltage feed-forward, cycle-by-cycle current limiting and simplified loop compensation.
7.2 Functional Block Diagram
BYPASS
SWITCH
(6V to 8.7V)
VCC
VIN
7V SERIES
REGULATOR
REFERENCE
ENABLE
+
-
UVLO
5V
1.25V
LOGIC
1.25V
UVLO
HYSTERESIS
CLK
(20 PA)
RT/SYNC
OSC
DRIVER
45 PA
Max Duty
Limit
0
S
Q
R
Q
OUT
5V
COMP
GND
5k
1.25V
PWM
100 k:
FB
+
-
LOGIC
1.4V
50 k:
SS
CS
2 k:
0.5V
SS
10 PA
SS
+
-
CLK + LEB
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7.3 Feature Description
7.3.1 High Voltage Start-Up Regulator
The LM5022-Q1 contains an internal high-voltage start-up regulator that allows the VIN pin to be connected
directly to line voltages as high as 60 V. The regulator output is internally current limited to 35 mA (typical). When
power is applied, the regulator is enabled and sources current into an external capacitor, CF, connected to the
VCC pin. The recommended capacitance range for CF is 0.1 µF to 100 µF. When the voltage on the VCC pin
reaches the rising threshold of 5 V, the controller output is enabled. The controller will remain enabled until VCC
falls below 4.7 V. In applications using a transformer, an auxiliary winding can be connected through a diode to
the VCC pin. This winding should raise the VCC pin voltage to above 7.5 V to shut off the internal startup
regulator. Powering VCC from an auxiliary winding improves conversion efficiency while reducing the power
dissipated in the controller. The capacitance of CF must be high enough that it maintains the VCC voltage greater
than the VCC UVLO falling threshold (4.7 V) during the initial start-up. During a fault condition when the
converter auxiliary winding is inactive, external current draw on the VCC line should be limited such that the
power dissipated in the start-up regulator does not exceed the maximum power dissipation capability of the
controller.
An external start-up or other bias rail can be used instead of the internal start-up regulator by connecting the
VCC and the VIN pins together and feeding the external bias voltage (7.5 V to 14 V) to the two pins.
7.3.2 Input Undervoltage Detector
The LM5022-Q1 contains an input undervoltage lockout (UVLO) circuit. UVLO is programmed by connecting the
UVLO pin to the center point of an external voltage divider from VIN to GND. The resistor divider must be
designed such that the voltage at the UVLO pin is greater than 1.25 V when VIN is in the desired operating
range. If the under voltage threshold is not met, all functions of the controller are disabled and the controller
remains in a low power standby state. UVLO hysteresis is accomplished with an internal 20 µA current source
that is switched on or off into the impedance of the set-point divider. When the UVLO threshold is exceeded, the
current source is activated to instantly raise the voltage at the UVLO pin. When the UVLO pin voltage falls below
the 1.25 V threshold the current source is turned off, causing the voltage at the UVLO pin to fall. The UVLO pin
can also be used to implement a remote enable / disable function. If an external transistor pulls the UVLO pin
below the 1.25 V threshold, the converter will be disabled. This external shutdown method is shown in Figure 12.
VIN
VIN
RUV2
LM5022
UVLO
ON/OFF
RUV1
2N7000 or
Equivalent
GND
Figure 12. Enable/Disable Using UVLO
10
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Feature Description (continued)
7.3.3 Error Amplifier
An internal high gain error amplifier is provided within the LM5022-Q1. The amplifier’s non-inverting input is
internally set to a fixed reference voltage of 1.25 V. The inverting input is connected to the FB pin. In nonisolated applications such as the boost converter the output voltage, VO, is connected to the FB pin through a
resistor divider. The control loop compensation components are connected between the COMP and FB pins. For
most isolated applications the error amplifier function is implemented on the secondary side of the converter and
the internal error amplifier is not used. The internal error amplifier is configured as an open drain output and can
be disabled by connecting the FB pin to ground. An internal 5-kΩ pullup resistor between a 5-V reference and
COMP can be used as the pull-up for an opto-coupler in isolated applications.
7.3.4 Current Sensing and Current Limiting
The LM5022-Q1 provides a cycle-by-cycle over current protection function. Current limit is accomplished by an
internal current sense comparator. If the voltage at the current sense comparator input exceeds 0.5 V, the
MOSFET gate drive will be immediately terminated. A small RC filter, located near the controller, is
recommended to filter noise from the current sense signal. The CS input has an internal MOSFET which
discharges the CS pin capacitance at the conclusion of every cycle. The discharge device remains on an
additional 65 ns after the beginning of the new cycle to attenuate leading edge ringing on the current sense
signal.
The LM5022-Q1 current sense and PWM comparators are very fast, and may respond to short duration noise
pulses. Layout considerations are critical for the current sense filter and sense resistor. The capacitor associated
with the CS filter must be located very close to the device and connected directly to the pins of the controller (CS
and GND). If a current sense transformer is used, both leads of the transformer secondary should be routed to
the sense resistor and the current sense filter network. The current sense resistor can be located between the
source of the primary power MOSFET and power ground, but it must be a low inductance type. When designing
with a current sense resistor all of the noise sensitive low-power ground connections should be connected
together locally to the controller and a single connection should be made to the high current power ground
(sense resistor ground point).
7.3.5 PWM Comparator and Slope Compensation
The PWM comparator compares the current ramp signal with the error voltage derived from the error amplifier
output. The error amplifier output voltage at the COMP pin is offset by 1.4 V and then further attenuated by a 3:1
resistor divider. The PWM comparator polarity is such that 0 V on the COMP pin will result in a zero duty cycle at
the controller output. For duty cycles greater than 50%, current mode control circuits can experience subharmonic oscillation. By adding an additional fixed-slope voltage ramp signal (slope compensation) this
oscillation can be avoided. Proper slope compensation damps the double pole associated with current mode
control (see Control Loop Compensation) and eases the design of the control loop compensator. The LM5022Q1 generates the slope compensation with a sawtooth-waveform current source with a slope of 45 µA × ƒSW,
generated by the clock (see Figure 13). This current flows through an internal 2-kΩ resistor to create a minimum
compensation ramp with a slope of 100 mV × ƒSW (typical). The slope of the compensation ramp increases when
external resistance is added for filtering the current sense (RS1) or in the position RS2. As shown in Figure 13 and
the Functional Block Diagram, the sensed current slope and the compensation slope add together to create the
signal used for current limiting and for the control loop itself.
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Feature Description (continued)
LM5022
ISW
45 PA
0
RS1
RS2
CS
2 k:
0.5V
RSNS
CSNS
+
Current
Limit
VCL
Figure 13. Slope Compensation
In peak current mode control the optimal slope compensation is proportional to the slope of the inductor current
during the power switch off-time. For boost converters the inductor current slope while the MOSFET is off is (VO VIN) / L. This relationship is combined with the requirements to set the peak current limit and is used to select
RSNS and RS2 in Application and Implementation.
7.3.6 Soft Start
The soft-start feature allows the power converter output to gradually reach the initial steady state output voltage,
thereby reducing start-up stresses and current surges. At power on, after the VCC and input under-voltage
lockout thresholds are satisfied, an internal 10-µA current source charges an external capacitor connected to the
SS pin. The capacitor voltage will ramp up slowly and will limit the COMP pin voltage and the switch current.
7.3.7 MOSFET Gate Driver
The LM5022-Q1 provides an internal gate driver through the OUT pin that can source and sink a peak current of
1 A to control external, ground-referenced N-channel MOSFETs.
7.3.8 Thermal Shutdown
Internal thermal shutdown circuitry is provided to protect the LM5022-Q1 in the event that the maximum junction
temperature is exceeded. When activated, typically at 165°C, the controller is forced into a low power standby
state, disabling the output driver and the VCC regulator. After the temperature is reduced (typical hysteresis is
25°C) the VCC regulator will be re-enabled and the LM5022-Q1 will perform a soft start.
7.4 Device Functional Modes
7.4.1 Oscillator, Shutdown, and SYNC
A single external resistor, RT, connected between the RT/SYNC and GND pins sets the LM5022-Q1 oscillator
frequency. To set the switching frequency, ƒSW, RT can be calculated from:
RT
(1 - 8 ´ 10
=
-8
´ fSW
fSW ´ 5.77 ´ 10
)
-11
where
•
•
12
fSW is in Hz
RT is in Ω
(1)
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Device Functional Modes (continued)
The LM5022-Q1 can also be synchronized to an external clock. The external clock must have a higher frequency
than the free running oscillator frequency set by the RT resistor. The clock signal should be capacitively coupled
into the RT/SYNC pin with a 100-pF capacitor as shown in Figure 14. A peak voltage level greater than 3.8 V at
the RT/SYNC pin is required for detection of the sync pulse. The sync pulse width should be set between 15 ns
to 150 ns by the external components. The RT resistor is always required, whether the oscillator is free running
or externally synchronized. The voltage at the RT/SYNC pin is internally regulated to 2 V, and the typical delay
from a logic high at the RT/SYNC pin to the rise of the OUT pin voltage is 120 ns. RT should be located very
close to the device and connected directly to the pins of the controller (RT/SYNC and GND).
LM5022
EXTERNAL
CLOCK
CSS
RT/SYNC
100 pF
RT
15 ns to 150 ns
EXTERNAL
CLOCK
120 ns
(Typical)
OUT PIN
Figure 14. SYNC Operation
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The most common circuit controlled by the LM5022-Q1 is a non-isolated boost regulator. The boost regulator
steps up the input voltage and has a duty ratio D of:
D=
VO - VIN + VD
VO + VD
where
•
VD is the forward voltage drop of the output diode
(2)
The following is a design procedure for selecting all the components for the boost converter circuit shown in
Figure 15. The application is "in-cabin" automotive, meaning that the operating ambient temperature ranges from
–20°C to 85°C. This circuit operates in continuous conduction mode (CCM), where inductor current stays above
0 A at all times, and delivers an output voltage of 40 V ±2% at a maximum output current of 0.5A. Additionally,
the regulator must be able to handle a load transient of up to 0.5 A while keeping VO within ±4%. The voltage
input comes from the battery/alternator system of an automobile, where the standard range 9 V to 16 V and
transients of up to 32 V must not cause any malfunction.
8.2 Typical Application
VIN = 9V to 16V
CIN1,2
L1
VO = 40V
D1
CINX
Q1
CO1,2
RS1
VIN
OUT
RT
CS
RT
RS2
UVLO
CSS
RUV1
LM5022
RUV2
COX
SS
RSNS
CCS
GND
CF
VCC
COMP
RFB2
FB
R1
C2
RFB1
C1
Figure 15. LM5022-Q1 Typical Application
8.2.1 Design Requirements
For typical low-side controller applications, use the parameters listed in Table 1.
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Typical Application (continued)
Table 1. Design Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Minimum input voltage
9 V to 16 V
Minimum output voltage
40 V
Output current
500 mA
Switching frequency
500 kHz
Table 2. BOM for Example Circuit
ID
PART NUMBER
TYPE
SIZE
PARAMETERS
QTY
VENDOR
U1
LM5022-Q1
Low-Side Controller
10-pin VSSOP
60V
1
TI
Q1
Si4850EY
MOSFET
SO-8
60V, 31mΩ, 27nC
1
Vishay
D1
CMSH2-60M
Schottky Diode
SMA
60V, 2A
1
Central Semi
L1
SLF12575T-M3R2
Inductor
12.5 x 12.5 x 7.5 mm
33µH, 3.2A, 40mΩ
1
TDK
Cin1, Cin2
C4532X7R1H475M
Capacitor
1812
4.7µF, 50V, 3mΩ
2
TDK
Co1, Co2
C5750X7R2A475M
Capacitor
2220
4.7µF,100V, 3mΩ
2
TDK
Cf
C2012X7R1E105K
Capacitor
0805
1µF, 25V
1
TDK
Cinx
Cox
C2012X7R2A104M
Capacitor
0805
100nF, 100V
2
TDK
C1
VJ0805A561KXXAT
Capacitor
0805
560pF 10%
1
Vishay
C2
VJ0805Y124KXXAT
Capacitor
0805
120nF 10%
1
Vishay
Css
VJ0805Y103KXXAT
Capacitor
0805
10nF 10%
1
Vishay
Ccs
VJ0805Y102KXXAT
Capacitor
0805
1nF 10%
1
Vishay
R1
CRCW08053011F
Resistor
0805
3.01kΩ 1%
1
Vishay
Rfb1
CRCW08056490F
Resistor
0805
649Ω 1%
1
Vishay
Rfb2
CRCW08052002F
Resistor
0805
20kΩ 1%
1
Vishay
Rs1
CRCW0805101J
Resistor
0805
100Ω 5%
1
Vishay
Rs2
CRCW08053571F
Resistor
0805
3.57kΩ 1%
1
Vishay
Rsns
ERJL14KF10C
Resistor
1210
100mΩ, 1%, 0.5W
1
Panasonic
Rt
CRCW08053322F
Resistor
0805
33.2kΩ 1%
1
Vishay
Ruv1
CRCW08052611F
Resistor
0805
2.61kΩ 1%
1
Vishay
Ruv2
CRCW08051002F
Resistor
0805
10kΩ 1%
1
Vishay
8.2.2 Detailed Design Procedure
8.2.2.1 Switching Frequency
The selection of switching frequency is based on the tradeoffs between size, cost, and efficiency. In general, a
lower frequency means larger, more expensive inductors and capacitors will be needed. A higher switching
frequency generally results in a smaller but less efficient solution, as the power MOSFET gate capacitances must
be charged and discharged more often in a given amount of time. For this application, a frequency of 500 kHz
was selected as a good compromise between the size of the inductor and efficiency. PCB area and component
height are restricted in this application. Following the equation given for RT in Equation 1, a 33.2-kΩ 1% resistor
should be used to switch at 500 kHz.
8.2.2.2 MOSFET
Selection of the power MOSFET is governed by tradeoffs between cost, size, and efficiency. Breaking down the
losses in the MOSFET is one way to determine relative efficiencies between different devices. For this example,
the SO-8 package provides a balance of a small footprint with good efficiency.
Losses in the MOSFET can be broken down into conduction loss, gate charging loss, and switching loss.
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Conduction, or I2R loss, PC, is approximately:
PC = D x
IO
1-D
2
x RDSON x 1.3
(3)
The factor 1.3 accounts for the increase in MOSFET on resistance due to heating. Alternatively, the factor of 1.3
can be ignored and the maximum on resistance of the MOSFET can be used.
Gate charging loss, PG, results from the current required to charge and discharge the gate capacitance of the
power MOSFET and is approximated as:
PG = VCC × QG × fSW
(4)
QG is the total gate charge of the MOSFET. Gate charge loss differs from conduction and switching losses
because the actual dissipation occurs in the LM5022-Q1 and not in the MOSFET itself. If no external bias is
applied to the VCC pin, additional loss in the LM5022-Q1 IC occurs as the MOSFET driving current flows through
the VCC regulator. This loss, PVCC, is estimated as:
PVCC = (VIN – VCC) × QG × fSW
(5)
Switching loss, PSW, occurs during the brief transition period as the MOSFET turns on and off. During the
transition period both current and voltage are present in the channel of the MOSFET. The loss can be
approximated as:
PSW = 0.5 × VIN × [IO / (1 – D)] × (tR + tF) × ƒSW
where
•
•
tR is the rise time of the MOSFET
tF is the fall time of the MOSFET
(6)
For this example, the maximum drain-to-source voltage applied across the MOSFET is VO plus the ringing due to
parasitic inductance and capacitance. The maximum drive voltage at the gate of the high side MOSFET is VCC,
or 7 V typical. The MOSFET selected must be able to withstand 40V plus any ringing from drain to source, and
be able to handle at least 7V plus ringing from gate to source. A minimum voltage rating of 50VD-S and 10VG-S
MOSFET will be used. Comparing the losses in a spreadsheet leads to a 60VD-S rated MOSFET in SO-8 with an
RDSON of 22 mΩ (the maximum vallue is 31 mΩ), a gate charge of 27 nC, and rise and falls times of 10 ns and
12 ns, respectively.
8.2.2.3 Output Diode
The boost regulator requires an output diode D1 (see Figure 15) to carrying the inductor current during the
MOSFET off-time. The most efficient choice for D1 is a Schottky diode due to low forward drop and near-zero
reverse recovery time. D1 must be rated to handle the maximum output voltage plus any switching node ringing
when the MOSFET is on. In practice, all switching converters have some ringing at the switching node due to the
diode parasitic capacitance and the lead inductance. D1 must also be rated to handle the average output current,
IO.
The overall converter efficiency becomes more dependent on the selection of D1 at low duty cycles, where the
boost diode carries the load current for an increasing percentage of the time. This power dissipation can be
calculating by checking the typical diode forward voltage, VD, from the I-V curve on the diode's datasheet and
then multiplying it by IO. Diode datasheets will also provide a typical junction-to-ambient thermal resistance, RθJA,
which can be used to estimate the operating die temperature of the Schottky. Multiplying the power dissipation
(PD = IO × VD) by RθJA gives the temperature rise. The diode case size can then be selected to maintain the
Schottky diode temperature below the operational maximum.
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In this example a Schottky diode rated to 60 V and 1 A will be suitable, as the maximum diode current will be 0.5
A. A small case such as SOD-123 can be used if a small footprint is critical. Larger case sizes generally have
lower RθJA and lower forward voltage drop, so for better efficiency the larger SMA case size will be used.
8.2.2.4 Boost Inductor
The first criterion for selecting an inductor is the inductance itself. In fixed-frequency boost converters this value
is based on the desired peak-to-peak ripple current, ΔiL, which flows in the inductor along with the average
inductor current, IL. For a boost converter in CCM IL is greater than the average output current, IO. The two
currents are related by the following expression:
IL = IO / (1 – D)
(7)
As with switching frequency, the inductance used is a tradeoff between size and cost. Larger inductance means
lower input ripple current, however because the inductor is connected to the output during the off-time only there
is a limit to the reduction in output ripple voltage. Lower inductance results in smaller, less expensive magnetics.
An inductance that gives a ripple current of 30% to 50% of IL is a good starting point for a CCM boost converter.
Minimum inductance should be calculated at the extremes of input voltage to find the operating condition with the
highest requirement:
VIN x D
L1 =
fSW x 'iL
(8)
By calculating in terms of amperes, volts, and megahertz, the inductance value will come out in micro henries.
In order to ensure that the boost regulator operates in CCM a second equation is needed, and must also be
evaluated at the corners of input voltage to find the minimum inductance required:
D(1-D) x VIN
L2 =
IO x fSW
(9)
By calculating in terms of volts, amps and megahertz the inductance value will come out in µH.
For this design ΔiL will be set to 40% of the maximum IL. Duty cycle is evaluated first at VIN(MIN) and at VIN(MAX).
Second, the average inductor current is evaluated at the two input voltages. Third, the inductor ripple current is
determined. Finally, the inductance can be calculated, and a standard inductor value selected that meets all the
criteria.
1. Inductance for Minimum Input Voltage
DVIN(MIN) = (40 – 9 + 0.5) / (40 + 0.5) = 78% IL-VIN(MIN) = 0.5 / (1 – 0.78) = 2.3 A ΔiL = 0.4 × 2.3 A = 0.92 A
L1-VIN(MIN) =
L2-VIN(MIN) =
(10)
9 x 0.78
= 15.3 PH
0.5 x 0.92
(11)
0.78 x 0.22 x 9
= 6.2 PH
0.5 x 0.5
(12)
2. Inductance for Maximum Input Voltage
DVIN(MAX) = (40 – 16 + 0.5) / (40 + 0.5) = 60% IL-VIN(MIAX) = 0.5 / (1 – 0.6) = 1.25A ΔiL = 0.4 × 1.25 A = 0.5 A
L1-VIN(MAX) =
L2-VIN(MAX) =
16 x 0.6
= 38.4 PH
0.5 x 0.5
(13)
(14)
0.6 x 0.4 x 16
= 15.4 PH
0.5 x 0.5
(15)
Maximum average inductor current occurs at VIN(MIN), and the corresponding inductor ripple current is 0.92 AP-P.
Selecting an inductance that exceeds the ripple current requirement at VIN(MIN) and the requirement to stay in
CCM for VIN(MAX) provides a tradeoff that allows smaller magnetics at the cost of higher ripple current at
maximum input voltage. For this example, a 33-µH inductor will satisfy these requirements.
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The second criterion for selecting an inductor is the peak current carrying capability. This is the level above
which the inductor will saturate. In saturation the inductance can drop off severely, resulting in higher peak
current that may overheat the inductor or push the converter into current limit. In a boost converter, peak current,
IPK, is equal to the maximum average inductor current plus one half of the ripple current. First, the current ripple
must be determined under the conditions that give maximum average inductor current:
VIN x D
'iL =
fSW x L
(16)
Maximum average inductor current occurs at VIN(MIN). Using the selected inductance of 33 µH yields the
following:
ΔiL = (9 × 0.78) / (0.5 × 33) = 425 mAP-P
(17)
The highest peak inductor current over all operating conditions is therefore:
IPK = IL + 0.5 × ΔiL = 2.3 + 0.213 = 2.51 A
(18)
Hence an inductor must be selected that has a peak current rating greater than 2.5 A and an average current
rating greater than 2.3A. One possibility is an off-the-shelf 33 µH ±20% inductor that can handle a peak current
of 3.2 A and an average current of 3.4 A. Finally, the inductor current ripple is recalculated at the maximum input
voltage:
ΔiL-VIN(MAX) = (16 × 0.6) / (0.5 × 33) = 0.58AP-P
(19)
8.2.2.5 Output Capacitor
The output capacitor in a boost regulator supplies current to the load during the MOSFET on-time and also filters
the AC portion of the load current during the off-time. This capacitor determines the steady state output voltage
ripple, ΔVO, a critical parameter for all voltage regulators. Output capacitors are selected based on their
capacitance, CO, their equivalent series resistance (ESR) and their RMS or AC current rating.
The magnitude of ΔVO is comprised of three parts, and in steady state the ripple voltage during the on-time is
equal to the ripple voltage during the off-time. For simplicity the analysis will be performed for the MOSFET
turning off (off-time) only. The first part of the ripple voltage is the surge created as the output diode D1 turns on.
At this point inductor/diode current is at the peak value, and the ripple voltage increase can be calculated as:
ΔVO1 = IPK × ESR
(20)
The second portion of the ripple voltage is the increase due to the charging of CO through the output diode. This
portion can be approximated as:
ΔVO2 = (IO / CO) × (D / ƒSW)
(21)
The final portion of the ripple voltage is a decrease due to the flow of the diode/inductor current through the
output capacitor’s ESR. This decrease can be calculated as:
ΔVO3 = ΔiL × ESR
(22)
The total change in output voltage is then:
ΔVO = ΔVO1 + ΔVO2 – ΔVO3
(23)
The combination of two positive terms and one negative term may yield an output voltage ripple with a net rise or
a net fall during the converter off-time. The ESR of the output capacitor(s) has a strong influence on the slope
and direction of ΔVO. Capacitors with high ESR such as tantalum and aluminum electrolytic create an output
voltage ripple that is dominated by ΔVO1 and ΔVO3, with a shape shown in Figure 16. Ceramic capacitors, in
contrast, have very low ESR and lower capacitance. The shape of the output ripple voltage is dominated by
ΔVO2, with a shape shown in Figure 17.
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ÂvO
VO
ID
Figure 16. ΔVO Using High ESR Capacitors
ÂvO
VO
ID
Figure 17. ΔVO Using Low ESR Capacitors
For this example the small size and high temperature rating of ceramic capacitors make them a good choice.
The output ripple voltage waveform of Figure 17 is assumed, and the capacitance will be selected first. The
desired ΔVO is ±2% of 40V, or 0.8VP-P. Beginning with the calculation for ΔVO2, the required minimum
capacitance is:
CO-MIN = (IO / ΔVO) x (DMAX / fSW) CO-MIN = (0.5 / 0.8) x (0.77 / 5 x 105) = 0.96 µF
(24)
The next higher standard 20% capacitor value is 1 µF, however to provide margin for component tolerance and
load transients two capacitors rated 4.7 µF each will be used. Ceramic capacitors rated 4.7 µF ±20% are
available from many manufacturers. The minimum quality dielectric that is suitable for switching power supply
output capacitors is X5R, while X7R (or better) is preferred. Careful attention must be paid to the DC voltage
rating and case size, as ceramic capacitors can lose 60% or more of their rated capacitance at the maximum DC
voltage. This is the reason that ceramic capacitors are often de-rated to 50% of their capacitance at their working
voltage. The output capacitors for this example will have a 100V rating in a 2220 case size.
The typical ESR of the selected capacitors is 3 mΩ each, and in parallel is approximately 1.5 mΩ. The worstcase value for ΔVO1 occurs during the peak current at minimum input voltage:
ΔVO1 = 2.5 × 0.0015 = 4 mV
(25)
The worst-case capacitor charging ripple occurs at maximum duty cycle:
ΔVO2 = (0.5 / 9.4 × 10-6) x (0.77 / 5 × 105) = 82 mV
(26)
Finally, the worst-case value for ΔVO3 occurs when inductor ripple current is highest, at maximum input voltage:
ΔVO3 = 0.58 × 0.0015 = 1 mV (negligible)
(27)
The output voltage ripple can be estimated by summing the three terms:
ΔVO = 4 mV + 82 mV - 1 mV = 85 mV
(28)
The RMS current through the output capacitor(s) can be estimated using the following, worst-case equation:
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IO-RMS = 1.13 x IL x D x (1 - D)
(29)
The highest RMS current occurs at minimum input voltage. For this example the maximum output capacitor RMS
current is:
IO-RMS(MAX) = 1.13 × 2.3 × (0.78 x 0.22)0.5 = 1.08 ARMS
(30)
These 2220 case size devices are capable of sustaining RMS currents of over 3A each, making them more than
adequate for this application.
8.2.2.6 VCC Decoupling Capacitor
The VCC pin should be decoupled with a ceramic capacitor placed as close as possible to the VCC and GND
pins of the LM5022-Q1. The decoupling capacitor should have a minimum X5R or X7R type dielectric to ensure
that the capacitance remains stable over voltage and temperature, and be rated to a minimum of 470 nF. One
good choice is a 1-µF device with X7R dielectric and 1206 case size rated to 25 V.
8.2.2.7 Input Capacitor
The input capacitors to a boost regulator control the input voltage ripple, ΔVIN, hold up the input voltage during
load transients, and prevent impedance mismatch (also called power supply interaction) between the LM5022-Q1
and the inductance of the input leads. Selection of input capacitors is based on their capacitance, ESR, and RMS
current rating. The minimum value of ESR can be selected based on the maximum output current transient,
ISTEP, using the following expression:
(1-D) x 'vIN
ESRMIN =
2 x ISTEP
(31)
For this example the maximum load step is equal to the load current, or 0.5A. The maximum permissible ΔVIN
during load transients is 4%P-P. ΔVIN and duty cycle are taken at minimum input voltage to give the worst-case
value:
ESRMIN = [(1 – 0.77) × 0.36] / (2 × 0.5) = 83 mΩ
(32)
The minimum input capacitance can be selected based on ΔVIN, based on the drop in VIN during a load transient,
or based on prevention of power supply interaction. In general, the requirement for greatest capacitance comes
from the power supply interaction. The inductance and resistance of the input source must be estimated, and if
this information is not available, they can be assumed to be 1 µH and 0.1 Ω, respectively. Minimum capacitance
is then estimated as:
CMIN =
2 x LS x VO x IO
2
VIN x RS
(33)
As with ESR, the worst-case, highest minimum capacitance calculation comes at the minimum input voltage.
Using the default estimates for LS and RS, minimum capacitance is:
CMIN =
2 x 1P x 40 x 0.5
2
9 x 0.1
= 4.9 PF
(34)
The next highest standard 20% capacitor value is 6.8 µF, but because the actual input source impedance and
resistance are not known, two 4.7 µF capacitors will be used. In general, doubling the calculated value of input
capacitance provides a good safety margin. The final calculation is for the RMS current. For boost converters
operating in CCM this can be estimated as:
IRMS = 0.29 × ΔiL(MAX)
(35)
From the inductor section, maximum inductor ripple current is 0.58 A, hence the input capacitor(s) must be rated
to handle 0.29 × 0.58 = 170 mARMS.
The input capacitors can be ceramic, tantalum, aluminum, or almost any type, however the low capacitance
requirement makes ceramic capacitors particularly attractive. As with the output capacitors, the minimum quality
dielectric used should X5R, with X7R or better preferred. The voltage rating for input capacitors need not be as
conservative as the output capacitors, as the need for capacitance decreases as input voltage increases. For this
example, the capacitor selected will be 4.7 µF ±20%, rated to 50 V, in the 1812 case size. The RMS current
rating of these capacitors is over 2A each, more than enough for this application.
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8.2.2.8 Current Sense Filter
Parasitic circuit capacitance, inductance and gate drive current create a spike in the current sense voltage at the
point where Q1 turns on. In order to prevent this spike from terminating the on-time prematurely, every circuit
should have a low-pass filter that consists of CCS and RS1, shown in Figure 15. The time constant of this filter
should be long enough to reduce the parasitic spike without significantly affecting the shape of the actual current
sense voltage. The recommended range for RS1 is between 10 Ω and 500 Ω, and the recommended range for
CCS is between 100 pF and 2.2 nF. For this example, the values of RS1 and CCS will be 100Ω and 1 nF,
respectively.
8.2.2.9 RSNS, RS2 and Current Limit
The current sensing resistor RSNS is used for steady state regulation of the inductor current and to sense
overcurrent conditions. The slope compensation resistor is used to ensure control loop stability, and both
resistors affect the current limit threshold. The RSNS value selected must be low enough to keep the power
dissipation to a minimum, yet high enough to provide good signal-to-noise ratio for the current sensing circuitry.
RSNS, and RS2 should be set so that the current limit comparator, with a threshold of 0.5 V, trips before the
sensed current exceeds the peak current rating of the inductor, without limiting the output power in steady state.
For this example the peak current, at VIN(MIN), is 2.5 A, while the inductor itself is rated to 3.2 A. The threshold for
current limit, ILIM, is set slightly between these two values to account for tolerance of the circuit components, at a
level of 3 A. The required resistor calculation must take into account both the switch current through RSNS and
the compensation ramp current flowing through the internal 2 kΩ, RS1 and RS2 resistors. RSNS should be selected
first because it is a power resistor with more limited selection. The following equation should be evaluated at
VIN(MIN), when duty cycle is highest:
RSNS =
RSNS =
L x fSW x VCL
(VO ± VIN) x 3 x D + L x fSW x ILIM
33 x 0.5 x 0.5
(40 - 9) x 3 x 0.78 + 33 x 0.5 x 3
(36)
= 0.068:
where
•
•
L is in µH
fSW in MHz
(37)
The closest 5% value is 100 mΩ. Power dissipation in RSNS can be estimated by calculating the average current.
The worst-case average current through RSNS occurs at minimum input voltage/maximum duty cycle and can be
calculated as:
PCS =
IO 2
1-D
x RSNS x D
(38)
(39)
PCS = [(0.5 / 0.22)2 x 0.1] × 0.78 = 0.4W
For this example a 0.1 Ω ±1%, thick-film chip resistor in a 1210 case size rated to 0.5W will be used.
With RSNS selected, RS2 can be determined using the following expression:
VCL - IILIM x RSNS
RS2 =
RS2 =
45P x D
- 2000 - RS1
(40)
0.5 - 3 x 0.1
- 2000 - 100 = 3598:
45P x 0.78
(41)
The closest 1% tolerance value is 3.57 kΩ.
8.2.2.10 Control Loop Compensation
The LM5022-Q1 uses peak current-mode PWM control to correct changes in output voltage due to line and load
transients. Peak current-mode provides inherent cycle-by-cycle current limiting, improved line transient response,
and easier control loop compensation.
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The control loop is comprised of two parts. The first is the power stage, which consists of the pulse width
modulator, output filter, and the load. The second part is the error amplifier, which is an op-amp configured as an
inverting amplifier. Figure 18 shows the regulator control loop components.
L
+ C
O
D
VIN
+
-
RO
RSNS
RC
+
C2
R1
RFB2
+
RFB1
C1
VREF
+
-
Figure 18. Power Stage and Error Amplifier
One popular method for selecting the compensation components is to create Bode plots of gain and phase for
the power stage and error amplifier. Combined, they make the overall bandwidth and phase margin of the
regulator easy to determine. Software tools such as Excel, MathCAD, and Matlab are useful for observing how
changes in compensation or the power stage affect system gain and phase.
The power stage in a CCM peak current mode boost converter consists of the DC gain, APS, a single low
frequency pole, ƒLFP, the ESR zero, ƒZESR, a right-half plane zero, ƒRHP, and a double pole resulting from the
sampling of the peak current. The power stage transfer function (also called the Control-to-Output transfer
function) can be written:
æ
s öæ
s ö
ç1 +
÷ ç1 ÷
wZESR ø è
wRHP ø
è
GPS = APS ´
æ
s öæ
s
s2 ö
÷
+
ç1 +
÷ çç 1 +
Qn ´ wn w2n ø÷
wLEP ø è
è
where
•
APS =
the DC gain is defined as:
(42)
(1 - D) x RO
2 x RSNS
where
(43)
RO = VO / IO
(44)
The system ESR zero is:
ZZESR =
1
RC x C O
(45)
The low frequency pole is:
ZLEP =
22
1
0.5 x (RO + ESR) x CO
(46)
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The right-half plane zero is:
VIN 2
RO x
VO
ZRHP =
L
(47)
The sampling double pole quality factor is:
1
Qn =
S -D + 0.5 + (1 - D)
Se
Sn
(48)
The sampling double corner frequency is:
ωn = π × fSW
(49)
The natural inductor current slope is:
Sn = RSNS × VIN / L
(50)
The external ramp slope is:
Se = 45 µA × (2000 + RS1 + RS2)] x ƒSW
(51)
In the equation for APS, DC gain is highest when input voltage and output current are at the maximum. In this the
example those conditions are VIN = 16 V and IO = 500 mA.
60
180
45
120
POWER STAGE PHASE (°)
POWER STAGE GAIN (dB)
DC gain is 44 dB. The low frequency pole fP = 2πωP is at 423 Hz, the ESR zero fZ = 2πωZ is at 5.6 MHz, and the
right-half plane zero ƒRHP = 2πωRHP is at 61 kHz. The sampling double-pole occurs at one-half of the switching
frequency. Proper selection of slope compensation (via RS2) is most evident the sampling double pole. A wellselected RS2 value eliminates peaking in the gain and reduces the rate of change of the phase lag. Gain and
phase plots for the power stage are shown in Figure 19 and Figure 20.
30
15
0
-15
-30
100
1k
10k
100k
1M
60
0
-60
-120
-180
100
1k
10k
100k
1M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 19. Power Stage Gain and Phase
Figure 20. Power Stage Gain and Phase
The single pole causes a roll-off in the gain of –20 dB/decade at lower frequency. The combination of the RHP
zero and sampling double pole maintain the slope out to beyond the switching frequency. The phase tends
towards –90° at lower frequency but then increases to –180° and beyond from the RHP zero and the sampling
double pole. The effect of the ESR zero is not seen because its frequency is several decades above the
switching frequency. The combination of increasing gain and decreasing phase makes converters with RHP
zeroes difficult to compensate. Setting the overall control loop bandwidth to 1/3 to 1/10 of the RHP zero
frequency minimizes these negative effects, but requires a compromise in the control loop bandwidth. If this loop
were left uncompensated, the bandwidth would be 89 kHz and the phase margin -54°. The converter would
oscillate, and therefore is compensated using the error amplifier and a few passive components.
The transfer function of the compensation block, GEA, can be derived by treating the error amplifier as an
inverting op-amp with input impedance ZI and feedback impedance ZF. The majority of applications will require a
Type II, or two-pole one-zero amplifier, shown in Figure 18. The LaPlace domain transfer function for this Type II
network is given by the following:
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GEA =
ZF
1
x
=
ZI RFB2 (C1 + C2)
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s x R1 x C2 +1
s x R1 x C1 x C2
s
+1
C1 + C2
(52)
Many techniques exist for selecting the compensation component values. The following method is based upon
setting the mid-band gain of the error amplifier transfer function first and then positioning the compensation zero
and pole:
1. Determine the desired control loop bandwidth: The control loop bandwidth, ƒ0dB, is the point at which the total
control loop gain (H = GPS × GEA) is equal to 0 dB. For this example, a low bandwidth of 10 kHz, or
approximately 1/6th of the RHP zero frequency, is chosen because of the wide variation in input voltage.
2. Determine the gain of the power stage at ƒ0dB: This value, A, can be read graphically from the gain plot of
GPS or calculated by replacing the ‘s’ terms in GPS with ‘2 πf0dB’. For this example the gain at 10 kHz is
approximately 16 dB.
3. Calculate the negative of A and convert it to a linear gain: By setting the mid-band gain of the error amplifier
to the negative of the power stage gain at f0dB, the control loop gain will equal 0 dB at that frequency. For this
example, –16 dB = 0.15V/V.
4. Select the resistance of the top feedback divider resistor RFB2: This value is arbitrary, however selecting a
resistance between 10 kΩ and 100 kΩ will lead to practical values of R1, C1 and C2. For this example, RFB2
= 20 kΩ 1%.
5. Set R1 = A × RFB2: For this example: R1 = 0.15 × 20000 = 3 kΩ
6. Select a frequency for the compensation zero, ƒZ1: The suggested placement for this zero is at the low
frequency pole of the power stage, ƒLFP = ωLFP / 2π. For this example, ƒZ1 = ƒLFP = 423 Hz
7. Set
C2 =
1
:
2S x R1 x fZ1
For this example, C2 = 125 nF
8. Select a frequency for the compensation pole, ƒP1: The suggested placement for this pole is at one-fifth of
the switching frequency. For this example, ƒP1 = 100 kHz
9. Set
C1 =
C2
:
2S x C2 x R1 x fP1 -1
For this example, C1 = 530 pF
10. Plug the closest 1% tolerance values for RFB2 and R1, then the closest 10% values for C1 and C2 into GEA
and model the error amp: The open-loop gain and bandwidth of the LM5022-Q1’s internal error amplifier are
75 dB and 4 MHz, respectively. Their effect on GEA can be modeled using the following expression:
OPG =
2S x GBW
2S x GBW
s+
ADC
ADC is a linear gain, the linear equivalent of 75 dB is approximately 5600V/V. C1 = 560 pF 10%, C2 = 120 nF
10%, R1 = 3.01 kΩ 1%
11. Plot or evaluate the actual error amplifier transfer function:
GEA-ACTUAL =
24
GEA x OPG
1 + GEA x OPG
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OVERALL LOOP GAIN (dB)
60
40
20
0
-20
-40
-60
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 21. Overall Loop Gain and Phase
OVERALL LOOP PHASE (°)
180
120
60
0
-60
-120
-180
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 22. Overall Loop Gain and Phase
12. Plot or evaluate the complete control loop transfer function: The complete control loop transfer function is
obtained by multiplying the power stage and error amplifier functions together. The bandwidth and phase
margin can then be read graphically or evaluated numerically. The bandwidth of this example circuit at VIN =
16 V is 10.5 kHz, with a phase margin of 66°.
13. Re-evaluate at the corners of input voltage and output current: Boost converters exhibit significant change in
their loop response when VIN and IO change. With the compensation fixed, the total control loop gain and
phase should be checked to ensure a minimum phase margin of 45° over both line and load.
8.2.2.11 Efficiency Calculations
A reasonable estimation for the efficiency of a boost regulator controlled by the LM5022-Q1 can be obtained by
adding together the loss is each current carrying element and using the equation:
K=
PO
PO + Ptotal-loss
(53)
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The following shows an efficiency calculation to complement the circuit design from Device Functional Modes.
Output power for this circuit is 40 V x 0.5 A = 20 W. Input voltage is assumed to be 13.8 V, and the calculations
used assume that the converter runs in CCM. Duty cycle for VIN = 13.8V is 66%, and the average inductor
current is 1.5 A.
8.2.2.11.1 Chip Operating Loss
This term accounts for the current drawn at the VIN pin. This current, IIN, drives the logic circuitry and the power
MOSFETs. The gate driving loss term from MOSFET is included in the chip operating loss. For the LM5022-Q1,
IIN is equal to the steady state operating current, ICC, plus the MOSFET driving current, IGC. Power is lost as this
current passes through the internal linear regulator of the LM5022-Q1.
IGC = QG × ƒSW IGC = 27 nC × 500 kHz = 13.5 mA
(54)
ICC is typically 3.5 mA, taken from the Electrical Characteristics table. Chip Operating Loss is then:
PQ = VIN × (IQ + IGC) PQ = 13.8 × (3.5 m + 13.5m) = 235 mW
(55)
8.2.2.11.2 MOSFET Switching Loss
PSW = 0.5 × VIN × IL × (tR + tF) x fSW PSW = 0.5 × 13.8 × 1.5 × (10 ns + 12 ns) x 5 × 105 = 114 mW
(56)
8.2.2.11.3 MOSFET and RSNS Conduction Loss
PC = D × (IL2 × (RDSON × 1.3 + RSNS)) PC = 0.66 × (1.52 × (0.029 + 0.1)) = 192 mW
(57)
8.2.2.11.4 Output Diode Loss
The average output diode current is equal to IO, or 0.5 A. The estimated forward drop, VD, is 0.5 V. The output
diode loss is therefore:
PD1 = IO × VD PD1 = 0.5 × 0.5 = 0.25 W
(58)
8.2.2.11.5 Input Capacitor Loss
This term represents the loss as input ripple current passes through the ESR of the input capacitor bank. In this
equation ‘n’ is the number of capacitors in parallel. The 4.7 µF input capacitors selected have a combined ESR
of approximately 1.5 mΩ, and ΔiL for a 13.8V input is 0.55A:
PCIN =
IIN-RMS2 x ESR
n
IIN-RMS = 0.29 x ΔiL = 0.29 × 0.55 = 0.16 A PCIN = [0.162 × 0.0015] / 2 = 0.02 mW (negligible)
(59)
(60)
8.2.2.11.6 Output Capacitor Loss
This term is calculated using the same method as the input capacitor loss, substituting the output capacitor RMS
current for VIN = 13.8 V. The output capacitors' combined ESR is also approximately 1.5 mΩ.
IO-RMS = 1.13 × 1.5 × (0.66 x 0.34)0.5 = 0.8 A PCO = [0.8 × 0.0015] / 2 = 0.6 mW
(61)
8.2.2.11.7 Boost Inductor Loss
The typical DCR of the selected inductor is 40 mΩ.
PDCR = IL2 × DCR PDCR = 1.52 × 0.04 = 90 mW
(62)
Core loss in the inductor is estimated to be equal to the DCR loss, adding an additional 90 mW to the total
inductor loss.
8.2.2.11.8 Total Loss
PLOSS = Sum of All Loss Terms = 972 mW
(63)
8.2.2.11.9 Efficiency
η = 20 / (20 + 0.972) = 95%
26
(64)
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8.2.3 Application Curves
10V/DIV
VO
SW
10V/DIV
1 és/DIV
VIN = 9-V
Figure 23. Efficiency vs. Load Current
IO = 0.5-A
Figure 24. SW Node Voltage
10V/DIV
VO
VO
SW
50 mV/DIV
10V/DIV
1 és/DIV
VIN = 16-V
1 és/DIV
IO = 0.5-A
VIN = 9-V
IO = 0.5-A
Figure 25. SW Node Voltage
Figure 26. Output Voltage Ripple (AC Coupled)
200 mA/DIV
IO
VO
VO
2V/DIV
50 mV/DIV
400 és/DIV
1 és/DIV
VIN = 16-V
IO = 0.5-A
VIN = 9-V
IO = 50mA 500mA
Figure 27. Output Voltage Ripple (AC Coupled)
Figure 28. Load Transient Response
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200 mA/DIV
IO
VO
1V/DIV
1 ms/DIV
VIN = 16-V
IO = 50mA - 500mA
Figure 29. Load Transient Response
9 Power Supply Recommendations
LM5022-Q1 is a power management device. The power supply for the device can be any DC voltage source
within the specified input range.
10 Layout
10.1 Layout Guidelines
To produce an optimal power solution with the LM5022-Q1, good layout and design of the PCB are as critical as
component selection. The following are the several guidelines in order to create a good layout of the PCB, as
based on Figure 15.
1. Using a low ESR ceramic capacitor, place CINX as close as possible to the VIN and GND pins of the
LM5022-Q1.
2. Using a low ESR ceramic capacitor, place COX close to the load as possible of the LM5022-Q1
3. Using a low ESR ceramic capacitor place CF close to the VCC and GND pins of the LM5022-Q1
4. Minimize the loop area formed by the output capacitor connections (Co1, Co2 ), by D1 and Rsns. Making
sure the cathode of D1 and Rsns are position next to each other and place Co1(+ )and Co1( -) close to D1
cathode and Rsns (-) respectively.
5. Rsns (+) should be connected to the CS pin with a separate trace made as short as possible. This trace
should be routed away from the inductor and the switch node (where D1, Q1, and L1 connect).
6. Minimize the trace length to the FB pin by positioning RFB1 and RFB2 close to the LM5022-Q1
7. Route the VOUT sense path away from noisy node and connect it as close as possible to the positive
side of COX.
10.1.1 Filter Capacitors
The low-value ceramic filter capacitors are most effective when the inductance of the current loops that they filter,
is minimized. Place CINX as close as possible to the VIN and GND pins of the LM5022-Q1. Place COX close to
the load, and CF next to the VCC and GND pins of the LM5022-Q1.
10.1.2 Sense Lines
The top of RSNS should be connected to the CS pin with a separate trace, made as short as possible. Route this
trace away from the inductor and the switch node (where D1, Q1, and L1 connect). For the voltage loop, keep
RFB1and RFB2 close to the LM5022-Q1 and run a trace, as close as possible, to the positive side of COX to RFB2.
As with the CS line, the FB line should be routed away from the inductor and the switch node. These measures
minimize the length of high impedance lines and reduce noise pickup.
28
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Layout Guidelines (continued)
10.1.3 Compact Layout
1. Parasitic inductance can be reduced by keeping the power path components close together. As described
above in point 4 in the Layout Guidelines, keep the high slew-rate current loops as tight as possible. Short,
thick traces or copper pours (shapes) are best
2. The switch node should be just large enough to connect all the components together without excessive
heating from the current it carries. The LM5022-Q1 (boost converter) operates in two distinct cycles whose
high current paths are shown in Figure 30:
+
-
Figure 30. Boost Converter Current Loops
The dark grey, inner loops represent the high current paths during the MOSFET on-time. The light grey, outer
loop represents the high current path during the off-time.
10.1.4 Ground Plane and Shape Routing
The diagram of Figure 30 is useful for analyzing the flow of continuous current vs. the flow of pulsating currents.
The circuit paths with current flow during both the on-time and off-time are considered to be continuous current,
while those that carry current during the on-time or off-time only are pulsating currents. Preference in routing
should be given to the pulsating current paths, as these are the portions of the circuit most likely to emit EMI.
The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as any
other circuit path. The continuous current paths on the ground net can be routed on the system ground plane
with less risk of injecting noise into other circuits. The path between the input source, input capacitor and the
MOSFET and the path between the output capacitor and the load are examples of continuous current paths. In
contrast, the path between the grounded side of the power switch and the negative output capacitor terminal
carries a large high slew-rate pulsating current. This path should be routed with a short, thick shape, preferably
on the component side of the PCB. Too keep the parasitic inductance low, multiple vias in parallel should be
placed on the negative pads of the input and output capacitors to connect the component side to the ground
plane. Vias should not be placed directly at the grounded side of the MOSFET (or RSNS) as they tend to inject
noise into the ground plane. A second pulsating current loop is the gate drive loop formed by the OUT and VCC
pins, Q1, RSNS and capacitor CF. These loops must be kept small.
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10.2 Layout Example
Figure 31. Typical Top Layer Overlay of the LM5022 Evaluation Board
Figure 32. Typical Bottom Layer Overlay of the LM5022 Evaluation Board
30
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Design Support
WEBENCH software uses an iterative design procedure and accesses comprehensive databases of
components. For more details, go to www.ti.com/webench.
11.2 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.3 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.5 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM5022QDGSRQ1
ACTIVE
VSSOP
DGS
10
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
5Q22
LM5022QDGSTQ1
ACTIVE
VSSOP
DGS
10
250
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
5Q22
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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