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LM5032
SNVS344B – MARCH 2005 – REVISED DECEMBER 2014
LM5032 High-Voltage Dual Interleaved Current Mode Controller
1 Features
3 Description
•
•
•
•
•
•
•
The LM5032 dual current mode PWM controller
contains all the features needed to control either two
independent forward dc/dc converters or a single high
current converter comprised of two interleaved power
stages. The two controller channels operate 180° out
of phase thereby reducing input ripple current. The
LM5032 includes a startup regulator that operates
over a wide input range up to 100 V and compound
(bipolar + CMOS) gate drivers that provide a robust
2.5-A peak sink current. The adjustable maximum
PWM duty cycle reduce stress on the primary side
MOSFET switches. Additional features include
programmable line undervoltage lockout, cycle-bycycle current limit, hiccup mode fault operation with
adjustable response time, PWM slope compensation,
soft-start, and a 2-MHz capable oscillator with
synchronization capability.
1
•
•
•
•
•
•
•
Two Independent PWM Current Mode Controllers
Integrated High-Voltage Startup Regulator
Compound 2.5-A Main Output Gate Drivers
Single Resistor Oscillator Setting to 2 MHz
Synchronizable Oscillator
Programmable Maximum Duty Cycle
Maximum Duty Cycle Fold-Back at High-Line
Voltage
Adjustable Timer for Hiccup Mode Current
Limiting
Integrated Slope Compensation
Adjustable Line Undervoltage Lockout
Independently Adjustable Soft-Start (Each
Regulator)
Direct Interface with Opto-Coupler Transistor
Thermal Shutdown
TSSOP 16-Pin Package
PART NUMBER
LM5032
PACKAGE
BODY SIZE (NOM)
TSSOP (16)
5.00 mm x 4.40 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
2 Applications
•
•
•
Device Information(1)
Telecommunication Power Converters
Industrial Power Converters
42-V Automotive Systems
Typical Application Circuit
VCC
VPWR
36V to 75V
Input
3.3V
VIN
LM5032
UVLO
CS1
OUT1
RES
Sync
ERROR AMP
& ISOLATION
RT
COMP1
DCL
2.5V
CS2
VCC
OUT2
SS2
ERROR AMP
& ISOLATION
SS1
COMP2
GND1
GND2
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM5032
SNVS344B – MARCH 2005 – REVISED DECEMBER 2014
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
5
5
8
Absolute Maximum Ratings ......................................
ESD Ratings ............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
7.3 Feature Description................................................. 12
7.4 Device Functional Modes........................................ 17
8
Application and Implementation ........................ 18
8.1 Application Information............................................ 18
8.2 Typical Application ................................................. 23
9 Power Supply Recommendations...................... 27
10 Layout................................................................... 27
10.1 Layout Guidelines ................................................. 27
10.2 Layout Example .................................................... 28
11 Device and Documentation Support ................. 28
Detailed Description ............................................ 10
11.1 Trademarks ........................................................... 28
11.2 Electrostatic Discharge Caution ............................ 28
11.3 Glossary ................................................................ 28
7.1 Overview ................................................................. 10
7.2 Functional Block Diagram ....................................... 11
12 Mechanical, Packaging, and Orderable
Information ........................................................... 28
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (April 2013) to Revision B
•
Added Pin Configuration and Functions section, Handling Rating table, Feature Description section, Device
Functional Modes, Application and Implementation section, Power Supply Recommendations section, Layout
section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information
section ................................................................................................................................................................................... 1
Changes from Original (April 2013) to Revision A
•
2
Page
Page
Changed layout of National Data Sheet to TI format ........................................................................................................... 27
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SNVS344B – MARCH 2005 – REVISED DECEMBER 2014
5 Pin Configuration and Functions
PW Package
16-Pin TSSOP
Top View
1
2
3
4
5
6
7
8
VIN
RT/SYNC
COMP1
CS1
DCL
COMP2
SS1
CS2
UVLO
SS2
VCC
RES
OUT1
OUT2
GND1
GND2
16
15
14
13
12
11
10
9
Pin Functions
PIN
I/O
DESCRIPTION
APPLICATIONS INFORMATION
NO.
NAME
1
VIN
P
Input Supply
Input to the startup regulator. The operating input range is 13 V to
100 V with transient capability to 105 V.
2
COMP1
I
PWM Control, Controller 1
The COMP1 input provides voltage feedback to the PWM
comparator inverting input of Controller 1 through a 3:1 divider. The
OUT1 duty cycle increases as the COMP1 voltage increases. An
internal 5-KΩ pull-up resistor to 5.0-V provides bias current to an
opto-coupler transistor.
3
CS1
I
Current Sense Input, Controller 1
Input for current mode control and the current limit sensing. If the
CS1 pin exceeds 0.5V the OUT1 pulse is terminated producing
cycle-by-cycle current limiting. External resistance connected to CS1
will adjust (increase) PWM slope compensation. This pin's voltage
must not exceed 1.25V.
4
SS1
I
Soft-start, Controller 1
An internal 50-µA current source charges an external capacitor to
set the soft-start rate. During a current limit restart sequence, the
internal current source is reduced to 1 µA to increase the delay
before retry. Forcing SS1 below 0.5 V shuts off Controller 1.
5
UVLO
I
VIN Under-Voltage Lockout
An external resistor divider sets the input voltage threshold to enable
the LM5032. The UVLO comparator reference voltage is 1.25 V. A
switched 20-µA current source provides adjustable UVLO hysteresis.
The UVLO pin voltage also controls the maximum duty cycle as
described in the Feature Description section.
6
VCC
P
Start-up regulator output
Output of the 7.7-V high-voltage start-up regulator. Current limit is a
minimum of 19 mA.
7
OUT1
O
Main Gate Driver, Controller 1
Gate driver output to the primary side switch for Controller 1. OUT1
swings between VCC and GND1 at a frequency equal to half the
oscillator frequency.
8
GND1
G
Ground, Controller 1
Ground connection for Controller 1 including gate driver, PWM
controller, soft-start and support functions.
9
GND2
G
Ground, Controller 2
Ground connection for Controller 2 including the gate driver, PWM
controller and soft-start.
10
OUT2
O
Main Gate Driver, Controller 2
Gate driver output to the primary side switch for Controller 2. OUT2
swings between VCC and GND2 at a frequency equal to half the
oscillator frequency.
11
RES
I
Hiccup mode restart adjust
An external capacitor sets the time delay before forced restart during
a sustained period of cycle-by-cycle current limiting. The hiccup
mode comparator threshold is 2.55 V.
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SNVS344B – MARCH 2005 – REVISED DECEMBER 2014
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Pin Functions (continued)
PIN
I/O
DESCRIPTION
APPLICATIONS INFORMATION
NO.
NAME
12
SS2
I
Soft-start, Controller 2
An internal 50-µA current source charges an external capacitor to
set the soft-start rate. During a current limit restart sequence, the
internal current source is reduced to 1 µA to increase the delay
before retry. Forcing SS2 below 0.5 V shuts off Controller 2.
13
CS2
I
Current Sense Input, Controller 2
Input for current mode control and the current limit sensing. If the
CS2 pin exceeds 0.5 V the OUT2 pulse is terminated producing
cycle-by-cycle current limiting. External resistance connected to CS2
will adjust (increase) PWM slope compensation. This pin's voltage
must not exceed 1.25V.
14
COMP2
I
PWM Control, Controller 2
The COMP2 input provides voltage feedback to the PWM
comparator inverting input of Controller 2 through a 3:1 divider. The
OUT2 duty cycle increases as the COMP2 voltage increases. An
internal 5kΩ pull-up resistor to 5.0 V provides bias current to the
opto-coupler transistor.
15
DCL
I
Duty Cycle Limit
An external resistor sets the maximum allowed duty cycle at OUT1
and OUT2.
16
RT/SYNC
I
Oscillator Adjust and Synchronizing
input
An external resistor sets the oscillator frequency. This pin also
accepts ac-coupled synchronization pulses from an external source.
6 Specifications
6.1 Absolute Maximum Ratings (1) (2)
MIN
MAX
UNIT
VIN to GND
–0.3
105
V
VCC to GND
–0.3
16
V
RT/SYNC, RES and DCL to GND
–0.3
5.5
V
CS Pins to GND
–0.3
1.25
V
All other inputs to GND
–0.3
7
V
Junction temperature
150
°C
Lead Temperature (Soldering 4 sec), (3)
260
°C
150
°C
Storage temperature, Tstg
(1)
(2)
(3)
–55
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Recommended Operating Conditions are
conditions under which operation of the device is intended to be functional. For specifications and test conditions, see the Electrical
Characteristics.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
For detailed information on soldering plastic TSSOP packages, refer to the Packaging Data Book available from Texas Instruments.
6.2 ESD Ratings
V(ESD)
(1)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
VALUE
UNIT
±2000
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VIN Voltage
MIN
MAX
UNIT
13
100
V
External Voltage Applied to VCC
8
15
V
Operating Junction Temperature
–40
125
°C
4
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6.4 Thermal Information
LM5032
THERMAL METRIC (1)
PW
UNIT
16 PINS
RθJA
Junction-to-ambient thermal resistance
96.8
RθJC(top)
Junction-to-case (top) thermal resistance
30.3
RθJB
Junction-to-board thermal resistance
42.4
ψJT
Junction-to-top characterization parameter
1.7
ψJB
Junction-to-board characterization parameter
41.8
(1)
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
6.5 Electrical Characteristics
MIN and MAX limits apply –40°C ≤ TJ ≤ 125°C. VIN = 48 V, VCC = 10 V externally applied, RT = RDCL = 42.2kΩ, UVLO = 1.5
V, TJ = 25°C, unless otherwise stated, see (1)and see (2).
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
8
UNIT
STARTUP REGULATOR (VIN, VCC Pins)
VCCReg
VCC voltage
Ext. supply disconnected.
7.4
7.7
ICC(Lim)
VCC current limit
VCC = 0V.
19
22
VCC UVT
VCC Under-voltage threshold
(VCC increasing)
Ext. supply disconnected, VIN =11V.
VCC decreasing
V
mA
VCC VCC 300 mV 100 mV
5.5
V
6.2
6.9
V
IIN
Startup regulator current
VIN = 90V, UVLO = 0V
500
600
µA
ICCIn
Supply current into VCC from
external source
Output loads = open, VCC = 10V
4.3
7
mA
1.22
1.25
1.28
V
16
20
24
µA
0.45
0.5
0.55
V
UVLO
UVLO
Under-voltage threshold
IHYST
Hysteresis current
CURRENT SENSE INPUT (CS1, CS2 Pins)
CS
Current Limit Threshold
CS delay to output
CS1 (CS2) taken from zero to 1.0V. Time for
OUT1 (OUT2) to fall to 90% of VCC. Output
load = 0 pF.
Leading edge blanking time at
CS1 (CS2)
CS1 (CS2) sink impedance
(clocked)
RCS
Internal pull-down FET on.
40
ns
50
ns
30
Equivalent input resistance at CS CS taken from 0.2V to 0.5V, internal FET off.
55
42
Ω
kΩ
CURRENT LIMIT RESTART (RES Pin)
ResTh
Threshold
2.4
2.55
2.7
V
Charge source current
15
20
25
µA
Discharge sink current
7.5
10
12.5
µA
Current source (normal
operation)
35
50
65
µA
Current source during a current
limit restart
0.7
1
1.3
µA
SOFT-START (SS1, SS2 Pins)
ISS
VSS
Open circuit voltage
5
V
OSCILLATOR (RT/SYNC Pin)
FS1
Frequency 1 (at OUT1, OUT2)
RT = 42.2 kΩ
183
200
217
kHz
FS2
Frequency 2 (at OUT1, OUT2)
RT = 13.7 kΩ
530
600
670
kHz
(1)
(2)
All electrical characteristics having room temperature limits are tested during production with TA = 25°C. All hot and cold limits are
specified by correlating the electrical characteristics to process and temperature variations and applying statistical process control.
Typical specifications represent the most likely parametric norm at 25°C operation
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Electrical Characteristics (continued)
MIN and MAX limits apply –40°C ≤ TJ ≤ 125°C. VIN = 48 V, VCC = 10 V externally applied, RT = RDCL = 42.2kΩ, UVLO = 1.5
V, TJ = 25°C, unless otherwise stated, see(1)and see(2).
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
DC voltage
TYP
MAX
2
Input Sync threshold
2.6
3.3
UNIT
V
3.7
V
PWM CONTROLLER (COMP1, COMP2, Duty Cycle Limit Pins)
Delay to output
VCOMP
COMP1 (COMP2) open circuit
voltage
ICOMP
COMP1 (COMP2) short circuit
current
COMP1 (COMP2) set to 2V. CS1 (CS2)
stepped from 0 to 0.4V. Time for OUT1
(OUT2) to fall to 90% of VCC. Output load = 0
pF.
COMP1 (COMP2) = 0V
0.6
COMP1 (COMP2) to PWM1
(PWM2) gain
50
ns
5
V
1
1.4
0.33
Minimum duty cycle
SS1 (SS2) = 0V
Maximum duty cycle 1
UVLO pin = 1.30V, RDCL = RT, COMP1
(COMP2) = open
76%
Maximum duty cycle 2
UVLO pin = 3.75V, RDCL = RT, COMP1
(COMP2) = open
20%
Maximum duty cycle 3
UVLO pin = 1.30V, RDCL = RT/4, COMP1
(COMP2) = open
20%
Maximum duty cycle 4
UVLO pin = 2.50V, RDCL = RT, COMP1
(COMP2) = open
50%
Maximum duty cycle 5
UVLO pin = 1.30V, RDCL = RT/2, COMP1
(COMP2) = open
40%
Slope compensation
Delta increase at PWM comparator to CS1
(CS2)
Channel mismatch
CS1 (CS2) = 0.25V
Soft-start to COMP offset
SS1 (SS2) = 0.8V
mA
V/V
0%
90
mV
7%
0
V
VCC0.2
V
MAIN OUTPUT DRIVERS (OUT1, OUT2)
Output high voltage
IOUT = 50mA (source)
VCC-1
Output low voltage
IOUT = 100 mA (sink)
0.3
Rise time
CLOAD = 1 nF
12
ns
Fall time
CLOAD = 1 nF
1
V
10
ns
Peak source current
1.5
A
Peak sink current
2.5
A
165
°C
20
°C
THERMAL SHUTDOWN
TSD
Shutdown temperature
Hysteresis
6
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UVLO
VIN
UVT
VCC
tVCC
1.5V
SS1
1.5V
COMP1
t1
OUT1
SS2
COMP2
1.5V
1.5V
OUT2
Figure 1. Startup Sequence
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6.6 Typical Characteristics
600
14
VCC Pin = Open
12
400
300
UVLO = 0V
200
VCC Pin = 10V
100
VCC pin = open,
INPUT CURRENT IIN (mA)
INPUT CURRENT IIN (PA)
500
OUT1 & OUT2 load = 2200 pF
10
8
Outputs frequency = 200 kHz
UVLO > 1.25V
6
4
VCC pin = open, Driver Outputs Open
2
0
VCC pin = 10V, Driver Outputs
open or loaded
0
0
20
40
60
80
100
0
20
VOLTAGE AT VIN (V)
40
60
80
100
VOLTAGE AT VIN (V)
Figure 2. IIN vs VIN
Figure 3. IIN vs VIN
100
8
7
Output Drivers @ 1 MHz
80
UVLO > 1.25V
VCC Pin Unloaded
6
500 kHz
40
1 MHz
200 kHz
50 kHz
5
VCC (V)
ICC (mA)
60
4
3
20
2
0
9
8
10
11
12
13
14
15
1
APPLIED VCC VOLTAGE (V)
0
OUT1, 2 = Open
OUT1, 2 load = 2200 pF
2
0
4
6
8
10
12
14
VOLTAGE AT VIN (V)
Figure 4. ICC vs Externally Applied VCC
Figure 5. VCC vs VIN
10
OSCILLATOR FREQUENCY (MHz)
8
7
6
VCC (V)
5
4
3
2
1
0.1
0.01
0
0
5
10
15
20
1
25
10
100
1000
RT (k:)
ICC (mA)
Figure 6. VCC vs ICC (Externally Loaded)
8
1.0
Figure 7. Oscillator Frequency vs RT Resistor
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Typical Characteristics (continued)
100
80
80
MAXIMUM DUTY CYCLE (%)
MAXIMUM DUTY CYCLE (%)
100
UVLO Pin = 1.26V
60
40
20
60
40
20
0
0
0
0.2
0.4
0.6
0.8
0
1.0
RDCL/RT
Figure 8. User Defined Maximum Duty Cycle vs RDCL
Resistor
5.0
Figure 9. Maximum Duty Cycle vs. UVLO Voltage
100
FREQUENCY @ OUT1, OUT2 (kHz)
210
R1 = 150 k:
MAXIMUM DUTY CYCLE (%)
1.0
2.0
3.0
4.0
1.25V
VOLTAGE AT UVLO PIN (V)
R2 = 10 k:
Figure 23
80
60
40
20
208
206
204
202
200
198
196
194
RT = 42.2k
192
190
0
0
20
40
60
-50
80
0
50
100
150
VOLTAGE AT VIN (V)
TEMPERATURE (oC)
Figure 10. Maximum Duty Cycle vs. VIN (Figure 24)
Figure 11. Frequency vs. Temperature
510
55
45
|
|
1.10
1.0
0.9
-50
CURRENT LIMIT THRESHOLD
@ CS1, CS2 (mV)
SOFT-START CURRENT (PA)
508
50
506
504
502
500
498
496
494
492
490
0
50
100
150
-50
TEMPERATURE (oC)
0
50
100
150
TEMPERATURE (oC)
Figure 12. Soft-Start Pin Current vs Temperature
Figure 13. Current Limit Threshold at CS1, CS2 vs
Temperature
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7 Detailed Description
7.1 Overview
The LM5032 contains all the features necessary to implement two independently regulated current mode dc/dc
converters, or a single high current converter comprised of two parallel interleaved channels using the Forward
converter topology. The two controllers operate 180° out of phase from a common oscillator, thereby reducing
input ripple current. Each regulator channel contains a complete PWM controller, current sense input, soft-start
circuit, and gate driver output. Common to both channels are the startup and VCC regulator, line under-voltage
lockout, 2 MHz capable oscillator, maximum duty cycle control, and the hiccup mode fault protection circuit.
The gate driver outputs (OUT1, OUT2) are designed to drive N-channel MOSFETs. Their compound
configuration reduces the turn-off-time, thereby reducing switching losses. Additional features include thermal
shutdown, slope compensation, and the oscillator synchronization capability.
10
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7.2 Functional Block Diagram
7.7V SERIES
REGULATOR
VI
N
VCC
VCC
6
VCC Disable
THERMAL
SHUTDOWN
BIAS VOLTAGE
GENERATOR
RT/
SYNC
VCC
5.0V
VCC UVT
CLK1, 2
UserMaxDC1, 2
OSCILLATOR
& RAMP
GENERATOR
20 PA
UVLO
SLOPE1, 2
DCL
5.0V
MaxDC1
1.25V
Ramp1
ILim1,2
20 PA
RES
UVLO
Drivers Off
RAMP1, 2
-VUVLO
LOGIC
Clk1,2
10 PA
DC1,2
2.55V
MaxDC2
R
S
Vref
Q
Ramp2
Restart
Latch
Q
Support Functions
Restart
5.0V
45 PA
SLOPE1
VCC
Drivers Off
5k
PWM
Comp 1
COMP1
10k
CLK1
5k
R
Q
S
Q
Driver Enable
Current
Limit
2k
SS1
Driver
Logic
GND1
PWM1
MaxDC1
UserMaxDC1
CS1
OUT1
Restart
PWM
Latch
DC1
CLK1
ILim1
OUT1+50 ns
42k
0.5V
VCC UVT
Controller 1
5.0V
45 PA
SLOPE2
5k
PWM
Comp 2
COMP2
VCC
PWM
Latch
10k
R
Drivers Off
Q
PWM2
CLK2
S
Current
Limit
2k
SS2
Driver Enable
MaxDC2
UserMaxDC2
ILim2
5.0V Restart
Latch
GND2
DC2
CLK2
42k
OUT2+50 ns
VCC UVT
SS1 1 PA
Driver
Logic
Q
5k
CS2
OUT2
Restart
0.5V
Controller 2
Driver Enable
5.0V
5.0V
49 PA
49 PA
Logic
SS1
Restart 5.0V
Latch
1 PA
Logic
SS2
SS2
Restart
1k
Drivers Off
Soft-start
1
1k
Soft-start
2
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7.3 Feature Description
7.3.1 Line Undervoltage Lock Out, UVLO, Shutdown
The LM5032 contains a line under-voltage lockout circuit (UVLO) designed to enable the VCC regulator and
output drivers when the system voltage (VPWR) exceeds the desired level (see Figure 14). VPWR is the voltage
normally applied to the transformer primary, and usually connected to the VIN pin (see the schematic on Page
1). The threshold at the UVLO comparator is 1.25V. An external resistor divider connected from VPWR to ground
provides 1.25V at the UVLO pin when VPWR is increased to the desired turn-on threshold. When VPWR is below
the threshold the VCC regulator and output drivers are disabled, and the internal 20 µA current source is off.
When VPWR reaches the threshold, the comparator output switches low to enable the internal circuits and the 20
µA current source. The 20 µA flows into the external divider’s junction, raising the voltage at UVLO, thereby
providing hysteresis. Internally the voltage at UVLO also drives the Maximum Duty Cycle Limiter circuit
(described below), which may influence the values chosen for the UVLO pin resistors. At maximum VPWR, the
voltage at UVLO should not exceed 6V. Refer to the Applications Information section for a procedure to calculate
the resistors values.
The LM5032 controller can be shutdown by forcing the UVLO pin below 1.25V with an external switch. When the
UVLO pin is low, the outputs and the VCC regulator are disabled, and the LM5032 enters a low power mode. If
VCC pin is not powered from an external source, the current into VIN drops to a nominal 500 µA. If the VCC pin
is powered from an external source, the current into VIN is nominally 50 µA, and the current into the VCC pin is
approximately 4.3 mA. To disable one regulator without affecting the other, see the description of the Soft-start
section.
VCC
LM5032
7.6V/6.2V
THERMAL
SHUTDOWN
VPWR
VCC Disable
20 PA
R1
UVLO
Drivers Off
UVLO
1.25V
R2
Max . Duty
Cycle Limiter
Figure 14. Drivers Off and VCC Disable
7.3.2 Startup Regulator, VIN, VCC
The high voltage startup regulator is integral to the LM5032. The input pin VIN can be connected directly to a
voltage between 13V and 100V, with transient capability to 105V. The startup regulator provides bias voltages to
the series pass VCC regulator and the UVLO circuit. The VCC regulator is disabled until the voltage at the UVLO
pin (described above) exceeds 1.25V. For applications where VPWR exceeds 100V the internal startup regulator
can be powered from an external startup regulator or other available low voltage source. See the Applications
Information section for details.
The VCC under-voltage threshold circuit (UVT) monitors the VCC regulator output. When the series pass
regulator is enabled and the internal VCC voltage increases to > 7.6V, the UVT comparator activates the PWM
controller and output drivers via the Drivers Off signal. The UVT comparator has built-in hysteresis, with the lower
threshold nominally set to 6.2V. See Figure 1 and Figure 14.
When enabled, the VCC regulated output is 7.7V ±4% with current limited to a minimum of 19 mA (typically 22
mA). The regulator’s output impedance is ≊ 6Ω.
The VCC pin requires a capacitor to ground for stability, as well as to provide the surge currents to the external
MOSFETs via the gate driver outputs. The capacitor should be physically close to the VCC and GND pins.
In most applications it is necessary to power VCC from an external source as the average current required at the
output drivers may exceed the current capability of the internal regulator and/or the thermal capability of the
LM5032 package (see Figure 4). Normally the external source is derived from the converter’s power stage once
the LM5032 outputs are active. Refer to the Applications Information section for more information.
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Feature Description (continued)
7.3.3 Drivers Off, VCC Disable
Referring to Figure 14, Drivers Off and VCC Disable are internal signals which, when active disable portions of the
LM5032. If the UVLO pin is below 1.25V, or if the thermal shutdown activates, the VCC Disable line switches high
to disable the VCC regulator. UVLO also activates the Drivers Off signal to disable the output drivers, connect the
SS1, SS2, COMP1, COMP2 and RES pins to ground, and enable the 50 µA Soft-start current sources.
If the VCC voltage falls below the under-voltage threshold of 6.2V , the UVT comparator activates only the Drivers
Off signal. The output drivers are disabled but the VCC regulator is not disabled. Additionally, the CS1, CS2, SS1,
SS2, COMP1, COMP2 and RES pins are internally grounded, and the 50 µA Soft-start current sources are
enabled.
7.3.4 Oscillator
The oscillator frequency is set with an external resistor RT connected between the RT/SYNC and GND1 pins.
The resistor value is calculated from:
RT =
17100
FS
- 0.001(FS - 400)
(1)
where FS is the desired oscillator frequency in kHz (maximum of 2 MHz), and RT is in kΩ. See Figure 7. The two
gate driver outputs (OUT1 and OUT2) switch at half the oscillator frequency and 180° out of phase with each
other. The voltage at the RT/SYNC pin is internally regulated at 2.0V. The RT resistor should be located as close
as possible to the LM5032 with short direct connections to the pins.
The LM5032 can be synchronized to an external clock by applying a narrow clock pulse to the RT/SYNC pin. See
the Applications Information section for details on this procedure. The RT resistor is always required, whether the
oscillator is free running or externally synchronized.
7.3.5 PWM Comparator/Slope Compensation
The PWM comparator of each controller compares a slope compensated current ramp signal with the loop error
voltage derived from the COMP pin. The COMP voltage is typically controlled by an external error
amplifier/optocoupler feedback circuit to regulate the converter output voltage. Internally, the voltage at the
COMP pin passes through two level shifting diodes and a gain reducing 3:1 resistor divider (see Figure 15). The
compensated current ramp signal is a combination of the current waveform at the CS pin, and an internally
generated ramp derived from the internal clock. At duty cycles greater than 50% current mode control circuits are
prone to subharmonic oscillation. By adding a small fixed ramp to the external current sense signal oscillations
can be avoided. The internal ramp has an amplitude of 45 µA and is sourced into an internal 2kΩ resistor, and a
42 kΩ resistor in parallel with the external impedance at the CS pin. The ramp current also flows through the
external impedance connected to the CS pin and thus, the amount of slope compensation can be adjusted by
varying the external circuit at the CS pin.
The output of the PWM comparator provides the pulse width information to the output drivers. This comparator is
optimized for speed in order to achieve minimum controllable duty cycles. The comparator’s output duty cycle is
0% for VCOMP ≤1.5V, and increases as VCOMP increases.
If either Soft-start pin is pulled low (internally or externally) the corresponding COMP pin is pulled down with it,
forcing the output duty cycle to zero. When the Soft-start pin voltage increases, the COMP pin is allowed to
increase. An internal 5 kΩ resistor connected from COMP to an internal 5.0V supply provides a pull-up for the
COMP pin and bias current to the collector of the opto-coupler transistor.
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Feature Description (continued)
V PWR
Current
Sense
Power
Transformer
V OUT
Slope Comp.
45 PA
Load
LM5032
30
PWM
Comparator
LEB
RF
2k
CS1
CF
42k
10k
5k
RCS
V REF
Error
Amplifier
COMP1
|
5.0V
5k
Figure 15. Typical Feedback Network
7.3.6 Cycle-by-Cycle Current Limit
Each CS pin is designed to accept a signal representative of its transformer primary current. If the voltage at CS
exceeds 0.5V the current sense comparator terminates the present main output driver (OUT pin) pulse. If the
high current fault persists, the controller operates with constant peak switch current in a cycle-by-cycle current
limit mode, and a Hiccup Mode Current Limit Restart cycle begins (see below).
Each CS pin is internally connect to ground through a 30Ω resistor during the main output off time to discharge
external filter capacitance. The discharge device remains on for an additional 50 ns after the main output driver
switches high to blank leading edge transients in the current sensing circuit. Discharging the CS pin filter each
cycle and blanking leading edge spikes reduces the filter requirement which improves the current sense
response time.
The current sense comparators are fast and respond to short duration noise pulses. The external circuitry at
each CS pin should include an R-C filter to suppress noise. Layout considerations are critical for the current
sense filter and the sense resistor. Refer to the Applications Information section for PC board layout guidelines.
7.3.7 Hiccup Mode Current Limit Restart
If cycle-by-cycle current limiting continues in either or both controllers for a sufficient period of time, the Current
Limit Restart circuit disables both regulators and initiates a soft-start sequence after a programmable delay. The
duration of cycle-by-cycle current limiting before turn-off occurs is programmed by the value of the external
capacitor at the RES pin. The dwell time before output switching resumes is programmed by the value of the
Soft-start capacitor(s). The circuit is detailed in Figure 16 and the timing is shown in Figure 17. A description of
this circuit’s operation is as follows:
a) No current limit detected:
The 10 µA discharge current source at RES is enabled pulling the RES pin to ground.
b) Current limit repeatedly detected at both CS inputs:
The 20 µA current source at RES is enabled continuously to charge the RES pin capacitor as shown in
Figure 17. The current limit comparators also terminate the PWM output pulses to provide a cycle-by-cycle
current limiting. When the voltage on the RES capacitor reaches the 2.55V restart comparator threshold, the
comparator sets the Restart Latch which produces the following restart sequence:
• The SS1 and SS2 pin charging currents are reduced from 50µA to 1 µA.
• An internal MOSFET is turned on to discharge the RES pin capacitor.
• The internal MOSFETs at SS1 and SS2 are turned on to discharge the Soft-start capacitors.
• COMP1 and COMP2 follow SS1 and SS2 respectively and reduce the PWM duty cycles to zero.
• When the voltages at the SS pins fall below 200mV, the internal MOSFETs at the SS pins are turned off
allowing the SS pins to be charged by the 1µA current sources.
• When either SS pin reaches ≊1.5V its PWM controller produces the first pulse of a soft-start sequence which
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Feature Description (continued)
resets the Restart Latch. The SS charging currents are increased to 50 µA and the soft-start sequence
continues at the normal rate.
If the overload condition still exists, the voltage at RES begins to increase again and repeat the restart cycle as
shown in Figure 17. If the overload condition has been cleared, the RES pin is held at ground by the 10 µA
current source.
c) Current limit repeatedly detected at one of the two CS inputs:
In this condition the RES pin capacitor is charged by the 20 µA current source once each clock cycle of the
current limited regulator, and discharged by the 10 µA current source once each clock cycle of the unaffected
regulator. The voltage at the RES pin increases one fourth as fast as in case b) described above. The current
limited regulator operates in a cycle-by-cycle current limit mode until the voltage at RES reaches the 2.55V
threshold. When the Restart Comparator output switches high the Restart Latch is set, both SS pin capacitors
are discharged to disable the regulator channels, and a restart sequence begins as described in case b) above.
To determine the value of the RES pin capacitor, see the Applications Information section.
CS1
Current
Limit
Current
Sense Circuit
5.0V
Restart
Current
Source
Logic
0.5V
Clk1, Clk2
Current
Limit
Current
Sense Circuit
SS1
COMP1
DC1
PWM #1
To Output
Drivers
C RES
2.55V
S
Voltage
Feedback
10 PA
RES
0.5V
CS2
Voltage
Feedback
20 PA
DC2
PWM #2
COMP2
Drivers Off
SS2
1PA
R
Restart
Comparator
Q
Restart
Latch
49 PA
SS1
SS1
SS
C SS1
200 mV
Logic
200 mV
Logic
Drivers Off
Soft-start #1
1PA
49 PA
SS2
SS2
SS
C SS2
LM5032
Soft-start #2
Figure 16. Current Limit Restart Circuit
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Feature Description (continued)
2.55V
Current Limit Detected at
CS1 and/or CS2
RES
0V
5.0V
SS1
and
SS2
50 PA
1 PA
#1.5V
OUT1
OUT2
t1
t2
t3
Figure 17. Current Limit Restart Timing
7.3.8 Soft-Start
Each soft-start circuit allows the corresponding regulator to gradually reach a steady state operating point,
thereby reducing startup current surges and output overshoot. Upon turn-on, both SS pins are internally held at
ground. When VCC increases past its under-voltage threshold (UVT), the SS pins are released and internal 50
µA current sources charge the external capacitors. The voltage at each COMP pin follows the SS pin, and when
COMP reaches ≊1.5V, the output pulses commence at a low duty cycle. The voltage at the SS pins continues to
increase and saturates at ≊5.0V, The voltage at each COMP pin increases to the value required for regulation
where it is controlled by its voltage feedback loop (see Figure 1).
If the internal Drivers Off line is activated (see Drivers Off, VCC Disable), both SS pins are internally grounded.
The SS pins pull the COMP pins to ground while the Driver Off signal disables the output drivers. When the
event which activated the Drivers Off line is cleared and Vcc exceeds its under-voltage threshold, the SS pins
are released. The internal 50 µA current sources then charge the external soft-start capacitors allowing each
regulator’s output duty cycle to increase.
If the Current Limit Restart threshold is reached due to repeated over-current detections, both SS pins (and the
COMP pins) are pulled to ground. The output drivers are disabled, and the 50 µA SS pin current sources are
reduced to 1 µA. After a short propagation delay the SS pins and the COMP pins are released, and the external
capacitors are charged up at a slow rate. When the COMP voltage reaches ≊ 1.5V, the output drivers are
enabled, and the current sources at the SS pins are increased to 50 µA. The output duty cycle then increases to
the value required for regulation.
To shutdown one regulator without affecting the other, ground the appropriate SS pin. This forces the COMP pin
to ground, reducing the output duty cycle to zero for that regulator. Releasing the SS pin allows normal operation
to resume.
7.3.9 Output Duty Cycle
The output driver’s duty cycle for each controller is normally controlled by comparing the voltage provided to the
COMP input by the external voltage feedback circuit with the current information at the CS pin. However, the
maximum duty cycle during transient or fault conditions may be intentionally limited by two other circuits, both of
which are common to the two controller channels.
User Defined Maximum Duty Cycle. The maximum allowed duty cycle can be set with the RDCL resistor
connected from the DCL pin to GND1, according to the following equation:
Maximum User Duty Cycle = 80% x RDCL/RT
(2)
RT is the oscillator frequency programming resistor connected to the RT/SYNC pin. The value of the RDCL resistor
must be calculated after the RT resistor is selected. See Figure 8. Referring to the block diagram of the voltage at
the DCL pin is compared to the Ramp1 and Ramp2 signals, creating the UserMaxDC1 and UserMaxDC2 timing
signals. These signal are provided to the two 4-input AND gates to limit the PWM duty cycle of both channels.
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Feature Description (continued)
Line Voltage Maximum Duty Cycle. The voltage at the UVLO pin, normally proportional to the voltage at VPWR,
further limits the maximum duty cycle at high input voltages. Referring to Figure 10, when the UVLO pin is below
1.25V, the outputs are disabled. At UVLO = 1.25V the maximum allowed duty cycle is 80% (or less if limited by
the DCL resistor). As the UVLO pin voltage increases with VPWR, the maximum duty cycle decreases, reaching a
minimum of 10% at ≊4.5V. Referring to the UVLO voltage, after passing through an inverting gain stage, is
compared to the Ramp1 and Ramp2 signals generated by the oscillator. The output of these comparators are the
MaxDC1 and MaxDC2 timing signals. These signals are provided to the two 4-input AND gates which limit the
PWM pulses delivered to the output drivers.
Resulting Output Duty Cycle. The controller duty cycle is determined by the four signals into the 4-input AND
gates in (UserMaxDC, MaxDC, PWM and CLK). The output driver pulsewidth is equal to the least of these four
pulses. Whichever input of the AND gate transitions high-to-low first terminates the output driver’s on-time.
7.3.10 Driver Outputs
OUT1, the primary switch driver for Controller 1 is designed to drive the gate of an N-channel MOSFET with 1.5A
sourcing current and 2.5A sinking current. The peak output levels are VCC and GND1. The ground return path
for Controller 1 is GND1. The corresponding pins for Controller 2 are OUT2 and GND2.
OUT1 and OUT2 are compound gate drivers with CMOS and Bipolar output transistors as shown in Figure 18.
The parallel MOS and Bipolar devices provide a faster turn-off of the primary switch thereby reducing switching
losses. The outputs switch at one-half the oscillator frequency with the rising edges at OUT1 and OUT2 180° out
of phase with each other. The on-time of OUT1 and OUT2 is determined by their respective duty cycle control.
LM5032
VCC
PWM
OUT
GND
Figure 18. Compound Gate Driver
7.3.11 Thermal Shutdown
The LM5032 should be operated so the junction temperature does not exceed 125°C. If a junction temperature
transient reaches 165°C (typical), the Thermal Shutdown circuit activates the VCC Disable and Drivers Off lines
(see Figure 14). The VCC regulator and the four output drivers are disabled, the SS1, SS2, and RES pins are
grounded, and the soft-start current is set to 50 µA. This puts the LM5032 in a low power state helping to prevent
catastrophic failures from accidental device overheating. When the junction temperature reduces below 145°C
(typical hysteresis = 20°C), the VCC regulator is enabled and a startup sequence is initiated (Figure 1).
7.4 Device Functional Modes
Normal device operating mode is described above in sections Line Undervoltage Lock Out, UVLO, Shutdown
through Cycle-by-Cycle Current Limit, and sections Soft-Start to Thermal Shutdown. Under overcurrent fault
conditions, the device operate in Hiccup Mode, as detailed above in the Hiccup Mode Current Limit Restart
section.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 VIN
The voltage applied to the VIN pin, normally the same as the system voltage applied to the power transformer’s
primary (VPWR), can vary in the range of 13 to 100V with transient capability to 105V. The current into VIN
depends primarily on the output driver capacitive loads, the switching frequency, and any external load at VCC. If
the power dissipation associated with the VIN current exceeds the package capability, an external voltage should
be applied to VCC (see Figure 2 & Figure 3) to reduce power in the internal start-up regulator. It is recommended
the circuit of Figure 19 be used to suppress transients which may occur at the input supply, in particular where
VIN is operated close to the maximum operating rating of the LM5032.
When all internal bias currents for the LM5032 and output driver currents are supplied through VIN and the
internal VCC regulator, the required input current (IIN) is shown in Figure 2 & Figure 3. In most applications, upon
turn-on, IIN increases with VIN as shown in Figure 2 until the UVLO threshold is reached. After the outputs are
enabled and the external VCC supply voltage is active, the current into VIN then drops to a nominal 120 µA.
V PWR
50
VIN
0.1 PF
LM5032
Figure 19. Input Transient Protection
8.1.2 For Applications > 100 V
For applications where the system input voltage (VPWR) exceeds 100V, VIN can be powered from an external
start-up regulator as shown in Figure 20, or from any other low voltage source as shown in Figure 21.
Connecting VIN and VCC together allows the LM5032 to be operated with VIN below 13V. The voltage at VCC
must not exceed 15V. The voltage source at the right side of Figure 20 is typically derived from the power stage,
and becomes active once the LM5032’s outputs are active.
VPWR
9V
0.1
VIN
VCC
C1
8V - 15V (from
power stage)
LM5032
Figure 20. Start-up Regulator for VPWR >100V
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Application Information (continued)
8V-15V
Start-up
Voltage
VIN
VCC
8V - 15V
C1
LM5032
Figure 21. Bypassing the Internal Start-up Regulator
8.1.3 UVLO
The under-voltage lockout threshold (UVLO) is internally set at 1.25V at the UVLO pin. With two external
resistors as shown in Figure 22, the LM5032 is enabled when VPWR exceeds the programmed threshold voltage.
When VPWR is above the threshold, the internal 20 µA current source is enabled to raise the voltage at the UVLO
pin, providing hysteresis. R1 and R2 are determined from the following equations:
R1 = VHYS/20 µA
R2 =
(3)
1.25 x R1
VPWR - 1.25
(4)
where VHYS is the desired UVLO hysteresis at VPWR, and VPWR in the second equation is the turn-on voltage. For
example, if the LM5032 is to be enabled when VPWR reaches 20V, and disabled when VPWR is decreased to 17V,
R1 calculates to 150 kΩ, and R2 calculates to 10 kΩ. The voltage at UVLO should not exceed 6V at any time.
V PWR
LM5032
20 PA
R1
UVLO
Enable VCC Regulator
and Output Drivers
1.25V
R2
Max. Duty
Cycle Limiter
Figure 22. UVLO Circuit
The LM5032 can be remotely shutdown by taking the UVLO pin below 1.25V with an external open collector or
open drain device, as shown in Figure 23. The outputs, and the VCC regulator, are disabled, and the LM5032
enters a low power mode. To shut down one regulator without affecting the other, see the Soft-start section.
V PWR
LM5032
20 PA
R1
UVLO
1.25V
R2
Shutdown
Control
Max. Duty
Cycle Limiter
Figure 23. Shutdown Control
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Application Information (continued)
8.1.4 VCC
The capacitor at VCC provides not only regulator noise filtering and stability, but also prevents VCC from
dropping to the lower under-voltage threshold level (UVT = 6.2V) when the output drivers source current surges
to the external MOSFET gates. Additionally, the capacitor provides a necessary time delay during startup. The
time delay allows the internal circuitry of the LM5032 and associated external circuitry to stabilize before VCC
reaches the upper UVT threshold level (7.6V), at which time the outputs are enabled and the soft-start sequence
begins. VCC is nominally regulated at 7.7V. The delay to the UVT level (Figure 1) is calculated from the
following:
tVCC =
C1 x 7.6V
ICC(Lim)
(5)
where C1 is the capacitor at VCC and ICC(Lim) is the VCC regulator’s current limit. If the capacitor is 0.1 µF, the
nominal ICC(Lim) of 22 mA provides a delay of approximately 35 µs. The capacitor value should range between 0.1
µF and 25 µF. Experimentation with the final design may be necessary to determine the optimum value for the
VCC capacitor.
The average VCC regulator current required to drive the external MOSFETs is a function of the MOSFET gate
capacitance and the switching frequency (see Figure 4). To ensure VCC does not droop below the lower UVT
threshold, an external supply should be diode connected to VCC to provide the required current, as shown in
Figure 24. The applied VCC voltage must be between 8V and 15V. Providing the VCC voltage higher than the
7.7V regulation level with an external supply shuts off the internal regulator, reducing power dissipation within the
IC. Internally there is a diode from the VCC regulator output to VIN. Typically the applied voltage is derived from
an auxiliary winding on the power transformer, or on the output inductor.
V PWR
VIN
VCC
LM5032
8V - 15V (from
external source)
C1
GND1
GND2
Figure 24. External Power to VCC
8.1.5 Oscillator, Sync Input
The oscillator frequency is generally selected in conjunction with the system magnetic components, and any
other aspects of the system which may be affected by the frequency. The RT resistor at the RT/SYNC pin sets
the frequency according to Equation 1. Each output (OUT1 and OUT2) switches at one-half the oscillator
frequency. If the required frequency tolerance is critical in a particular application, the tolerance of the external
resistor and the frequency tolerance specified in the Electrical Characteristics table must be considered when
selecting the RT resistor.
If the LM5032 is to be synchronized to an external clock, that signal must be coupled into the RT/SYNC pin
through a 100 pF capacitor. The external synchronizing frequency must be at least 4% higher than the free
running frequency set by the RT resistor and no higher than twice the free running frequency. The RT/SYNC pin
voltage is nominally regulated at 2.0V and the external pulse amplitude should lift the pin to between 3.8V and
5.0V on the low-to-high transition. The synchronization pulse width should be between 15 and 150 ns. The RT
resistor is always required, whether the oscillator is free running or externally synchronized.
8.1.6 Voltage Feedback, COMP1, COMP2
Each COMP pin is designed to accept a voltage feedback signal from the respective regulated output via an
error amplifier and (typically) an opto-coupler. A typical configuration is shown in Figure 15. VOUT is compared to
a reference by the error amplifier which has an appropriate frequency compensation network. The amplifier’s
output drives the opto-coupler, which in turn drives the COMP pin.
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Application Information (continued)
When the LM5032’s two controller channels are configured to provide a single high current output, COMP1 and
COMP2 are typically connected together, and to the feedback signal from the optocoupler.
8.1.7 Current Sense, CS1, CS2
Each CS pin receives an input signal representative of its transformer’s primary current, either from a current
sense transformer or from a resistor in series with the source of the primary switch, as shown in Figure 25 and
Figure 26. In both cases the sensed current creates a ramping voltage across R1, and the RF/CF filter
suppresses noise and transients. R1, RF and CF should be as physically close to the LM5032 as possible, and
the ground connection from the current sense transformer, or R1, should be a dedicated track to the appropriate
GND pin. The current sense components must provide >0.5V at the CS pin when an over-current condition
exists.
Power
Current
Sense
VPWR
VIN
Transformer
RF
CS1
LM5032
CF
R1
GND1
Q1
OUT1
Figure 25. Current Sense Using a Current Sense Transformer
Power
Transformer
VPWR
VIN
Q1
OUT1
LM5032
RF
CS1
CF
R1
GND1
Figure 26. Current Sense Using a Source Sense Resistor (R1)
8.1.8 Hiccup Mode Current Limit Restart
This circuit’s operation is described in the Functional Description. Also see Figure 16 and Figure 17. In the case
of continuous current limit detection at both CS pins, the time required to reach the 2.55V RES pin threshold is:
t1 =
CRES x 2.55V
20 PA
= 1.275 x 105 x CRES
(6)
For example, if CRES = 0.1 µF the time t1 in Figure 18 is approximately 12.75 ms.
In the case of continuous current limit detection at one CS pin only, the time to reach the 2.55V threshold is
increased by a factor of four, or:
t1 = 5.1 x 105 x CRES
(7)
The time t2 in Figure 17 is set by the capacitor at each SS pin and the internal 1 µA current source, and is equal
to:
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Application Information (continued)
CSS x 1.5V
t2 =
1 PA
= 1.5 x 106 x CSS
(8)
If CSS = 0.1 µF t2 is ≊150 ms. Time t3 is set by the internal 50 µA current source, and is equal to:
CSS x 3.5V
t3 =
50 PA
= 7 x 104 x CSS
(9)
The time t2 provides a periodic dwell time for the converter in the event of a sustained overload or short circuit.
This results in lower average input current and lower power dissipated within the circuit components. It is
recommended that the ratio of t2/(t1 + t3) be in the range of 5 to 10 to make good use of this feature.
If the application requires no delay from the first detection of a current limit condition, so that t1 is effectively
zero, the RES pin can be left open (no external capacitor). If it is desired to disable the hiccup mode current limit
operation then the RES pin should be connected to ground.
8.1.9 Soft-Start
The capacitors at SS1 and SS2 determine the time required for each regulator’s output duty cycle to increase
from zero to its final value for regulation. The minimum acceptable time is dependent on the output capacitance
and the response of each feedback loop to the COMP pin. If the Soft-start time is too quick, the output could
significantly overshoot its intended voltage before the feedback loop has a chance to regulate the PWM
controller.
After power is applied and VCC has passed its upper UVT threshold (≊7.6V), the voltage at each SS pin ramps up
as its external capacitor is charged up by an internal 50 µA current source (see Figure 1). The voltage at the
COMP pins follow the SS pins. When both have reached ≊1.5V, PWM pulses appear at the driver outputs with
very low duty cycle. The voltage at each SS pin continues to increase to ≊5.0V. The voltage at each COMP pin,
and the PWM duty cycle, increase to the value required for regulation as determined by its feedback loop. The
time t1 in Figure 1 is calculated from:
CSS x 1.5V
t1 =
50 PA
= 3 x 104 x CSS
(10)
With a 0.1 µF capacitor at SS, t1 is ≊3 ms.
If the Hiccup Mode Current Limit Restart circuit activates due to repeated current limit detections at CS1 and/or
CS2, both SS1 and SS2 are internally grounded (see the section on Hiccup Mode Current Limit Restart). After a
short propagation delay, the SS pins are released and the external SS pin capacitors are charged by internal 1
µA current sources. The slow charge rate provides a rest or dwell time for the converter power stage (t2 in
Figure 17), reducing the average input current and component temperature rise while in an overload condition.
When the voltage at the SS and COMP pins reach ≊1.5V, the first pulse out of either PWM comparator switches
the internal SS pin current sources to 50 µA. The voltages at the SS and COMP pins then increase more quickly,
increasing the duty cycle at the output drivers. The rest time t2 is the time required for SS to reach 1.5V:
CSS x 1.5V
t2 =
1 PA
= 1.5 x 106 x CSS
(11)
With a 0.1 µF capacitor at SS, t2 is ≊150 ms.
Experimentation with the startup sequence and over-current restart condition is usually necessary to determine
the appropriate value for the SS capacitors.
To shutdown one regulator without affecting the other, ground the appropriate SS pin with an open collector or
open drain device as shown in Figure 27. The SS pin forces the COMP pin to ground which reduces the PWM
duty cycle to zero for that regulator. Releasing the SS pin allows normal operation to resume.
When the LM5032’s two controller channels are configured to provide a single high current output, SS1 and SS2
are typically connected together, requiring a single capacitor for the two pins.
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Application Information (continued)
To Output
Drivers
LM5032
Opto Coupler
COMP1
PWM
Controller #1
SS1
Shutdown
Control
Softstart
#1
C SS1
PWM
Controller # 2
COMP2
Softstart
#2
SS2
Opto Coupler
CSS2
Figure 27. Shutting Down One Regulator Channel
8.1.10 Line Voltage Dependent Maximum Duty Cycle
As VPWR increases and the voltage at UVLO follows, the maximum allowed duty cycle decreases according to
the graph of Figure 9. Using values from the example above (R1 = 150 kΩ, R2 = 10 kΩ in Figure 22), the
maximum duty cycle varies as shown in Figure 10. If it is desired to increase the slope of the ramp in Figure 10,
Figure 28 shows a suggested configuration. After the LM5032 is enabled, Z1 clamps the voltage across R1B,
and UVLO increases with VPWR at a rate determined by the ratio R2/(R1A + R2).
V PWR
R1A
LM5032
20 PA
R1B
Z1
UVLO
1.25V
R2
Max. Duty
Cycle Limiter
Figure 28. Altering the Slope of Duty Cycle vs. VPWR
8.1.11 User Defined Max Duty Cycle
The maximum allowed duty cycle at OUT1 and OUT2 can be set with a resistor from DCL to GND1. See
Figure 8 and Equation 2. The default maximum duty cycle (80%) determined by the internal clock signals can be
selected by setting RDCL = RT. The oscillator frequency setting resistor (RT) must be determined before RDCL is
selected. The DCL pin should not be left open.
8.2 Typical Application
Figure 29 shows an example of an LM5032-controlled 200-W interleaved regulator which provides a single
regulated 48-V output. The interleaving of two power stages to a single output reduces the ripple voltage across
both input and output capacitors, and improves the power stage efficiency compared to a single-stage design.
Since the two interleaved control blocks are used to regulate a single combined output, the two soft-start pins
SS1 and SS2 are connected to a single soft-start capacitor, and the two COMP1 and COMP2 pins are
connected together to a single error amplifier.
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Typical Application (continued)
Figure 29. Evaluation Module Schematic
8.2.1 Design Requirements
DESIGN PARAMETERS
VALUE
Input voltage range, VIN
18 V to 45 V
Output voltage, VOUT
48 V
Output current, IOUT
4A
Output ripple voltage, VRIPPLE(OUT)
< 2% (960 mVpp)
Switching frequency, FSW (per
phase)
123 kHz
8.2.2 Detailed Design Procedure
8.2.2.1 Oscillator Frequency and Maximum Duty Cycle
The LM5032 oscillator frequency should be set at twice the target switching frequency of each interleaved power
stage, for example, Fosc = 2 x Fsw.
From Equation 1, the required value of resistor on the RT pin (R9 in Figure 29) is calculated as follows:
RT =
17100
17100
- 0.001´ (Fosc - 400) =
- 0.001´ (246 - 400) = 69.67k W
Fosc
246
(12)
The nearest E96 value of 69.8 kΩ is used.
The maximum duty cycle is set to 80% (see the Output Duty Cycle section) by choosing the same value resistor
on the DCL pin, so R30 is also set to 69.8 kΩ.
8.2.2.2 Power Stage Design
8.2.2.2.1 Boost Inductor Selection
Maximum and minimum operating duty cycles are calculated at maximum and minimum input voltage, where Vd
is the boost diode forward voltage drop, and Vsw(on) is the voltage drop across the boost MOSFET plus current
sense element:
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Dmax =
Dmin =
VOUT + Vd - VIN(MIN)
VOUT + Vd - Vsw(on)
VOUT + Vd - VIN(max)
VOUT + Vd - Vsw(on)
=
48 + 0.7 - 18
= 63.7%
48 + 0.7 - 0.5
(13)
48 + 0.7 - 45
=
= 7.7%
48 + 0.7 - 0.5
(14)
The highest average inductor current in each phase is calculated at highest load and minimum input voltage,
where each phase is assumed to carry 50% of the total load current:
L(avg) =
50% ´ Iout
0.5 ´ 4
=
= 5.51A
1 - Dmax
1 - 0.637
(15)
Allowing the peak-to-peak inductor current ripple to be 100% of the average:
DIL(pk -pk) = IL(avg) = 5.51A
(16)
And the peak inductor current IL(peak) will be:
IL(peak) = IL(avg) +
DIL(pk -pk)
2
= 8.26A
(17)
Knowing the switching frequency, maximum duty cycle and target peak-peak ripple current, the required
inductance can be calculated:
Lmin =
(VIN(min) - VSW(on) ´ Dmax
fSW ´ DIL(pk -pk)
=
(18 - 0.5) ´ 0.637
= 16.4mH
123kHz ´ 5.51
(18)
Off-the-shelf available inductors of 15 µH were used, resulting in slightly higher peak-peak inductor ripple current,
and slightly higher inductor peak current.
8.2.2.2.2 Output Capacitor Selection
The output ripple across the boost output capacitor can be approximated from the following equation, where
COUT is the value of output capacitance, and ESR is the equivalent-series-resistance of the output capacitance.
For this design, the chosen electrolytic output capacitors are 150 μF with 160-mΩ ESR, so the net capacitance is
300 μF and net ESR is 80 mΩ.
é IOUT(max) ´ (1 - Dmin ) ù
é 4 ´ (1 - 0.077) ù
DVOUT = ê
ú + ëéIL(peak) ´ ESR ûù = ê
ú + [8.26 ´ 0.08] = 711mV
ë 2 ´ 123k ´ 300m û
ë 2 ´ fSW ´ COUT û
(19)
This meets the target 2% specification. However, the extra ceramic output capacitors will also absorb a
significant percentage of the switching frequency ripple, so the resulting output peak-to-peak ripple voltage
should be lower than the value calculated above, and should be comfortably less than the 2% specification.
8.2.2.2.3 Boost MOSFET Selection
The boost MOSFET should be rated for at least the rated output voltage plus some margin for voltage ringing. A
60-V device was selected. Since the boost inductor value was chosen to achieve peak-to-peak ripple current
equal to 100% of the average current, the RMS MOSFET current at maximum load and minimum Vin is:
ISW(rms) =
Dmax
´ DIL(pk -pk)2 + 3 ´ Ipeak 2 - 3 ´ Ipeak ´ DIL(pk -pk) =
3
0.637
´ 5.512 + 3 ´ 8.262 - 3 ´ 8.26 ´ 5.51 = 4.57A
3
(20)
The chosen 60-V rated MOSFET SUD50N06-9L has 9.3-mΩ Rds(on), resulting in approximately 200-mW
conduction loss.
8.2.2.2.4 Boost Diode Selection
The boost diode must have a reverse voltage rating of at least VOUT, plus some margin for ringing. Thus a 60-V
rated part was selected. Since fast reverse recovery is important, a Schottky device can be used at this voltage
rating. A common-cathode dual-diode MBR1560 was selected, with each diode connected to one or other of the
interleaved phases.
8.2.2.3 UVLO Setting
To ensure start-up below the required minimum system input voltage of 18 V, the UVLO divider resistors R6 and
R8 are set to 17.4 kΩ and 2 kΩ, respectively. This sets the input UVLO turn-on level to:
Vin(on) =
R6 + R8
17.4 + 2
´ VUVLO =
´ 1.25 = 12.125V
R8
2
(21)
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This gives plenty margin to the required 16-V minimum. Resistor R7 in series with the UVLO pin increases the
effective UVLO hysteresis.
8.2.2.4 VIN, VCC, Startup
To reduce the power dissipation in the internal startup regulator on the VIN pin, a separate external switching
regulator is used. This consists of U2 (LM5009) plus associated circuitry C5, C6, R1, R2, C7, D1, L1, R3 and R4.
This buck regulator is designed to generate a 10-V regulated supply voltage for the VCC of U1 LM5032. See the
LM5009 device datasheet, SNVS402, for detailed design information.
Since the LM5032 internal VIN regulator is not used in this design, the LM5032 VIN and VCC pins are shorted
together.
8.2.2.5 Soft-Start and Overload
Since the two soft-start pins SS1 and SS2 are connected to a single soft-start capacitor, C12, the combined
charging current of both soft-start pins charges the single soft-start capacitor. The soft-start delay to
commencement of first PWM switching can be calculated from:
t ss _ delay =
1.5V ´ CSS 1.5 ´ 0.1mF
=
= 1.5ms
100mA
100mA
(22)
Thereafter, the soft-start ramp time will depend on the power stage design and the operating conditions (input
voltage and output load).
8.2.2.6 Current Sense
In order to improve the efficiency, a lower value current sense shunt resistance is used. To enable this lower
value, the normal operating range of the CS1/CS2 pins is reduced by adding an external DC offset to the
CS1/CS2 pins, as shown in Figure 30.
U3 VREF
2.0V
U1
R23
10k
Q1
1k
CS1
R10
R12-R16
0.022
Figure 30. Current Sense DC Offset Circuit
This circuit uses the 2.048-V reference U3 to add a typical offset of 185 mV to both current-sense pins. This
reduces the active range of the internal cycle-by-cycle current-limit comparator to 315 mV, allowing the currentsense shunt to be decreased to 66% of the value that would be otherwise required.
From the power stage design calculations, the peak inductor current in each power stage was approximately 9 A
at max load and minimum Vin. Allowing for tolerances, and providing some margin for output overload, the
current-sense shunt resistors are chosen for a peak current limit of approximately 15 A:
R12 / R13 =
(0.5 - 0.185) V
= 21mW
15A
(23)
A standard value of 20 mΩ was used.
8.2.3 Application Curves
Figure 31 shows the measured efficiency as a function of load current and input voltage.
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Figure 32 illustrates the switching nodes of the two interleaved phases, and the resulting output ripple at twice
the switching frequency. This was measured at Vin of 24 V, where duty cycle is approx. 50% and maximum
ripple-cancellation occurs.
Figure 31. Efficiency vs. Load and Vin
Figure 32. Output Ripple at 24-V Vin
9 Power Supply Recommendations
The VCC pin requires a local decoupling capacitor to ground for stability of the internal regulator from the VIN
pin. This decoupling capacitor also provides the current pulses to drive the gates of the external MOSFETs
through the driver output pins. The decoupling capacitor should be placed close to the VCC and GND1/GNS2
pins, and should be tracked directly to the pins.
The two ground pins (GND1 and GND2) must be connected together with a short direct connection.
10 Layout
10.1 Layout Guidelines
The LM5032 Current Sense and PWM comparators are very fast, and respond to short duration noise pulses.
The components at the CS, COMP, SS, DCL, UVLO, and the RT/SYNC pins should be as physically close as
possible to the IC, thereby minimizing noise pickup in the PC board tracks.
Layout considerations are critical for the current sense filter. If current sense transformers are used, both leads of
each transformer secondary should be routed to the sense filter components and to the IC pins. The ground side
of each transformer should be connected via a dedicated PC board track to its appropriate GND pin, rather than
through the ground plane.
If the current sense circuits employ sense resistors in the drive transistor sources, low inductance resistors
should be used. In this case, all the noise sensitive low current ground tracks should be connected in common
near the IC, and then a single connection made to the power ground (sense resistor ground point). The outputs
of the LM5032 should have short direct paths to the power MOSFETs in order to minimize inductance in the PC
board traces.
The two ground pins (GND1, GND2) must be connected together with a short direct connection to avoid jitter due
to relative ground bounce in the operation of the two regulators.
If the internal dissipation of the LM5032 produces high junction temperatures during normal operation, the use of
wide PC board traces can help conduct heat away from the IC. Judicious positioning of the PC board within the
end product, along with use of any available air flow (forced or natural convection) can help reduce the junction
temperatures.
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10.2 Layout Example
From EA2
From CS2
From VIN
VIN
From EA1
From CS1
COMP1
CS1
To GD1
RT/SYNC
DCL
COMP2
SS1 LM5032
CS2
UVLO
SS2
VCC
RES
OUT1
OUT2
GND1
GND2
To GD2
Top-side copper
Bottom-side copper
Figure 33. Layout Example
11 Device and Documentation Support
11.1 Trademarks
All trademarks are the property of their respective owners.
11.2 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.3 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM5032MTC/NOPB
ACTIVE
TSSOP
PW
16
92
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
LM5032
MTC
LM5032MTCX/NOPB
ACTIVE
TSSOP
PW
16
2500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
LM5032
MTC
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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