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LM5164-Q1
SNVSB51A – SEPTEMBER 2018 – REVISED MARCH 2019
LM5164-Q1 100-V Input, 1-A synchronous buck DC/DC converter with ultra-low IQ
1 Features
3 Description
•
The LM5164-Q1 synchronous buck converter is
designed to regulate over a wide input voltage range,
minimizing the need for external surge suppression
components. A minimum controllable on-time of 50 ns
facilitates large step-down conversion ratios, enabling
the direct step-down from a 48-V nominal input to
low-voltage rails for reduced system complexity and
solution cost. The LM5164-Q1 operates during input
voltage dips as low as 6 V, at nearly 100% duty cycle
if needed, making it an excellent choice for highperformance 48-V battery automotive applications
and MHEV/EV systems.
1
•
•
•
•
AEC-Q100-qualified for automotive applications
– Device temperature grade 1: –40°C to +125°C,
ambient temperature range
Designed for reliable and rugged applications
– Wide input voltage range of 6 V to 100 V
– Junction temperature range: –40°C to +150°C
– Fixed 3-ms internal soft-start timer
– Peak and valley current-limit protection
– Input UVLO and thermal shutdown protection
Suited for scalable automotive power supplies
– Low minimum on- and off-times of 50 ns
– Adjustable switching frequency up to 1 MHz
– Diode emulation for high light-load efficiency
– 10.5-µA no-load input quiescent current
– 3-µA shutdown quiescent current
– Optimized for CISPR 25 EMI standard
Integration reduces solution size and cost
– COT mode control architecture
– Integrated 0.725-Ω NFET buck switch supports
wide duty cycle range
– Integrated 0.34-Ω NFET synchronous rectifier
eliminates external Schottky diode
– 1.2-V internal voltage reference
– No loop compensation components
– Internal VCC bias regulator and boot diode
Create a custom design using WEBENCH® power
designer
With integrated high-side and low-side power
MOSFETs, the LM5164-Q1 delivers up to 1-A of
output current. A constant on-time (COT) control
architecture provides nearly constant switching
frequency with excellent load and line transient
response. Additional features of the LM5164-Q1
include ultra-low IQ and diode emulation mode
operation for high light-load efficiency, innovative
peak and valley overcurrent protection, integrated
VCC bias supply and bootstrap diode, precision
enable and input UVLO, and thermal shutdown
protection with automatic recovery. An open-drain
PGOOD indicator provides sequencing, fault
reporting, and output voltage monitoring.
The LM5164-Q1 is qualified to automotive AEC-Q100
grade 1 and is available in a 8-pin SO PowerPAD™
package. Its 1.27-mm pin pitch provides adequate
spacing for high-voltage applications.
Device Information(1)
PART NUMBER
LM5164-Q1
2 Applications
Automotive 48-V mild hybrid ECU bias supplies
Automotive DC/DC converters
Automotive HVAC compressors and PTC heaters
Typical Application
LO
68 µH
U1
VIN = 6 V...100 V
VIN
EN/UVLO
RON
BST
GND
100
VOUT = 12 V
IOUT = 1 A
CBST
2.2 nF RFB1
448 k:
90
PGOOD
80
COUT
47 µF
FB
RRON
100 k:
4.89 mm × 3.90 mm
Typical Application Efficiency, VOUT = 12 V
SW
LM5164-Q1
CIN
2.2 µF
BODY SIZE (NOM)
SO PowerPAD (8)
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
RFB2
49.9 k:
Efficiency (%)
•
•
•
PACKAGE
70
60
50
VIN = 14V
VIN = 24V
VIN = 48V
VIN = 72V
40
*VOUT tracks VIN if VIN < 12 V
30
0.001
0.01
0.1
1
Load (A)
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM5164-Q1
SNVSB51A – SEPTEMBER 2018 – REVISED MARCH 2019
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
5
5
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics .............................................
Detailed Description .............................................. 9
7.1
7.2
7.3
7.4
Overview .................................................................. 9
Functional Block Diagram ....................................... 10
Feature Description................................................. 10
Device Functional Modes........................................ 15
8
Application and Implementation ........................ 16
8.1 Application Information............................................ 16
8.2 Typical Application .................................................. 16
9 Power Supply Recommendations...................... 22
10 Layout................................................................... 23
10.1 Layout Guidelines ................................................. 23
10.2 Layout Example .................................................... 25
11 Device and Documentation Support ................. 26
11.1
11.2
11.3
11.4
11.5
11.6
11.7
Device Support......................................................
Related Documentation .......................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
26
26
27
27
27
27
27
12 Mechanical, Packaging, and Orderable
Information ........................................................... 27
4 Revision History
Changes from Original (September 2018) to Revision A
•
2
Page
First release of production-data data sheet............................................................................................................................ 1
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SNVSB51A – SEPTEMBER 2018 – REVISED MARCH 2019
5 Pin Configuration and Functions
DDA Package
8-Pin SO PowerPAD
Top View
GND
SW
VIN
BST
EP
EN/UVLO
PGOOD
RON
FB
Pin Functions
PIN
NO.
NAME
I/O (1)
DESCRIPTION
1
GND
G
Ground connection for internal circuits.
2
VIN
P/I
Regulator supply input pin to high-side power MOSFET and internal bias regulator. Connect
directly to the input supply of the buck converter with short, low impedance paths.
3
EN/UVLO
I
Precision enable and undervoltage lockout (UVLO) programming pin. If the EN/UVLO voltage is
below 1.1 V, the converter is in the shutdown mode with all functions disabled. If the UVLO voltage
is greater than 1.1 V and below 1.5 V, the converter is in standby mode with the internal VCC
regulator operational and no switching. If the EN/UVLO voltage is above 1.5 V, the start-up
sequence begins.
4
RON
I
On-time programming pin. A resistor between this pin and GND sets the buck switch on-time.
5
FB
I
Feedback input of voltage regulation comparator.
6
PGOOD
O
Power good indicator. This pin is an open-drain output pin. Connect to a source voltage through an
external pullup resistor between 10 kΩ to 100 kΩ
7
BST
P/I
Bootstrap gate-drive supply. Required to connect a high-quality 2.2-nF 50-V X7R ceramic capacitor
between BST and SW to bias the internal high-side gate driver.
8
SW
P
Switching node that is internally connected to the source of the high-side NMOS buck switch and
the drain of the low-side NMOS synchronous rectifier. Connect to the switching node of the power
inductor.
—
EP
—
Exposed pad of the package. No internal electrical connection. Solder the EP to the GND pin and
connect to a large copper plane to reduce thermal resistance.
(1)
G = Ground, I = Input, O = Output, P = Power
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SNVSB51A – SEPTEMBER 2018 – REVISED MARCH 2019
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6 Specifications
6.1 Absolute Maximum Ratings
Over the recommended operating junction temperature range of –40°C to +150°C (unless otherwise noted) (1)
Input voltage
Bootstrap
capacitor
MIN
MAX
VIN to GND
–0.3
100
EN to GND
–0.3
100
FB to GND
–0.3
5.5
RON to GND
–0.3
5.5
1.5
2.5
BST to GND
–0.3
105.5
BST to SW
–0.3
5.5
SW to GND
–1.5
100
External BST to SW capacitance
Output voltage
SW to GND (20-ns transient)
UNIT
V
nF
V
–3
PGOOD to GND
–0.3
14
Operating junction temperature, TJ
–40
150
°C
Storage temperature, Tstg
–65
150
°C
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
VALUE
Electrostatic
discharge
V(ESD)
(1)
Human body model (HBM), per AEC-Q100-002
HBM ESD Classification Level 2, all pins (1)
±2000
Charged device model (CDM), per AEC-Q100-011
CDM ESD Classification level C4B. All pins except 1, 4, 5, and 8
±500
Charged device model (CDM), per AEC-Q100-011
CDM ESD Classification level C4B. Pins 1, 4, 5, and 8
±750
UNIT
V
V
AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
Over the recommended operating junction temperature range of –40°C to +150°C (unless otherwise noted)(1)
MIN
MAX
UNIT
100
V
Switch node voltage
100
V
Enable voltage
100
V
1.25
A
1000
kHz
VIN
Input voltage
VSW
VEN/UVLO
ILOAD
Load current
FSW
Switching frequency
CBST
External BST to SW capacitance
tON
Programmable on-time
4
NOM
6
1
2.2
50
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nF
10000
ns
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6.4 Thermal Information
LM5164-Q1
THERMAL METRIC (1)
DDA (SOIC)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
41.1
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
37.3
°C/W
RθJB
Junction-to-board thermal resistance
30.6
°C/W
ΨJT
Junction-to-top characterization parameter
6.7
°C/W
ΨJB
Junction-to-board characterization parameter
24.4
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
2.4
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
6.5 Electrical Characteristics
Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over the full –40°C to 150°C junction
temperature range unless otherwise indicated. VIN = 24 V and VEN/UVLO = 2 V unless otherwise stated.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
3
15
µA
SUPPLY CURRENT
IQ-SHUTDOWN
VIN shutdown current
VEN = 0 V
IQ-SLEEP1
VIN sleep current
VEN = 2.5 V, VFB = 1.5 V
10.5
25
µA
IQ-ACTIVE
VIN active current
VEN = 2.5 V
600
880
µA
VSD-RISING
Shutdown threshold
VEN/UVLO rising
1.1
V
VSD-FALLING
Shutdown threshold
VEN/UVLO falling
0.45
VEN-RISING
Enable threshold
VEN/UVLO rising
1.45
1.5
1.55
V
VEN-FALLING
Enable threshold
VEN/UVLO falling
1.35
1.4
1.44
V
FB regulation voltage
VFB falling
1.181
1.2
1.218
V
tON1
On-time1
VVIN = 6 V, RRON = 75 kΩ
5000
ns
tON2
On-time2
VVIN = 6 V, RRON = 25 kΩ
650
ns
tON3
On-time3
VVIN = 12 V, RRON = 75 kΩ
2550
ns
tON4
On-time4
VVIN = 12 V, RRON = 25 kΩ
830
ns
VPG-UTH
FB upper threshold for PGOOD high
to low
VFB rising
1.105
1.14
1.175
V
VPG-LTH
FB lower threshold for PGOOD high to
VFB falling
low
1.055
1.08
1.1
V
VPG-HYS
PGOOD upper and lower threshold
hysteresis
VFB falling
60
mV
RPG
PGOOD pulldown resistance
VFB = 1 V
30
Ω
Gate drive UVLO
VBST rising
2.7
EN/UVLO
V
FEEDBACK
VREF
TIMING
PGOOD
BOOTSTRAP
VBST-UV
3.4
V
POWER SWITCHES
RDSON-HS
High-side MOSFET RDSON
ISW = –100 mA
0.725
Ω
RDSON-LS
Low-side MOSFET RDSON
ISW = 100 mA
0.33
Ω
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Electrical Characteristics (continued)
Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over the full –40°C to 150°C junction
temperature range unless otherwise indicated. VIN = 24 V and VEN/UVLO = 2 V unless otherwise stated.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
1.75
3
4.75
ms
1.5
1.75
A
1.75
SOFT START
tSS
Internal soft-start time
CURRENT LIMIT
IPEAK1
Peak current limit threshold (HS)
1.25
IPEAK2
Peak current limit threshold (LS)
1.25
1.5
IDELTA-ILIM
Min of (IPEAK1 or IPEAK2) minus IVALLEY
200
300
IVALLEY
Valley current limit threshold
0.95
1.2
A
mA
1.4
A
THERMAL SHUTDOWN
TSD
Thermal shutdown threshold
TSD-HYS
Thermal shutdown hysteresis
6
TJ rising
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175
°C
10
°C
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6.6 Typical Characteristics
At TA = 25°C, VOUT = 12 V, LO = 68 µH, RRON = 105 kΩ, unless otherwise specified. See Figure 12.
100
100
90
90
Efficiency (%)
Efficiency (%)
80
70
60
80
50
70
VIN = 14V
VIN = 24V
VIN = 48V
VIN = 72V
40
30
0.001
VIN = 14V
VIN = 24V
VIN = 48V
VIN = 72V
60
0.01
0.1
1
0
0.2
0.4
Load (A)
Figure 1. Conversion Efficiency (Log Scale)
0.8
1
Figure 2. Conversion Efficiency (Linear Scale)
20
20
Sleep
Shutdown
18
Sleep
Shutdown
18
16
Quiescent Current (uA)
16
14
Current (PA)
0.6
Load (A)
12
10
8
6
4
14
12
10
8
6
4
2
2
0
-50
0
-25
0
25
50
75
100
Junction Temperature (qC)
125
150
0
10
20
30
D005
Figure 3. VIN Shutdown and Sleep Supply Current vs
Temperature
40
50
60
Input Voltage (V)
70
80
90
100
D006
Figure 4. VIN Shutdown and Sleep Supply Current vs Input
Voltage
600
725
590
700
Active Current (PA)
Active Current (PA)
580
675
650
625
600
570
560
550
540
530
520
575
550
-50
510
500
-25
0
25
50
75
100
Junction Temperature (qC)
125
150
0
10
D007
Figure 5. VIN Active Current vs Temperature
20
30
40
50
60
Input Voltage (V)
70
80
90
100
D008
Figure 6. VIN Active Current vs Input Voltage
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Typical Characteristics (continued)
At TA = 25°C, VOUT = 12 V, LO = 68 µH, RRON = 105 kΩ, unless otherwise specified. See Figure 12.
1.21
1.4
1
RDSON (:)
FB Regulation Threshold (V)
1.2
1.205
1.2
0.8
0.6
0.4
1.195
0.2
-25
0
25
50
75
100
Junction Temperature (qC)
125
0
-50
150
-25
25
50
75
100
Junction Temperature (qC)
125
150
D010
Figure 7. Feedback Comparator Threshold vs Temperature
Figure 8. MOSFETs On-State Resistance vs Temperature
1.6
7
1.5
6
1.3
1.2
4
3
2
1
Peak Current
Valley Current
1
-50
RRT = 105 k:
RRT = 43.2 k:
5
1.4
1.1
0
-25
0
25
50
75
100
Junction Temperature (qC)
125
150
0
10
D011
Figure 9. Peak and Valley Current Limit vs Temperature
8
0
D009
ON-Time (Ps)
Current Limit (A)
1.19
-50
High-Side FET
Low-Side FET
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20
30
40
50
60
Input Voltage (V)
70
80
90
100
D012
Figure 10. COT On-Time vs VIN
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7 Detailed Description
7.1 Overview
The LM5164-Q1 is an easy-to-use, ultra-low IQ constant on-time (COT) synchronous step-down buck regulator.
With integrated high-side and low-side power MOSFETs, the LM5164-Q1 is a low-cost, highly efficient buck
converter that operates from a wide input voltage of 6 V to 100 V, delivering up to 1-A DC load current. The
LM5164-Q1 is available an 8-pin SO Power PAD package with 1.27-mm pin pitch for adequate spacing in highvoltage applications.. This constant on-time (COT) converter is ideal for low-noise, high-current, and fast load
transient requirements, operating with a predictive on-time switching pulse. Over the input voltage range, input
voltage feedforward is employed to achieve a quasi-fixed switching frequency. A controllable on-time as low as
50 ns permits high step-down ratios and a minimum forced off-time of 50 ns provides extremely high duty cycles
allowing VIN to drop close to VOUT before frequency foldback occurs. At light loads the device transitions into an
ultra-low IQ mode to maintain high efficiency and prevent draining battery cells connected to the input when the
system is in standby. The LM5164-Q1 implements a smart peak and valley current limit detection circuit to
ensure robust protection during output short circuit conditions. Control loop compensation is not required for this
regulator, reducing design time and external component count.
The LM5164-Q1 incorporates additional features for comprehensive system requirements, including an opendrain Power Good circuit for power-rail sequencing and fault reporting, internally-fixed soft start, monotonic startup into prebiased loads, precision enable for programmable line undervoltage lockout (UVLO), smart cycle-bycycle current limit for optimal inductor sizing, and thermal shutdown with automatic recovery. These features
enable a flexible and easy-to-use platform for a wide range of applications. The LM5164-Q1 supports a wide
range of end-equipment systems requiring a regulated output from a high input supply where the transient
voltage deviates from its DC level. Examples of such end equipment systems are 48-V automotive systems, high
cell-count battery-pack systems, 24-V industrial systems, and 48-V telecom and PoE voltage ranges. The pin
arrangement is designed for a simple layout requiring only a few external components.
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7.2 Functional Block Diagram
VIN
VIN
VDD
BIAS
REGULATOR
CIN
VDD UVLO
RUV1
EN/UVLO
±
+
RUV2
STANDBY
THERMAL
SHUTDOWN
1.5 V
±
+
BST
SHUTDOWN
LOGIC
0.4 V
VIN
CBST
RON
ON/OFF
TIMERS
CONSTANT
ON-TIME
CONTROL
LOGIC
RFB1
FB
FEEDBACK
COMPARATOR
VREF
VOUT
VCC
SLEEP
DETECT
±
+
RFB2
LO
SW
ZX DETECT
COUT
ZC
PGOOD
±
+
VOUT
RRON
DISABLE
PEAK/VALLEY
CURRENT LIMIT
FB
GND
±
+
PGOOD
0.9*VREF COMPARATOR
7.3 Feature Description
7.3.1 Control Architecture
The LM5164-Q1 step-down switching converter employs a constant on time (COT) control scheme. The COT
control scheme sets a fixed on-time tON of the high-side FET using a timing resistor (RON). The tON is adjusted as
Vin changes and is inversely proportion to input voltage to maintain a fixed frequency when in continuous
conduction mode (CCM). After expiration of tON, the high side FET remains off until the feedback pin is equal or
below the reference voltage of 1.2 V. In order to maintain stability, the feedback comparator requires a minimal
ripple voltage that is in phase with the inductor current during the off-time. Furthermore, this change in feedback
voltage during the off-time must be large enough to dominate any noise present at the feedback node. The
minimum recommended ripple voltage is 20 mV. Refer to Table 1 for different types of ripple injection schemes
that ensure stability over the full input voltage range.
During a rapid start-up or a positive load step, the regulator operates with minimum off-times until regulation is
achieved. This feature enables extremely fast load transient response with minimum output voltage undershoot.
When regulating the output in steady-state operation, the off-time automatically adjusts itself to produce the SWpin duty cycle required for output voltage regulation to maintain a fixed switching frequency. In CCM the
switching frequency FSW is programmed by the RRON resistor. Use Equation 1 to calculate the switching
frequency.
10
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Feature Description (continued)
VOUT (V) ˜ 2500
RRON (k:)
FSW (kHz)
(1)
Table 1. Ripple Generation Methods
TYPE 1
TYPE 2
TYPE 3
Lowest Cost
Reduced Ripple
Minimum Ripple
LO
VIN
VIN
LM5164
CIN
VOUT
LO
VIN
VIN
SW
EN/UVLO
LM5164
CBST
BST
RFB 1
CIN
RESR
VOUT
LO
VIN
SW
EN/UVLO
VIN
BST
RA
LM5164
CBST
CFF
RFB 1
CIN
RESR
VOUT
SW
EN/UVLO
BST
CA
CBST
RFB1
CB
RON
FB
GND
PGOOD
RESR
RESR
20mV ˜ VOUT
t
VFB1 ˜ 'IL(nom)
VOUT
t
2 ˜ VIN ˜ FSW ˜ COUT
FB
RON
RFB2
RRON
COUT
RFB2
RRON
GND
RESR
(2)
(3)
CFF t
VOUT
2 ˜ VIN ˜ FSW ˜ COUT
2S ˜ FSW
1
˜ (RFB1 || RFB2 )
RON
FB
GND
PGOOD
RRON
PGOOD
20mV
t
'IL(nom)
RESR t
COUT
CA t
COUT
RFB2
10
FSW ˜ (RFB1 || RFB2 )
(7)
(4) R C d
A A
VIN-nom
(5)
VOUT ˜ t ON
@VIN-nom
20mV
(8)
t TR-settling
(6) CB t 3 ˜ R
FB1
(9)
Table 1 presents 3 different methods for generating appropriate voltage ripple at the feedback node. Type-1
ripple generation method uses a single resistor, RESR in series with the output capacitor. The generated voltage
ripple has two components, capacitive ripple caused by the inductor ripple current charging and discharging the
output capacitor and resistive ripple caused by the inductor ripple current flowing into the output capacitor and
through series resistance RESR. The capacitive ripple component is out of phase with the inductor current and
does not decrease monotonically during the off-time. The resistive ripple component is in phase with the inductor
current and decreases monotonically during the off-time. The resistive ripple must exceed the capacitive ripple at
VOUT for stable operation. If this condition is not satisfied, unstable switching behavior is observed in COT
converters, with multiple on-time bursts in close succession followed by a long off time. Equation 2 and
Equation 3 define the value of the series resistance RESR to ensure sufficient in-phase ripple at the feedback
node.
Type-2 ripple generation uses a CFF capacitor in addition to the series resistor. As the output voltage ripple is
directly AC-coupled by CFF to the feedback node, the RESR and ultimately the output voltage ripple are reduced
by a factor of VOUT / VFB1.
Type-3 ripple generation uses an RC network consisting of RA and CA, and the switch node voltage to generate a
triangular ramp that is in-phase with the inductor current. This triangular wave is the AC-coupled into the
feedback node with capacitor CB. Because this circuit does not use output voltage ripple, it is suited for
applications where low output voltage ripple is critical. TI application note AN-1481 Controlling Output ripple and
achieving ESR independence in constant on-time (COT) regulator designs provides additional details on this
topic.
Diode emulation mode (DEM) prevents negative inductor current, and pulse skipping maintains highest efficiency
at light load currents by decreasing the effective switching frequency. DEM operation occurs when the
synchronous power MOSFET switches off as inductor valley current reaches zero. Here, the load current is less
than half of the peak-to-peak inductor current ripple in CCM. Turning off the low-side MOSFET at zero current
reduces switching loss, and preventing negative current conduction reduces conduction loss. Power conversion
efficiency is thus higher in a DEM converter than an equivalent forced-PWM CCM converter. With DEM
operation, the duration that both power MOSFETs remain off progressively increases as load current decreases.
When this idle duration exceeds 15 μs, the converter transitions into an ultra-low IQ mode, consuming only 10-μA
quiescent current from the input.
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Feature Description (continued)
7.3.2 Internal VCC Regulator and Bootstrap Capacitor
The LM5164-Q1 contains an internal linear regulator that is powered from VIN with a nominal output of 5 V,
eliminating the need for an external capacitor to stabilize the linear regulator. The internal VCC regulator supplies
current to internal circuit blocks including the synchronous FET driver and logic circuits. The input pin (VIN) can
be connected directly to line voltages up to 100 V. As the power MOSFET has a low total gate charge, use a low
bootstrap capacitor value to reduce the stress on the internal regulator. It is required to select a high-quality 2.2nF 50-V X7R ceramic bootstrap capacitor as specified in the Absolute Maximum Ratings. Selecting a higher
value capacitance stresses the internal VCC regulator and damages the device. A lower capacitance than
required may not be sufficient to drive the internal gate of the power MOSFET. An internal diode connects from
the VCC regulator to the BST pin to replenish the charge in the high-side gate drive bootstrap capacitor when the
SW voltage is low.
7.3.3 Regulation Comparator
The feedback voltage at FB is compared to an internal 1.2-V reference. The LM5164-Q1 voltage regulation loop
regulates the output voltage by maintaining the FB voltage equal to the internal reference voltage, VREF. A
resistor divider programs the ratio from output voltage VOUT to FB. For a target VOUT setpoint, calculate RFB2
based on the selected RFB1 using Equation 10.
1.2 V
˜ RFB1
VOUT 1.2 V
RFB2
(10)
TI recommends selecting RFB1 in the range of 100 kΩ to 1 MΩ for most applications. A larger RFB1 consumes
less DC current, which is mandatory if light-load efficiency is critical. RFB1 larger than 1 MΩ is not recommended
as the feedback path becomes more susceptible to noise. It is important to route the feedback trace away from
the noisy area of the PCB and keep the feedback resistors close to the FB pin.
7.3.4 Internal Soft Start
The LM5164-Q1 employs an internal soft-start control ramp that allows the output voltage to gradually reach a
steady-state operating point, thereby reducing start-up stresses and current surges. The soft-start feature
produces a controlled, monotonic output voltage start-up. The soft-start time is internally set to 3 ms.
7.3.5 ON-Time Generator
The on-time of the LM5164-Q1 high-side FET is determined by the RRON resistor and is inversely proportional to
the input voltage, VIN. The inverse relationship with VIN results in a nearly constant frequency as VIN is varied.
Calculate the on-time using Equation 11.
t ON
V
RRON k:
VIN V ˜ 2.5
(11)
Determine the RRON resistor using Equation 12 to set a specific switching frequency in CCM.
RRON (k:)
VOUT (V) ˜ 2500
FSW (kHz)
(12)
Select RRON for a minimum on-time (at maximum VIN) greater than 50 ns for proper operation. In addition to this
minimum on-time, the maximum frequency for this device is limited to 1 MHz.
7.3.6 Current Limit
The LM5164-Q1 manages overcurrent conditions with cycle-by-cycle current limiting of the peak inductor current.
The current sensed in the high-side MOSFET is compared every switching cycle to the current limit threshold
(1.5 A). To protect the converter from potential current runaway conditions, the LM5164-Q1 includes a fold-back
valley current limit feature, set at 1.2 A, that is enabled if a peak current limit is detected. As shown in Figure 11,
if the peak current in the high-side MOSFET exceeds 1.5 A (typical), the present cycle is immediately terminated
12
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Feature Description (continued)
regardless of the programmed on-time (tON), the high-side MOSFET is turned off and the fold-back valley current
limit is activated. The low-side MOSFET remains on until the inductor current drops below this fold-back valley
current limit, after which the next on-pulse is initiated. This method folds back the switching frequency to prevent
overheating and limits the average output current to less than 1.5 A to ensure proper short-circuit and heavy-load
protection of the LM5164-Q1.
vFB
VREF
iL
Peak ILIM
IAVG(ILIM)
Valley ILIM
IAVG1
t
tON
< tON
tSW
> tSW
Figure 11. Current Limit Timing Diagram
Current is sensed after a leading-edge blanking time following the high-side MOSFET turnon transition. The
propagation delay of the current limit comparator is 100 ns. During high step-down conditions when the on-time
is less than 100 ns, a back-up peak current limit comparator in the low-side FET also set at 1.5 A will enable the
fold-back valley current limit set at 1.2 A. This innovative current limit scheme enables ultra-low duty-cycle
operation permitting large step down voltage conversions while ensuring robust protection of the converter.
7.3.7 N-Channel Buck Switch and Driver
The LM5164-Q1 integrates an N-channel buck switch and associated floating high-side gate driver. The gatedriver circuit works in conjunction with an external bootstrap capacitor and an internal high-voltage bootstrap
diode. A high-quality 2.2-nF, 50-V X7R ceramic capacitor connected between the BST and SW pins provides the
voltage to the high-side driver during the buck switch on-time. See Internal VCC Regulator and Bootstrap
Capacitor for limitations. During the off-time, the SW pin is pulled down to approximately 0 V, and the bootstrap
capacitor charges from the internal VCC through the internal bootstrap diode. The minimum off-timer, set to 50
ns (typical), ensures a minimum time each cycle to recharge the bootstrap capacitor. When the on-time is less
than 300ns, the minimum off-timer is forced to 250 ns to ensure that the BST capacitor is charged in a single
cycle. This is vital during wake up from sleep mode when the BST capacitor is most likely discharged.
7.3.8 Synchronous Rectifier
The LM5164-Q1 provides an internal low-side synchronous rectifier N-channel MOSFET. This MOSFET provides
a low-resistance path for the inductor current to flow when the high-side MOSFET is turned off.
The synchronous rectifier operates in a diode emulation mode. Diode emulation enables the regulator to operate
in a pulse-skipping mode during light load conditions. This mode leads to a reduction in the average switching
frequency at light loads. Switching losses and FET gate driver losses, both of which are proportional to switching
frequency, are significantly reduced at very light loads and efficiency is improved. This pulse-skipping mode also
reduces the circulating inductor current and losses associated with conventional CCM at light loads.
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Feature Description (continued)
7.3.9 Enable/Undervoltage Lockout (EN/UVLO)
The LM5164-Q1 contains a dual-level EN/UVLO circuit. When the EN/UVLO voltage is below 1.1 V (typical), the
converter is in a low-current shutdown mode and the input quiescent current (IQ) is dropped down to 3 µA. When
the voltage is greater than 1.1 V but less than 1.5 V (typical), the converter is in standby mode. In standby mode
the internal bias regulator is active while the control circuit is disabled. When the voltage exceeds the rising
threshold of 1.5 V (typical), normal operation begins. Install a resistor divider from VIN to GND to set the
minimum operating voltage of the regulator. Use Equation 13 and Equation 14 to calculate the input UVLO
turnon and turnoff voltages, respectively.
VIN(on)
VIN(off)
§
RUV1 ·
1.5 V ˜ ¨ 1
¸
© RUV2 ¹
(13)
§
RUV1 ·
1.4 V ˜ ¨ 1
¸
© RUV2 ¹
(14)
TI recommends selecting RUV1 in the range of 1 MΩ for most applications. A larger RUV1 consumes less DC
current, which is mandatory if light-load efficiency is critical. If input UVLO is not required, the power-supply
designer can either drive EN/UVLO as an enable input driven by a logic signal or connect it directly to VIN. If
EN/UVLO is directly connected to VIN, the regulator begins switching as soon as the internal bias rails are
active.
7.3.10 Power Good (PGOOD)
The LM5164-Q1 provides a PGOOD flag pin to indicate when the output voltage is within the regulation level.
Use the PGOOD signal for start-up sequencing of downstream converters or for fault protection and output
monitoring. PGOOD is an open-drain output that requires a pullup resistor to a DC supply not greater than 14 V.
The typical range of pullup resistance is 10 kΩ to 100 kΩ. If necessary, use a resistor divider to decrease the
voltage from a higher voltage pullup rail. When the FB voltage exceeds 95% of the internal reference VREF, the
internal PGOOD switch turns off and PGOOD can be pulled high by the external pullup. If the FB voltage falls
below 90% of VREF, an internal 25-Ω PGOOD switch turns on and PGOOD is pulled low to indicate that the
output voltage is out of regulation. The rising edge of PGOOD has a built-in deglitch delay of 5 µs.
7.3.11 Thermal Protection
The LM5164-Q1 includes an internal junction temperature monitor to protect the device in the event of a higher
than normal junction temperature. If the junction temperature exceeds 175°C (typical), thermal shutdown occurs
to prevent further power dissipation and temperature rise. The LM5164-Q1 initiates a restart sequence when the
junction temperature falls to 165°C, based on a typical thermal shutdown hysteresis of 10°C. This is a nonlatching protection, and, as such, the device cycles into and out of thermal shutdown if the fault persists.
14
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7.4 Device Functional Modes
7.4.1 Shutdown Mode
EN/UVLO provides ON and OFF control for the LM5164-Q1. When VEN/UVLO is below approximately 1.1 V, the
device is in shutdown mode. Both the internal linear regulator and the switching regulator are off. The quiescent
current in shutdown mode drops to 3 µA at VIN = 24 V. The LM5164-Q1 also employs internal bias rail
undervoltage protection. If the internal bias supply voltage is below its UV threshold, the regulator remains off.
7.4.2 Active Mode
The LM5164-Q1 is in active mode when VEN/UVLO is above the precision enable threshold and the internal bias
rail is above its UV threshold. In COT active mode, the LM5164-Q1 is in one of three modes depending on the
load current:
1. CCM with fixed switching frequency when load current is above half of the peak-to-peak inductor current
ripple
2. Pulse skipping and diode emulation mode (DEM) when the load current is less than half of the peak-to-peak
inductor current ripple in CCM operation
3. Current limit CCM with peak and valley current limit protection when an overcurrent condition is applied at
the output.
7.4.3 Sleep Mode
Control Architecture gives a brief introduction to the LM5164-Q1 diode emulation (DEM) feature. The converter
enters DEM during light-load conditions when the inductor current decays to zero and the synchronous MOSFET
is turned off to prevent negative current in the system. In the DEM state, the load current is lower than half the
peak-to-peak inductor current ripple and the switching frequency decreases when the load is further decreased
as the device operates in a pulse skipping mode. A switching pulse is set when VFB drops below 1.2 V.
As the frequency of operation decreases and VFB remains above 1.2 V (VREF) with the output capacitor sourcing
the load current for greater than 15 µs, the converter enters an ultra-low IQ sleep mode to prevent draining the
input power supply. The input quiescent current (IQ) required by the LM5164-Q1 decreases to 10 µA in sleep
mode, improving the light-load efficiency of the regulator. In this mode all internal controller circuits are turned off
to ensure very low current consumption by the device. Such low IQ renders the LM5164-Q1 as the best option to
extend operating lifetime for off-battery applications. The FB comparator and internal bias rail are active to detect
when the FB voltage drops below the internal reference VREF and the converter transitions out of sleep mode into
active mode. There is a 9 µs wake-up delay from sleep to active states.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM5164-Q1 requires only a few external components to step down from a wide range of supply voltages to a
fixed output voltage. Several features are integrated to meet system design requirements, including precision
enable, input voltage UVLO, internal soft start, programmable switching frequency, and a PGOOD indicator.
To expedite and streamline the process of designing of a LM5164-Q1-based converter, a comprehensive
LM5164-Q1 Quickstart calculator is available for download to assist the designer with component selection for a
given application. This tool is complemented by the availability of an evaluation module (EVM), numerous
PSPICE models, as well as TI's WEBENCH® Power Designer.
8.2 Typical Application
For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation and test results of
an LM5164-Q1-powered implementation, see TI Designs reference design library.
The schematic of a 12-V, 1-A COT converter is shown in Figure 12.
VIN
SW
LM5164-Q1
CIN
2.2 PF
VOUT = 12 V
IOUT = 1 A
LO
68 PH
U1
VIN = 15 V...100 V
EN/UVLO
CBST
2.2 nF
CA
RA
453 k: 3.3 nF
BST
RON
FB
GND
PGOOD
CB
56 pF
RRON
100 k:
RFB1
453 k:
COUT
47 PF
RFB2
49.9 k:
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Figure 12. Typical Application VIN(nom) = 48 V, VOUT = 12 V, IOUT(max) = 1 A, FSW(nom) = 300 kHz
NOTE
This and subsequent design examples are provided herein to showcase the LM5164-Q1
converter in several different applications. Depending on the source impedance of the
input supply bus, an electrolytic capacitor may be required at the input to ensure stability,
particularly at low input voltage and high output current operating conditions. See Power
Supply Recommendations for more detail.
16
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8.2.1 Design Requirements
The target full-load efficiency is 92% based on a nominal input voltage of 48 V and an output voltage of 12 V.
The required input voltage range is 15 V to 100 V. The LM5164-Q1 delivers a fixed 12-V output voltage. The
switching frequency is set by resistor RRON at 300 kHz. The output voltage soft-start time is 3 ms. The required
components are listed in Table 2. Refer to the LM5164-Q1EVM-041 EVM user's guide for more detail.
Table 2. List of Components
COUNT
REF DES
VALUE
2
CIN
2.2 µF
Capacitor, Ceramic, 2.2µF, 100V, X7R, 10%
DESCRIPTION
CGA6N3X7R2A225K230AB
PART NUMBER
TDK
MANUFACTURER
1
COUT
22 µF
Capacitor, Ceramic, 22µF, 25V, X7R, 10%
TMK325B7226KMHT
Taiyo Yuden
1
CA
3300 pF
Capacitor, Ceramic, 3300pF, 16V, X7R, 10%
CGA3E2X7R2A332K080AA
TDK
1
CB
56 pF
Capacitor, Ceramic, 56pF, 50V, X7R, 10%
C0603C560J5GACTU
Kemet
1
CBST
2.2 nF
Capacitor, Ceramic, 2200pF, 50V, X7R, 10%
GCM155R71H222KA37D
MuRata
1
LF
68 µH
Inductor, 68 µH, 170 mΩ, >1.8A
MSS1246T-683MLB
Coilcraft
1
RRON
100 kΩ
Resistor, Chip, 100 k, 1%, 0.1 W, 0603
RG1608P-1053-B-T5
Susumu Co Ltd
1
RFB1
453 kΩ
Resistor, Chip, 453 k, 1%, 0.1 W, 0603
RT0603BRD07448KL
Yageo
1
RFB2
49.9 kΩ
Resistor, Chip, 49.9 k, 1%, 0.1 W, 0603
RG1608P-4992-B-T5
Susumu Co Ltd
1
RA
453 kΩ
Resistor, Chip, 453 k, 1%, 0.1W, 0603
RT0603BRD07453KL
Yageo
1
U1
Wide VIN synchronous buck converter
LM5164QDDARQ1
TI
8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM5164-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 Switching Frequency (RRON)
The switching frequency of LM5164-Q1 is set by the on-time programming resistor placed at RON. As shown by
Equation 15, a standard 100 kΩ, 1% resistor sets the switching frequency at 300 kHz.
RRON (k:)
VOUT (V) ˜ 2500
FSW (kHz)
(15)
Note that at very low duty cycles, the 50 ns minimum controllable on-time of the high-side MOSFET, tON(min),
limits the maximum switching frequency. In CCM, tON(min) limits the voltage conversion step-down ratio for a given
switching frequency. Calculate the minimum controllable duty cycle using Equation 16.
DMIN
t ON(min) ˜ FSW
(16)
Ultimately, the choice of switching frequency for a given output voltage affects the available input voltage range,
solution size and efficiency. The maximum supply voltage for a given tON(min) before switching frequency
reduction occurs is given by Equation 17.
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VIN(max)
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VOUT
t ON(min) ˜ FSW
(17)
8.2.2.3 Buck Inductor (LO)
The inductor ripple current (assuming CCM operation) and peak inductor current are given respectively by
Equation 18 and Equation 19.
'IL
VOUT §
VOUT ·
˜ ¨1
¸
FSW ˜ LO ©
VIN ¹
IL(peak)
IOUT(max)
(18)
'IL
2
(19)
For most applications, choose an inductance such that the inductor ripple current, ΔIL, is between 30% and 50%
of the rated load current at nominal input voltage. Calculate the inductance using Equation 20.
LO
§
VOUT
˜ ¨1
¨
V
IN(nom)
©
VOUT
FSW ˜ 'IL
·
¸
¸
¹
(20)
Choosing a 68-μH inductor in this design results in 447-mA peak-to-peak ripple current at a nominal input voltage
of 48 V, equivalent to 45% of the 1-A rated load current.
Check the inductor data sheet to make sure the saturation current of the inductor is well above the current limit
setting of the LM5164-Q1. Ferrite-core inductors have relatively lower core losses and are preferred at high
switching frequencies, but exhibit a hard saturation characteristic – the inductance collapses abruptly when the
saturation current is exceeded. This results in an abrupt increase in inductor ripple current, higher output voltage
ripple, and reduced efficiency in turn compromising reliability. Note that inductor saturation current levels
generally decrease as the core temperature increases.
8.2.2.4 Output Capacitor (COUT)
Select a ceramic output capacitor to limit the capacitive voltage ripple at the converter output. This is the
sinusoidal ripple voltage that is generated from the triangular inductor current ripple flowing into and out of the
capacitor. Select an output capacitance using Equation 21 to limit the voltage ripple component to 0.5% of the
output voltage.
COUT t
8 ˜ FSW
'IL
˜ VOUT(ripple)
(21)
Substituting ΔIL(nom) of 447 mA gives COUT greater than 3.1 μF. With voltage coefficients of ceramic capacitors
taken in consideration, a 22-µF, 25-V rated capacitor with X7R dielectric is selected.
8.2.2.5 Input Capacitor (CIN)
An input capacitor is necessary to limit the input ripple voltage while providing AC current to the buck power
stage at every switching cycle. To minimize the parasitic inductance in the switching loop, position the input
capacitors as close as possible to the VIN and GND pins of the LM5164-Q1. The input capacitors conduct a
square-wave current of peak-to-peak amplitude equal to the output current. It follows that the resultant capacitive
component of AC ripple voltage is a triangular waveform. Together with the ESR-related ripple component, the
peak-to-peak ripple voltage amplitude is given by Equation 22.
VIN(ripple)
IOUT ˜ D ˜ 1 D
FSW ˜ CIN
IOUT ˜ RESR
(22)
The input capacitance required for a load current, based on an input voltage ripple specification (ΔVIN), is given
by Equation 23:
18
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CIN t
IOUT ˜ D ˜ 1 D
FSW ˜ VIN(ripple)
IOUT ˜ RESR
(23)
The recommended high-frequency input capacitance is 2.2 µF or higher. Ensure the input capacitor is a highquality X7S or X7R ceramic capacitor with sufficient voltage rating for CIN. Based on the voltage coefficient of
ceramic capacitors, choose a voltage rating of twice the maximum input voltage. Additionally, some bulk
capacitance is required if the LM5164-Q1 is not located within approximately 5 cm from the input voltage source.
This capacitor provides parallel damping to the resonance associated with parasitic inductance of the supply
lines and high-Q ceramics. See Power Supply Recommendations for more detail.
8.2.2.6 Type 3 Ripple Network
A Type 3 ripple generation network uses an RC filter consisting of RA and CA across SW and VOUT to generate a
triangular ramp that is in phase with the inductor current. This triangular ramp is then AC-coupled into the
feedback node using capacitor CB as shown in Figure 12. Type 3 ripple injection is suited for applications where
low output voltage ripple is crucial.
Calculate RA and CA using Equation 24 and Equation 25 to provide the required ripple amplitude at the FB pin.
CA t
10
FSW ˜ RFB1 RFB2
(24)
For the feedback resistor values given in Figure 12, Equation 24 dictates a minimum CA of 742 pF. In this design,
a 3300 pF capacitance is chosen. This is done to keep RA within practical limits between 100 kΩ and 1 MΩ when
using Equation 25.
R A CA t
VIN(nom)
VOUT ˜ t ON(nom)
20mV
(25)
Based on CA set at 3.3 nF, RA is calculated to be 453 kΩ to provide a 20-mV ripple voltage at FB. The general
recommendation for a Type 3 network is to calculate RA and CA to get 20 mV of ripple at typical operating
conditions, while ensuring a 12-mV minimum ripple voltage on FB at minimum VIN.
While the amplitude of the generated ripple does not affect the output voltage ripple, it impacts the output
regulation as it reflects as a DC error of approximately half the amplitude of the generated ripple. For example, a
converter circuit with Type 3 network that generates a 40-mV ripple voltage at the feedback node has
approximately 10-mV worse load regulation scaled up through the FB divider to VOUT than the same circuit that
generates a 20-mV ripple at FB. Calculate the coupling capacitance CB using Equation 26.
t TR-settling
CB t
3 ˜ RFB1
where
•
tTR-settling is the desired load transient response settling time
(26)
CB calculates to 56 pF based on a 75-µs settling time. This value avoids excessive coupling capacitor discharge
by the feedback resistors during sleep intervals when operating at light loads. To avoid capacitance fall-off with
DC bias, use a C0G or NP0 dielectric capacitor for CB.
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8.2.3 Application Curves
100
100
90
90
Efficiency (%)
Efficiency (%)
80
70
60
50
70
VIN = 14V
VIN = 24V
VIN = 48V
VIN = 72V
40
30
0.001
80
VIN = 14V
VIN = 24V
VIN = 48V
VIN = 72V
60
0.01
0.1
1
0
Load (A)
0.2
0.4
0.6
0.8
1
Load (A)
Figure 13. Conversion Efficiency (Log Scale)
Figure 14. Conversion Efficiency (Linear Scale)
12.4
12.3
Output Voltage (V)
12.2
12.1
12
11.9
VIN = 15V
VIN = 24V
VIN = 36V
VIN = 48V
VIN = 60V
11.8
11.7
11.6
0
0.2
0.4
0.6
Output Current (A)
0.8
1
VIN = 24 V
Figure 15. Load and Line Regulation Performance
VIN = 24 V
IOUT = 0 A
Figure 16. Load Step Response
VIN = 24 V
Figure 17. No-Load Start-up with VIN
20
IOUT = 0.25 A to 1 A at 0.1 A/μs
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IOUT = 1 A (Resistive)
Figure 18. Full-Load Start-up with VIN
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VIN = 24 V
IOUT = 0 A
Figure 19. No-Load Start-up and Shutdown with EN/UVLO
VIN = 24 V
IOUT = 0 A
VIN = 24 V
Figure 20. Full-Load Start-up and Shutdown with EN/UVLO
VIN = 24 V
Figure 21. Pre-bias Start-up with EN/UVLO
VIN = 24 V
Load = Short to 0 A
Figure 23. Short Circuit Recovery
IOUT = 1 A (Resistive)
Load = 0 A to Short
Figure 22. Short Circuit Applied
VIN = 24 V
Load = 0 A to Short to 0 A
Figure 24. No Load to Short Circuit/Short Circuit Recovery
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VIN = 24 V
IOUT = 0 A
VIN = 24 V
Figure 25. No-Load Switching
IOUT = 1 A
Figure 26. Full-Load Switching
9 Power Supply Recommendations
The LM5164-Q1 buck converter is designed to operate from a wide input voltage range between 6 V and 100 V.
The characteristics of the input supply must be compatible with the Absolute Maximum Ratings and
Recommended Operating Conditions tables. In addition, the input supply must be capable of delivering the
required input current to the fully-loaded regulator. Estimate the average input current with Equation 27.
VOUT ˜ IOUT
VIN ˜ K
IIN
where
•
η is the efficiency
(27)
If the converter is connected to an input supply through long wires or PCB traces with a large impedance, take
special care to achieve stable performance. The parasitic inductance and resistance of the input cables may
have an adverse affect on converter operation. The parasitic inductance in combination with the low-ESR
ceramic input capacitors form an underdamped resonant circuit. This circuit can cause overvoltage transients at
VIN each time the input supply is cycled ON and OFF. The parasitic resistance causes the input voltage to dip
during a load transient. If the converter is operating close to the minimum input voltage, this dip can cause false
UVLO fault triggering and a system reset. The best way to solve such issues is to reduce the distance from the
input supply to the regulator and use an aluminum electrolytic input capacitor in parallel with the ceramics. The
moderate ESR of the electrolytic capacitor helps to damp the input resonant circuit and reduce any voltage
overshoots. A 10-μF electrolytic capacitor with a typical ESR of 0.5 Ω provides enough damping for most input
circuit configurations.
An EMI input filter is often used in front of the regulator that, unless carefully designed, can lead to instability as
well as some of the effects mentioned above. The application report Simple Success with Conducted EMI for
DC-DC Converters (SNVA489) provides helpful suggestions when designing an input filter for any switching
regulator.
22
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10 Layout
10.1 Layout Guidelines
PCB layout is a critical portion of good power supply design. There are several paths that conduct high slew-rate
currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise and EMI or
degrade the power supply performance.
1. To help eliminate these problems, bypass the VIN pin to GND with a low-ESR ceramic bypass capacitor with
a high-quality dielectric. Place CIN as close as possible to the LM5164-Q1 VIN and GND pins. Grounding for
both the input and output capacitors should consist of localized top-side planes that connect to the GND pin
and GND PAD.
2. Minimize the loop area formed by the input capacitor connections to the VIN and GND pins.
3. Locate the inductor close to the SW pin. Minimize the area of the SW trace or plane to prevent excessive
capacitive coupling.
4. Tie the GND pin directly to the power pad under the device and to a heat-sinking PCB ground plane.
5. Use a ground plane in one of the middle layers as a noise shielding and heat dissipation path.
6. Have a single-point ground connection to the plane. Route the ground connections for the feedback, softstart, and enable components to the ground plane. This prevents any switched or load currents from flowing
in analog ground traces. If not properly handled, poor grounding results in degraded load regulation or erratic
output voltage ripple behavior.
7. Make VIN, VOUT and ground bus connections as wide as possible. This reduces any voltage drops on the
input or output paths of the converter and maximizes efficiency.
8. Minimize trace length to the FB pin. Place both feedback resistors, RFB1 and RFB2, close to the FB pin. Place
CFF (if needed) directly in parallel with RFB1. If output setpoint accuracy at the load is important, connect the
VOUT sense at the load. Route the VOUT sense path away from noisy nodes and preferably through a layer on
the other side of a grounded shielding layer.
9. The RON pin is sensitive to noise. Thus, locate the RRON resistor as close as possible to the device and
route with minimal lengths of trace. The parasitic capacitance from RON to GND must not exceed 20 pF.
10. Provide adequate heat sinking for the LM5164-Q1 to keep the junction temperature below 150°C. For
operation at full rated load, the top-side ground plane is an important heat-dissipating area. Use an array of
heat-sinking vias to connect the exposed pad to the PCB ground plane. If the PCB has multiple copper
layers, these thermal vias must also be connected to inner layer heat-spreading ground planes.
10.1.1 Compact PCB Layout for EMI Reduction
Radiated EMI generated by high di/dt components relates to pulsing currents in switching converters. The larger
area covered by the path of a pulsing current, the more electromagnetic emission is generated. The key to
minimizing radiated EMI is to identify the pulsing current path and minimize the area of that path.
The critical switching loop of the buck converter power stage in terms of EMI is denoted in Figure 27. The
topological architecture of a buck converter means that a particularly high di/dt current path exists in the loop
comprising the input capacitor and the integrated MOSFETs of the LM5164-Q1, and it becomes mandatory to
reduce the parasitic inductance of this loop by minimizing the effective loop area.
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Layout Guidelines (continued)
VIN
VIN
2
CIN
LM5164
High
di/dt
loop
BST
High-side
NMOS
gate driver
Q1
LO
8
Low-side
NMOS
gate driver
SW
VOUT
CO
Q2
1
GND
GND
Figure 27. DC/DC Buck Converter With Power Stage Circuit Switching Loop
The input capacitor provides the primary path for the high di/dt components of the high-side MOSFET's current.
Placing a ceramic capacitor as close as possible to the VIN and GND pins is the key to EMI reduction. Keep the
trace connecting SW to the inductor as short as possible and just wide enough to carry the load current without
excessive heating. Use short, thick traces or copper pours (shapes) for current conduction path to minimize
parasitic resistance. Place the output capacitor close to the VOUT side of the inductor, and connect the capacitor's
return terminal to the GND pin and exposed PAD of the LM5164-Q1.
10.1.2 Feedback Resistors
Reduce noise sensitivity of the output voltage feedback path by placing the resistor divider close to the FB pin,
rather than close to the load. This reduces the trace length of FB signal and noise coupling. The FB pin is the
input to the feedback comparator, and as such is a high impedance node sensitive to noise. The output node is a
low impedance node, so the trace from VOUT to the resistor divider can be long if a short path is not available.
Route the voltage sense trace from the load to the feedback resistor divider, keeping away from the SW node,
the inductor and VIN to avoid contaminating the feedback signal with switch noise, while also minimizing the trace
length. This is most important when high feedback resistances, greater than 100 kΩ, are used to set the output
voltage. Also, route the voltage sense trace on a different layer from the inductor, SW node and VIN, such that
there is a ground plane that separates the feedback trace from the inductor and SW node copper polygon. This
provides further shielding for the voltage feedback path from switching noise sources.
24
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10.2 Layout Example
Figure 28 shows an example layout for the PCB top layer of a 2-layer board with essential components placed
on the top side.
Type 3 ripple
injection
Connect BST cap
close to BST and SW
Place FB resistors very
close to FB & GND pins
PGOOD
connection
Thermal vias under
LM5164 PAD
Place resistor R8
close to the RON pin
VOUT
connection
Optional RC
GND
connection snubber to
reduce SW
node ringing
Connect ceramic EN/UVLO
input cap close to connection
VIN and GND
Figure 28. LM5164-Q1 Single-Sided PCB Layout Example
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
• LM5164-Q1 Quickstart Calculator
• LM5164-Q1 Simulation Models
• For TI's Reference Design Library, visit TIDesigns
• For TI's WEBENCH Design Environment, visit the WEBENCH® Design Center
11.1.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM5164-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Related Documentation
For related documentation see the following:
• LM5164-Q1EVM-041 EVM user's guide
• Selecting an ideal ripple generation network for your COT buck converter
• White Papers:
– Valuing wide VIN, low-EMI synchronous buck circuits for cost-effective, demanding applications
– An overview of conducted EMI specifications for power supplies
– An overview of radiated EMI specifications for power supplies
• TI Designs:
– TIDA-01395 24-V AC Power stage with wide VIN converter and battery gauge for smart thermostat
– TIDA-010030 Accurate gauging and 50-μA standby current, 13S, 48-V li-ion battery pack reference design
• Power House Blogs:
– Use a low-quiescent-current switcher for high-voltage conversion
• Behind the Wheel Blogs:
– How a DC/DC converter package and pinout design can enhance automotive EMI performance
• AN-2162: Simple success with conducted EMI from DC/DC converters
• Automotive cranking simulator user's guide
• Powering drones with a wide VIN DC/DC converter
• Using new thermal metrics
• Semiconductor and IC package thermal metrics
26
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11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.5 Trademarks
PowerPAD, E2E are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.6 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM5164QDDARQ1
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 150
L5164Q
LM5164QDDATQ1
ACTIVE SO PowerPAD
DDA
8
250
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 150
L5164Q
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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