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LMH2100
SNWS020C – NOVEMBER 2007 – REVISED OCTOBER 2015
LMH2100 50-MHz to 4-GHz 40-dB Logarithmic Power Detector for CDMA and WCDMA
1 Features
3 Description
•
•
•
•
•
•
•
•
The LMH2100 is a 40-dB RF power detector intended
for use in CDMA and WCDMA applications. The
device has an RF frequency range from 50 MHz to
4 GHz. It provides an accurate temperature and
supply compensated output voltage that relates
linearly to the RF input power in dBm. The circuit
operates with a single supply from 2.7 V to 3.3 V.
1
Supply Voltage: 2.7 V to 3.3 V
Output Voltage: 0.3 V to 2 V
40-dB Linear in dB Power Detection Range
Shutdown
Multi-Band Operation from 50 MHz to 4 GHz
0.5-dB Accurate Temperature Compensation
External Configurable Output Filter Bandwidth
0.4-mm Pitch DSBGA Package
2 Applications
•
•
•
•
UMTS/CDMA/WCDMA RF Power Control
GSM/GPRS RF Power Control
PA Modules
IEEE 802.11b, g (WLAN)
The LMH2100 has an RF power detection range from
−45 dBm to −5 dBm, and is ideally suited for direct
use in combination with a 30-dB directional coupler.
Additional low-pass filtering of the output signal can
be realized by means of an external resistor and
capacitor. Typical Application: Output RC Low Pass
Filter shows a detector with an additional output low
pass filter. The filter frequency is set with RS and CS.
Typical Application: Feedback (R)C Low Pass Filter
shows a detector with an additional feedback low
pass filter. Resistor RP is optional and will lower the
Trans impedance gain (RTRANS). The filter frequency
is set with CP//CTRANS and RP//RTRANS.
The device is active for Enable = High; otherwise it is
in a low power-consumption shutdown mode. To save
power and prevent discharge of an external filter
capacitance, the output (OUT) is high-impedance
during shutdown.
Device Information(1)
PART NUMBER
LMH2100
PACKAGE
BODY SIZE (MAX)
DSBGA (6)
1.274 mm × 0.874 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application: Output RC Low Pass Filter
RF
COUPLER
Typical Application: Feedback (R)C Low Pass
Filter
ANTENNA
PA
RF
50 :
COUPLER
VDD
RFIN
2
1
50 :
6
OUT
4
3
5
RS
REF
GND
VDD
+
CS
LMH2100
EN
ANTENNA
PA
ADC
RFIN
2
1
6
OUT
LMH2100
EN
4
3
5
+
RP
REF
CP
ADC
-
GND
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LMH2100
SNWS020C – NOVEMBER 2007 – REVISED OCTOBER 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
6.7
7
1
1
1
2
3
4
Absolute Maximum Ratings ..................................... 4
ESD Ratings.............................................................. 4
Recommended Operating Ratings ........................... 4
Thermal Information .................................................. 5
2.7-V DC and AC Electrical Characteristics.............. 5
Timing Requirements .............................................. 11
Typical Characteristics ............................................ 11
Detailed Description ............................................ 23
7.1 Overview ................................................................. 23
7.2 Functional Block Diagram ....................................... 23
7.3 Feature Description................................................. 23
7.4 Device Functional Modes........................................ 29
8
Application and Implementation ........................ 30
8.1 Application Information............................................ 30
8.2 Typical Applications ............................................... 33
9 Power Supply Recommendations...................... 39
10 Layout................................................................... 40
10.1 Layout Guidelines ................................................ 40
10.2 Layout Example .................................................... 42
11 Device and Documentation Support ................. 43
11.1
11.2
11.3
11.4
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
43
43
43
43
12 Mechanical, Packaging, and Orderable
Information ........................................................... 43
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision B (March 2013) to Revision C
•
Added Device Information and Pin Configuration and Functions sections, ESD Ratings table, Feature Description,
Device Functional Modes, Application and Implementation, Power Supply Recommendations, Layout, Device and
Documentation Support, and Mechanical, Packaging, and Orderable Information sections. ................................................ 1
Changes from Revision A (March 2013) to Revision B
•
2
Page
Page
Changed layout of National Data Sheet to TI format ........................................................................................................... 42
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5 Pin Configuration and Functions
YFQ Package
6-Pin DSBGA
Top View
A1
VDD
A2
OUT
B1
RFIN
B2
REF
C1
GND
C2
EN
Pin Functions
PIN
NUMBER
NAME
I/O
A1
VDD
Power Supply
A2
OUT
Output
B1
RFIN
Analog Input
B2
REF
Reference Output
C1
GND
GND
C2
EN
Logic Input
DESCRIPTION
Positive supply voltage
Ground referenced detector output voltage (linear in dB)
RF input signal to the detector, internally terminated with 50 Ω.
Reference output, for differential output measurement (without pedestal). Connected
to inverting input of output amplifier.
Power ground
The device is enabled for EN = High, and brought to a low-power shutdown mode for
EN = Low.
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SNWS020C – NOVEMBER 2007 – REVISED OCTOBER 2015
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2)
MIN
MAX
UNIT
3.6
V
RF input, input power
10
dBm
RF input, DC voltage
400
mV
Enable input voltage
VSS – 0.4 < VEN < VDD + 0.4
V
(3)
150
°C
Maximum lead temperature (soldering,10 sec)
260
°C
150
°C
Supply voltage, VDD to GND
Junction temperature
−65
Storage temperature, Tstg
(1)
(2)
(3)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. For ensured
specifications and the test conditions, see the 2.7-V DC and AC Electrical Characteristics.
If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
The maximum power dissipation is a function of TJ(MAX), RθJA. The maximum allowable power dissipation at any ambient temperature is
PD = (TJ(MAX) – TA)/RθJA. All numbers apply for packages soldered directly into a PC board.
6.2 ESD Ratings
VALUE
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
V(ESD)
(1)
(2)
Electrostatic discharge
(1)
UNIT
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±2000
Machine model
±200
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
NOM
MAX
2.7
Temperature range
–40
85
RF frequency range
50
4000
MHz
–45
–58
–5
–18
dBm
dBV
RF input power range (2)
(1)
(2)
4
3.3
UNIT
Supply voltage
V
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. For ensured
specifications and the test conditions, see the 2.7-V DC and AC Electrical Characteristics.
Power in dBV = dBm + 13 when the impedance is 50 Ω.
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6.4 Thermal Information
LMH2100
THERMAL METRIC (1)
YFQ (DSBGA)
UNIT
6 PINS
RθJA
Junction-to-ambient thermal resistance
RθJC(top)
RθJB
(2)
133.7
°C/W
Junction-to-case (top) thermal resistance
1.7
°C/W
Junction-to-board thermal resistance
22.6
°C/W
ψJT
Junction-to-top characterization parameter
5.7
°C/W
ψJB
Junction-to-board characterization parameter
22.2
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
The maximum power dissipation is a function of TJ(MAX), RθJA. The maximum allowable power dissipation at any ambient temperature is
PD = (TJ(MAX) - TA)/RθJA. All numbers apply for packages soldered directly into a PC board.
6.5 2.7-V DC and AC Electrical Characteristics
Unless otherwise specified, all limits are ensured at TA = 25°C, VDD = 2.7 V, RF input frequency ƒ = 1855 MHz CW
(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes. (1).
(2)
TYP (3)
Active mode: EN = High, no signal present
at RFIN
6.3
7.1
Active mode: EN = High, no signal present
at RFIN
TA = –40°C to +85°C
5
PARAMETER
TEST CONDITIONS
MIN
(2)
UNIT
7.9
mA
MAX
SUPPLY INTERFACE
IDD
Supply current
Shutdown: EN = Low, no signal present at
RFIN.
9.2
0.5
0.9
Shutdown: EN = Low, no signal present at
RFIN.
TA = –40°C to +85°C
1.9
EN = Low: PIN = 0 dBm (4)
TA = –40°C to +85°C
10
µA
LOGIC ENABLE INTERFACE
VLOW
EN logic low input level (Shutdown TA = –40°C to +85°C
Mode)
VHIGH
EN logic high input level
TA = –40°C to +85°C
IEN
Current into EN pin
TA = –40°C to +85°C
0.6
V
60
nA
56.4
Ω
1.1
V
RF INPUT INTERFACE
RIN
(1)
(2)
(3)
(4)
Input resistance
46.7
51.5
2.7-V DC and AC Electrical Characteristics values apply only for factory testing conditions at the temperature indicated. Factory testing
conditions result in very limited self-heating of the device such that TJ = TA. No specification of parametric performance is indicated in
the electrical tables under conditions of internal self-heating where TJ > TA.
All limits are ensured by test or statistical analysis.
Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary
over time and will also depend on the application and configuration. The typical values are not tested and are not specified on shipped
production material.
All limits are ensured by design and measurements which are performed on a limited number of samples. Limits represent the mean
±3–sigma values.
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2.7-V DC and AC Electrical Characteristics (continued)
Unless otherwise specified, all limits are ensured at TA = 25°C, VDD = 2.7 V, RF input frequency ƒ = 1855 MHz CW
(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.(1).
PARAMETER
TEST CONDITIONS
MIN
(2)
TYP (3)
MAX
(2)
UNIT
OUTPUT INTERFACE
From positive rail, sourcing,
VREF = 0 V, IOUT = 1 mA
VOUT
Output voltage swing
15.3
From positive rail, sourcing,
VREF = 0 V, IOUT = 1 mA
TA = –40°C to +85°C
28.9
mV
From negative rail, sinking,
VREF = 2.7 V, IOUT = 1 mA
13.1
From negative rail, sinking,
VREF = 2.7 V, IOUT = 1 mA
TA = –40°C to +85°C
IOUT
Output short circuit current
5.8
Sourcing, VREF = 0 V, VOUT = 2.6 V
TA = –40°C to +85°C
5.2
Sinking, VREF = 2.7 V, VOUT = 0.1 V
6.2
Sinking, VREF = 2.7 V, VOUT = 0.1 V
TA = –40°C to +85°C
5.4
Small signal bandwidth
No RF input signal. Measured from REF
input current to VOUT
RTRANS
Output amp transimpedance gain
No RF input signal, from IREF to VOUT, DC
SR
Slew rate
ROUT
Output impedance
IOUT,SD
(5)
Output leakage current in
shutdown mode
22.3
28.3
Sourcing, VREF = 0 V, VOUT = 2.6 V
BW
23.9
7.3
mA
8.3
416
40.7
43.3
Positive, VREF from 2.7 V to 0 V
3.4
3.9
Positive, VREF from 2.7 V to 0 V
TA = –40°C to +85°C
3.3
Negative, VREF from 0 V to 2.7 V
3.8
Negative, VREF from 0 V to 2.7 V
TA = –40°C to +85°C
3.7
No RF input signal, EN = High, DC
measurement
kHz
46.7
V/µs
4.4
0.2
kΩ
1.8
Ω
No RF input signal, EN = High, DC
measurement
4
EN = Low, VOUT = 2 V
TA = –40°C to +85°C
100
nA
RF DETECTOR TRANSFER
VOUT,MAX
VOUT,MIN
(5)
6
Maximum output voltage
PIN= −5 dBm (5)
Minimum output voltage (pedestal)
ƒ = 50 MHz, MIN and MAX at TA = –40°C
to +85°C
1.69
1.77
1.82
ƒ = 900 MHz, MIN and MAX at TA = –40°C
to +85°C
1.67
1.78
1.83
ƒ = 1855 MHz, MIN and MAX at TA =
–40°C to +85°C
1.57
1.65
1.70
ƒ = 2500 MHz, MIN and MAX at TA =
–40°C to +85°C
1.47
1.55
1.60
ƒ = 3000 MHz, MIN and MAX at TA =
–40°C to +85°C
1.38
1.46
1.51
ƒ = 3500 MHz, MIN and MAX at TA =
–40°C to +85°C
1.25
1.34
1.40
ƒ = 4000 MHz, MIN and MAX at TA =
–40°C to +85°C
1.16
1.25
1.30
No input signal
207
266
324
No input signal, TA = –40°C to +85°C
173
365
V
mV
All limits are ensured by design and measurements which are performed on a limited number of samples. Limits represent the mean
±3–sigma values. The typical value represents the statistical mean value.
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2.7-V DC and AC Electrical Characteristics (continued)
Unless otherwise specified, all limits are ensured at TA = 25°C, VDD = 2.7 V, RF input frequency ƒ = 1855 MHz CW
(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.(1).
(2)
TYP (3)
ƒ = 50 MHz, MIN and MAX at TA = –40°C
to +85°C
1.38
1.44
1.49
ƒ = 900 MHz, MIN and MAX at TA = –40°C
to +85°C
1.34
1.43
1.46
ƒ = 1855 MHz, MIN and MAX at TA =
–40°C to +85°C
1.27
1.32
1.36
ƒ = 2500 MHz, MIN and MAX at TA =
–40°C to +85°C
1.19
1.23
1.27
ƒ = 3000 MHz, MIN and MAX at TA =
–40°C to +85°C
1.11
1.16
1.19
ƒ = 3500 MHz, MIN and MAX at TA =
–40°C to +85°C
1
1.05
1.1
ƒ = 4000 MHz, MIN and MAX at TA =
–40°C to +85°C
0.91
0.97
1.01
ƒ = 50 MHz
39.6
40.9
42.1
ƒ = 900 MHz
37.0
38.2
39.4
ƒ = 1855 MHz
34.5
35.5
36.5
ƒ = 2500 MHz
32.7
33.7
34.6
ƒ = 3000 MHz
31.1
32.1
33.1
ƒ = 3500 MHz
29.7
30.7
31.6
f = 4000 MHz
28.5
29.4
30.3
ƒ = 50 MHz
–50.2
−49.5
–48.8
ƒ = 900 MHz
–53.6
−52.7
–51.8
ƒ = 1855 MHz
–53.2
−52.3
–51.4
ƒ = 2500 MHz
–52.4
−51.2
–50.1
ƒ = 3000 MHz
–51.2
−50.1
–48.9
ƒ = 3500 MHz
–49.1
−47.8
–46.4
ƒ = 4000 MHz
–47.3
−46.1
–44.9
PARAMETER
ΔVOUT
KSLOPE
PINT
TEST CONDITIONS
Output voltage range
PIN from −45 dBm to −5 dBm (5)
Logarithmic slope (5)
Logarithmic intercept (5)
en
Output referred noise (6)
vN
(5)
Output referred noise
MIN
PIN = −10 dBm at 10 kHz
1.5
Integrated over frequency band, 1 kHz to
6.5 kHz
100
(6)
PIN = −10 dBm, ƒ = 1800 MHz
TA = –40°C to +85°C
UNIT
V
mV/dB
dBm
µV/√Hz
150
PIN = −10 dBm, ƒ = 1800 MHz
Power supply rejection ratio (6)
(2)
µVRMS
Integrated over frequency band, 1 kHz to
6.5 kHz
TA = –40°C to +85°C
PSRR
MAX
60
55
dB
This parameter is ensured by design and/or characterization and is not tested in production.
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2.7-V DC and AC Electrical Characteristics (continued)
Unless otherwise specified, all limits are ensured at TA = 25°C, VDD = 2.7 V, RF input frequency ƒ = 1855 MHz CW
(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.(1).
PARAMETER
(2)
TYP (3)
ƒ = 50 MHz
–0.2
0.12
ƒ = 50 MHz, MIN and MAX at TA = –40°C
to +85°C
–0.8
ƒ = 900 MHz
–0.4
TEST CONDITIONS
MIN
MAX
(2)
UNIT
POWER MEASUREMENT PERFORMANCE
ƒ = 900 MHz, MIN and MAX at TA = –40°C
to +85°C
ELC
Log conformance error (5)
−40 dBm ≤ PIN ≤ −10 dBm
–0.3
ƒ = 1855 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.7
ƒ = 2500 MHz
–0.2
ƒ = 2500 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.8
ƒ = 3000 MHz
–0.1
ƒ = 3000 MHz, MIN and MAX at TA =
–40°C to +85°C
-0.036
ƒ = 3500 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.048
ƒ = 4000 MHz, MIN and MAX at TA =
–40°C to +85°C
E1
8
dB
Variation over temperature
−40 dBm ≤ PIN ≤ −10 dBm
Measurement Error for a 1-dB
Input power step (5)
−40 dBm ≤ PIN ≤ −10 dBm
0.2
0.3
-0.03
0.3
0.4
0.04
0.8
1.1
0.13
1.6
0.35
3.3
3.5
0.65
4.6
–1
4.9
ƒ = 50 MHz, MIN and MAX at TA = –40°C
to +85°C
–0.63
0.43
ƒ = 900 MHz, MIN and MAX at TA = –40°C
to +85°C
–0.94
0.30
ƒ = 1855 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.71
0.33
ƒ = 2500 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.88
0.35
ƒ = 3000 MHz, MIN and MAX at TA =
–40°C to +85°C
–1.03
0.37
ƒ = 3500 MHz, MIN and MAX at TA =
–40°C to +85°C
–1.10
0.33
ƒ = 4000 MHz, MIN and MAX at TA =
–40°C to +85°C
–1.12
0.33
ƒ = 50 MHz, MIN and MAX at TA = –40°C
to +85°C
–0.064
0.066
ƒ = 900 MHz, MIN and MAX at TA = –40°C
to +85°C
–0.123
0.051
ƒ = 1855 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.050
0.067
ƒ = 2500 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.058
0.074
ƒ = 3000 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.066
0.069
ƒ = 3500 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.082
0.066
ƒ = 4000 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.098
0.072
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dB
1.8
–1
ƒ = 4000 MHz
EVOT
–0.06
1
ƒ = 3500 MHz
(5)
1.3
–1
ƒ = 1855 MHz
1.2
dB
dB
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2.7-V DC and AC Electrical Characteristics (continued)
Unless otherwise specified, all limits are ensured at TA = 25°C, VDD = 2.7 V, RF input frequency ƒ = 1855 MHz CW
(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.(1).
PARAMETER
E10 dB
ST
ST
ST
Measurement Error for a 10-dB
Input power step (5)
−40 dBm ≤ PIN ≤ −10 dBm
Temperature sensitivity
–40°C < TA < 25°C
−40 dBm ≤ PIN ≤ −10 dBm (5)
Temperature sensitivity
25°C < TA < 85°C
−40 dBm ≤ PIN ≤ −10 dBm (5)
Temperature sensitivity
−40°C < TA < 25°C (5)
PIN = −10 dBm
TEST CONDITIONS
MIN
(2)
TYP (3)
MAX
(2)
ƒ = 50 MHz, MIN and MAX at TA = –40°C
to +85°C
–0.40
0.27
ƒ = 900 MHz, MIN and MAX at TA = –40°C
to +85°C
–0.58
0.22
ƒ = 1855 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.29
0.20
ƒ = 2500 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.28
0.24
ƒ = 3000 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.38
0.29
ƒ = 3500 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.60
0.40
ƒ = 4000 MHz, MIN and MAX at TA =
–40°C to +85°C
–0.82
0.43
ƒ = 50 MHz, MIN and MAX at TA = –40°C
to +85°C
–6.5
8.6
ƒ = 900 MHz, MIN and MAX at TA = –40°C
to +85°C
–4.7
14.5
ƒ = 1855 MHz, MIN and MAX at TA =
–40°C to +85°C
–5.1
11.0
ƒ = 2500 MHz, MIN and MAX at TA =
–40°C to +85°C
–4.3
13.6
ƒ = 3000 MHz, MIN and MAX at TA =
–40°C to +85°C
–1.5
15.8
ƒ = 3500 MHz, MIN and MAX at TA =
–40°C to +85°C
0.1
16.9
ƒ = 4000 MHz, MIN and MAX at TA =
–40°C to +85°C
0.5
17.3
ƒ = 50 MHz, MIN at TA = –40°C to +85°C
–10.5
0.5
ƒ = 900 MHz, MIN at TA = –40°C to +85°C
–10.5
2.6
ƒ = 1855 MHz, MIN at TA = –40°C to
+85°C
–11.3
3.4
ƒ = 2500 MHz, MIN at TA = –40°C to
+85°C
–10.6
5.8
ƒ = 3000 MHz, MIN at TA = –40°C to
+85°C
–11.2
6.1
ƒ = 3500 MHz, MIN at TA = –40°C to
+85°C
–12.9
5.5
ƒ = 4000 MHz, MIN at TA = –40°C to
+85°C
–17.8
5.5
ƒ = 50 MHz, MAX at TA = –40°C to +85°C
–5.4
8.6
ƒ = 900 MHz, MAX at TA = –40°C to +85°C
UNIT
dB
mdB/°C
mdB/°C
0.3
14.5
ƒ = 1855 MHz, MAX at TA = –40°C to
+85°C
–3.1
11.0
ƒ = 2500 MHz, MAX at TA = –40°C to
+85°C
–1.6
13.6
ƒ = 3000 MHz, MAX at TA = –40°C to
+85°C
0.9
15.8
ƒ = 3500 MHz, MAX at TA = –40°C to
+85°C
2.5
16.9
ƒ = 4000 MHz, MAX at TA = –40°C to
+85°C
2.7
17.3
mdB/°C
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2.7-V DC and AC Electrical Characteristics (continued)
Unless otherwise specified, all limits are ensured at TA = 25°C, VDD = 2.7 V, RF input frequency ƒ = 1855 MHz CW
(Continuous Wave, unmodulated). Maximum and minimum limits apply at the temperature extremes.(1).
PARAMETER
ST
PMAX
PMIN
DR
10
Temperature sensitivity
25°C < TA < 85°C (5)
PIN = −10 dBm
Maximum input power for ELC = 1
dB (5)
Minimum input power for ELC = 1
dB (5)
Dynamic range for ELC = 1 dB (5)
TEST CONDITIONS
MIN
(2)
TYP (3)
MAX
(2)
ƒ = 50 MHz, MIN and MAX at TA = –40°C
to +85°C
–10.5
0.5
ƒ = 900 MHz, MIN and MAX at TA = –40°C
to +85°C
–10.5
2.6
ƒ = 1855 MHz, MIN and MAX at TA =
–40°C to +85°C
–11.3
3.3
ƒ = 2500 MHz, MIN and MAX at TA =
–40°C to +85°C
–10.6
5.4
ƒ = 3000 MHz, MIN and MAX at TA =
–40°C to +85°C
–11.2
6.1
ƒ = 3500 MHz, MIN and MAX at TA =
–40°C to +85°C
–12.9
4.4
ƒ = 4000 MHz, MIN and MAX at TA =
–40°C to +85°C
–17.8
–1.1
ƒ = 50 MHz, MIN at TA = –40°C to +85°C
–9.2
–7.4
ƒ = 900 MHz, MIN at TA = –40°C to +85°C
–10.5
–8.6
ƒ = 1855 MHz, MIN at TA = –40°C to
+85°C
–8.2
–6.5
ƒ = 2500 MHz, MIN at TA = –40°C to
+85°C
-7.3
–5.6
ƒ = 3000 MHz, MIN at TA = –40°C to
+85°C
–6.3
–4.4
ƒ = 3500 MHz, MIN at TA = –40°C to
+85°C
–6.9
–1.9
ƒ = 4000 MHz, MIN at TA = –40°C to
+85°C
–11.1
–7.2
mdB/°C
dBm
ƒ = 50 MHz, MAX at TA = –40°C to +85°C
–38.9
–38.1
ƒ = 900 MHz, MAX at TA = –40°C to +85°C
–43.1
–42.3
ƒ = 1855 MHz, MAX at TA = –40°C to
+85°C
–42.2
–41
ƒ = 2500 MHz, MAX at TA = –40°C to
+85°C
–40.6
-38.9
ƒ = 3000 MHz, MAX at TA = –40°C to
+85°C
–38.7
–37
ƒ = 3500 MHz, MAX at TA = –40°C to
+85°C
–35.9
–34.7
ƒ = 4000 MHz, MAX at TA = –40°C to
+85°C
–33.5
–32
dBm
ƒ = 50 MHz, MIN at TA = –40°C to +85°C
29.5
31.6
ƒ = 900 MHz, MIN at TA = –40°C to +85°C
33.3
35.2
ƒ = 1855 MHz, MIN at TA = –40°C to
+85°C
34.2
36.5
ƒ = 2500 MHz, MIN at TA = –40°C to
+85°C
34.1
36.1
ƒ = 3000 MHz, MIN at TA = –40°C to
+85°C
33.4
35.5
ƒ = 3500 MHz, MIN at TA = –40°C to
+85°C
28.5
35.1
ƒ = 4000 MHz, MIN at TA = –40°C to
+85°C
22.7
26.3
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UNIT
dB
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6.6 Timing Requirements
MIN
NOM
MAX
8.2
9.8
Turnon time, no signal at PIN, Low-High transition EN, VOUT to 90%
tON
Turnon time, no signal at PIN, Low-High transition EN, VOUT to 90%
TA = –40°C to +85°C
12
Rise time (1), PIN = no signal to 0 dBm, VOUT from 10% to 90%
tR
(1)
µs
2
Rise time (1), PIN = no signal to 0 dBm, VOUT from 10% to 90%
TA = –40°C to +85°C
12
Fall time (1), PIN = no signal to 0 dBm, VOUT from 90% to 10%
tF
UNIT
µs
2
Fall time (1), PIN = no signal to 0 dBm, VOUT from 90% to 10%
TA = –40°C to +85°C
12
µs
This parameter is ensured by design and/or characterization and is not tested in production.
6.7 Typical Characteristics
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
10
85°C
-40°C
10
8
8
8
8
ISUPPLY (mA)
SUPPLY CURRENT (mA)
10
10
25°C
66
4
4
66
2
2
2
0
0
2.5
2.8
3.1
25°C
-40°C
4
4
2
0
2.2
85°C
0
3.4
650
700
SUPPLY VOLTAGE (V)
Figure 1. Supply Current vs Supply Voltage
850
900
Figure 2. Supply Current vs Enable Voltage
45
-38
-40°C
LOGARITHMIC INTERCEPT (dBm)
LOGARITHMIC SLOPE (mV/dB)
750
800
VENABLE (mV)
25°C
40
85°C
35
30
25
10M
100M
1G
-40°C
-42
-46
25°C
85°C
-50
-54
10M
10G
FREQUENCY (Hz)
100M
1G
10G
FREQUENCY (Hz)
Figure 3. Log Slope vs Frequency
Figure 4. Log Intercept vs Frequency
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Typical Characteristics (continued)
2.0
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
25°C
85°C
85°C
25°C
0.2
-2.0
-40°C
0.0
-55
-45
-35
-25
-15
-5
2.0
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
-40°C
25°C
0.0
-55
2.0
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
-40°C
85°C
0.2
-2.0
-40°C
0.0
-55
-45
-35
-25
-15
-5
2.5
2.0
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
2.0
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
85°C
0.0
-55
-2.0
-2.5
5
-1.5
-2.0
-45
-35
-25
-15
-5
-2.5
5
2.0
2.5
1.8
2.0
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
25°C
-40°C
85°C
85°C
0.2
0.0
-55
-40°C
25°C
-45
-35
-25
-2.0
-15
-5
-2.5
5
RF INPUT POWER (dBm)
Figure 9. Mean Output Voltage and Log Conformance Error
vs
RF Input Power at 3000 MHz
12
-40°C
Figure 8. Mean Output Voltage and Log Conformance Error
vs
RF Input Power at 2500 MHz
25°C
-45
-35
-25
-15
-5
RF INPUT POWER (dBm)
85°C
85°C
-40°C
-55
VOUT (V)
2.5
-40°C
25°C
25°C
0.0
LOG CONFORMANCE ERROR (dB)
VOUT (V)
2.0
0.2
-5
RF INPUT POWER (dBm)
1.8
-40°C
-15
2.0
0.2
-2.5
5
Figure 7. Mean Output Voltage and Log Conformance Error
vs
RF Input Power at 1855 MHz
85°C
-25
1.8
RF INPUT POWER (dBm)
25°C
-35
Figure 6. Mean Output Voltage and Log Conformance Error
vs
RF Input Power at 900 MHz
VOUT (V)
2.5
LOG CONFORMANCE ERROR (dB)
VOUT (V)
2.0
1.8
85°C
-45
-2.5
5
RF INPUT POWER (dBm)
Figure 5. Mean Output Voltage and Log Conformance Error
vs
RF Input Power at 50 MHz
25°C
-2.0
-40°C
RF INPUT POWER (dBm)
25°C
85°C
85°C
0.2
-2.5
5
25°C
LOG CONFORMANCE ERROR (dB)
-40°C
2.5
LOG CONFORMANCE ERROR (dB)
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
2.0
1.8
LOG CONFORMANCE ERROR (dB)
2.5
VOUT (V)
2.0
1.8
LOG CONFORMANCE ERROR (dB)
VOUT (V)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
Figure 10. Mean Output Voltage and Log Conformance Error
vs
RF Input Power at 3500 MHz
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Typical Characteristics (continued)
2.0
2.5
1.8
2.0
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
25°C
-40°C
85°C
85°C
0.2
-40°C
0.0
-55
25°C
-45
-35
-25
-15
-5
RF INPUT POWER (dBm)
-2.0
LOG CONFORMANCE ERROR (dB)
VOUT (V)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
-2.5
5
Figure 11. Mean Output Voltage and Log Conformance Error
vs
RF Input Power at 4000 MHz
Figure 12. Log Conformance Error (Mean ±3 sigma) vs
RF Input Power at 50 MHz
Figure 13. Log Conformance Error (Mean ±3 sigma) vs
RF Input Power at 900 MHz
Figure 14. Log Conformance Error (Mean ±3 sigma) vs
RF Input Power at 1855 MHz
Figure 15. Log Conformance Error (Mean ±3 sigma) vs
RF Input Power at 2500 MHz
Figure 16. Log Conformance Error (Mean ±3 sigma) vs
RF Input Power at 3000 MHz
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Typical Characteristics (continued)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
Figure 18. Log Conformance Error (Mean ±3 sigma) vs
RF Input Power at 4000 MHz
1.5
1.5
1.5
1.0
1.5
1.0
1.0
0.5
0.5
TEMPERATURE DRIFT (dB)
TEMPERATURE DRIFT (dB)
Figure 17. Log Conformance Error (Mean ±3 sigma) vs
RF Input Power at 3500 MHz
-40°C
0.0
0.0
-0.5
-0.5
-1.0
85°C
-1.0
-1.5
1.0
0.5
0.5
-40°C
0.0
0.0
-0.5
-0.5
-1.0
85°C
-1.0
-1.5
-1.5
-1.5
-55
-45
-35
-25
-15
-5
-55
5
1.5
1.5
1.5
1.0
1.5
1.0
-40°C
0.0
0.0
-0.5
-0.5
-1.0
85°C
-1.0
-1.5
-25
-15
-5
5
1.0
0.5
0.5
85°C
0.0
0.0
-0.5
-0.5
-1.0
-40°C
-1.0
-1.5
-1.5
-1.5
-55
14
-35
Figure 20. Mean Temperature Drift Error vs
RF Input Power at 900 MHz
TEMPERATURE DRIFT (dB)
TEMPERATURE DRIFT (dB)
Figure 19. Mean Temperature Drift Error vs
RF Input Power at 50 MHz
1.0
0.5
0.5
-45
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
-45
-35
-25
-15
-5
5
-55
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
Figure 21. Mean Temperature Drift Error vs
RF Input Power at 1855 MHz
Figure 22. Mean Temperature Drift Error vs
RF Input Power at 2500 MHz
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Typical Characteristics (continued)
1.5
1.5
1.5
1.0
1.5
1.0
1.0
0.5
0.5
TEMPERATURE DRIFT (dB)
TEMPERATURE DRIFT (dB)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
85°C
0.0
0.0
-0.5
-0.5
-1.0
-40°C
-1.0
-1.5
-1.5
-55
-45
-35
-25
-15
-5
1.0
0.5
0.5
85°C
0.0
0.0
-0.5
-0.5
-1.0
-40°C
-1.0
-1.5
-1.5
5
-55
RF INPUT POWER (dBm)
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
Figure 23. Mean Temperature Drift Error vs
RF Input Power at 3000 MHz
Figure 24. Mean Temperature Drift Error vs
RF Input Power at 3500 MHz
TEMPERATURE DRIFT (dB)
1.5
1.5
1.0
1.0
0.5
0.5
85°C
0.0
0.0
-0.5
-0.5
-1.0
-1.0
-1.5
-40°C
-1.5
-55
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
Figure 25. Mean Temperature Drift Error vs
RF Input Power at 4000 MHz
Figure 26. Temperature Drift Error (Mean ±3 sigma) vs
RF Input Power at 50 MHz
Figure 27. Temperature Drift Error (Mean ±3 sigma) vs
RF Input Power at 900 MHz
Figure 28. Temperature Drift Error (Mean ±3 sigma) vs
RF Input Power at 1855 MHz
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Typical Characteristics (continued)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
Figure 29. Temperature Drift Error (Mean ±3 sigma) vs
RF Input Power at 2500 MHz
Figure 30. Temperature Drift Error (Mean ±3 sigma) vs
RF Input Power at 3000 MHz
Figure 31. Temperature Drift Error (Mean ±3 sigma) vs
RF Input Power at 3500 MHz
Figure 32. Temperature Drift Error (Mean ±3 sigma) vs
RF Input Power at 4000 MHz
0.3
0.3
0.3
0.2
0.3
0.2
ERROR (dB)
ERROR (dB)
0.0
0.0
-0.1
-0.1
-0.2
85°C
-0.2
-0.3
-45
-35
-25
-15
-5
0.0
0.0
-0.1
-0.1
-0.2
85°C
-0.3
5
-55
RF INPUT POWER (dBm)
Figure 33. Error for 1 dB Input Power Step vs
RF Input Power at 50 MHz
16
0.1
0.1
-0.2
-0.3
-0.3
-55
-40°C
0.2
-40°C
0.2
0.1
0.1
-45
-35
-25
-15
-5
RF INPUT POWER (dBm)
5
Figure 34. Error for 1 dB Input Power Step vs
RF Input Power at 900 MHz
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Typical Characteristics (continued)
0.3
0.3
0.3
0.2
0.3
0.2
0.2
0.1
0.1
0.2
0.1
0.1
ERROR (dB)
ERROR (dB)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
85°C
0.0
0.0
-0.1
-0.1
-0.2
-40°C
-0.2
-0.3
-40°C
0.0
0.0
-0.1
-0.1
-0.2
85°C
-0.2
-0.3
-0.3
-0.3
-55
-45
-35
-25
-15
-5
5
-55
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
Figure 35. Error for 1 dB Input Power Step vs
RF Input Power at 1855 MHz
Figure 36. Error for 1 dB Input Power Step vs
RF Input Power at 2500 MHz
0.3
0.3
0.3
0.2
0.2
0.1
0.1
-40°C
ERROR (dB)
ERROR (dB)
0.3
0.2
0.0
0.0
-0.1
-0.1
-0.2
85°C
-40°C
0.0
0.0
-0.1
-0.1
-0.2
85°C
-0.2
-0.3
-0.2
-0.3
-0.3
-0.3
-55
-45
-35
-25
-15
-5
RF INPUT POWER (dBm)
-55
5
1.00
0.3
0.2
0.75
1.00
0.50
0.75
0.50
0.25
0.25
0.00
0.00
-0.25
-0.25
-0.50
-0.75
-0.50
-1.00
-40°C
ERROR (dB)
0.1
0.1
0.0
0.0
-0.1
-0.1
-0.2
-35
-25
-15
-5
5
Figure 38. Error for 1 dB Input Power Step vs
RF Input Power at 3500 MHz
0.3
0.2
-45
RF INPUT POWER (dBm)
Figure 37. Error for 1 dB Input Power Step vs
RF Input Power at 3000 MHz
ERROR (dB)
0.2
0.1
0.1
85°C
-0.2
-0.3
-40°C
85°C
-0.75
-0.3
-1.00
-55
-45
-35
-25
-15
-5
5
-60
RF INPUT POWER (dBm)
Figure 39. Error for 1 dB Input Power step vs
RF Input Power at 4000 MHz
-50
-40
-30
-20
-10
RF INPUT POWER (dBm)
0
Figure 40. Error for 10 dB Input Power Step vs
RF Input Power at 50 MHz
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Typical Characteristics (continued)
1.00
1.00
0.75
1.00
0.50
0.75
0.50
0.25
0.25
0.00
0.00
-0.25
-0.25
-0.50
-0.75
-0.50
-1.00
0.75
1.00
0.50
0.75
0.50
0.25
0.25
0.00
0.00
-0.25
-0.25
-0.50
-0.75
-0.50
-1.00
-40°C
ERROR (dB)
ERROR (dB)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
85°C
-0.75
-40°C
85°C
-0.75
-1.00
-60
-50
-40
-30
-20
-10
-1.00
0
-60
RF INPUT POWER (dBm)
1.00
1.00
0.75
1.00
0.50
0.75
0.50
0.25
0.25
0.00
0.00
-0.25
-0.25
-0.50
-0.75
-0.50
-1.00
0.75
1.00
0.50
0.75
0.50
0.25
0.25
0.00
0.00
-0.25
-0.25
-0.50
-0.75
-0.50
-1.00
-40°C
85°C
-40°C
85°C
-0.75
-0.75
-1.00
-1.00
-60
-50
-40
-30
-20
-10
-60
0
Figure 43. Error for 10 dB Input Power Step vs
RF Input Power at 2500 MHz
-40
-30
-20
-10
0
Figure 44. Error for 10 dB Input Power Step vs
RF Input Power at 3000 MHz
1.00
1.00
0.75
1.00
0.50
0.75
0.50
0.25
0.25
0.00
0.00
-0.25
-0.25
-0.50
-0.75
-0.50
-1.00
-40°C
ERROR (dB)
ERROR (dB)
-50
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
85°C
-0.75
0.75
1.00
0.50
0.75
0.50
0.25
0.25
0.00
0.00
-0.25
-0.25
-0.50
-0.75
-0.50
-1.00
-40°C
85°C
-0.75
-1.00
-1.00
-60
-50
-40
-30
-20
-10
-60
0
-50
-40
-30
-20
-10
0
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
Figure 45. Error for 10 dB Input Power Step vs
RF Input Power at 3500 MHz
18
0
Figure 42. Error for 10 dB Input Power Step vs
RF Input Power at 1855 MHz
ERROR (dB)
ERROR (dB)
Figure 41. Error for 10 dB Input Power Step vs
RF Input Power at 900 MHz
-50
-40
-30
-20
-10
RF INPUT POWER (dBm)
Figure 46. Error for 10 dB Input Power step vs
RF Input Power at 4000 MHz
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Typical Characteristics (continued)
20
20
15
20
10
15
10
5
5
00
-5
-5
-10
15
20
10
15
10
5
5
00
-5
-5
-10
-15
-10
-20
SENSITIVITY (mdB/°C)
SENSITIVITY (mdB/°C)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
-40°C
85°C
-15
-15
-10
-20
85°C
-15
-20
-55
-45
-35
-25
-15
-5
-20
5
-55
RF INPUT POWER (dBm)
-35
-25
-15
-5
5
Figure 48. Mean Temperature Sensitivity vs
RF Input Power at 900 MHz
20
15
20
10
15
10
5
5
00
-5
-5
-10
15
20
10
15
10
5
5
00
-5
-5
-10
SENSITIVITY (mdB/°C)
20
-40°C
85°C
-15
-10
-20
-45
RF INPUT POWER (dBm)
Figure 47. Mean Temperature Sensitivity vs
RF Input Power at 50 MHz
SENSITIVITY (mdB/°C)
-40°C
-40°C
85°C
-15
-10
-20
-15
-15
-20
-20
-55
-45
-35
-25
-15
-5
-55
5
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
Figure 49. Mean Temperature Sensitivity vs
RF Input Power at 1855 MHz
Figure 50. Mean Temperature Sensitivity vs
RF Input Power at 2500 MHz
20
15
20
10
15
10
5
5
00
-5
-5
-10
-15
-10
-20
-40°C
SENSITIVITY (mdB/°C)
SENSITIVITY (mdB/°C)
20
85°C
-15
15
20
10
15
10
5
5
00
-5
-5
-10
-15
-10
-20
-40°C
85°C
-15
-20
-20
-55
-45
-35
-25
-15
-5
5
-55
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
Figure 51. Mean Temperature Sensitivity vs
RF Input Power at 3000 MHz
Figure 52. Mean Temperature Sensitivity vs
RF Input Power at 3500 MHz
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Typical Characteristics (continued)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
20
15
-40°C
SENSITIVITY (mdB/° C)
SENSITIVITY (mdB/°C)
20
15
20
10
15
10
5
5
00
-5
-5
-10
85°C
-15
-10
-20
-15
-40°C
10
5
0
-5
-10
85°C
-15
-20
-55
-45
-35
-25
-15
-5
RF INPUT POWER (dBm)
-20
-55
5
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
Figure 53. Mean Temperature Sensitivity vs
RF Input power at 4000 MHz
Figure 54. Temperature Sensitivity (Mean ±3 sigma) vs
RF Input Power at 50 MHz
20
20
15
15
-40°C
10
SENSITIVITY (mdB/°C)
SENSITIVITY (mdB/°C)
-40°C
5
0
-5
-10
5
0
-5
-10
85°C
85°C
-15
-20
-55
10
-15
-45
-35
-25
-15
-5
-20
5
-55
-45
-25
-15
-5
5
RF INPUT POWER (dBm)
Figure 55. Temperature Sensitivity (Mean ±3 sigma) vs
RF Input Power at 900 MHz
Figure 56. Temperature Sensitivity (Mean ±3 sigma) vs
RF Input Power at 1855 MHz
20
20
-40°C
15
15
10
10
SENSITIVITY (mdB/°C)
SENSITIVITY (mdB/° C)
-40°C
5
0
-5
-10
5
0
-5
-10
85°C
85°C
-15
-15
-20
-55
20
-35
RF INPUT POWER (dBm)
-45
-35
-25
-15
-5
-20
5
-55
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
Figure 57. Temperature Sensitivity (Mean ±3 sigma) vs
RF Input Power at 2500 MHz
Figure 58. Temperature Sensitivity (Mean ±3 sigma) vs
RF Input Power at 3000 MHz
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Typical Characteristics (continued)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
20
20
-40°C
15
15
10
10
SENSITIVITY (mdB/°C)
SENSITIVITY (mdB/° C)
-40°C
5
0
-5
-10
5
0
-5
-10
85°C
-15
-15
85°C
-20
-55
-45
-35
-25
-15
-5
-20
5
-55
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
Figure 59. Temperature Sensitivity (Mean ±3 sigma) vs
RF Input Power at 3500 MHz
Figure 60. Temperature Sensitivity (mean ±3 sigma) vs.
RF Input Power at 4000 MHz
IS-95
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
WCDMA 64 CH
CW
IS-95
WCDMA 64 ch
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
0.2
-2.0
0.0
-2.5
5
-55
-45
-35
-25
-15
-5
2.5
CW
1.8
VOUT (V)
2.0
LOG CONFORMANCE ERROR (dB)
VOUT (V)
2.0
2.5
CW
CW
WCDMA 64 ch
IS-95
-2.0
0.0
-55
-45
-35
-25
-15
1.5
2.5
1.0
2.0
1.5
1.0
0.5
0.5
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
-5
-2.5
5
RF INPUT POWER (dBm)
Figure 61. Output Voltage and Log Conformance Error vs
RF Input Power for Various Modulation Types at 900 MHz
Figure 62. Output Voltage and Log Conformance Error vs
RF Input Power for Various Modulation Types at 1855 MHz
100
10
9
VOLTAGE NOISE (µV/ Hz)
R
75
RF INPUT IMPEDANCE (Ö)
WCDMA 64 CH
0.2
RF INPUT POWER (dBm)
50
25
0
X
-25
-50
-75
-100
10M
2.0
IS-95
1.6
2.0
1.4
1.8
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
LOG CONFORMANCE ERROR (dB)
2.0
1.8
8
7
6
5
4
3
2
1
100M
1G
0
10
10G
FREQUENCY (Hz)
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 63. RF Input Impedance vs
Frequency (Resistance and Reactance)
Figure 64. Output Noise Spectrum vs Frequency
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Typical Characteristics (continued)
Unless otherwise specified, VDD = 2.7 V, TA = 25°C, measured on a limited number of samples.
100k
100k
270
GAIN
TRANSIMPEDANCE GAIN
70
60
PSRR (dB)
50
40
30
20
225
270
180
225
180
135
135
90
90
45
045
-45
0
-90
10k
10k
PHASE
1k
1k
PHASE (°)
80
-45
10
100
0
10
100
1k
10k
100k
100
1M
-90
100
1k
Figure 65. Power Supply Rejection Ratio vs Frequency
10M
60
25°C
60
50
85°C
SINKING OUTPUT CURRENT (mA)
SOURCING OUTPUT CURRENT (mA)
1M
Figure 66. Output Amplifier Gain and Phase vs Frequency
60
-40°C
50
40
40
30
30
20
20
10
100
25°C
60
50
85°C
-40°C
50
40
40
30
30
20
20
10
100
0
0
0.0
0.5
1.0
1.5
2.0
VOUT (V)
2.5
0.0
3.0
0.5
1.0
1.5
2.0
2.5
3.0
VOUT (V)
Figure 67. Sourcing Output Current vs Output Voltage
Figure 68. Sinking Output Current vs Output Voltage
2.70
0.08
2.70
0.08
-40°C
2.68
2.68
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
100k
FREQUENCY (Hz)
FREQUENCY (Hz)
85°C
25°C
2.66
2.66
-40°C
2.64
2.64
2.62
2.62
2.60
0.06
0.06
25°C
85°C
0.04
0.04
0.02
0.02
0.00
2.60
0
1
2
3
4
0.00
5
0
SOURCING CURRENT (mA)
Figure 69. Output Voltage vs Sourcing Current
22
10k
1
2
3
4
SINKING CURRENT (mA)
5
Figure 70. Output Voltage vs Sinking Current
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7 Detailed Description
7.1 Overview
The LMH2100 is a versatile logarithmic RF power detector suitable for use in power measurement systems. The
LMH2100 is particularly well suited for CDMA and UMTS applications. It produces a DC voltage that is a
measure for the applied RF power.
The core of the LMH2100 is a progressive compression LOG detector consisting of four gain stages. Each of
these saturating stages has a gain of approximately 10 dB and therefore realizes about 10 dB of the detector
dynamic range. The five diode cells perform the actual detection and convert the RF signal to a DC current. This
DC current is subsequently supplied to the transimpedance amplifier at the output, that converts it into an output
voltage. In addition, the amplifier provides buffering of and applies filtering to the detector output signal. To
prevent discharge of filtering capacitors between OUT and GND in shutdown, a switch is inserted at the amplifier
input that opens in shutdown to realize a high impedance output of the device.
7.2 Functional Block Diagram
B2 REF
VDD A1
RTRANS
en
en
EN C2
I/I
VREF
RFIN B1
GN
C1
D
V-V
10 dB
10 dB
10 dB
A2 OUT
+
+
-
en
10 dB
RIN
7.3 Feature Description
7.3.1 Characteristics of the LMH2100
The LMH2100 is a logarithmic RF power detector with approximately 40-dB dynamic range. This dynamic range
plus its logarithmic behavior make the LMH2100 ideal for various applications such as wireless transmit power
control for CDMA and UMTS applications. The frequency range of the LMH2100 is from 50 MHz to 4 GHz, which
makes it suitable for various applications.
The LMH2100 transfer function is accurately temperature compensated. This makes the measurement accurate
for a wide temperature range. Furthermore, the LMH2100 can easily be connected to a directional coupler
because of its 50-Ω input termination. The output range is adjustable to fit the ADC input range. The detector can
be switched into a power saving shutdown mode for use in pulsed conditions.
7.3.2 Accurate Power Measurement
The power measurement accuracy achieved with a power detector is not only determined by the accuracy of the
detector itself, but also by the way it is integrated into the application. In many applications some form of
calibration is employed to improve the accuracy of the overall system beyond the intrinsic accuracy provided by
the power detector. For example, for LOG-detectors calibration can be used to eliminate part to part spread of
the LOG-slope and LOG-intercept from the overall power measurement system, thereby improving its power
measurement accuracy.
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Feature Description (continued)
This section shows how calibration techniques can be used to improve the accuracy of a power measurement
system beyond the intrinsic accuracy of the power detector itself. The main focus of the section is on power
measurement systems using LOG-detectors, specifically the LMH2100, but the more generic concepts can also
be applied to other power detectors. Other factors influencing the power measurement accuracy, such as the
resolution of the ADC reading the detector output signal will not be considered here since they are not
fundamentally due to the power detector.
7.3.2.1
LOG-Conformance Error
Probably the simplest power measurement system that can be realized is obtained when the LOG-detector
transfer function is modelled as a perfect linear-in-dB relationship between the input power and output voltage:
VOUT,MOD = FDET,MOD(PIN) = KSLOPE(PIN ± PINTERCEPT)
(1)
in which KSLOPE represents the LOG-slope and PINTERCEPT the LOG-intercept. The estimator based on this model
implements the inverse of the model equation, that is:
PEST = FEST(VOUT) =
VOUT
+ PINTERCEPT
KSLOPE
(2)
The resulting power measurement error, the LOG-conformance error, is thus equal to:
ELCE = PEST - PIN =
=
VOUT
- (PIN - PINTERCEPT )
KSLOPE
VOUT - VOUT,MOD
KSLOPE
(3)
The most important contributions to the LOG-conformance error are generally:
• The deviation of the actual detector transfer function from an ideal Logarithm (the transfer function is
nonlinear in dB).
• Drift of the detector transfer function over various environmental conditions, most importantly temperature;
KSLOPE and PINTERCEPT are usually determined for room temperature only.
• Part-to-part spread of the (room temperature) transfer function.
The latter component is conveniently removed by means of calibration, that is, if the LOG slope and LOGintercept are determined for each individual detector device (at room temperature). This can be achieved by
measurement of the detector output voltage - at room temperature - for a series of different power levels in the
LOG-linear range of the detector transfer function. The slope and intercept can then be determined by means of
linear regression.
An example of this type of error and its relationship to the detector transfer function is depicted in Figure 71.
24
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2.0
2.5
1.8
2.0
1.6
2.0
1.8
1.4
1.6
1.4
1.2
1.2
1.0
1.0
0.8
0.6
0.8
0.4
0.2
0.6
0.0
0.4
1.5
2.5
2.0
1.0
1.5
1.0
0.5
0.5
0.0
0.0
-0.5
-1.0
-0.5
-1.5
-2.0
-1.0
-2.5
-1.5
-40°C
25°C
85°C
85°C
25°C
0.2
-2.0
-40°C
0.0
-55
-45
-35
-25
-15
-5
LOG CONFORMANCE ERROR (dB)
VOUT (V)
Feature Description (continued)
-2.5
5
RF INPUT POWER (dBm)
Figure 71. LOG-Conformance Error and LOG-Detector Transfer Function
In the center of the detector's dynamic range, the LOG-conformance error is small, especially at room
temperature; in this region the transfer function closely follows the linear-in-dB relationship while KSLOPE and
PINTERCEPT are determined based on room temperature measurements. At the temperature extremes the error in
the center of the range is slightly larger due to the temperature drift of the detector transfer function. The error
rapidly increases toward the top and bottom end of the detector's dynamic range; here the detector saturates and
its transfer function starts to deviate significantly from the ideal LOG-linear model. The detector dynamic range is
usually defined as the power range for which the LOG conformance error is smaller than a specified amount.
Often an error of ±1 dB is used as a criterion.
7.3.2.2 Temperature Drift Error
A more accurate power measurement system can be obtained if the first error contribution, due to the deviation
from the ideal LOG-linear model, is eliminated. This is achieved if the actual measured detector transfer function
at room temperature is used as a model for the detector, instead of the ideal LOG-linear transfer function used in
the previous section.
The formula used for such a detector is:
VOUT,MOD = FDET(PIN,TO)
where
•
TO represents the temperature during calibration (room temperature).
(4)
The transfer function of the corresponding estimator is thus the inverse of this:
-1
PEST = FDET[VOUT(T),T0]
(5)
In this expression VOUT(T) represents the measured detector output voltage at the operating temperature T.
The resulting measurement error is only due to drift of the detector transfer function over temperature, and can
be expressed as:
-1
EDRIFT (T,T0) = PEST - PIN = FDET[VOUT(T),T0] - PIN
-1
-1
= FDET[VOUT(T),T0] - FDET[VOUT(T),T)]
(6)
Unfortunately, the (numeric) inverse of the detector transfer function at different temperatures makes this
expression rather impractical. However, since the drift error is usually small VOUT(T) is only slightly different from
VOUT(TO). This means that we can apply the following approximation:
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Feature Description (continued)
EDRIFT(T,T0) | EDRIFT(T0,T0)
w
-1
-1
+ (T - T0) w7 {FDET[VOUT(T),T0] - FDET[VOUT(T),T]}
(7)
This expression is easily simplified by taking the following considerations into account:
• The drift error at the calibration temperature E(TO,TO) equals zero (by definition).
• The estimator transfer FDET(VOUT,TO) is not a function of temperature; the estimator output changes over
temperature only due to the temperature dependence of VOUT.
• The actual detector input power PIN is not temperature dependent (in the context of this expression).
• The derivative of the estimator transfer function to VOUT equals approximately 1/KSLOPE in the LOG-linear
region of the detector transfer function (the region of interest).
Using this, we arrive at:
w -1
FDET[VOUT(T),T0]
w7
-1
w VOUT(T)
w
= (T ± T0)
FDET[VOUT(T),T0]
wVOUT
w7
EDRIFT (T,T0) | (T ± T0)
|
VOUT(T) ± VOUT(T0)
KSLOPE
(8)
This expression is very similar to the expression of the LOG-conformance error determined previously. The only
difference is that instead of the output of the ideal LOG-linear model, the actual detector output voltage at the
calibration temperature is now subtracted from the detector output voltage at the operating temperature.
Figure 72 depicts an example of the drift error.
TEMPERATURE DRIFT (dB)
1.5
1.5
1.0
1.0
0.5
0.5
-40°C
0.0
0.0
-0.5
-0.5
-1.0
85°C
-1.0
-1.5
-1.5
-55
-45
-35
-25
-15
-5
5
RF INPUT POWER (dBm)
Figure 72. Temperature Drift Error of the LMH2100 at ƒ = 1855 MHz
In agreement with the definition, the temperature drift error is zero at the calibration temperature. Further, the
main difference with the LOG-conformance error is observed at the top and bottom end of the detection range;
instead of a rapid increase the drift error settles to a small value at high and low input power levels due to the
fact that the detector saturation levels are relatively temperature independent.
In a practical application it may not be possible to use the exact inverse detector transfer function as the
algorithm for the estimator. For example it may require too much memory and/or too much factory calibration
time. However, using the ideal LOG-linear model in combination with a few extra data points at the top and
bottom end of the detection range - where the deviation is largest - can already significantly reduce the power
measurement error.
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Feature Description (continued)
7.3.2.2.1 Temperature Compensation
A further reduction of the power measurement error is possible if the operating temperature is measured in the
application. For this purpose, the detector model used by the estimator should be extended to cover the
temperature dependency of the detector.
Since the detector transfer function is generally a smooth function of temperature (the output voltage changes
gradually over temperature), the temperature is in most cases adequately modeled by a first-order or secondorder polynomial (see Equation 9).
VOUT,MOD = FDET(PIN,T0)[1 + (T-T0)TC1(PIN)
+ (T-T0)2TC2(PIN) + O(T3)]
(9)
The required temperature dependence of the estimator, to compensate for the detector temperature dependence
can be approximated similarly:
-1
PEST = FDET[VOUT(T),T0]{1 + (T-T0)S1[VOUT(T)] +
+ (T-T0)2S2[VOUT(T)] + O(T3)}
-1
| FDET[VOUT(T),T0]{1 + (T-T0)S1[VOUT(T)]}
(10)
The last approximation results from the fact that a first-order temperature compensation is usually sufficiently
accurate. The remainder of this section will therefore concentrate on first-order compensation. For second and
higher-order compensation a similar approach can be followed.
Ideally, the temperature drift could be completely eliminated if the measurement system is calibrated at various
temperatures and input power levels to determine the Temperature Sensitivity S1. In a practical application,
however that is usually not possible due to the associated high costs. The alternative is to use the average
temperature drift in the estimator, instead of the temperature sensitivity of each device individually. In this way it
becomes possible to eliminate the systematic (reproducible) component of the temperature drift without the need
for calibration at different temperatures during manufacturing. What remains is the random temperature drift,
which differs from device to device. Figure 73 illustrates the idea. The graph at the left schematically represents
the behavior of the drift error versus temperature at a certain input power level for a large number of devices.
ERROR
ERROR
+3s
+3s
T
T
MEAN
MEAN
-3s
-3s
AFTER
TEMPERATURE
CORRECTION
Figure 73. Elimination of the Systematic Component from the Temperature Drift
The mean drift error represents the reproducible - systematic - part of the error, while the mean ± 3 sigma limits
represent the combined systematic plus random error component. Obviously the drift error must be zero at
calibration temperature T0. If the systematic component of the drift error is included in the estimator, the total drift
error becomes equal to only the random component, as illustrated in the graph at the right of Figure 73. A
significant reduction of the temperature drift error can be achieved in this way only if:
• The systematic component is significantly larger than the random error component (otherwise the difference
is negligible).
• The operating temperature is measured with sufficient accuracy.
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Feature Description (continued)
It is essential for the effectiveness of the temperature compensation to assign the appropriate value to the
temperature sensitivity S1. Two different approaches can be followed to determine this parameter:
• Determination of a single value to be used over the entire operating temperature range.
• Division of the operating temperature range in segments and use of separate values for each of the
segments.
Also for the first method, the accuracy of the extracted temperature sensitivity increases when the number of
measurement temperatures increases. Linear regression to temperature can then be used to determine the two
parameters of the linear model for the temperature drift error: the first order temperature sensitivity S1 and the
best-fit (room temperature) value for the power estimate at T0: FDET[VOUT(T),T0]. Note that to achieve an overall over all temperatures - minimum error, the room temperature drift error in the model can be non-zero at the
calibration temperature (which is not in agreement with the strict definition).
The second method does not have this drawback but is more complex. In fact, segmentation of the temperature
range is a form of higher-order temperature compensation using only a first-order model for the different
segments: one for temperatures below 25°C, and one for temperatures above 25°C. The mean (or typical)
temperature sensitivity is the value to be used for compensation of the systematic drift error component.
Figure 75 shows the temperature drift error without and with temperature compensation using two segments.
With compensation the systematic component is completely eliminated; the remaining random error component
is centered around zero. Note that the random component is slightly larger at −40°C than at 85°C.
Figure 74. Temperature Drift Error without Temperature
Compensation
Figure 75. Temperature Drift Error without with
Temperature Compensation
In a practical power measurement system, temperature compensation is usually only applied to a small power
range around the maximum power level for two reasons:
• The various communication standards require the highest accuracy in this range to limit interference.
• The temperature sensitivity itself is a function of the power level it becomes impractical to store a large
number of different temperature sensitivity values for different power levels.
The 2.7-V DC and AC Electrical Characteristics in the datasheet specifies the temperature sensitivity for the
aforementioned two segments at an input power level of −10 dBm (near the top-end of the detector dynamic
range). The typical value represents the mean which is to be used for calibration.
7.3.2.2.2 Differential Power Errors
Many third generation communication systems contain a power control loop through the base station and mobile
unit that requests both to frequently update the transmit power level by a small amount (typically 1 dB). For such
applications it is important that the actual change of the transmit power is sufficiently close to the requested
power change.
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Feature Description (continued)
The error metrics in the datasheet that describe the accuracy of the detector for a change in the input power are
E1 dB (for a 1-dB change in the input power) and E10 dB (for a 10-dB step, or ten consecutive steps of 1 dB). Since
it can be assumed that the temperature does not change during the power step the differential error equals the
difference of the drift error at the two involved power levels:
E1dB(PIN,T)= EDRIFT(PIN+1dB,T) - EDRIFT(PIN,T)
E10dB(PIN,T)= EDRIFT(PIN+10dB,T) - EDRIFT(PIN,T)
(11)
It should be noted that the step error increases significantly when one (or both) power levels in the above
expression are outside the detector dynamic range. For E10 dB this occurs when PIN is less than 10 dB below the
maximum input power of the dynamic range, PMAX.
7.4 Device Functional Modes
7.4.1 Shutdown
To save power, the LMH2100 can be brought into a low-power shutdown mode. The device is active for EN =
HIGH (VEN>1.1 V) and in the low-power shutdown mode for EN = LOW (VEN < 0.6 V). In this state the output of
the LMH2100 is switched to a high impedance mode. Using the shutdown function, care must be taken not to
exceed the absolute maximum ratings. Forcing a voltage to the enable input that is 400 mV higher than VDD or
400 mV lower than GND will damage the device and further operations is not ensured. The absolute maximum
ratings can also be exceeded when the enable EN is switched to HIGH (from shutdown to active mode) while the
supply voltage is low (off). This should be prevented at all times. A possible solution to protect the part is to add
a resistor of 100 kΩ in series with the enable input.
7.4.1.1 Output Behavior in Shutdown
In order to save power, the LMH2100 can be used in pulsed mode, such that it is active to perform the power
measurement only during a fraction of the time. During the remaining time the device is in low-power shutdown.
Applications using this approach usually require that the output value is available at all times, also when the
LMH2100 is in shutdown. The settling time in active mode, however, should not become excessively large. This
can be realized by the combination of the LMH2100 and a low pass output filter (see Figure 81).
In active mode, the filter capacitor CS is charged to the output voltage of the LMH2100, which in this mode has a
low output impedance to enable fast settling. During shutdown-mode, the capacitor should preserve this voltage.
Discharge of CS through any current path should therefore be avoided in shutdown. The output impedance of the
LMH2100 becomes high in shutdown, such that the discharge current cannot flow from the capacitor top plate,
through RS, and the LMH2100 devices's OUT pin to GND. This is realized by the internal shutdown mechanism
of the output amplifier and by the switch depicted in Figure 85. Additionally, it should be ensured that the ADC
input impedance is high as well, to prevent a possible discharge path through the ADC.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 Functionality and Application of RF Power Detectors
This section describes the functional behavior of RF power detectors and their typical application. Based on a
number of key electrical characteristics of RF power detectors, Functionality of RF Power Detectors discusses
the functionality of RF power detectors in general and of the LMH2100 LOG detector in particular. Subsequently,
Typical Applications describes two important applications of the LMH2100 detector.
8.1.1.1 Functionality of RF Power Detectors
An RF power detector is a device that produces a DC output voltage in response to the RF power level of the
signal applied to its input. A wide variety of power detectors can be distinguished, each having certain properties
that suit a particular application. This section provides an overview of the key characteristics of power detectors,
and discusses the most important types of power detectors. The functional behavior of the LMH2100 is
discussed in detail.
8.1.1.1.1 Key Characteristics of RF Power Detectors
Power detectors are used to accurately measure the power of a signal inside the application. The attainable
accuracy of the measurement is therefore dependent upon the accuracy and predictability of the detector transfer
function from the RF input power to the DC output voltage.
Certain key characteristics determine the accuracy of RF detectors and they are classified accordingly:
• Temperature Stability
• Dynamic Range
• Waveform Dependency
• Transfer Shape
Generally, the transfer function of RF power detectors is slightly temperature dependent. This temperature drift
reduces the accuracy of the power measurement, because most applications are calibrated at room temperature.
In such systems, the temperature drift significantly contributes to the overall system power measurement error.
The temperature stability of the transfer function differs for the various types of power detectors. Generally,
power detectors that contain only one or few semiconductor devices (diodes, transistors) operating at RF
frequencies attain the best temperature stability.
The dynamic range of a power detector is the input power range for which it creates an accurately reproducible
output signal. What is considered accurate is determined by the applied criterion for the detector accuracy; the
detector dynamic range is thus always associated with certain power measurement accuracy. This accuracy is
usually expressed as the deviation of its transfer function from a certain predefined relationship, such as ”linear in
dB" for LOG detectors and ”square-law" transfer (from input RF voltage to DC output voltage) for Mean-Square
detectors. For LOG-detectors, the dynamic range is often specified as the power range for which its transfer
function follows the ideal linear-in-dB relationship with an error smaller than or equal to ±1 dB. Again, the
attainable dynamic range differs considerably for the various types of power detectors.
According to its definition, the average power is a metric for the average energy content of a signal and is not
directly a function of the shape of the signal in time. In other words, the power contained in a 0-dBm sine wave is
identical to the power contained in a 0-dBm square wave or a 0-dBm WCDMA signal; all these signals have the
same average power. Depending on the internal detection mechanism, though, power detectors may produce a
slightly different output signal in response to the aforementioned waveforms, even though their average power
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Application Information (continued)
level is the same. This is due to the fact that not all power detectors strictly implement the definition formula for
signal power, being the mean of the square of the signal. Most types of detectors perform some mixture of peak
detection and average power detection. A waveform independent detector response is often desired in
applications that exhibit a large variety of waveforms, such that separate calibration for each waveform becomes
impractical.
The shape of the detector transfer function from the RF input power to the DC output voltage determines the
required resolution of the ADC connected to it. The overall power measurement error is the combination of the
error introduced by the detector, and the quantization error contributed by the ADC. The impact of the
quantization error on the overall transfer's accuracy is highly dependent on the detector transfer shape, as shown
in Figure 76 and Figure 77.
2
2
ÂV1
VOUT (V)
VOUT (V)
ÂV
ÂV
ÂV2
0
-60
ÂP
ÂP
RF INPUT POWER (dBm)
0
0
-60
Figure 76. Convex Detector Transfer Function
ÂP
ÂP
RF INPUT POWER (dBm)
0
Figure 77. Linear Transfer Function
Figure 76 and Figure 77 shows two different representations of the detector transfer function. In both graphs the
input power along the horizontal axis is displayed in dBm, since most applications specify power accuracy
requirements in dBm (or dB). The figure on the left shows a convex detector transfer function, while the transfer
function on the right hand side is linear (in dB). The slope of the detector transfer function — the detector
conversion gain – is of key importance for the impact of the quantization error on the total measurement error. If
the detector transfer function slope is low, a change, ΔP, in the input power results only in a small change of the
detector output voltage, such that the quantization error will be relatively large. On the other hand, if the detector
transfer function slope is high, the output voltage change for the same input power change will be large, such
that the quantization error is small. The transfer function on the left has a very low slope at low input power
levels, resulting in a relatively large quantization error. Therefore, to achieve accurate power measurement in this
region, a high-resolution ADC is required. On the other hand, for high input power levels the quantization error
will be very small due to the steep slope of the curve in this region. For accurate power measurement in this
region, a much lower ADC resolution is sufficient. The curve on the right has a constant slope over the power
range of interest, such that the required ADC resolution for a certain measurement accuracy is constant. For this
reason, the LOG-linear curve on the right will generally lead to the lowest ADC resolution requirements for
certain power measurement accuracy.
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Application Information (continued)
8.1.1.1.2 Types of RF Power Detectors
Three different detector types are distinguished based on the four characteristics previously discussed:
• Diode Detector
• (Root) Mean Square Detector
• Logarithmic Detectors
8.1.1.1.2.1 Diode Detector
A diode is one of the simplest types of RF detectors. As depicted in Figure 78, the diode converts the RF input
voltage into a rectified current. This unidirectional current charges the capacitor. The RC time constant of the
resistor and the capacitor determines the amount of filtering applied to the rectified (detected) signal.
Z0
D
VREF
RS
CS
VOUT
Figure 78. Diode Detector
The advantages and disadvantages can be summarized as follows:
• The temperature stability of the diode detectors is generally very good, since they contain only one
semiconductor device that operates at RF frequencies.
• The dynamic range of diode detectors is poor. The conversion gain from the RF input power to the output
voltage quickly drops to very low levels when the input power decreases. Typically a dynamic range of 20 dB
to 25 dB can be realized with this type of detector.
• The response of diode detectors is waveform dependent. As a consequence of this dependency for example
its output voltage for a 0-dBm WCDMA signal is different than for a 0-dBm unmodulated carrier. This is due to
the fact that the diode measures peak power instead of average power. The relation between peak power and
average power is dependent on the wave shape.
• The transfer shape of diode detectors puts high requirements on the resolution of the ADC that reads their
output voltage. Especially at low input power levels a very high ADC resolution is required to achieve
sufficient power measurement accuracy (See Figure 76).
8.1.1.1.2.2 (Root) Mean Square Detector
This type of detector is particularly suited for the power measurements of RF modulated signals that exhibits
large peak to average power ratio variations. This is because its operation is based on direct determination of the
average power and not – like the diode detector – of the peak power.
The advantages and disadvantages can be summarized as follows:
• The temperature stability of (R)MS detectors is almost as good as the temperature stability of the diode
detector; only a small part of the circuit operates at RF frequencies, while the rest of the circuit operates at
low frequencies.
• The dynamic range of (R)MS detectors is limited. The lower end of the dynamic range is limited by internal
device offsets.
• The response of (R)MS detectors is highly waveform independent. This is a key advantage compared to other
types of detectors in applications that employ signals with high peak-to-average power variations. For
example, the (R)MS detector response to a 0-dBm WCDMA signal and a 0-dBm unmodulated carrier is
essentially equal.
• The transfer shape of R(MS) detectors has many similarities with the diode detector and is therefore subject
to similar disadvantages with respect to the ADC resolution requirements (see Figure 77).
8.1.1.1.2.3 Logarithmic Detectors
The transfer function of a logarithmic detector has a linear in dB response, which means that the output voltage
changes linearly with the RF power in dBm. This is convenient since most communication standards specify
transmit power levels in dBm as well.
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Application Information (continued)
The advantages and disadvantages can be summarized as follows:
• The temperature stability of the LOG detector transfer function is generally not as good as the stability of
diode and R(MS) detectors. This is because a significant part of the circuit operates at RF frequencies.
• The dynamic range of LOG detectors is usually much larger than that of other types of detectors.
• Since LOG detectors perform a kind of peak detection their response is wave form dependent, similar to
diode detectors.
• The transfer shape of LOG detectors puts the lowest possible requirements on the ADC resolution (See
Figure 77).
8.2 Typical Applications
RF power detectors can be used in a wide variety of applications. The first example shows the LMH2100 in a
Figure 79, the second application measures the Figure 88.
8.2.1 Application With Transmit Power Control Loop
The key benefit of a transmit power control loop circuit is that it makes the transmit power insensitive to changes
in the Power Amplifier (PA) gain control function, such as changes due to temperature drift. When a control loop
is used, the transfer function of the PA is eliminated from the overall transfer function. Instead, the overall
transfer function is determined by the power detector. The overall transfer function accuracy depends thus on the
RF detector accuracy. The LMH2100 is especially suited for this application, due to the accurate temperature
stability of its transfer function.
Figure 79 shows a block diagram of a typical transmit power control system. The output power of the PA is
measured by the LMH2100 through a directional coupler. The measured output voltage of the LMH2100 is
filtered and subsequently digitized by the ADC inside the baseband chip. The baseband adjusts the PA output
power level by changing the gain control signal of the RF VGA accordingly. With an input impedance of 50 Ω, the
LMH2100 can be directly connected to a 30 dB directional coupler without the need for an additional external
attenuator. The setup can be adjusted to various PA output ranges by selection of a directional coupler with the
appropriate coupling factor.
B
A
S
E
B
A
N
D
RF
COUPLER
PA
VGA
ANTENNA
50 :
GAIN
RS
OUT RFIN
ADC
CS
LMH2100
EN
LOGIC
GND
Figure 79. Transmit Power Control System
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Typical Applications (continued)
8.2.1.1 Design Requirements
Some of the design requirements for this logarithmic RMS power detector include:
Table 1. Design Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Supply voltage
2.7 V
RF input frequency (unmodulated continuous wave)
1855 MHz
Minimum power level
0 dBm
Maximum power level
–5 dBm
Maximum output voltage
2V
8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Detector Interfacing
For optimal performance of the LMH2100, it is important that all its pins are connected to the surrounding
circuitry in the appropriate way. This section discusses guidelines and requirements for the electrical connection
of each pin of the LMH2100 to ensure proper operation of the device. Starting from a block diagram, the function
of each pin is elaborated. Subsequently, the details of the electrical interfacing are separately discussed for each
pin. Special attention will be paid to the output filtering options and the differences between single ended and
differential interfacing with an ADC.
8.2.1.2.1.1 Concept of Power Measurements
Power measurement systems generally consists of two clearly distinguishable parts with different functions:
1. A power detector device, that generates a DC output signal (voltage) in response to the power level of the
(RF) signal applied to its input.
2. An “estimator” that converts the measured detector output signal into a (digital) numeric value representing
the power level of the signal at the detector input.
A sketch of this conceptual configuration is depicted in Figure 80.
FEST
MODEL
PIN
FDET
VOUT
PEST
PARAMETERS
Figure 80. Generic Concept of a Power Measurement System
The core of the estimator is usually implemented as a software algorithm, receiving a digitized version of the
detector output voltage. Its transfer FEST from detector output voltage to a numerical output should be equal to
the inverse of the detector transfer FDET from (RF) input power to DC output voltage. If the power measurement
system is ideal, that is, if no errors are introduced into the measurement result by the detector or the estimator,
the measured power PEST - the output of the estimator - and the actual input power PIN should be identical. In
that case, the measurement error E, the difference between the two, should be identically zero:
E = PEST - PIN { 0
œ PEST = FEST[FDET(PIN)] = PIN
-1
œ FEST(VOUT) = FDET(VOUT)
34
(12)
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From the expression above it follows that one would design the FEST transfer function to be the inverse of the
FDET transfer function.
In practice the power measurement error will not be zero, due to the following effects:
• The detector transfer function is subject to various kinds of random errors that result in uncertainty in the
detector output voltage; the detector transfer function is not exactly known.
• The detector transfer function might be too complicated to be implemented in a practical estimator.
The function of the estimator is then to estimate the input power PIN, that is, to produce an output PEST such that
the power measurement error is - on average - minimized, based on the following information:
1. Measurement of the not completely accurate detector output voltage VOUT
2. Knowledge about the detector transfer function FDET, for example the shape of the transfer function, the
types of errors present (part-to-part spread, temperature drift) etc.
Obviously the total measurement accuracy can be optimized by minimizing the uncertainty in the detector output
signal (select an accurate power detector), and by incorporating as much accurate information about the detector
transfer function into the estimator as possible.
The knowledge about the detector transfer function is condensed into a mathematical model for the detector
transfer function, consisting of:
• A formula for the detector transfer function.
• Values for the parameters in this formula.
The values for the parameters in the model can be obtained in various ways. They can be based on
measurements of the detector transfer function in a precisely controlled environment (parameter extraction). If
the parameter values are separately determined for each individual device, errors like part-to-part spread are
eliminated from the measurement system.
Errors may occur when the operating conditions of the detector (for example, the temperature) become
significantly different from the operating conditions during calibration (for example, room temperature). Examples
of simple estimators for power measurements that result in a number of commonly used metrics for the power
measurement error are discussed in LOG-Conformance Error, the Temperature Drift Error, the Temperature
Compensation and Temperature Drift Error.
8.2.1.2.1.2 RF Input
RF parts typically use a characteristic impedance of 50 Ω. To comply with this standard the LMH2100 has an
input impedance of 50 Ω. Using a characteristic impedance other then 50 Ω will cause a shift of the logarithmic
intercept with respect to the value given in the 2.7-V DC and AC Electrical Characteristics. This intercept shift
can be calculated according to Equation 13.
2 RSOURCE ·
PINT-SHIFT = 10 LOG §
© RSOURCE + 50¹
(13)
The intercept will shift to higher power levels for RSOURCE > 50 Ω, and will shift to lower power levels for RSOURCE
< 50 Ω.
8.2.1.2.1.3 Output and Reference
The possible filtering techniques that can be applied to reduce ripple in the detector output voltage are discussed
in Filtering. In addition two different topologies to connect the LMH2100 to an ADC are elaborated.
8.2.1.2.1.3.1 Filtering
The output voltage of the LMH2100 is a measure for the applied RF signal on the RF input pin. Usually, the
applied RF signal contains AM modulation that causes low frequency ripple in the detector output voltage. CDMA
signals for instance contain a large amount of amplitude variations. Filtering of the output signal can be used to
eliminate this ripple. The filtering can either be realized by a low pass output filter or a low pass feedback filter.
Those two techniques are depicted in Figure 81 and Figure 82.
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VDD
VDD
RFIN
2
1
6
OUT
RS
RFIN
+
CS
ADC
LMH2100
EN
4
5
3
REF
1
2
4
ADC
CP
REF
5
3
+
RP
LMH2100
EN
-
OUT
6
-
GND
GND
Figure 82. Low Pass Feedback Filter
Figure 81. Low Pass Output Filter
Depending on the system requirements one of the these filtering techniques can be selected. The low pass
output filter has the advantage that it preserves the output voltage when the LMH2100 is brought into shutdown.
This is elaborated in Output Behavior in Shutdown. In the feedback filter, resistor RP discharges capacitor CP in
shutdown and therefore changes the output voltage of the device.
A disadvantage of the low pass output filter is that the series resistor RS limits the output drive capability. This
may cause inaccuracies in the voltage read by an ADC when the ADC input impedance is not significantly larger
than RS. In that case, the current flowing through the ADC input induces an error voltage across filter resistor RS.
The low pass feedback filter doesn’t have this disadvantage.
Note that adding an external resistor between OUT and REF reduces the transfer gain (LOG-slope and LOGintercept) of the device. The internal feedback resistor sets the gain of the transimpedance amplifier.
The filtering of the low pass output filter is realized by resistor RS and capacitor CS. The −3 dB bandwidth of this
filter can then be calculated by: ƒ−3 dB = 1 / 2πRSCS. The bandwidth of the low pass feedback filter is determined
by external resistor RP in parallel with the internal resistor RTRANS, and external capacitor CP in parallel with
internal capacitor CTRANS (see Figure 85). The −3 dB bandwidth of the feedback filter can be calculated by ƒ−3 dB
= 1 / 2π (RP//RTRANS) (CP + CTRANS). The bandwidth set by the internal resistor and capacitor (when no external
components are connected between OUT and REF) equals ƒ−3 dB = 1 / 2π RTRANS CTRANS = 450 kHz.
8.2.1.2.1.4 Interface to the ADC
The LMH2100 can be connected to the ADC with a single-ended or a differential topology. The single ended
topology connects the output of the LMH2100 to the input of the ADC and the reference pin is not connected. In
a differential topology, both the output and the reference pins of the LMH2100 are connected to the ADC. The
topologies are depicted in Figure 83 and Figure 84.
VDD
VDD
RFIN
2
1
6
4
3
5
2
+
RP
LMH2100
EN
RFIN
OUT
ADC
CP
REF
1
6
OUT
RP
LMH2100
EN
-
4
3
5
+
REF
CP
ADC
-
GND
GND
Figure 84. Differential Application
Figure 83. Single-Ended
The differential topology has the advantage that it is compensated for temperature drift of the internal reference
voltage. This can be explained by looking at the transimpedance amplifier of the LMH2100 (Figure 85).
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REF
CTRANS
RTRANS
IDET
VREF
OUT
+
+
-
Figure 85. Output Stage of the LMH2100
It can be seen that the output of the amplifier is set by the detection current IDET multiplied by the resistor RTRANS
plus the reference voltage VREF:
VOUT = IDET RTRANS + VREF
(14)
IDET represents the detector current that is proportional to the RF input power. The equation shows that
temperature variations in VREF are also present in the output VOUT. In case of a single ended topology the output
is the only pin that is connected to the ADC. The ADC voltage for single ended is thus:
VADC = IDET RTRANS + VREF
(15)
A differential topology also connects the reference pin, which is the value of reference voltage VREF. The ADC
reads VOUT – VREF:
VADC = VOUT – VREF = IDET RTRANS
(16)
Equation 16 does not contain the reference voltage VREF anymore. Temperature variations in this reference
voltage are therefore not measured by the ADC.
8.2.1.3 Application Curves
2.0
2.0
1855 MHz
2.0
1.2
1.2
50 MHz
2500 MHz
0.8
0.8
RFIN = -15 dBm
1.2
RFIN = -25 dBm
0.8
RFIN = -35 dBm
3000 MHz
0.4
0.4
0.4
0.0
RFIN = -45 dBm
4000 MHz
0.0
-60
1.6
900 MHz
VOUT (V)
VOUT (V)
1.6
1.6
RFIN = - 5 dBm
-50
-40
-30
-20
-10
0
10
RF INPUT POWER (dBm)
0.0
10M
100M
1G
10G
FREQUENCY (Hz)
Figure 86. Output Voltage vs RF Input Power
Figure 87. Output Voltage vs Frequency
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8.2.2 Application With Voltage Standing Wave Ratio Measurement
Transmission in RF systems requires matched termination by the proper characteristic impedance at the
transmitter and receiver side of the link. In wireless transmission systems though, matched termination of the
antenna can rarely be achieved. The part of the transmitted power that is reflected at the antenna bounces back
toward the PA and may cause standing waves in the transmission line between the PA and the antenna. These
standing waves can attain unacceptable levels that may damage the PA. A Voltage Standing Wave Ratio
(VSWR) measurement is used to detect such an occasion. It acts as an alarm function to prevent damage to the
transmitter.
VSWR is defined as the ratio of the maximum voltage divided by the minimum voltage at a certain point on the
transmission line:
1+ _*_
VSWR =
1 - _*_
(17)
Where Γ = VREFLECTED / VFORWARD denotes the reflection coefficient.
This means that to determine the VSWR, both the forward (transmitted) and the reflected power levels have to
be measured. This can be accomplished by using two LMH2100 RF power detectors according to Figure 88. A
directional coupler is used to separate the forward and reflected power waves on the transmission line between
the PA and the antenna. One secondary output of the coupler provides a signal proportional to the forward power
wave, the other secondary output provides a signal proportional to the reflected power wave. The outputs of both
RF detectors that measure these signals are connected to a micro-controller or baseband that calculates the
VSWR from the detector output signals.
RF
COUPLER
ANTENNA
PA
MICRO
CONTROLLER
VDD
RFIN
REVERSE
POWER
EN
2
1
6
OUT
ADC1
LMH2100
4
3
5
RP1
CP1
REF
GND
RFIN
TRANSMITTED
POWER
EN
2
1
6
OUT
ADC2
LMH2100
4
3
5
RP2
CP2
REF
GND
Figure 88. VSWR Application
38
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9 Power Supply Recommendations
The LMH2100 is designed to operate from an input voltage supply range between 2.7 V to 3.3 V. This input
voltage must be well regulated. Enable voltage levels lower than 400 below GND could lead to incorrect
operation of the device. Also, the resistance of the input supply rail must be low enough to ensure correct
operation of the device.
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10 Layout
10.1
Layout Guidelines
As with any other RF device, careful attention must be paid to the board layout. If the board layout is not properly
designed, unwanted signals can easily be detected or interference will be picked up. This section gives
guidelines for proper board layout for the LMH2100.
Electrical signals (voltages and currents) need a finite time to travel through a trace or transmission line. RF
voltage levels at the generator side and at the detector side can therefore be different. This is not only true for
the RF strip line, but for all traces on the PCB. Signals at different locations or traces on the PCB will be in a
different phase of the RF frequency cycle. Phase differences in, for example, the voltage across neighboring
lines, may result in crosstalk between lines due to parasitic capacitive or inductive coupling. This crosstalk is
further enhanced by the fact that all traces on the PCB are susceptible to resonance. The resonance frequency
depends on the trace geometry. Traces are particularly sensitive to interference when the length of the trace
corresponds to a quarter of the wavelength of the interfering signal or a multiple thereof.
10.1.1 Supply Lines
Because the PSRR of the LMH2100 is finite, variations of the supply can result in some variation at the output.
This can be caused among others by RF injection from other parts of the circuitry or the on/off switching of the
PA.
10.1.1.1 Positive Supply (VDD)
In order to minimize the injection of RF interference into the LMH2100 through the supply lines, the phase
difference between the PCB traces connecting to VDD and GND should be minimized. A suitable way to achieve
this is to short both connections for RF. This can be done by placing a small decoupling capacitor between the
VDD and GND. It should be placed as close as possible to the VDD and GND pins of the LMH2100 as indicated
in Figure 91. Be aware that the resonance frequency of the capacitor itself should be above the highest RF
frequency used in the application, because the capacitor acts as an inductor above its resonance frequency.
Low frequency supply voltage variations due to PA switching might result in a ripple at the output voltage. The
LMH2100 has a PSRR of 60 dB for low frequencies.
10.1.1.2 Ground (GND)
The LMH2100 needs a ground plane free of noise and other disturbing signals. It is important to separate the RF
ground return path from the other grounds. This is due to the fact that the RF input handles large voltage swings.
A power level of 0 dBm will cause a voltage swing larger than 0.6 VPP, over the internal 50-Ω input resistor. This
will result in a significant RF return current toward the source. It is therefore recommended that the RF ground
return path not be used for other circuits in the design. The RF path should be routed directly back to the source
without loops.
10.1.2 RF Input Interface
The LMH2100 is designed to be used in RF applications, having a characteristic impedance of 50Ω. To achieve
this impedance, the input of the LMH2100 needs to be connected via a 50Ω transmission line. Transmission lines
can be easily created on PCBs using microstrip or (grounded) coplanar waveguide (GCPW) configurations. This
section will discuss both configurations in a general way. For more details about designing microstrip or GCPW
transmission lines, a microwave designer handbook is recommended.
10.1.3 Microstrip Configuration
One way to create a transmission line is to use a microstrip configuration. A cross section of the configuration is
shown in Figure 89, assuming a two-layer PCB.
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Layout Guidelines (continued)
METAL CONDUCTOR
W
FR4 PCB
H
GROUND PLANE
Figure 89. Microstrip Configuration
A conductor (trace) is placed on the topside of a PCB. The bottom side of the PCB has a fully copper ground
plane. The characteristic impedance of the microstrip transmission line is a function of the width W, height H, and
the dielectric constant εr.
Characteristics such as height and the dielectric constant of the board have significant impact on transmission
line dimensions. A 50-Ω transmission line may result in impractically wide traces. A typical 1.6-mm thick FR4
board results in a trace width of 2.9 mm, for instance. This is impractical for the LMH2100 because the pad width
of the 6-Bump DSBGA package is 0.24 mm. The transmission line has to be tapered from 2.9 mm to 0.24 mm.
Significant reflections and resonances in the frequency transfer function of the board may occur due to this
tapering.
10.1.4 GCPW Configuration
A transmission line in a (grounded) coplanar waveguide (GCPW) configuration will give more flexibility in terms of
trace width. The GCPW configuration is constructed with a conductor surrounded by ground at a certain
distance, S, on the top side. Figure 90 shows a cross section of this configuration. The bottom side of the PCB is
a ground plane. The ground planes on both sides of the PCB should be firmly connected to each other by
multiple vias. The characteristic impedance of the transmission line is mainly determined by the width W and the
distance S. In order to minimize reflections, the width W of the center trace should match the size of the package
pad. The required value for the characteristic impedance can subsequently be realized by selection of the proper
gap width S.
METAL CONDUCTOR
S
W
S
FR4 PCB
H
GROUND PLANE
Figure 90. GCPW Configuration
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Layout Guidelines (continued)
10.1.5 Reference (REF)
The Reference pin can be used to compensate for temperature drift of the internal reference voltage as
described in Interface to the ADC. The REF pin is directly connected to the inverting input of the transimpedance
amplifier. Thus, RF signals and other spurious signals couple directly through to the output. Introduction of RF
signals can be prevented by connecting a small capacitor between the REF pin and ground. The capacitor
should be placed close to the REF pin as depicted in Figure 91.
10.1.6 Output (OUT)
The OUT pin is sensitive to crosstalk from the RF input, especially at high power levels. The ESD diode between
OUT and VDD may rectify the crosstalk, but may add an unwanted inaccurate DC component to the output
voltage.
The board layout should minimize crosstalk between the detectors input RFIN and the detectors output. Using an
additional capacitor connected between the output and the positive supply voltage (VDD pin) or GND can
prevent this. For optimal performance this capacitor should be placed as close as possible to the OUT pin of the
LMH2100.
10.2 Layout Example
DECOUPLING
CAPACITOR
GND
GND
TRANSMISSION LINE
VDD
OUT
RFIN
REF
GND
EN
CROSSTALK FILTER
CAPACITOR
GND
GND
Figure 91. Recommended LMH2100 Board Layout
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11 Device and Documentation Support
11.1 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.2 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.3 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
11.4 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LMH2100TM/NOPB
ACTIVE
DSBGA
YFQ
6
250
RoHS & Green
SNAGCU
Level-1-260C-UNLIM
-40 to 85
J
LMH2100TMX/NOPB
ACTIVE
DSBGA
YFQ
6
3000
RoHS & Green
SNAGCU
Level-1-260C-UNLIM
-40 to 85
J
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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