LMH6321
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SNOSAL8C – APRIL 2006 – REVISED MARCH 2013
LMH6321 300 mA High Speed Buffer with Adjustable Current Limit
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FEATURES
DESCRIPTION
•
•
•
•
•
•
•
•
•
The LMH6321 is a high speed unity gain buffer that
slews at 1800 V/µs and has a small signal bandwidth
of 110 MHz while driving a 50Ω load. It can drive
±300 mA continuously and will not oscillate while
driving large capacitive loads.
1
2
High Slew Rate 1800 V/μs
Wide Bandwidth 110 MHz
Continuous Output Current ±300 mA
Output Current Limit Tolerance ±5 mA ±5%
Wide Supply Voltage Range 5V to ±15V
Wide Temperature Range −40°C to +125°C
Adjustable Current Limit
High Capacitive Load Drive
Thermal Shutdown Error Flag
APPLICATIONS
•
•
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Line Driver
Pin Driver
Sonar Driver
Motor Control
The LMH6321 features an adjustable current limit.
The current limit is continuously adjustable from 10
mA to 300 ma with a ±5 mA ±5% accuracy. The
current limit is set by adjusting an external reference
current with a resistor. The current can be easily and
instantly adjusted, as needed by connecting the
resistor to a DAC to form the reference current. The
sourcing and sinking currents share the same current
limit.
The LMH6321 is available in a space saving 8-pin SO
PowerPAD or a 7-pin DDPAK power package. The
SO PowerPAD package features an exposed pad on
the bottom of the package to increase its heat sinking
capability. The LMH6321 can be used within the
feedback loop of an operational amplifier to boost the
current output or as a stand alone buffer.
CONNECTION DIAGRAM
EF
1
8
2
7
NC
G=1
V
G=1
4
6 7
5
VOUT
V
GND
EF
CL
5
3
-
4
2
+
6
V
-
V
1
3
GND
VOUT
VIN
VIN
CL
+
A. V− pin is connected to tab on back of each package.
Figure 1. 8-Pin SO PowerPAD
Figure 2. 7-Pin DDPAK(A)
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2006–2013, Texas Instruments Incorporated
LMH6321
SNOSAL8C – APRIL 2006 – REVISED MARCH 2013
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ABSOLUTE MAXIMUM RATINGS (1) (2)
ESD Tolerance
(3)
Human Body Model
2.5 kV
Machine Model
250V
Supply Voltage
36V (±18V)
Input to Output Voltage
(4)
±5V
Input Voltage
±VSUPPLY
Output Short-Circuit to GND
(5)
Continuous
−65°C to +150°C
Storage Temperature Range
Junction Temperature (TJMAX)
Lead Temperature
+150°C
(Soldering, 10 seconds)
260°C
(6)
Power Dissipation
CL Pin to GND Voltage
(1)
(2)
(3)
(4)
(5)
(6)
±1.2V
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For specifications and the test conditions, see the
Electrical Characteristics Table.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
Human Body Model is 1.5 kΩ in series with 100 pF. Machine Model is 0Ω in series with 200 pF.
If the input-output voltage differential exceeds ±5V, internal clamping diodes will turn on. The current through these diodes should be
limited to 5 mA max. Thus for an input voltage of ±15V and the output shorted to ground, a minimum of 2 kΩ should be placed in series
with the input.
The maximum continuous current must be limited to 300mA. See APPLICATION HINTS for more details.
The maximum power dissipation is a function of TJ(MAX), θJA, and TA. The maximum allowable power dissipation at any ambient
temperature is PD = TJ(MAX)–TA)/θJA. See THERMAL MANAGEMENT of APPLICATION HINTS.
OPERATING RATINGS
−40°C to +125°C
Operating Temperature Range
Operating Supply Range
5V to ±16V
Thermal Resistance (θJA)
SO PowerPAD Package
(1)
Thermal Resistance (θJC)
180°C/W
DDPAK Package
4°C/W
Thermal Resistance (θJA)
(1)
2
80°C/W
Soldered to PC board with copper foot print equal to DAP size. Natural convection (no air flow). Board material is FR-4.
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SNOSAL8C – APRIL 2006 – REVISED MARCH 2013
±15V ELECTRICAL CHARACTERISTICS
The following specifications apply for Supply Voltage = ±15V, VCM = 0, RL ≥ 100 kΩ and RS = 50Ω, CL open, unless otherwise
noted. Boldface limits apply for TA = TJ = TMIN to TMAX; all other limits TA = TJ = 25°C.
Symbol
AV
Parameter
Voltage Gain
Min
Typ
RL = 1 kΩ, VIN = ±10V
Conditions
0.99
0.98
0.995
RL = 50Ω, VIN = ±10V
0.86
0.84
0.92
Max
Units
V/V
V/V
VOS
Input Offset Voltage
RL = 1 kΩ, RS = 0V
±4
±35
±52
mV
IB
Input Bias Current
VIN = 0V, RL = 1 kΩ, RS = 0V
±2
±15
±17
μA
R.IN
Input Resistance
R.L = 50Ω
250
kΩ
CIN
Input Capacitance
3.5
pF
RO
Output Resistance
IO = ±10 mA
5
IS
Power Supply Current
RL = ∞, VIN = 0
11
14.5
16.5
14.9
18.5
20.5
750 µA into
CL Pin
VO1
VO2
VO3
VEF
TSH
Positive Output Swing
IO = 300 mA, RS = 0V, VIN = ±VS
Negative Output Swing
IO = 300 mA, RS = 0V, VIN = ±VS
Positive Output Swing
RL = 1 kΩ, RS = 0V, VIN = ±VS
Negative Output Swing
RL = 1 kΩ, RS = 0V, VIN = ±VS
Positive Output Swing
RL = 50Ω, RS = 0V, VIN = ±VS
Negative Output Swing
RL = 50Ω, RS = 0V, VIN = ±VS
Error Flag Output Voltage
RL = ∞, VIN = 0,
EF pulled up with 5 kΩ
to +5V
Thermal Shutdown Temperature
−13.4
11.6
11.2
−11.9
Normal
5.00
During
Thermal
Shutdown
0.25
10
3
PSSR
Power Supply Rejection Ratio
RL = 1 kΩ, VIN = 0V,
VS = ±5V to ±15V
V
−12.9
−12.6
V
12.2
Hysteresis
EF pulled up with 5 kΩ to +5V
−10.3
−9.8
13.4
168
Supply Current at Thermal
Shutdown
Slew Rate
−11.3
13.1
12.9
mA
11.9
Measure Quantity is Die (Junction)
Temperature
ISH
SR
11.2
10.8
Ω
−10.9
−10.6
V
V
°C
Positive
58
54
66
Negative
58
54
64
VIN = ±11V, RL = 1 kΩ
2900
VIN = ±11V, RL = 50Ω
1800
mA
dB
V/μs
BW
−3 dB Bandwidth
VIN = ±20 mVPP, RL = 50Ω
110
MHz
LSBW
Large Signal Bandwidth
VIN = 2 VPP, RL = 50Ω
48
MHz
HD2
nd
2
Harmonic Distortion
VO = 2 VPP, f = 100 kHz
VO = 2 VPP, f = 1 MHz
RL = 50Ω
−59
RL = 100Ω
−70
RL = 50Ω
−57
RL = 100Ω
−68
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dBc
3
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±15V ELECTRICAL CHARACTERISTICS (continued)
The following specifications apply for Supply Voltage = ±15V, VCM = 0, RL ≥ 100 kΩ and RS = 50Ω, CL open, unless otherwise
noted. Boldface limits apply for TA = TJ = TMIN to TMAX; all other limits TA = TJ = 25°C.
Symbol
HD3
Parameter
3rd Harmonic Distortion
Conditions
VO = 2 VPP, f = 100 kHz
VO = 2 VPP, f = 1 MHz
Min
Typ
RL = 50Ω
−59
RL = 100Ω
−70
RL = 50Ω
−62
RL = 100Ω
−73
Max
dBc
en
Input Voltage Noise
f ≥ 10 kHz
in
Input Current Noise
f ≥ 10 kHz
ISC1
Output Short Circuit Current
Source (1)
VO = 0V,
Program Current
into CL = 25 µA
Sourcing
VIN = +3V
4.5
4.5
10
15.5
15.5
Sinking
VIN = −3V
4.5
4.5
10
15.5
15.5
VO = 0V
Program Current
into CL = 750 µA
Sourcing
VIN = +3V
280
273
295
308
325
Sinking
VIN = −3V
280
275
295
310
325
ISC2
Units
2.8
nV/√Hz
2.4
pA/√Hz
Output Short Circuit Current
Source
RS = 0V, VIN = +3V (1) (2)
320
300
570
750
920
Output Short Circuit Current Sink
RS = 0V, VIN = −3V (1) (2)
300
305
515
750
910
±0.5
±4.0
±8.0
mA
mA
mA
V/I Section
CLVOS
Current Limit Input Offset Voltage
RL = 1 kΩ, GND = 0V
CLIB
Current Limit Input Bias Current
RL = 1 kΩ
CL
CMRR
Current Limit Common Mode
Rejection Ratio
RL = 1 kΩ, GND = −13 to +14V
(1)
(2)
4
−0.5
−0.8
−0.2
60
56
69
mV
μA
dB
VIN = + or −4V at TJ = −40°C.
For the condition where the CL pin is left open the output current should not be continuous, but instead, should be limited to low duty
cycle pulse mode such that the RMS output current is less than or equal to 300 mA.
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SNOSAL8C – APRIL 2006 – REVISED MARCH 2013
±5V ELECTRICAL CHARACTERISTICS
The following specifications apply for Supply Voltage = ±5V, VCM = 0, RL ≥ 100 kΩ and RS = 50Ω, CL Open, unless otherwise
noted. Boldface limits apply for TA = TJ = TMIN to TMAX; all other limits TA = TJ = 25°C.
Symbol
AV
Parameter
Voltage Gain
Min
Typ
RL = 1 kΩ, VIN = ±3V
Conditions
0.99
0.98
0.994
RL = 50Ω, VIN = ±3V
0.86
0.84
0.92
Max
V/V
VOS
Offset Voltage
RL = 1 kΩ, RS = 0V
IB
Input Bias Current
RIN
Input Resistance
CIN
Input Capacitance
RO
Output Resistance
IOUT = ±10 mA
5
IS
Power Supply Current
RL = ∞, VIN = 0V
10
13.5
14.7
±2.5
±35
±50
mV
VIN = 0V, RL = 1 kΩ, RS = 0V
±2
±15
±17
μA
RL = 50Ω
250
kΩ
3.5
pF
14
17.5
19.5
750 μA into CL Pin
VO1
VO2
VO3
PSSR
ISC1
ISC2
Positive Output Swing
IO = 300 mA, RS = 0V, VIN = ±VS
Negative Output Swing
IO = 300 mA, RS = 0V, VIN = ±VS
Positive Output Swing
RL = 1 kΩ, RS = 0V, VIN = ±VS
Negative Output Swing
RL = 1 kΩ, RS = 0V, VIN = ±VS
Positive Output Swing
RL = 50Ω, RS = 0V, VIN = ±VS
Negative Output Swing
RL = 50Ω, RS = 0V, VIN = ±VS
Power Supply Rejection Ratio
RL = 1 kΩ, VIN = 0,
VS = ±5V to ±15V
Output Short Circuit Current
Output Short Circuit Current
Source
Slew Rate
1.3
0.9
−0.5
−0.1
3.5
−3.5
2.8
2.5
−3.1
−2.9
3.1
−3.0
V
V
V
V
−2.6
−2.4
58
54
66
Negative
58
54
64
VO = 0V, Program Current Sourcing
into CL = 25 μA
VIN = +3V
4.5
4.5
9
14.0
15.5
Sinking
VIN = −3V
4.5
4.5
9
14.0
15.5
VO = 0V, Program Current Sourcing
into CL = 750 μA
VIN = +3V
275
270
290
305
320
Sinking
VIN = −3V
275
270
290
310
320
300
470
300
400
RS = 0V, VIN = +3V (1) (2)
mA
1.9
−1.3
3.2
2.9
Ω
Positive
V
dB
mA
mA
Output Short Circuit Current Sink RS = 0V, VIN = −3V
SR
Units
(1) (2)
VIN = ±2 VPP, RL = 1 kΩ
450
VIN = ±2 VPP, RL = 50Ω
210
V/μs
BW
−3 dB Bandwidth
VIN = ±20 mVPP, RL = 50Ω
90
MHz
LSBW
Large Signal Bandwidth
VIN = 2 VPP, RL = 50Ω
39
MHz
TSD
Thermal Shutdown
Temperature
170
Hysteresis
10
°C
V/I Section
(1)
(2)
For the condition where the CL pin is left open the output current should not be continuous, but instead, should be limited to low duty
cycle pulse mode such that the RMS output current is less than or equal to 300 mA.
VIN = + or −4V at TJ = −40°C.
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±5V ELECTRICAL CHARACTERISTICS (continued)
The following specifications apply for Supply Voltage = ±5V, VCM = 0, RL ≥ 100 kΩ and RS = 50Ω, CL Open, unless otherwise
noted. Boldface limits apply for TA = TJ = TMIN to TMAX; all other limits TA = TJ = 25°C.
Symbol
Parameter
Conditions
CLVOS
Current Limit Input Offset
Voltage
RL = 1 kΩ, GND = 0V
CLIB
Current Limit Input Bias Current
RL = 1 kΩ, CL = 0V
CL
CMRR
Current Limit Common Mode
Rejection Ratio
RL = 1 kΩ, GND = −3V to +4V
6
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Min
Typ
Max
Units
2.7
+5
±5.0
mV
−0.5
−0.6
−0.2
60
56
65
μA
dB
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SNOSAL8C – APRIL 2006 – REVISED MARCH 2013
TYPICAL PERFORMANCE CHARACTERISTICS
Overshoot
vs.
Capacitive Load
Slew Rate
60
3000
UNDERSHOOT
SLEW RATE (V/Ps)
OVERSHOOT (%)
RL = 1 k:
2600
50
40
OVERSHOOT
30
20
2200
1800
RL = 50:
1400
1000
VIN = 100 mVPP
10
600
RL = OPEN
VS = ±15V
0
10
200
100
1k
10k
0
4
CL (pF)
Slew Rate
Small Signal Step Response
RL = 1 k:
2200
1800
INPUT SIGNAL
RL = 50:
1400
1000
600
8
12
20
16
20
VIN = 200 mVPP
(100 mV/DIV)
OUTPUT SIGNAL
VS = ±15V
SLEW RATE (V/Ps)
16
Figure 4.
2600
4
12
Figure 3.
3000
200
0
8
SUPPLY VOLTAGE (±V)
RL = 1 k:
VS = ±5V
TIME (10 ns/DIV)
24
Figure 5.
Figure 6.
Small Signal Step Response
Input Offset Voltage of Amplifier
vs.
Supply Voltage
VIN = 200 mVPP
RL = 1 k:
VS = ±15V
INPUT OFFSET VOLTAGE (mV)
10
(100 mV/DIV)
INPUT SIGNAL
OUTPUT SIGNAL
INPUT AMPLITUDE (VPP)
25°C
85°C
9
8
125°C
-40°C
7
6
TIME (10 ns/DIV)
3
5
7
9
11
13
15
SUPPLY VOLTAGE (±V)
Figure 7.
Figure 8.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
OUTPUT SIGNAL
VIN = 200 mVPP
RL = 50:
VS = ±5V
(100 mV/DIV)
Small Signal Step Response
INPUT SIGNAL
OUTPUT SIGNAL
INPUT SIGNAL
(100 mV/DIV)
Small Signal Step Response
RL = 50:
VS = ±15V
TIME (10 ns/DIV)
Figure 9.
Figure 10.
Large Signal Step Response—Leading Edge
Large Signal Step Response—Leading Edge
OUTPUT SIGNAL
RL = 1 k:
VS = ±15V
INPUT SIGNAL
VIN = 20 VPP
RL = 50:
VS = ±15V
(5V/DIV)
VIN = 20 VPP
(5V/DIV)
INPUT SIGNAL
OUTPUT SIGNAL
TIME (10 ns/DIV)
TIME (5 ns/DIV)
TIME (5 ns/DIV)
Large Signal Step Response — Trailing Edge
Large Signal Step Response — Trailing Edge
VS = ±15V
INPUT SIGNAL
VIN = 20 VPP
RL = 50:
VS = ±15V
(5V/DIV)
RL = 1 k:
(5V/DIV)
INPUT SIGNAL
VIN = 20 VPP
OUTPUT SIGNAL
Figure 12.
OUTPUT SIGNAL
Figure 11.
TIME (5 ns/DIV)
TIME (5 ns/DIV)
Figure 13.
8
VIN = 200 mVPP
Figure 14.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Large Signal Step Response
Large Signal Step Response
RL = 50:
VS = ±5V
VS = ±15V
VOUT (2V/DIV)
VOUT (0.5V/DIV)
RL = 1 k:
TIME (20 ns/DIV)
TIME (20 ns/DIV)
Figure 15.
Figure 16.
Large Signal Step Response
Large Signal Step Response
RL = 1 k:
VS = ±5V
VS = ±15V
VOUT (2V/DIV)
VOUT (0.5V/DIV)
RL = 50:
TIME (20 ns/DIV)
TIME (20 ns/DIV)
Figure 17.
Figure 18.
Harmonic Distortion with 50Ω Load
Harmonic Distortion with 100Ω Load
-20
-20
VS = ±15V
f = 1 MHz
-30
-40
HD2 and HD3 (dBc)
HD2 and HD3 (dBc)
-30
VS = ±15V
f = 1 MHz
HD2
-50
-60
HD3
-40
HD2
-50
-60
-70
-70
HD3
-80
-80
0
5
10
15
20
25
30
0
5
10
15
20
25
30
OUTPUT AMPLITUDE (VPP)
OUTPUT AMPLITUDE (VPP)
Figure 19.
Figure 20.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Noise
vs.
Frequency
Harmonic Distortion with 50Ω Load
10000
-30
VS = ±15V
1000
HD2
-45
100
NOISE
HD2 & HD3 (dBc)
-35 R = 50:
L
f = 100 kHz
-40
-50
10
-55
CURRENT pA/ Hz)
HD3
-60
VOLTAGE nV/ Hz)
1.0
-65
-70
0
5
10
15
20
0.1
1.0
25
10
Figure 22.
Gain
vs.
Frequency
Gain
vs.
Frequency
5
0
0
-5
-5
GAIN (dB)
GAIN (dB)
Figure 21.
-10
-15
-10
-20
VS = ±5V
-25
100k
1M
10M
100M
VS = ±15V
RL = 50:
-25
100k
1M
1G
10M
100M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 23.
Figure 24.
Gain
vs.
Frequency
Gain
vs.
Frequency
5
5
0
0
-5
-5
GAIN (dB)
GAIN (dB)
100k
-15
RL = 50:
-10
-15
1G
-10
-15
VS = ±5V
VS = ±15V
RL = 1 k:
-20
100k
10
10k
FREQUENCY (Hz)
5
-20
1k
100
OUTPUT VOLTAGE (V)
1M
10M
100M
1G
RL = 1 k:
-20
100k
1M
10M
100M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 25.
Figure 26.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Supply Current
vs.
Supply Voltage
Output Impedance
vs.
Sourcing Current
14
5.2
125°C
85°C
25°C
10
5
OUTPUT IMPEDANCE (:)
SUPPLY CURRENT (mA)
VS = ±5V
-40°C
12
-40°C
8
6
4
4.8
125°C
4.6
85°C
25°C
4.4
2
0
1
3
5
7
9
11
13
15
17
4.2
19
5
7
SUPPLY VOLTAGE (±V)
9
11
13
15
17
19
SOURCING CURRENT (mA)
Figure 27.
Figure 28.
Output Impedance
vs.
Sinking Current
Output Impedance
vs.
Sourcing Current
5.6
5
VS = ±15V
VS = ±5V
5.4
5.2
125°C
5
4.8
OUTPUT IMPEDANCE (:)
OUTPUT IMPEDANCE (:)
-40°C
4.8
-40°C
4.6
25°C
4.4
4.2
125°C
85°C
25°C
4.6
4
5
7
9
11
13
15
17
19
5
7
9
11
13
15
17
19
SINKING CURRENT (mA)
SOURCING CURRENT (mA)
Figure 29.
Figure 30.
Output Impedance
vs.
Sinking Current
Output Short Circuit Current—Sourcing vs.
Program Current
5.2
400
VS = ±15V
5
-40°C
4.8
25°C
4.6
VS = ±15V
OUTPUT CURRENT (mA)
OUTPUT IMPEDANCE (:)
85°C
4.4
4.2
7
9
11
200
125°C
-40°C
85°C
25°C
100
85°C
125°C
5
300
13
15
17
19
0
25
125 225 325 425 525 625 725 825
PROGRAM CURRENT (PA)
SINKING CURRENT (mA)
Figure 31.
Figure 32.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Output Short Circuit Current—Sinking vs.
Program Current
Output Short Circuit Current—Sourcing vs.
Program Current
400
400
VS = ±5V
300
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
VS = ±15V
125°C
-40°C
85°C
200
25°C
100
0
25
300
125°C
-40°C
85°C
200
25°C
100
0
25
125 225 325 425 525 625 725 825
125 225 325 425 525 625 725 825
PROGRAM CURRENT (PA)
PROGRAM CURRENT (PA)
Figure 33.
Figure 34.
Output Short Circuit Current—Sinking vs.
Program Current
Positive Output Swing
vs.
Sourcing Current
4
400
125°C
VS = ±5V
3.5
OUTPUT SWING (V)
OUTPUT CURRENT (mA)
85°C
300
125°C
-40°C
85°C
200
25°C
100
3
2.5
25°C
2
-40°C
1.5
1
VS = ±5V
+
VIN = V
0.5
0
25
CL = OPEN
0
125 225 325 425 525 625 725 825
0
100
PROGRAM CURRENT (PA)
200
300
Figure 35.
Figure 36.
Negative Output Swing
vs.
Sinking Current
Positive Output Swing
vs.
Sourcing Current
0
14
125°C
-0.5
+
VIN = V
OUTPUT SWING (V)
OUTPUT SWING (V)
13
-1.5
-40°C
-2
25°C
-2.5
-3.5
-4
85°C
VS = ±5V
-400
-40°C
11
9
-300
-200
-100
0
0
100
200
300
400
500
SOURCING CURRENT (mA)
SINKING CURRENT (mA)
Figure 37.
12
25°C
12
125°C
CL = OPEN
-500
CL = OPEN
10
-
VIN = V
500
VS = ±15V
85°C
-1
-3
400
SOURCING CURRENT (mA)
Figure 38.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Negative Output Swing
vs.
Sinking Current
Output Short Circuit Current—Sourcing vs.
Supply Voltage
-9
1000
VS = ±15V
OUTPUT SWING (V)
-10
-40°C
25°C
CL = OPEN
-11
25°C
-40°C
-12
-13
800
OUTPUT CURRENT (mA)
VIN = V
-
600
85°C
400
125°C
200
125°C
VIN = +3
85°C
-14
-500
-400
-300
-200
CL = OPEN
-100
0
0
2
4
6
SINKING CURRENT (mA)
10
12
14
Figure 39.
Figure 40.
Output Short Circuit Current—Sinking vs.
Supply Voltage
Positive Output Swing
vs.
Supply Voltage
800
16
18
15
-40°C
RL = 50:
25°C
13
600
85°C
125°C
400
-40°C
200
OUTPUT SWING (V)
OUTPUT CURRENT (mA)
8
SUPPLY VOLTAGE (±V)
11
125°C
9
85°C
7
-40°C
25°C
5
VIN = -3V
CL = OPEN
0
2
4
6
8
10
12
14
16
3
18
5
7
SUPPLY VOLTAGE (±V)
9
11
13
15
SUPPLY VOLTAGE (±V)
Figure 41.
Figure 42.
Positive Output Swing
vs.
Supply Voltage
Negative Output Swing
vs.
Supply Voltage
15
-3
RL = 50:
RL = 1 k:
13
-5
OUTPUT SWING (V)
OUTPUT SWING (V)
-40°C
11
125°C
9
85°C
-40°C
7
25°C
5
-7
25°C
85°C
-9
125°C
-11
-13
3
-15
5
7
9
11
13
15
5
7
9
11
13
SUPPLY VOLTAGE (±V)
SUPPLY VOLTAGE (±V)
Figure 43.
Figure 44.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Negative Output Swing
vs.
Supply Voltage
Input Offset Voltage of Amplifier
vs.
Common Mode Voltage
-3
15
VS = ±5V
INPUT OFFSET VOLTAGE (mV)
RL = 1 k:
OUTPUT SWING (V)
-5
-40°C
-7
25°C
85°C
-9
125°C
-11
-13
10
-40°C
5
25°C
0
125°C
85°C
-5
-15
5
7
9
11
13
15
2
Figure 46.
Input Offset Voltage of Amplifier vs.
Common Mode Voltage
Input Bias Current of Amplifier
vs.
Supply Voltage
3
0
INPUT BIAS CURRENT (PA)
125°C
15
-40°C
5
25°C
-40°C
-5
85°C
-15
125°C
-25
125°C
-2
85°C
-4
25°C
-40°C
-6
-8
-10
-12
-8
-4
0
4
8
12
5
3
7
9
11
13
15
SUPPLY VOLTAGE (±V)
COMMON MODE VOLTAGE (V)
Figure 47.
Figure 48.
Input Offset Voltage of V/I Section vs.
Common Mode Voltage
Input Offset Voltage of V/I Section vs.
Common Mode Voltage
5
4
VS = ±15V
INPUT OFFSET VOLTAGE (mV)
VS = ±5V
INPUT OFFSET VOLTAGE (mV)
1
COMMON MODE VOLTAGE (V)
VS = ±15V
4
3
25°C
-40°C
2
-40°C
1
85°C
0
-1
-2
-3
14
0
-1
Figure 45.
25
INPUT OFFSET VOLTGE (mV)
-2
-3
SUPPLY VOLTAGE (±V)
-2
-1
0
1
2
3
2
25°C
-40°C
0
-2
125°C
-4
85°C
-6
-8
-10
-12
-8
-4
0
4
8
COMMON MODE VOLTAGE (V)
COMMON MODE VOLTAGE (V)
Figure 49.
Figure 50.
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APPLICATION HINTS
BUFFERS
Buffers are often called voltage followers because they have largely unity voltage gain, thus the name has
generally come to mean a device that supplies current gain but no voltage gain. Buffers serve in applications
requiring isolation of source and load, i.e., high input impedance, low output impedance (high output current
drive). In addition, they offer gain flatness and wide bandwidth.
Most operational amplifiers, that meet the other given requirements in a particular application, can be configured
as buffers, though they are generally more complex and are, by and large, not optimized for unity gain operation.
The commercial buffer is a cost effective substitute for an op amp. Buffers serve several useful functions, either
in tandem with op amps or in standalone applications. As mentioned, their primary function is to isolate a high
impedance source from a low impedance load, since a high Z source can’t supply the needed current to the load.
For example, in the case where the signal source to an analog to digital converter is a sensor, it is recommended
that the sensor be isolated from the A/D converter. The use of a buffer ensures a low output impedance and
delivery of a stable output to the converter. In A/D converter applications buffers need to drive varying and
complex reactive loads.
Buffers come in two flavors: Open Loop and Closed Loop. While sacrificing the precision of some DC
characteristics, and generally displaying poorer gain linearity, open loop buffers offer lower cost and increased
bandwidth, along with less phase shift and propagation delay than do closed loop buffers. The LMH6321 is of the
open loop variety.
Figure 51 shows a simplified diagram of the LMH6321 topology, revealing the open loop complementary follower
design approach. Figure 52 shows the LMH6321 in a typical application, in this case, a 50Ω coaxial cable driver.
+
V
Q5
Q7
Q3
R1
D1 D3 D5 D7 D9 D11
R3
2:
Q1
VIN
VOUT
D2 D4 D6 D8D10 D12
R4
2:
Q2
R2
Q8
Q4
Q6
-
V
Figure 51. Simplified Schematic
SUPPLY BYPASSING
The method of supply bypassing is not critical for frequency stability of the buffer, and, for light loads, capacitor
values in the neighborhood of 1 nF to 10 nF are adequate. However, under fast slewing and large loads, large
transient currents are demanded of the power supplies, and when combined with any significant wiring
inductance, these currents can produce voltage transients. For example, the LMH6321 can slew typically at 1000
V/μs. Therefore, under a 50Ω load condition the load can demand current at a rate, di/dt, of 20 A/μs. This current
flowing in an inductance of 50 nH (approximately 1.5” of 22 gage wire) will produce a 1V transient. Thus, it is
recommended that solid tantalum capacitors of 5 μF to 10 μF, in parallel with a ceramic 0.1 μF capacitor be
added as close as possible to the device supply pins.
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V
R1
10 k:
TP1
EF
+
V
C2
0.1 PF
R3
R2
10 k: 1%
10 k: 1%
+
EF
INPUT
CL
VOUT
LMH6321
CIN
-
V
C1
1 nF
GND
50: COAXIAL
CABLE
|
VIN
VCL
R4
50:
OUTPUT
R6
50:
C3
0.1 PF
V
-
Figure 52. 50Ω Coaxial Cable Driver with Dual Supplies
For values of capacitors in the 10 μF to 100 μF range, ceramics are usually larger and more costly than
tantalums but give superior AC performance for bypassing high frequency noise because of their very low ESR
(typically less than 10 MΩ) and low ESL.
LOAD IMPEDANCE
The LMH6321 is stable under any capacitive load when driven by a 50Ω source. As shown by Figure 3 in
TYPICAL PERFORMANCE CHARACTERISTICS, worst case overshoot is for a purely capacitive load of about 1
nF. Shunting the load capacitance with a resistor will reduce the overshoot.
SOURCE INDUCTANCE
Like any high frequency buffer, the LMH6321 can oscillate with high values of source inductance. The worst case
condition occurs with no input resistor, and a purely capacitive load of 50 pF, where up to 100 nH of source
inductance can be tolerated. With a 50Ω load, this goes up to 200 nH. However, a 100Ω resistor placed in series
with the buffer input will ensure stability with a source inductances up to 400 nH with any load.
OVERVOLTAGE PROTECTION
(Refer to the simplified schematic in Figure 51).
If the input-to-output differential voltage were allowed to exceed the Absolute Maximum Rating of 5V, an internal
diode clamp would turn on and divert the current around the compound emitter followers of Q1/Q3 (D1 – D11 for
positive input), or around Q2/Q4 (D2 – D12 for negative inputs). Without this clamp, the input transistors Q1 – Q4
would zener, thereby damaging the buffer.
To limit the current through this clamp, a series resistor should be added to the buffer input (see R1 in Figure 52).
Although the allowed current in the clamp can be as high as 5 mA, which would suggest a 2 kΩ resistor from a
15V source, it is recommended that the current be limited to about 1 mA, hence the 10 kΩ shown.
The reason for this larger resistor is explained in the following: One way that the input/output voltage differential
can exceed the Abs Max value is under a short circuit condition to ground while driving the input with up to ±15V.
However, in the LMH6321 the maximum output current is set by the programmable Current Limit pin (CL). The
value set by this pin is specified to be accurate to 5 mA ±5%. If the input/output differential exceeds 5V while the
output is trying to supply the maximum set current to a shorted condition or to a very low resistance load, a
portion of that current will flow through the clamp diodes, thus creating an error in the total load current. If the
input resistor is too low, the error current can exceed the 5 mA ±5% budget.
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BANDWIDTH AND STABILITY
As can be seen in the schematic of Figure 52, a small capacitor is inserted in parallel with the series input
resistors. The reason for this is to compensate for the natural band-limiting effect of the 1st order filter formed by
this resistor and the input capacitance of the buffer. With a typical CIN of 3.5 pF (Figure 52), a pole is created at
fp2 = 1/(2πR1CIN) = 4.5 MHz
(1)
This will band-limit the buffer and produce further phase lag. If used in an op amp-loop application with an
amplifier that has the same order of magnitude of unity gain crossing as fp2, this additional phase lag will
produce oscillation.
The solution is to add a small feed-forward capacitor (phase lead) around the input resistor, as shown in
Figure 52. The value of this capacitor is not critical but should be such that the time constant formed by it and the
input resistor that it is in parallel with (RIN) be at least five times the time constant of RINCIN. Therefore,
C1 = (5RIN/R1)(CIN)
(2)
from Electrical Characteristics, RIN is 250 kΩ.
In the case of the example in Figure 52, RINCIN produces a time-constant of 870 ns, so C1 should be chosen to
be a minimum of 4.4 μs, or 438 pF. The value of C1 (1000 pF) shown in Figure 52 gives 10 μs.
OUTPUT CURRENT AND SHORT CIRCUIT PROTECTION
The LMH6321 is designed to deliver a maximum continuous output current of 300 mA. However, the maximum
available current, set by internal circuitry, is about 700 mA at room temperature. The output current is
programmable up to 300 mA by a single external resistor and voltage source.
The LMH6321 is not designed to safely output 700 mA continuously and should not be used this way. However,
the available maximum continuous current will likely be limited by the particular application and by the package
type chosen, which together set the thermal conditions for the buffer (see THERMAL MANAGEMENT) and could
require less than 300 mA.
The programming of both the sourcing and sinking currents into the load is accomplished with a single resistor.
Figure 53 shows a simplified diagram of the V to I converter and ISC protection circuitry that, together, perform
this task.
Referring to Figure 53, the two simplified functional blocks, labeled V/I Converter and Short Circuit Protection,
comprise the circuitry of the Current Limit Control.
The V/I converter consists of error amplifier A1 driving two PNP transistors in a Darlington configuration. The two
input connections to this amplifier are VCL (inverting input) and GND (non-inverting input). If GND is connected to
zero volts, then the high open loop gain of A1, as well as the feedback through the Darlington, will force CL, and
thus one end REXT to be at zero volts also. Therefore, a voltage applied to the other end of REXT will force a
current
IEXT = VPROG/REXT
(3)
into this pin. Via this pin, IOUT is programmable from 10 mA to 300 mA by setting IEXT from 25 μA to 750 µA by
means of a fixed REXT of 10 kΩ and making VCL variable from 0.25V to 7.5V. Thus, an input voltage VCL is
converted to a current IEXT. This current is the output from the V/I converter. It is gained up by a factor of two and
sent to the Short Circuit Protection block as IPROG. IPROG sets a voltage drop across RSC which is applied to the
non-inverting input of error amp A2. The other input is across RSENSE. The current through RSENSE, and hence the
voltage drop across it, is proportional to the load current, via the current sense transistor QSENSE. The output of
A2 controls the drive (IDRIVE) to the base of the NPN output transistor, Q3 which is, proportional to the amount
and polarity of the voltage differential (VDIFF ) between AMP2 inputs, that is, how much the voltage across RSENSE
is greater than or less than the voltage across RSC. This loop gains IEXT up by another 200, thus
ISC = 2 x 200 (IEXT) = 400 IEXT
(4)
Therefore, combining Equation 3 and Equation 4, and solving for REXT , we get
REXT = 400 VPROG/ISC
(5)
If the VCL pin is left open, the output short circuit current will default to about 700 mA. At elevated temperatures
this current will decrease.
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REXTERNAL
VCL
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IEXT 25 PA to 750 PA
VPROG
V/I CONVERTER
A1
GND
+
|
+
NPN OUTPUT
XTR
+
V
IOUT SENSE
XTR
IDRIVE
AMP2
QSENSE
TO INPUT
STAGE
R3
2:
SHORT CIRCUIT
PROTECTION
±VDIFF
RSENSE
200:
ILOAD
+
V
-
CONNECT TO GROUND
(FOR DUAL SUPPLIES)
OR MID RAIL FOR
SINGLE SUPPLY
RSC
400:
ISENSE
IPROG
50 PA to 1.5 mA
OUTPUT
TO LOWER OUTPUT STAGE
Only the NPN output ISC protection is shown. Depending on the polarity of VDIFF, AMP2 will turn IDRIVE either on or
off.
Figure 53. Simplified Diagram of Current Limit Control
THERMAL MANAGEMENT
Heatsinking
For some applications, a heat sink may be required with the LMH6321. This depends on the maximum power
dissipation and maximum ambient temperature of the application. To accomplish heat sinking, the tabs on
DDPAK and SO PowerPAD package may be soldered to the copper plane of a PCB for heatsinking (note that
these tabs are electrically connected to the most negative point in the circuit, i. e.,V−).
Heat escapes from the device in all directions, mainly through the mechanisms of convection to the air above it
and conduction to the circuit board below it and then from the board to the air. Natural convection depends on
the amount of surface area that is in contact with the air. If a conductive plate serving as a heatsink is thick
enough to ensure perfect thermal conduction (heat spreading) into the far recesses of the plate, the temperature
rise would be simply inversely proportional to the total exposed area. PCB copper planes are, in that sense, an
aid to convection, the difference being that they are not thick enough to ensure perfect conduction. Therefore,
eventually we will reach a point of diminishing returns (as seen in Figure 55). Very large increases in the copper
area will produce smaller and smaller improvement in thermal resistance. This occurs, roughly, for a 1 inch
square of 1 oz copper board. Some improvement continues until about 3 square inches, especially for 2 oz
boards and better, but beyond that, external heatsinks are required. Ultimately, a reasonable practical value
attainable for the junction to ambient thermal resistance is about 30 °C/W under zero air flow.
A copper plane of appropriate size may be placed directly beneath the tab or on the other side of the board. If
the conductive plane is placed on the back side of the PCB, it is recommended that thermal vias be used per
JEDEC Standard JESD51-5.
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Determining Copper Area
One can determine the required copper area by following a few basic guidelines:
1. Determine the value of the circuit’s power dissipation, PD
2. Specify a maximum operating ambient temperature, TA(MAX). Note that when specifying this parameter, it
must be kept in mind that, because of internal temperature rise due to power dissipation, the die
temperature, TJ, will be higher than TA by an amount that is dependent on the thermal resistance from
junction to ambient, θJA. Therefore, TA must be specified such that TJ does not exceed the absolute
maximum die temperature of 150°C.
3. Specify a maximum allowable junction temperature, TJ(MAX), which is the temperature of the chip at maximum
operating current. Although no strict rules exist, typically one should design for a maximum continuous
junction temperature of 100°C to 130°C, but no higher than 150°C which is the absolute maximum rating for
the part.
4. Calculate the value of junction to ambient thermal resistance, θJA
5. Choose a copper area that will ensure the specified TJ(MAX) for the calculated θJA. θJA as a function of copper
area in square inches is shown in Figure 54.
The maximum value of thermal resistance, junction to ambient θJA, is defined as:
θJA = (TJ(MAX) - TA(MAX) )/ PD(MAX)
where
•
•
•
TJ(MAX) = the maximum recommended junction temperature
TA(MAX) = the maximum ambient temperature in the user’s environment
PD(MAX) = the maximum recommended power dissipation
(6)
NOTE
The allowable thermal resistance is determined by the maximum allowable heat rise ,
TRISE = TJ(MAX) - TA(MAX) = (θJA) (PD(MAX)). Thus, if ambient temperature extremes force
TRISE to exceed the design maximum, the part must be de-rated by either decreasing PD
to a safe level, reducing θJA, further, or, if available, using a larger copper area.
Procedure
1. First determine the maximum power dissipated by the buffer, PD(MAX). For the simple case of the buffer
driving a resistive load, and assuming equal supplies, PD(MAX) is given by:
PD(MAX) = IS (2V+) + V+2/4RL
where
•
IS = quiescent supply current
(7)
2. Determine the maximum allowable die temperature rise,
TR(MAX) = TJ(MAX)-TA(MAX) = PD(MAX)θJA
(8)
3. Using the calculated value of TR(MAX) and PD(MAX) the required value for junction to ambient thermal
resistance can be found:
θJA = TR(MAX)/PD(MAX)
(9)
4. Finally, using this value for θJA choose the minimum value of copper area from Figure 54.
Example
Assume the following conditions:
V+ = V− = 15V, RL = 50Ω, IS = 15 mA TJ(MAX) = 125°C, TA(MAX) = 85°C.
1. From Equation 7
– PD(MAX) = IS (2V+) + V+2/4RL = (15 mA)(30V) + 15V2/200Ω = 1.58W
2. From Equation 8
– TR(MAX) = 125°C - 85°C = 40°C
3. From Equation 9
– θJA = 40°C/1.58W = 25.3°C/W
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Examining Figure 54, we see that we cannot attain this low of a thermal resistance for one layer of 1 oz copper.
It will be necessary to derate the part by decreasing either the ambient temperature or the power dissipation.
Other solutions are to use two layers of 1 oz foil, or use 2 oz copper (see Table 1), or to provide forced air flow.
One should allow about an extra 15% heat sinking capability for safety margin.
THERMAL RESISTANCE TJA (°C/W)
80
70
60
50
40
30
20
1
0
2
3
COPPER FOIL AREA (SQ. IN.)
Figure 54. Thermal Resistance (typ) for 7-L DDPAK Package Mounted on 1 oz. (0.036 mm) PC Board Foil
5
MAX POWER DISSIPATION (W)
4
3
2
1
TO-263 PACKAGE
PCB MOUNT
1 SQ. IN. COPPER
0
-40
-25
25
75
125
AMBIENT TEMPERATURE (°C)
Figure 55. Derating Curve for DDPAK package. No Air Flow
Table 1. θJA vs. Copper Area and PD for DDPAK. 1.0 oz cu Board. No Air Flow. Ambient Temperature =
24°C
θJA @ 1.0W
(°C/W)
θJA @ 2.0W
(°C/W)
1 Layer = 1”x2” cu Bottom
62.4
54.7
2 Layer = 1”x2” cu Top & Bottom
36.4
32.1
2 Layer = 2”x2” cu Top & Bottom
23.5
22.0
2 Layer = 2”x4” cu Top & Bottom
19.8
17.2
Copper Area
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As seen in the previous example, buffer dissipation in DC circuit applications is easily computed. However, in AC
circuits, signal wave shapes and the nature of the load (reactive, non-reactive) determine dissipation. Peak
dissipation can be several times the average with reactive loads. It is particularly important to determine
dissipation when driving large load capacitance.
A selection of thermal data for the SO PowerPAD package is shown in Table 2. The table summarizes θJA for
both 0.5 watts and 0.75 watts. Note that the thermal resistance, for both the DDPAK and the SO PowerPAD
package is lower for the higher power dissipation levels. This phenomenon is a result of the principle of Newtons
Law of Cooling. Restated in term of heatsink cooling, this principle says that the rate of cooling and hence the
thermal conduction, is proportional to the temperature difference between the junction and the outside
environment (ambient). This difference increases with increasing power levels, thereby producing higher die
temperatures with more rapid cooling.
Table 2. θJA vs. Copper Area and PD for SO PowerPAD. 1.0 oz cu Board. No Airflow. Ambient
Temperature = 22°C
θJA @ 0.5W
(°C/W)
θJA @ 0.75W
(°C/W)
1 Layer = 0.05 sq. in. (Bottom) + 3 Via Pads
141.4
138.2
1 Layer = 0.1 sq. in. (Bottom) + 3 Via Pads
134.4
131.2
1 Layer = 0.25 sq. in. (Bottom) + 3 Via Pads
115.4
113.9
1 Layer = 0.5 sq. in. (Bottom) + 3 Via Pads
105.4
104.7
1 Layer = 1.0 sq. in. (Bottom) + 3 Via Pads
100.5
100.2
2 Layer = 0.5 sq. in. (Top)/ 0.5 sq. in. (Bottom) + 33
Via Pads
93.7
92.5
2 Layer = 1.0 sq. in. (Top)/ 1.0 sq. in. (Bottom) + 53
Via Pads
82.7
82.2
Copper Area/Vias
ERROR FLAG OPERATION
The LMH6321 provides an open collector output at the EF pin that produces a low voltage when the Thermal
Shutdown Protection is engaged, due to a fault condition. Under normal operation, the Error Flag pin is pulled up
to V+ by an external resistor. When a fault occurs, the EF pin drops to a low voltage and then returns to V+ when
the fault disappears. This voltage change can be used as a diagnostic signal to alert a microprocessor of a
system fault condition. If the function is not used, the EF pin can be either tied to ground or left open. If this
function is used, a 10 kΩ, or larger, pull-up resistor (R2 in Figure 52) is recommended. The larger the resistor the
lower the voltage will be at this pin under thermal shutdown. Table 3 shows some typical values of VEF for 10 kΩ
and 100 kΩ.
Table 3. VEF vs. R2
R2( inFigure 52)
@ V+ = 5V
@V+ = 15V
10 kΩ
0.24V
0.55V
100 KΩ
0.036V
0.072V
SINGLE SUPPLY OPERATION
If dual supplies are used, then the GND pin can be connected to a hard ground (0V) (as shown in Figure 52).
However, if only a single supply is used, this pin must be set to a voltage of one VBE (∼0.7V) or greater, or more
commonly, mid rail, by a stiff, low impedance source. This precludes applying a resistive voltage divider to the
GND pin for this purpose. Figure 56 shows one way that this can be done.
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21
LMH6321
SNOSAL8C – APRIL 2006 – REVISED MARCH 2013
www.ti.com
+
V
LMH6321
+
GND
V
V
-
-
R1
OP AMP
+
R2
Figure 56. Using an Op Amp to Bias the GND Pin to ½ V+ for Single Supply Operation
In Figure 56, the op amp circuit pre-biases the GND pin of the buffer for single supply operation.
The GND pin can be driven by an op amp configured as a constant voltage source, with the output voltage set by
the resistor voltage divider, R1 and R2. It is recommended that These resistors be chosen so as to set the GND
pin to V+/2, for maximum common mode range.
SLEW RATE
Slew rate is the rate of change of output voltage for large-signal step input changes. For resistive load, slew rate
is limited by internal circuit capacitance and operating current (in general, the higher the operating current for a
given internal capacitance, the faster is the slew rate). Figure 57 shows the slew capabilities of the LMH6321
under large signal input conditions, using a resistive load.
3000
VS = ±15V
SLEW RATE (V/Ps)
2600
RL = 1 k:
2200
1800
RL = 50:
1400
1000
600
200
0
4
8
12
16
20
24
INPUT AMPLITUDE (VPP)
Figure 57. Slew Rate vs. Peak-to-Peak Input Voltage
However, when driving capacitive loads, the slew rate may be limited by the available peak output current
according to the following expression.
dv/dt = IPK/CL
22
(10)
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LMH6321
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SNOSAL8C – APRIL 2006 – REVISED MARCH 2013
and rapidly changing output voltages will require large output load currents. For example if the part is required to
slew at 1000 V/μs with a load capacitance of 1 nF the current demand from the LMH6321 would be 1A.
Therefore, fast slew rate is incompatible with large CL. Also, since CL is in parallel with the load, the peak current
available to the load decreases as CL increases.
Figure 58 illustrates the effect of the load capacitance on slew rate. Slew rate tests are specified for resistive
loads and/or very small capacitive loads, otherwise the slew rate test would be a measure of the available output
current. For the highest slew rate, it is obvious that stray load capacitance should be minimized. Peak output
current should be kept below 500 mA. This translates to a maximum stray capacitance of 500 pF for a slew rate
of 1000 V/μs.
10000
SLEW RATE (V/Ps)
1000
100
10
1
0.1
0.1
1
10
100
1000
CAPACITANCE (nF)
Figure 58. Slew Rate vs. Load Capacitance
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SNOSAL8C – APRIL 2006 – REVISED MARCH 2013
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REVISION HISTORY
Changes from Revision B (March 2013) to Revision C
•
24
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 23
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PACKAGE OPTION ADDENDUM
www.ti.com
6-Feb-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
LMH6321MR/NOPB
ACTIVE SO PowerPAD
DDA
8
95
Green (RoHS
& no Sb/Br)
SN
Level-3-260C-168 HR
-40 to 125
LMH63
21MR
LMH6321MRX/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
SN
Level-3-260C-168 HR
-40 to 125
LMH63
21MR
LMH6321TS/NOPB
ACTIVE
DDPAK/
TO-263
KTW
7
45
Pb-Free (RoHS
Exempt)
SN
Level-3-245C-168 HR
-40 to 125
LMH6321TS
LMH6321TSX/NOPB
ACTIVE
DDPAK/
TO-263
KTW
7
500
Pb-Free (RoHS
Exempt)
SN
Level-3-245C-168 HR
-40 to 125
LMH6321TS
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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