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LMH6629
SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
LMH6629 Ultra-Low Noise, High-Speed Operational Amplifier with Shutdown
1 Features
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1
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•
•
•
•
•
•
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3 Description
Specified for VS = 5 V, RL = 100 Ω, AV = 10V/V
WSON-8 Package, unless Specified. –3dB
Bandwidth 900 MHz
Input Voltage Noise 0.69 nV/√Hz
Input Offset Voltage Max. Over Temperature ±0.8
mV
Slew Rate 1600 V/ μs
HD2 @ f = 1 MHz, 2VPP −90 dBc
HD3 @ f = 1 MHz, 2VPP −94 dBc
Supply Voltage Range 2.7 V to 5.5 V
Typical Supply Current 15.5 mA
Selectable Min. Gain ≥4 or ≥10 V/V
Enable Time: 75 ns
Output Current ±250 mA
WSON-8 and SOT-23-5 Packages
2 Applications
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Instrumentation Amplifiers
Ultrasound Pre-amps
Wide-band Active Filters
Opto-Electronics
Medical Imaging Systems
Base-Station Amplifiers
Low-Noise Single Ended to Differential
Conversion
Trans-Impedance Amplifier
The LMH6629 is a high-speed, ultra-low noise
amplifier designed for applications requiring wide
bandwidth with high gain and low noise such as in
communication, test and measurement, optical and
ultrasound systems.
The LMH6629 operates on 2.7-V to 5.5-V supply with
an input common mode range that extends below
ground and outputs that swing to within 0.8 V of the
rails for ease of use in single supply applications.
Heavy loads up to ±250 mA can be driven by highfrequency large signals with the LMH6629's –3dB
bandwidth of 900 MHz and 1600 V/µs slew rate. The
LMH6629 (WSON-8 package only) has userselectable internal compensation for minimum gains
of 4 or 10 controlled by pulling the COMP pin low or
high, thereby avoiding the need for external
compensation capacitors required in competitive
devices. Compensation for the SOT-23-5 package is
internally set for a minimum stable gain of 10 V/V.
The WSON-8 package also provides the power-down
enable/disable feature.
The low-input noise (0.69 nV/√Hz and 2.6 pA/√Hz),
low distortion (HD2/HD3 = −90 dBc/−94 dBc) and
ultra-low DC errors (800 µV VOS maximum over
temperature, ±0.45 µV/°C drift) allow precision
operation in both ac- and dc-coupled applications.
The LMH6629 is fabricated in Texas Instruments'
proprietary SiGe process and is available in a 3 mm ×
3 mm 8-pin WSON package as well as the SOT-23-5
package.
Device Information(1)
PART NUMBER
LMH6629
PACKAGE
BODY SIZE (NOM)
SOT-23 (5)
2.90 mm × 1.60 mm
WSON (8)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
Transimpedance Amplifier
CF
0.6 pF
(1.2 pF (2) series)
+5VDC
D1
RF
1.2k
CD = 10 pF
ID
+5VDC
LMH
6629
+5VDC
RL
500
R2
2k
C1
0.1 PF
R1
3k
Vout = 3VDC - 1200 x ID
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LMH6629
SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
6.7
7
1
1
1
2
3
4
Absolute Maximum Ratings ...................................... 4
ESD Ratings.............................................................. 4
Recommended Operating Conditions....................... 4
Thermal Information .................................................. 4
Electrical Characteristics 5V .................................... 5
Electrical Characteristics 3.3V ................................. 8
Typical Performance Characteristics ...................... 11
Detailed Description ............................................ 19
7.1 Overview ................................................................. 19
7.2 Functional Block Diagram ....................................... 19
7.3 Feature Description................................................. 20
7.4 Device Functional Modes........................................ 31
8
Application and Implementation ........................ 32
8.1 Application Information............................................ 32
8.2 Typical Application .................................................. 32
9 Power Supply Recommendations...................... 34
10 Layout................................................................... 35
10.1 Layout Guidelines ................................................. 35
10.2 Layout Example .................................................... 36
11 Device and Documentation Support ................. 38
11.1
11.2
11.3
11.4
Documentation Support ........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
38
38
38
38
12 Mechanical, Packaging, and Orderable
Information ........................................................... 38
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision H (October 2014) to Revision I
Page
•
Updated ESD Ratings table. .................................................................................................................................................. 4
•
Revised paragraph beginning with "The optimum value of "CF in the Low-Noise Transimpedance Amplifier section ........ 29
•
Updated Related Documentation section ............................................................................................................................. 38
Changes from Revision G (March 2013) to Revision H
•
Added, updated, or renamed the following sections: Device Information Table, Pin Configuration and Functions,
Application and Implementation; Power Supply Recommendations; Layout; Device and Documentation Support;
Mechanical, Packaging, and Ordering Information................................................................................................................. 1
Changes from Revision F (March 2013) to Revision G
•
2
Page
Page
Changed layout of National Data Sheet to TI format ............................................................................................................. 1
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SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
5 Pin Configuration and Functions
5-Pin
Package DBV
Top View
OUT
V
-
1
5
+
V
2
+
PD
1
8
V
FB
2
7
OUT
IN-
3
6
COMP
IN+
4
5
V
DAP
-
+
+IN
8-Pin
Package NGQ08A
Top View
4
3
V-
-
-IN
Pin Functions
NAME
NUMBER
DBV
NGQ08A
I/O
COMP
6
I
FB
2
I/O
DESCRIPTION
Compensation
Feedback
-IN
4
3
I
Inverting input
+IN
3
4
I
Non-inverting input
OUT
1
7
O
Output
1
I
Power Down
PD
V-
2
5
I
Negative supply
+
5
8
I
Positive supply
V
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SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
(1) (2) (3)
Positive Supply Voltage
MIN
MAX
UNIT
−0.5
6.0
V
Differential Input Voltage
3
V
±10
mA
Analog Input Voltage
−0.5 to VS
V
Digital Input Voltage
−0.5 to VS
V
+150
°C
+150
°C
Input current
Junction Temperature
−65
Storage Temperature (Tstg)
(1)
(2)
(3)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
Electrical table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very
limited self-heating of the device such that TJ = TA. No guarantee of parametric performance is indicated in the electrical tables under
conditions of internal self-heating where TJ > TA.
6.2 ESD Ratings
VALUE
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
V(ESD)
(1)
(2)
Electrostatic discharge
(1)
UNIT
±2000
Machine model
±200
Charged-device model (CDM), per JEDEC specification JESD22C101 (2)
±750
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. Manufacturing with
less than 500-V HBM is possible with the necessary precautions. Pins listed as ±2000 V may actually have higher performance.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. Manufacturing with
less than 250-V CDM is possible with the necessary precautions. Pins listed as ±750 V may actually have higher performance.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
(1)
MIN
Supply Voltage (V+ - V−)
Operating Temperature Range
(1)
NOM
MAX
UNIT
2.7
5.5
V
−40
+125
°C
Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test
conditions, see the Electrical Characteristics.
6.4 Thermal Information
THERMAL METRIC (1)
RθJA
(1)
4
Junction-to-ambient thermal resistance
DBV
NGQ08A
5 PINS
8 PINS
179
71
UNIT
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
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SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
6.5 Electrical Characteristics 5V
The following specifications apply for single supply with VS = 5 V, RL = 100 Ω terminated to 2.5 V, gain = 10V/V, VO = 2VPP,
VCM = VS/2, COMP Pin = HI (WSON-8 package), unless otherwise noted. (1)
PARAMETER
TEST CONDITIONS
TA = 25°C
MIN (2)
TYP (3)
UNIT
MAX (2)
DYNAMIC PERFORMANCE
SSBW
LSBW
Small Signal
−3dB bandwidth
VO = 200 mVPP, WSON-8 package
900
VO = 200 mVPP, SOT-23-5 package
1000
AV= 4, VO = 200 mVPP,
COMP Pin = LO
Large signal −3dB VO = 2VPP
bandwidth
COMP Pin = LO, AV= 4, VO = 2VPP
0.1 dB bandwidth
Peaking
SR
Slew rate
tr/ tf
Rise/fall time
Ts
800
380
330
AV= 10, VO = 200 mVPP, SOT-23-5 package
190
MHz
95
VO = 200 mVPP, WSON-8 package
0
VO = 200 mVPP, SOT-23-5 package
2
AV= 10, 2 V step
MHz
190
AV= 10, VO = 200 mVPP, WSON-8 package
AV= 4, VO = 200 mVPP,
COMP Pin = LO
MHz
dB
1600
AV= 4, 2 V step, COMP Pin = LO
530
AV= 10, 2 V step, 10% to 90%,
WSON-8 package
0.90
AV= 10, 2 V step, 10% to 90%,
SOT-23-5 package
0.95
AV= 4, 2 V step, 10% to 90%,
COMP Pin = LO, (Slew Rate Limited)
2.8
Settling time
AV= 10, 1 V step, ±0.1%
42
Overload
recovery
VIN = 1 VPP
V/μs
ns
2
NOISE and DISTORTION
HD2
HD3
2nd Order
distortion
3rd Order
distortion
OIP3
Two-tone 3rd
order intercept
point
en
Noise voltage
in
Noise current
NF
Noise figure
(1)
(2)
(3)
fc = 1 MHz, VO = 2 VPP
−90
COMP Pin = LO, AV= 4, fc = 1 MHz,
VO = 2 VPP
−88
fc = 10 MHz, VO = 2 VPP
−70
COMP Pin = LO, fc = 10 MHz,
AV= 4 V, VO = 2 VPP
−65
fc = 1 MHz, VO = 2VPP
−94
COMP Pin = LO, AV= 4,
fc = 1 MHz, VO = 2 VPP
−87
fc = 10 MHz, VO = 2 VPP
−82
COMP Pin = LO, fc = 10 MHz, VO = 2VPP
−75
fc = 25 MHz, VO = 2 VPP composite
31
fc = 75 MHz, VO = 2 VPP composite
27
Input referred f > 1MHz
RS = RT = 50 Ω
dBc
dBc
dBm
0.69
nV/√Hz
2.6
pA/√Hz
8.0
dB
Electrical table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very
limited self-heating of the device such that TJ = TA. No guarantee of parametric performance is indicated in the electrical tables under
conditions of internal self-heating where TJ > TA.
All limits are ensured by testing or statistical analysis
Typical numbers are the most likely parametric norm.
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SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
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Electrical Characteristics 5V (continued)
The following specifications apply for single supply with VS = 5 V, RL = 100 Ω terminated to 2.5 V, gain = 10V/V, VO = 2VPP,
VCM = VS/2, COMP Pin = HI (WSON-8 package), unless otherwise noted.(1)
PARAMETER
TEST CONDITIONS
TA = 25°C
MIN (2)
TYP (3)
MAX (2)
UNIT
ANALOG I/O
CMVR
Input voltage
range
VO
Output voltage
range
0.89
-40°C ≤ TJ ≤ +125°C
IOUT
Linear output
current
VOS
Input offset
voltage
TcVOS
Input offset
voltage
temperature drift
See
(5)
IBI
Input bias current
See
(6)
IOS
Input offset
current
TCIOS
Input offset
voltage
temperature drift
CDIFF
RCM
Input capacitance
Input resistance
VO = 2.5 V
-40°C ≤ TJ ≤ +125°C
0.82 to 4.19
0.95
0.76
No Load
3.8
−0.30 to 3.8
CMRR > 70 dB, SOT-23-5 package
RL = 100 Ω to VS/2
CCM
−0.30
CMRR > 70 dB, WSON-8 package
0.85
(4)
4.1
250
-40°C ≤ TJ ≤ +125°C
mA
±780
±800
−15
-40°C ≤ TJ ≤ +125°C
−23
−37
±0.1
-40°C ≤ TJ ≤ +125°C
±1.8
±3.0
±2.8
Common Mode
1.7
Differential Mode (7)
4
Common Mode
450
µV
μV/°C
±0.45
(5)
V
4.0
±150
See
4.0
3.9
0.72 to 4.28
V
μA
μA
nA/°C
pF
kΩ
MISCELLANEOUS PARAMETERS
CMRR
Common mode
rejection ratio
PSRR
Power supply
rejection ratio
VCM from 0 V to 3.7 V, WSON-8
package
82
-40°C ≤ TJ ≤ +125°C
VCM from 0 V to 3.7 V, SOT-23-5 package
AVOL
Open loop gain
WSON-8 package
(5)
(6)
(7)
6
83
78
74
-40°C ≤ TJ ≤ +125°C
SOT-23-5 package
(4)
87
81
-40°C ≤ TJ ≤ +125°C
87
70
dB
78
72
78
The maximum continuous output current (IOUT) is determined by device power dissipation limitations. Continuous short circuit operation
at elevated ambient temperature can result in exceeding the maximum allowed junction temperature of 150°C
Drift determined by dividing the change in parameter at temperature extremes by the total temperature change.
Negative input current implies current flowing out of the device
Simulation results.
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SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
Electrical Characteristics 5V (continued)
The following specifications apply for single supply with VS = 5 V, RL = 100 Ω terminated to 2.5 V, gain = 10V/V, VO = 2VPP,
VCM = VS/2, COMP Pin = HI (WSON-8 package), unless otherwise noted.(1)
PARAMETER
TEST CONDITIONS
TA = 25°C
MIN (2)
TYP (3)
MAX (2)
UNIT
DIGITAL INPUTS/TIMING
VIL
Logic low-voltage
threshold
VIH
Logic high-voltage
PD and COMP pins, WSON-8 package
threshold
IIL
Logic Low-bias
current
PD and COMP pins = 0.8 V,
WSON-8 package (6)
IIH
Logic High-bias
current
PD and COMP pins = 2.5 V,
WSON-8 package (6)
Ten
Enable time
Tdis
Disable time
PD and COMP pins, WSON-8 package
0.8
V
2.5
−23
-40°C ≤ TJ ≤ +125°C
−16
-40°C ≤ TJ ≤ +125°C
−28
−19
−34
−38
−22
−14
−27
−29
75
WSON-8 package
µA
ns
80
POWER REQUIREMENTS
IS
Supply current
No Load, Normal Operation (PD
Pin = HI or open for WSON-8
package)
No Load, Shutdown (PD Pin =LO
for WSON-8 package)
15.5
-40°C ≤ TJ ≤ +125°C
18.2
1.1
-40°C ≤ TJ ≤ +125°C
16.7
2.0
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mA
1.85
7
LMH6629
SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
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6.6 Electrical Characteristics 3.3V
The following specifications apply for single supply with VS = 3.3 V, RL = 100 Ω terminated to 1.65 V, gain = 10V/V, VO = 1
VPP, VCM = VS/2, COMP Pin = HI (WSON-8 package), unless otherwise noted. (1)
PARAMETER
TEST CONDITIONS
TA = 25°C
MIN (2)
TYP (3)
MAX (2)
UNIT
DYNAMIC PERFORMANCE
VO = 200 mVPP, WSON-8 package
SSBW
LSBW
950
Large signal −3dB VO = 1VPP
bandwidth
COMP Pin = LO, AV= 4, VO = 1VPP
540
0.1 dB Bandwidth
Peaking
SR
Slew rate
tr/ tf
Rise/fall time
Ts
820
Small signal −3dB VO = 200 mVPP, SOT-23-5 package
bandwidth
COMP Pin = LO, AV= 4,
VO = 200 mVPP
730
320
AV= 10, VO = 200 mVPP,
WSON-8 package
330
AV= 10, VO = 200 mVPP,
SOT-23-5 package
190
COMP Pin = LO, AV= 4,
VO = 200 mVPP
85
VO = 200 mVPP, WSON-8 package
VO = 200 mVPP, SOT-23-5 package
AV= 10, 1.3V step
MHz
0
1.8
1100
COMP Pin = LO, AV= 4, 1.3V step
500
AV= 10, 1V step, 10% to 90%,
WSON-8 package
0.7
AV= 10, 1V step, 10% to 90%,
SOT-23-5 package
0.55
AV= 4, COMP Pin = LO, 1V step, 10% to 90% (Slew Rate
Limited)
1.3
Settling time
AV= 10, 1V step, ±0.1%
70
Overload
recovery
VIN = 1VPP
MHz
MHz
dB
V/µs
ns
2
NOISE and DISTORTION
HD2
HD3
2nd Order
distortion
3rd Order
distortion
OIP3
Two-tone 3rd
order intercept
point
en
Noise voltage
in
Noise current
NF
Noise figure
(1)
(2)
(3)
8
fc = 1MHz, VO = 1VPP
-82
COMP Pin = LO, AV= 4,
fc = 1MHz, VO = 1VPP
-88
fc = 10 MHz, VO = 1VPP
-67
COMP Pin = LO, fc = 10 MHz,
AV= 4V, VO = 1VPP
-74
fc = 1MHz, VO = 1VPP
-94
COMP Pin = LO, AV= 4,
fc = 1MHz, VO = 1VPP
-112
fc = 10 MHz, VO = 1VPP
-79
COMP pin = LO, fc = 10 MHz,
VO = 1VPP
-96
fc = 25 MHz, VO = 1VPP composite
30
fc = 75 MHz, VO = 1VPP composite
26
Input referred, f > 1MHz
RS = RT = 50 Ω
dBc
dBc
dBm
0.69
nV/√Hz
2.6
pA/√Hz
8.0
dB
Electrical table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very
limited self-heating of the device such that TJ = TA. No guarantee of parametric performance is indicated in the electrical tables under
conditions of internal self-heating where TJ > TA.
All limits are ensured by testing or statistical analysis.
Typical numbers are the most likely parametric norm.
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Product Folder Links: LMH6629
LMH6629
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Electrical Characteristics 3.3V (continued)
The following specifications apply for single supply with VS = 3.3 V, RL = 100 Ω terminated to 1.65 V, gain = 10V/V, VO = 1
VPP, VCM = VS/2, COMP Pin = HI (WSON-8 package), unless otherwise noted.(1)
PARAMETER
TEST CONDITIONS
TA = 25°C
MIN (2)
TYP (3)
UNIT
MAX (2)
ANALOG I/O
CMVR
Input voltage
range
VO
Output voltage
range
CMRR > 70 dB, WSON-8 package
-40°C ≤ TJ ≤ +125°C
IOUT
Linear output
current
VOS
Input offset
voltage
TcVOS
Input offset
voltage
temperature drift
See
(5)
IBI
Input bias current
See
(6)
Input offset
current
TCIOS
Input offset
voltage
temperature drift
CCM
CDIFF
RCM
Input capacitance
Input resistance
-40°C ≤ TJ ≤ +125°C
0.79 to 2.50
0.95
0.76
No load
2.1
-0.30 to 2.1
0.90
RL = 100 Ω to VS/2
IOS
-0.30
CMRR > 70 dB, SOT-23-5 package
0.70 to 2.60
2.5
2.4
230
±150
-40°C ≤ TJ ≤ +125°C
mA
±680
µV
±700
±1
µV/°C
-15
-23
±0.13
±1.8
-40°C ≤ TJ ≤ +125°C
-35
-40°C ≤ TJ ≤ +125°C
See
V
2.3
0.80
VO = 1.65 V (4)
2.4
µA
±3.0
(5)
±3.2
Common Mode
nA/°C
1.7
Differential Mode (7)
4
Common Mode
1
pF
MΩ
MISCELLANEOUS PARAMETERS
CMRR
Common mode
rejection ratio
PSRR
Power supply
rejection ratio
AVOL
Open loop gain
VCM from 0 V to 2.0 V, WSON-8
package
84
-40°C ≤ TJ ≤ +125°C
VCM from 0 V to 2.0 V,
SOT-23-5 package
87
82
-40°C ≤ TJ ≤ +125°C
WSON-8 package
(5)
(6)
(7)
84
dB
79
78
-40°C ≤ TJ ≤ +125°C
SOT-23-5 package
(4)
87
81
79
73
79
The maximum continuous output current (IOUT) is determined by device power dissipation limitations. Continuous short circuit operation
at elevated ambient temperature can result in exceeding the maximum allowed junction temperature of 150°C
Drift determined by dividing the change in parameter at temperature extremes by the total temperature change.
Negative input current implies current flowing out of the device.
Simulation results.
Submit Documentation Feedback
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SNOSB18I – APRIL 2010 – REVISED DECEMBER 2014
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Electrical Characteristics 3.3V (continued)
The following specifications apply for single supply with VS = 3.3 V, RL = 100 Ω terminated to 1.65 V, gain = 10V/V, VO = 1
VPP, VCM = VS/2, COMP Pin = HI (WSON-8 package), unless otherwise noted.(1)
PARAMETER
TEST CONDITIONS
TA = 25°C
MIN (2)
TYP (3)
MAX (2)
UNIT
DIGITAL INPUTS/TIMING
VIL
Logic low-voltage
threshold
0.8
PD and COMP pins, WSON-8 package
VIH
Logic high-voltage
threshold
IIL
Logic low-bias
current
PD and COMP pins = 0.8 V,
WSON-8 package (6)
IIH
Logic high-bias
current
PD and COMP pins = 2.0 V,
WSON-8 package (6)
Ten
Enable time
Tdis
Disable time
V
2.0
-17
-40°C ≤ TJ ≤ +125°C
-16
-40°C ≤ TJ ≤ +125°C
-23
-14
-28
-32
-22
-13
-27
-31
75
WSON-8 package
µA
ns
80
POWER REQUIREMENTS
IS
Supply current
No Load, Normal Operation (PD
Pin = HI or open for WSON-8
package)
13.7
-40°C ≤ TJ ≤ +125°C
No Load, Shutdown (PD Pin = LO
for WSON-8 package)
-40°C ≤ TJ ≤ +125°C
10
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14.9
16.0
0.89
mA
1.4
1.5
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6.7 Typical Performance Characteristics
1
1
0
0
Normalized Gain (dB)
Normalized Gain (dB)
Unless otherwise specified, VS = ±2.5V, Rf = 240 Ω, RL = 100 Ω, VO = 2 VPP, COMP pin = HI, AV = +10 V/V, WSON-8 and
SOT-23-5 packages (unless specifically noted).
Av = 100 V/V
-1
Av = -60 V/V
Av = -40 V/V
-2
Av = -20 V/V
-3
Av = -4 V/V
(COMP Pin = LO)
-4
Av = -60 V/V
Av = -40 V/V
-2
Av = -20 V/V
-3
Av = -4 V/V
(COMP Pin = LO)
-4
-5
100k
1M
10M
f (Hz)
100M
-5
100k
1G
Vo = 2 Vpp
0
0
Normalized Gain (dB)
1
Av = 200 V/V
Av = 100 V/V
-2
Av = 30 V/V
-3
Av = 4 V/V
(COMP Pin = LO)
Av = 10 V/V
-4
-5
100k
10M
f (Hz)
1M
10M
f (Hz)
Vs = ±1.5V
-1 Av = 200 V/V
-2 Av = 100 V/V
Av = 30 V/V
-3
Av = 10 V/V
-5
100k
1G
Vo = 2 Vpp
1M
10M
f (Hz)
Vo = 1 Vpp
Figure 3. Non-Inverting Frequency Response
1G
Av = 4 V/V
(COMP Pin = LO)
-4
100M
100M
Figure 2. Inverting Frequency Response
1
-1
1M
Vo = 1 Vpp
Figure 1. Inverting Frequency Response
Normalized Gain (dB)
Av = 100 V/V
-1
100M
1G
Vs = ±1.5 V
Figure 4. Non-Inverting Frequency Response
3
3
2
2
0
Normalized Gain (dB)
Normalized Gain (dB)
1
-1
-2
Av = 100 V/V
-3
-4
Av = 30 V/V
-5
-6
-7
Av = 4 V/V
(COMP Pin = LO)
0
-1
Av = 115 V/V
-2
Av = 80 V/V
-3
Av = 35 V/V
-8
-4
-9
-10
100k
1
Av = 10 V/V
Av = 10 V/V
1M
10M
100M
1G
-5
100k
10G
1M
10M
100M
1G
10G
f (Hz)
f (Hz)
Vo = 0.2 Vpp
Vo = 0.2 Vpp
Figure 5. Non-Inverting Frequency Response,
WSON-8 Package
Figure 6. Non-Inverting Frequency Response,
SOT-23-5 Package
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Typical Performance Characteristics (continued)
1
2
0
1
Vo = 0.2 Vpp
Normalized Gain (dB)
Normalized Gain (dB)
Unless otherwise specified, VS = ±2.5V, Rf = 240 Ω, RL = 100 Ω, VO = 2 VPP, COMP pin = HI, AV = +10 V/V, WSON-8 and
SOT-23-5 packages (unless specifically noted).
-1
Vo = 0.2 Vpp
-2
Vo = 2 Vpp
-3
-4
0
-1
-2
Vo = 2 Vpp
-3
-4
-5
100k
1M
10M
100M
-5
100k
1G
1M
f (Hz)
Figure 7. Non-Inverting Frequency Response
with Varying VO, WSON-8 Package
2
1
Normalized Gain (dB)
Normalized Gain (dB)
0
-1
-2
Vo = 0.2 Vpp
-3
Vo = 1 Vpp
-4
Vo = 0.2 Vpp
0
-1
-2
Vo = 1 Vpp
-3
-4
1M
10M
100M
-5
100k
1G
1M
Av = 10 V/V
Av=10 V/V
Vs = +/-1.5 V
100M
1G
10G
Vs= +/-1.5 V
Figure 10. Non-Inverting Frequency Response
with Varying VO, SOT-23-5 Package
4
4
2
2
0
0
Normalized Gain (dB)
Normalized Gain (dB)
Figure 9. Non-Inverting Frequency Response
with Varying VO, WSON-8 Package
-2
Vo = 80 mVpp
-4
-6
Vo = 0.8 Vpp
-8
-10
-2
Vo = 80 mVpp
-4
-6
Vo = 0.8 Vpp
-8
-10
1
10
f (MHz)
100
-12
0.1
1,000
COMP Pin = LO
Av = 4 V/V
Figure 11. Non-Inverting Frequency Response
with Varying VO, WSON-8 Package
12
10M
f (Hz)
f (Hz)
Av = 4 V/V
10G
Figure 8. Non-Inverting Frequency Response
with Varying VO, SOT-23-5 Package
1
-12
0.1
1G
Av=10 V/V
Av = 10 V/V
-5
100k
10M
100M
f (Hz)
1
10
f (MHz)
100
Vs = ±1.5 V
1,000
COMP Pin = LO
Figure 12. Non-Inverting Frequency Response
with Varying VO, WSON-8 Package
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Typical Performance Characteristics (continued)
Unless otherwise specified, VS = ±2.5V, Rf = 240 Ω, RL = 100 Ω, VO = 2 VPP, COMP pin = HI, AV = +10 V/V, WSON-8 and
SOT-23-5 packages (unless specifically noted).
2
2
1
0
Normalized Gain (dB)
Normalized Gain (dB)
0
-1
33 pF, Riso = 17 :
-2
-3
4.7 pF, Riso = 72 :
-4
-5
-1
-2
-5
-6
-7
-8
10
1,000
RL = 100 Ω || CL
RISO as noted (measured @ CL)
100
f (MHz)
1,000
RL = 100 Ω || CL, AV = 4 V/V
COMP Pin = LO
RISO as noted (measured @ CL)
Figure 13. Frequency Response with Cap. Loading
2
Figure 14. Frequency Response Cap. Loading,
WSON-8 Package
2
1 k: 1.5 k:
0
1.5 kÖ
1 kÖ
Normalized Gain (dB)
0
-2
240 :
750 :
-4
-2
240 Ö
750 Ö
-4
-6
-6
-8
100k
10 pF, Riso = 40 :
-4
-7
100
f (MHz)
4.7 pF, Riso = 72 :
-3
-6
-8
10
Normalized Gain (dB)
33 pF, Riso = 17 :
1
10 pF, Riso = 40 :
1M
10M
100M
-8
100k
1G
1M
10M
100M
1G
f (Hz)
f (Hz)
Figure 15. Frequency Response vs. Rf, WSON-8 Package
Figure 16. Frequency Response vs. Rf, SOT-23-5 Package
4
2
1.5 kÖ
1.5 k:
750 Ö
2
-2
Normalized Gain (dB)
Normalized Gain (dB)
0
511 :
240 :
750 :
-4
-6
-8
100k
0
-2
240 Ö
-6
1M
10M
100M
-8
100k
1G
1M
10M
100M
1G
f (Hz)
f (Hz)
VS = ±1.5 V
1 kÖ
-4
VS = ± 1.5 V
VO = 1 VPP
Figure 17. Frequency Response vs. Rf, WSON-8 Package
VO = 1 VPP
Figure 18. Frequency Response vs. Rf, SOT-23-5 Package
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Typical Performance Characteristics (continued)
Unless otherwise specified, VS = ±2.5V, Rf = 240 Ω, RL = 100 Ω, VO = 2 VPP, COMP pin = HI, AV = +10 V/V, WSON-8 and
SOT-23-5 packages (unless specifically noted).
-60
-50
HD2
HD2
-60
-70
-70
HD (dBc)
HD (dBc)
-80
HD3
-90
-80
HD3
-90
-100
-110
0.0
-100
0.5
1.0
1.5
2.0
Vo (Vpp)
2.5
-110
0.0
3.0
0.5
1.0
1.5
2.0
2.5
3.0
Vo (Vpp)
20 MHz
20 MHz
Figure 19. Distortion vs. Swing, WSON-8 Package
Figure 20. Distortion vs. Swing, SOT-23-5 Package
-85
-80
-85
-90
HD (dBc)
HD (dBc)
HD2
HD3
HD2
-90
-95
HD3
-95
-100
-100
1.0
1.5
2.0
2.5
-105
1.0
3.0
1.5
Vo (Vpp)
2.0
2.5
3.0
Vo (Vpp)
1 MHz
1 MHz
Figure 21. Distortion vs. Swing, WSON-8 Package
Figure 22. Distortion vs. Swing, SOT-23-5 Package
-65
-60
-70
-65
HD2
-70
HD2
-80
HD (dBc)
HD (dBc)
-75
HD3
HD3
-75
-80
-85
-85
-90
-95
0
-90
20
40
60
80
-95
0
100
Av (V/V)
1 MHz
40
60
80
100
Av (V/V)
Vout = 2 Vpp
1 MHz
Figure 23. Distortion vs. Gain, WSON-8 Package
14
20
Vout = 2 Vpp
Figure 24. Distortion vs. Gain, SOT-23-5 Package
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Typical Performance Characteristics (continued)
Unless otherwise specified, VS = ±2.5V, Rf = 240 Ω, RL = 100 Ω, VO = 2 VPP, COMP pin = HI, AV = +10 V/V, WSON-8 and
SOT-23-5 packages (unless specifically noted).
-40
-40
-50
-50
HD2
-60
HD2
HD (dBc)
HD (dBc)
-60
-70
HD3
-70
HD3
-80
-80
-90
-90
-100
0
10
20
30
f (MHz)
40
-100
0
50
10
20
30
40
50
f (MHz)
Vo = 2 Vpp
Vo = 2 Vpp
Figure 25. Distortion vs. Frequency, WSON-8 Package
Figure 26. Distortion vs. Frequency, SOT-23-5 Package
-30
1.3
1.2
-40
-50
en (nV/Hz)
IMD3 (dBc)
1.1
75 MHz
25 MHz
-60
1.0
0.9
0.8
-70
0.7
-80
0.5
1.0
1.5
2.0
2.5
3.0
0.6
0.1
3.5
Vo (Vpp Composite)
Figure 27. 3rd Order Intermodulation Distortion
vs. Output Voltage
1
10
100
f (kHz)
1,000
10,000
Figure 28. Input Noise Voltage vs. Frequency
12
in (pA/Hz)
10
8
6
4
2
0.1
1
10
100
f (kHz)
1,000
10,000
Figure 29. Input Noise Current vs. Frequency
Figure 30. PSRR vs. Frequency
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Typical Performance Characteristics (continued)
Unless otherwise specified, VS = ±2.5V, Rf = 240 Ω, RL = 100 Ω, VO = 2 VPP, COMP pin = HI, AV = +10 V/V, WSON-8 and
SOT-23-5 packages (unless specifically noted).
4.0
80
160
Gain, COMP HI
3.5
120
20
40
Vo from V+ (V)
80
125°C
Phase (°)
Gain (dB)
40 Gain, COMP LO
2.5
2.0
1.5
1M
10M
-40°C
1.0
Phase, COMP HI
0
100k
25°C
3.0
Phase, COMP LO
60
0.5
0
1G
100M
0.0
100
150
200
f (Hz)
250
300
350
400
Isource (mA)
Figure 31. Open Loop Gain/Phase Response
Figure 32. Output Source Current, WSON-8 Package
4.5
3.5
4.0
3.0
-40°C
3.5
3.0
Vo from V (V)
125°C
+
Vo from V- (V)
2.5
2.5
2.0
25°C
1.5
-40°C
2.0
25°C
1.5
1.0
1.0
0.5
0.5
0.0
100
125°C
150
200
250
300
350
0.0
100
400
Isink (mA)
150
200
250
300
350
400
Isource (mA)
Figure 33. Output Sink Current WSON-8 Package
Figure 34. Output Source Current, SOT-23-5 Package
3.5
3.0
-40°C
Vo (0.5V/DIV)
-
Vo from V (V)
2.5
2.0
25°C
1.5
1.0
0.5
125°C
0.0
100
150
200
250
300
350
400
TIME (10 ns/DIV)
Isink (mA)
Figure 35. Output Sink Current, SOT-23-5 Package
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Figure 36. Large Signal Step Response
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Typical Performance Characteristics (continued)
Vo (0.2V/Div)
Vo (0.5 V/DIV)
Unless otherwise specified, VS = ±2.5V, Rf = 240 Ω, RL = 100 Ω, VO = 2 VPP, COMP pin = HI, AV = +10 V/V, WSON-8 and
SOT-23-5 packages (unless specifically noted).
Time (4 ns/Div)
Time (10 ns/DIV)
Figure 37. Large Signal Step Response
Figure 38. Large Signal Step Response
Vo (0.05V/Div)
COMP Pin = LO
Vo (0.2 V/DIV)
AV= 4V/V
Time (4 ns/DIV)
Time (2 ns/Div)
Vs = 3.3 V
Figure 39. Large Signal Step Response
Figure 40. Small Signal Step Response, WSON-8 Package
1.0
3
2
0.5
0.0
1
0
PD\ (V)
Vo (V)
Vo (0.05 V/DIV)
PD\
Vo
-1
-0.5
-2
-1.0
Time (2 ns/DIV)
-3
Time (50 ns/DIV)
Vs = 3.3 V
Figure 41. Small Signal Step Response, WSON-8 Package
Figure 42. Turn-On Waveform, WSON-8 Package
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Typical Performance Characteristics (continued)
Unless otherwise specified, VS = ±2.5V, Rf = 240 Ω, RL = 100 Ω, VO = 2 VPP, COMP pin = HI, AV = +10 V/V, WSON-8 and
SOT-23-5 packages (unless specifically noted).
1.0
3
18.0
Vo
2
17.0
0.5
0
Is (mA)
0.0
PD\ (V)
Vo (V)
1
125°C
16.0
25°C
15.0
-40°C
-1
-0.5
PD\
14.0
-2
-1.0
13.0
2.5
-3
3.0
3.5
Time (50 ns/DIV)
Figure 43. Turn-Off Waveform, WSON-8 Package
4.5
5.0
5.5
Figure 44. Supply Current vs. Supply Voltage
25
400
350
125°C
300
20
85°C
IB (PA)
250
VOS ( V)
4.0
Vs (V)
200
150
15
25°C
25°C
100
10
50
0
-50
2.5
-40°C
-40°C
3.0
3.5
4.0
4.5
5.0
5
2.5
5.5
VS (V)
3.0
3.5
4.0
4.5
5.0
5.5
Vs (V)
Figure 45. Offset Voltage vs. Supply Voltage
(Typical Unit)
Figure 46. Input Bias Current vs. Supply Voltage
(Typical Unit)
1.0
125°C
IOS (PA)
0.5
25°C
0.0
-0.5
-40°C
-1.0
-1.5
2.5
3.0
3.5
4.0
4.5
5.0
5.5
Vs (V)
Figure 47. Input Offset Current vs. Supply Voltage (Typical Unit)
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7 Detailed Description
7.1 Overview
The LMH6629 is a high gain bandwidth, ultra low-noise voltage feedback operational amplifier. The excellent
noise and bandwidth enables applications such as medical diagnostic ultrasound, magnetic tape and disk
storage and fiberoptics to achieve maximum high frequency signal-to-noise ratios. The following discussion will
enable the proper selection of external components to achieve optimum system performance.
7.2 Functional Block Diagram
The LMH6629 (WSON-8 package only) has some additional features to allow maximum flexibility. As shown in
Figure 48, there are provisions for low-power shutdown and two internal compensation settings, which are
discussed in more detail in Compensation. Also provided is a feedback (FB) pin which allows the placement of
the feedback resistor directly adjacent to the inverting input (IN-) pin. This pin simplifies printed circuit board
layout and minimizes the possibility of unwanted interaction between the feedback path and other circuit
elements.
PD
FB
-IN
+IN
1
8
2
7
3
6
4
+
5
+
V
OUT
COMP
-
V
Figure 48. 8-Pin WSON Pinout Diagram
The WSON-8 package requires the bottom-side Die Attach Paddle (DAP) to be soldered to the circuit board for
proper thermal dissipation and to get the thermal resistance number specified. The DAP is tied to the V- potential
within the LMH6629 package. Thus, the circuit board copper area devoted to DAP heatsinking connection should
be at the V- potential as well. Please refer to the package drawing for the recommended land pattern and
recommended DAP connection dimensions.
PD
1
FB
2
+
8
V
7
OUT
6
COMP
5
V
DAP
IN-
3
IN+
4
V-
-
Figure 49. WSON–8 DAP(Top View)
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7.3 Feature Description
7.3.1 WSON-8 Control Pins and SOT-23-5 Comparison
The LMH6629 WSON-8 package has two digital control pins; PD and COMP pins. The PD pin, used for power
down, floats high (device on) when not driven. When the PD pin is pulled low, the amplifier is disabled and the
amplifier output stage goes into a high impedance state so the feedback and gain set resistors determine the
output impedance of the circuit. The other control pin, the COMP pin, allows control of the internal compensation
and defaults to the lower gain mode or logic 0.
The SOT-23-5 package has the following differences relative to the WSON-8 package:
1. No power down (shutdown) capability.
2. No COMP pin to set the minimum stable gain. SOT-23–5 package minimum stable gain is internally fixed to
be 10V/V.
3. No feedback (FB) pin.
From a performance point of view, the WSON-8 and the SOT-23-5 packages perform very similarly except in the
following areas:
1. SSBW, Peaking, and 0.1 dB Bandwidth: These differences are highlighted in the Typical Performance
Characteristics and the Electrical Characteristics 5V tables. Most notable differences are with small signal
(0.2 Vpp) and close to the minimum stable gain of 10V/V.
2. Distortion: It is possible to get slightly different distortion performance. The board layout and decoupling
capacitor return current routing strongly influence distortion performance.
3. Output Current: In heavy current applications, there will be differences between these package types
because of the difference in their respective Thermal Resistances (RθJA).
7.3.2 Compensation
The LMH6629 has two compensation settings that can be controlled by the COMP pin (WSON-8 package only).
The default setting is set through an internal pulldown resistor and places the COMP pin at the logic 0 state. In
this configuration the on-chip compensation is set to the maximum and bandwidth is reduced to enable stability
at gains as low as 4V/V.
When this pin is driven to the logic 1 state, the internal compensation is decreased to allow higher bandwidth at
higher gains. In this state, the minimum stable gain is 10V/V. Due to the reduced compensation, slew rate and
large signal bandwidth are significantly enhanced for the higher gains.
NOTE
As mentioned earlier, the SOT-23-5 package does not offer the two compensation settings
that the WSON-8 offers. The SOT-23-5 is internally set for a minimum gain of 10 V/V.
It is possible to externally compensate the LMH6629 for any of the following reasons, as shown in Figure 50.
• To operate the SOT-23-5 package (which does not offer the COMP pin) at closed loop gains < 10V/V.
• To operate the WSON-8 package at gains below the minimum stable gain of 4V /V when the COMP pin is
LO. NOTE: In this case, Figure 50 “Constraint 1” may be changed to ≥ 4 V/V instead of ≥ 10 V/V.
• To operate either package at low gain and need maximum slew rate (COMP pin HI).
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Feature Description (continued)
Rg
Rs
VIN
Rf
VIN
Rc
Rc
Cc
Cc
RT
Rf
R1
Rs
Rp
a) Non-inverting
COMP pin = HI
b) Inverting
R p = Rs || R T
R EQ = R f || R g
R g = R s + R1
Constraint 1: (1 +
Costraint 2:
R + REQ
Rf
)(1 + p
) t 10 V/V
Rg
Rc
1
d 90 MHz
2SCcRc
Figure 50. External Compensation
This circuit operates by increasing the Noise Gain (NG) beyond the minimum stable gain of the LMH6629 while
maintaining a positive loop gain phase angle at 0 dB. There are two constraints shown in Figure 50: “Constraint
1” ensures that NG has increased to at least 10 V/V when the loop gain approaches 0dB, and “Constraint 2”
places an upper limit on the feedback phase lead network frequency to make sure it is fully effective in the
frequency range when loop gain approaches 0dB. These two constraints allow one to estimate the “starting
value” for Rc and Cc which may need to be fine tuned for proper response.
Here is an example worked out for more clarification:
• Assume that the objective is to use the SOT-23-5 version of the LMH6629 for a closed loop gain of +3.7 V/V
using the technique shown in Figure 50.
• Selecting Rf = 249 Ω → Rg = 91 Ω → REQ= 66.6 Ω.
• For 50-Ω source termination (Rs= 50 Ω), select RT= 50 Ω → Rp = 25 Ω.
• Using “Constraint 1” (= 10V/V) allows one to compute Rc ≊ 56 Ω. Using “Constraint 2” (= 90 MHz) defines the
appropriate value of Cc ≊ 33 pF.
• The frequency response plot shown in Figure 51 is the measured response with Rc and Cc values computed
above and shows a -3 dB response of about 1 GHz.
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Feature Description (continued)
4.0
Normalized Gain (dB)
2.0
0.0
RA
RB
-2.0
Cf ¶
Rg
Rf
-4.0
-6.0
1
10
100
1,000
10,000
f (MHz)
Cf = 1.5 pF
RA = 33 Ω
RB = 91 Ω
Figure 51. SOT-23-5 Package Low Closed Loop Gain Operation
with External Compensation
For the Figure 51 measured results, a compensation capacitor (Cf') was used across Rf to compensate for the
summing node net capacitance due to the board and the SOT-23–5 LMH6629. The RA and RB combination
reduces the effective capacitance of Cf‘ by the ratio of 1+RB / RA, with the constraint that RB 1pF) to be used. The WSON-8 package does not need this
compensation across Rf due to its lower parasitics.
With the COMP pin HI (WSON-8 package only) or with the SOT-23–5 package, this circuit achieves high slew
rate and takes advantage of the LMH6629’s superior low-noise characteristics without sacrificing stability, while
enabling lower gain applications. It should be noted that the Rc, Cc combination does lower the input impedance
and increases noise gain at higher frequencies. With these values, the input impedance reduces by 3 dB at 490
MHz. The Noise Gain transfer function “zero” is given by Equation 1 and it has a 3-dB increase at 32.8 MHz with
these values:
External Compensation Noise Gain Increase:
1
Noise Gain " zero" #
2S (RC + R p + REQ ) CC
22
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Feature Description (continued)
7.3.3 Cancellation of Offset Errors Due to Input Bias Currents
The LMH6629 offers exceptional offset voltage accuracy. In order to preserve the low offset voltage errors, care
must be taken to avoid voltage errors due to input bias currents. This is important in both inverting and noninverting applications.
The non-inverting circuit is used here as an example. To cancel the bias current errors of the non-inverting
configuration, the parallel combination of the gain setting (Rg) and feedback (Rf) resistors should equal the
equivalent source resistance (Rseq) as defined in Figure 52. Combining this constraint with the non-inverting gain
equation also seen in Figure 52 allows both Rf and Rg to be determined explicitly from Equation 2:
Rf = AVRseq and Rg = Rf/(AV-1)
(2)
Figure 52. Non-Inverting Amplifier Configuration
When driven from a 0-Ω source, such as the output of an op amp, the non-inverting input of the LMH6629 should
be isolated with at least a 25-Ω series resistor.
As seen in Figure 53, bias current cancellation is accomplished for the inverting configuration by placing a
resistor (Rb) on the non-inverting input equal in value to the resistance seen by the inverting input (Rf || (Rg+Rs)).
Rb should to be no less than 25 Ω for optimum LMH6629 performance. A shunt capacitor (not shown) can
minimize the additional noise of Rb.
Figure 53. Inverting Amplifier Configuration
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Feature Description (continued)
7.3.4 Total Input Noise vs. Source Resistance
To determine maximum signal-to-noise ratios from the LMH6629, an understanding of the interaction between
the amplifier’s intrinsic noise sources and the noise arising from its external resistors is necessary. Figure 54
describes the noise model for the non-inverting amplifier configuration showing all noise sources. In addition to
the intrinsic input voltage noise (en) and current noise (in = in+ = in−) source, there is also thermal voltage noise (et
= √(4KTR)) associated with each of the external resistors.
Figure 54. Non-Inverting Amplifier Noise Model
Equation 3 provides the general form for total equivalent input voltage noise density (eni).
General Noise Equation:
(3)
Equation 4 is a simplification of Equation 3 that assumes Rf || Rg = Rseq for bias current cancellation:
Equation 4: Noise Equation with Rf || Rg = Rseq
(4)
Figure 55 schematically shows eni alongside VIN (the portion of VS source which reaches the non-inverting input
of Figure 52) and external components affecting gain (Av= 1 + Rf / Rg), all connected to an ideal noiseless
amplifier.
Rf
(Noiseless)
Rseq = RS || RT
Rg
(Noiseless)
VS
RS
(Noiseless)
eni
Noiseless Op
Amp
RT
(Noiseless)
Set RSeq = Rf || Rg for bias current offset
cancellation
Figure 55. Non-Inverting Amplifier Equivalent Noise Source Schematic
24
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Feature Description (continued)
Figure 56 illustrates the equivalent noise model using this assumption. Figure 57 is a plot of eni against
equivalent source resistance (Rseq) with all of the contributing voltage noise source of Equation 4. This plot gives
the expected eni for a given (Rseq) which assumes Rf||Rg = Rseq for bias current cancellation. The total equivalent
output voltage noise (eno) is eni*AV.
Figure 56. Noise Model with Rf||Rg = Rseq
As seen in Figure 57, eni is dominated by the intrinsic voltage noise (en) of the amplifier for equivalent source
resistances below 15 Ω. Between 15 Ω and 2.5 kΩ, eni is dominated by the thermal noise (et = √(4kT(2Rseq)) of
the equivalent source resistance Rseq. Incidentally, this is the range of Rseq values where the LMH6629 has the
best (lowest) Noise Figure (NF) for the case where Rseq = Rf || Rg.
Above 2.5 kΩ, eni is dominated by the amplifier’s current noise (in = √2 * inRseq). When Rseq = 190 Ω (that is, Rseq
= en/√2 * in), the contribution from voltage noise and current noise of LMH6629 is equal. For example, configured
with a gain of +10V/V giving a −3dB of 825 MHz and driven from Rseq = Rf || Rg = 20 Ω (eni = 1.07 nV√Hz from
Figure 57), the LMH6629 produces a total equivalent output noise voltage (eni * 10 V/V * √(1.57 * 825 MHz)) of
385 μVrms.
Voltage Noise Density (nV/ Hz)
1000
eni
100
10
et
in
1
en
0.1
1
10
100
1k
10k
100k
RSEQ (:)
RSEQ = RF || RG
Figure 57. Voltage Noise Density vs. Source Resistance
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Feature Description (continued)
If bias current cancellation is not a requirement, then Rf || Rg does not need to equal Rseq. In this case, according
to Equation 3, Rf || Rg should be as low as possible to minimize noise. Results similar to Equation 3 are obtained
for the inverting configuration of Figure 53 if Rseq is replaced by Rb and Rg is replaced by Rg + Rs. With these
substitutions, Equation 3 will yield an eni referred to the non-inverting input. Referring eni to the inverting input is
easily accomplished by multiplying eni by the ratio of non-inverting to inverting gains (1+Rg/ Rf).
7.3.5 Noise Figure
Noise Figure (NF) is a measure of the noise degradation caused by an amplifier.
General Noise Figure Equation:
(5)
Looking at the two parts of the NF expression (inside the log function) yields:
• Si/ So→ Inverse of the power gain provided by the amplifier
• No/ Ni→ Total output noise power, including the contribution of RS, divided by the noise power at the input
due to RS
• To simplify this, consider Na as the noise power added by the amplifier (reflected to its input port):
• Si/ So→ 1/ G
• No/ Ni→ G * (Ni+Na) / Ni (where G*(Ni +Na ) = No)
Substituting these two expressions into the NF expression:
NF = 10 log
§ G(Ni + Na) ·
¨ Ni ¸
G©
¹
1
§
©
= 10 log ¨1 +
Na·
Ni ¸
¹
(6)
The noise figure expression has simplified to depend only on the ratio of the noise power added by the amplifier
at its input (considering the source resistor to be in place but noiseless in getting Na) to the noise power
delivered by the source resistor (considering all amplifier elements to be in place but noiseless in getting Ni).
For a given amplifier with a desired closed loop gain, to minimize noise figure:
• Minimize Rf || Rg
• Choose the Optimum RS (ROPT)
ROPT is the point at which the NF curve reaches a minimum and is approximated by:
ROPT ≈ en/ in
26
(7)
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Feature Description (continued)
Figure 58 is a plot of NF vs RS with the circuit of Figure 52 (Rf = 240 Ω, AV = +10V/V). The NF curves for both
Unterminated (RT = open) and Terminated systems (RT = RS) are shown. Table 1 indicates NF for various source
resistances including RS = ROPT.
18
16
Terminated
Noise Figure (dB)
14
12
10
8
6
4
2
Unterminated
0
10
100
1k
10k
100k
RS (:)
f > 1 MHz
Figure 58. Noise Figure vs. Source Resistance
Table 1. Noise Figure for Various Rs
RS (Ω)
NF (TERMINATED) (dB)
NF (UNTERMINATED) (dB)
50
8
3.2
ROPT
4.1
(ROPT = 750 Ω)
1.1
(ROPT = 350 Ω)
7.3.6 Single-Supply Operation
The LMH6629 can be operated with single power supply as shown in Figure 59. Both the input and output are
capacitively coupled to set the DC operating point.
Figure 59. Single-Supply Operation
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7.3.7 Low-Noise Transimpedance Amplifier
Figure 60 implements a high-speed, single-supply, low-noise Transimpedance amplifier commonly used with
photo-diodes. The transimpedance gain is set by RF.
CF
0.6 pF
(1.2 pF (2) series)
+5VDC
D1
RF
1.2k
CD = 10 pF
ID
+5VDC
LMH
6629
+5VDC
RL
500
R2
2k
C1
0.1 PF
R1
3k
Vout = 3VDC - 1200 x ID
Figure 60. 200 MHz Transimpedance Amplifier Configuration
Figure 61 shows the Noise Gain (NG) and transfer function (I-V Gain). As with most Transimpedance amplifiers,
it is required to compensate for the additional phase lag (Noise Gain zero at fZ) created by the total input
capacitance ( CD (diode capacitance) + CCM (LMH6629 CM input capacitance) + CDIFF (LMH6629 DIFF input
capacitance) ) looking into RF. This is accomplished by placing CF across RF to create enough phase lead (Noise
Gain pole at fP) to stabilize the loop.
OP AMP OPEN
LOOP GAIN
GAIN (dB)
I-V GAIN (:)
NOISE GAIN (NG)
1 + sRF (CIN + CF)
1 + sRFCF
1+
CIN
CF
0 dB
FREQUENCY
fz #
1
2SRFCIN
fP =
1
GBWP
2SRFCF
Figure 61. Transimpedance Amplifier Noise Gain and Transfer Function
28
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The optimum value of CF is given by Equation 8 resulting in the I-V -3dB bandwidth shown in Equation 9, or
around 200 MHz in this case (assuming GBWP= 4GHz with COMP pin = HI for WSON-8 package). This CF
value is a “starting point” and CF needs to be tuned for the particular application as it is often less than 1 pF and
thus is easily affected by board parasitics. For maximum speed, the LMH6629 COMP pin should be HI (or use
the SOT-23 package).
Optimum CF Value:
CF =
CIN
2S(GBWP)RF
(8)
Resulting -3dB Bandwidth
f - 3 dB #
GBWP
2 S R F C IN
(9)
Equation 10 provides the total input current noise density (ini) equation for the basic Transimpedance
configuration and is plotted against feedback resistance (RF) showing all contributing noise sources in Figure 62.
The plot indicates the expected total equivalent input current noise density (ini) for a given feedback resistance
(RF). This is depicted in the schematic of Figure 63 where total equivalent current noise density (ini) is shown at
the input of a noiseless amplifier and noiseless feedback resistor (RF). The total equivalent output voltage noise
density (eno) is ini*RF.
Noise Equation for Transimpedance Amplifier:
(10)
Current Noise Density (pA/ Hz)
16
14
ini
12
10
8
it
en/RF
6
in
4
2
0
100
1k
10k
Rf (:)
Figure 62. Current Noise Density vs. Feedback Resistance
+5VDC
D1
CD = 10 pF
RF
(Noiseless)
ID
ini
Noiseless Op Amp
Figure 63. Transimpedance Amplifier Equivalent Input Source Model
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From Figure 62, it is clear that with LMH6629’s extremely low-noise characteristics, for RF < 2.5 kΩ, the noise
performance is entirely dominated by RF thermal noise. Only above this RF threshold, LMH6629’s input noise
current (in) starts being a factor and at no RF setting does the LMH6629 input noise voltage play a significant
role. This noise analysis has ignored the possible noise gain increase, due to photo-diode capacitance, at higher
frequencies.
7.3.8 Low-Noise Integrator
Figure 64 shows a deBoo integrator implemented with the LMH6629. Positive feedback maintains integration
linearity. The LMH6629’s low input offset voltage and matched inputs allow bias current cancellation and provide
for very precise integration. Keeping RG and RS low helps maintain dynamic stability.
VO #VIN
KO
KO = 1 +
;
sRSC
RF
RG
RB
VO
RS
+
VIN
C
R
-
50:
50:
RF
RG
R F = RB
RG = RS||R
Figure 64. Low-Noise Integrator
7.3.9 High-Gain Sallen-Key Active Filters
The LMH6629 is well suited for high-gain Sallen-Key type of active filters. Figure 65 shows the 2nd order SallenKey low-pass filter topology. Using component predistortion methods discussed in OA-21, Component PreDistortion for Sallen Key Filters (SNOA369), enables the proper selection of components for these highfrequency filters.
C1
R1
R2
VIN
C2
Vo
#
VIN
K
s
s2
1+
+ 2
Zp Qp Zp
1
= R1C1(1 - K ) + C2 (R1 + R2)
Zp Qp
VO
RF
RG
1
R1R2 C1C2
2 =
Zp
Figure 65. Low Pass Sallen-Key Active Filter Topology
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7.4 Device Functional Modes
With an industry-leading low noise voltage operating off a supply voltage as low as 2.7-V and a common mode
input voltage range that extends 0.3 V below V−, the LMH6629 finds applications in single supply, high
bandwidth, ultra-low noise applications. With a GBWP of 4GHz, the LMH6629 can operate at large gains and
deliver exceptional speed and low noise. Choose the WSON(8) package for the ultimate flexibility (including
Power Down and COMP pin which allows tailoring internal compensation to the operating gain conditions), or the
SOT23-5 package if Power Down is not needed and closed loop gain is ≥ 20dB.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The following discussion details some of the applications that can benefit from the LMH6629’s ultra-low noise,
wide bandwidth, and single supply capability.
Note that It is essential to use a low-noise / low-distortion device to drive a high resolution ADC. This will
minimize the impact on the quantization noise and to make sure that the driver’s distortion does not dominate the
acquired data.
Equation 11 demonstrates the converter noise expression and Equation 12 shows the converter noise
expression evaluated for the example depicted in Figure 66. Figure 67 shows a high-performance low-noise
equalizer for such applications as magnetic tape channels using the LMH6629. Figure 68 shows the circuit’s
simulated frequency response.
8.2 Typical Application
Many high-resolution data converters (ADC’s) require a differential input driver. In order to preserve the ADC’s
dynamic range, the analog input driver must have a noise floor which is lower than the ADC’s noise floor.
Figure 66 shows a ground referenced bipolar input (symmetrical swing around 0V) SE to differential converter
used to drive a high resolution ADC. The combination of LMH6629’s low noise and the converter architecture
reduces the impact on the ADC noise.
3Vpp
47, Rg
0.417V
Vset
2.32k
1%
R1
464
1%
R2
2.5V
470, KRg
5V
A2
LMH6629
2.5V
RO
VREF
VinVin+
CO
1k, Rf
ADC
RO
470, KRg
5V
600 mVpp
A1
LMH6629
GND
2.5V
3Vpp
200, Rin
VIN
Governing Expressions :
Vo _ diff
K=
Vin
R in =
Vset =
2R f
K
Example:
Vo _ CM = 2. 5V;
K = 10, Let R f = 1k : o Rin = 200:
Vo _ diff = 6 Vpp ;
Vset =
2 x 2.5V
= 0. 417 V
10 + 2
Vin = 600 mVpp
2 Vo _ CM
K +2
Figure 66. Low-Noise Single-Ended (SE) to Differential Converter
32
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Typical Application (continued)
8.2.1 Design Requirements
For an ADC with N bits, the quantization Signal-to-noise ratio (SNR) is 6.02* N + 1.76 in dB. For example, a 12bit ADC has a SNR of 74 dB (= 5000 V/V). Assuming a full-scale differential input of 2Vpp (0.707 V_RMS), the
quantization oise referred to the ADC’s input is ~140 μV_RMS (= 0.707 V_RMS / 5000 V/V) over the bandwidth
“visible” to the ADC. Assuming an ADC input bandwidth of 20 MHz, this translates to just 25 nV/RtHz (= 141
µV_RMS / SQRT(20 MHz * π/2)) noise density at the output of the driver. Using an amplifier to form the singleended (SE) to Differential converter / driver for such an application is challenging, especially when there is some
gain required. In addition, the input driver’s linearity (harmonic distortion) must also be high enough such that the
spurs that get through to the ADC input are below the ADC’s LSB threshold or -73 dBc (= 20*log (1/ 212)) or
lower in this case. Therefore, it is essential to use a low-noise / low-distortion device to drive a high resolution
ADC in order to minimize the impact on the quantization noise and to make sure that the driver’s distortion does
not dominate the acquired data.
8.2.2 Detailed Design Procedure
In the circuit depicted in Figure 66, the required gain dictates the resistor ratio “K”. With “K” and the driver output
CM voltage (VO_CM) known, VSET can be established. Reasonable values for Rf and Rg can be set to complete
the design.
In terms of output swing, with the LMH6629 output swing capability which requires ~0.85 V of headroom from
either rail, the maximum total output swing into the ADC is limited to 6.6 VPP (=(5 – 2 x 0.85V) x 2); that is true
with VO_CM set to mid-rail between V+ and V-. It should also be noted that the LMH6629’s input CMVR range
includes the lower rail (V-) and that is the reason there is great flexibility in setting Vo_CM by controlling VSET.
Another feature is that A1 and A2 inputs act like “virtual grounds” and thus do not see any signal swing. Note that
due to the converter’s biasing, the source, VIN, needs to sink a current equal to VSET / RIN.
The converter example shown in Figure 66 operates with a noise gain of 6 (=1+ K / 2) and thus requires that the
COMP pin to be tied low (WSON-8 package only). The 1st order approximated small signal bandwidth will be 280
MHz (=1.7 GHz / 6 V/V) which is computed using 1.7 GHz as the GBWP with COMP pin LO.
From a noise point of view, concentrating only on the dominant noise sources involved, here is the expression for
the expected differential noise density at the input of the ADC.
Converter Noise Expression:
>en (1 + K / 2)@ 2 . 23 + >(eRin _ thermal) K / 2)@2. 2 2 + >(eRg _ thermal ) K)@2
Vnoise #
(11)
3
2
en is the LMH6629 input noise voltage and eRin_thermal is the thermal noise of RIN. The “2 ” and the “2 ” multipliers
account for the different instances of each noise source (2 for en, and 1 for eRin_thermal).
Equation 11, evaluated for the circuit example of Figure 66, is shown in Equation 12:
>0.69 nV/RtHz x 6@2.2 3 + >1.82 nV/RtHz x 5@2 .2 2 + >0.88 nV/ RtHz x 10@2 = 23.4 nV/RtHz
Vnoise #
(12)
Because of the LMH6629’s low input noise voltage (en), noise is dominated by the thermal noise of RIN. It is
evident that the input resistor, RIN, can be reduced to lower the noise with lower input impedance as the tradeoff.
8.2.2.1 Low-Noise Magnetic Media Equalizer
Figure 67 shows a high-performance low-noise equalizer for such applications as magnetic tape channels using
the LMH6629. The circuit combines an integrator (used to limit noise) with a bandpass filter (used to boost the
response centered at a frequency or over a band of interest) to produce the low-noise equalization. The circuit’s
simulated frequency response is illustrated in Figure 68.
In this circuit, the bandpass filter center frequency is set by Equation 13:
´
¶C
=
1
2S LC
(13)
For higher selectivity, use high C values; for wider bandwidth, use high L values, while keeping the product of L
and C values the same to keep fc intact. The integrator’s -3dB roll-off is set by
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Typical Application (continued)
1
2SC1(R1 + R)
(14)
1
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