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LMP7718MME/NOPB

LMP7718MME/NOPB

  • 厂商:

    BURR-BROWN(德州仪器)

  • 封装:

    VSSOP8

  • 描述:

    LMP7718 DUAL CHANNEL, 88 MHZ, PR

  • 数据手册
  • 价格&库存
LMP7718MME/NOPB 数据手册
LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 88 MHz, Precision, Low Noise, 1.8V CMOS Input, Decompensated Operational Amplifier Check for Samples: LMP7717, LMP7718 FEATURES DESCRIPTION • • • • • • • The LMP7717 (single) and the LMP7718 (dual) low noise, CMOS input operational amplifiers offer a low input voltage noise density of 5.8 nV/√Hz while consuming only 1.15 mA (LMP7717) of quiescent current. The LMP7717/LMP7718 are stable at a gain of 10 and have a gain bandwidth (GBW) product of 88 MHz. The LMP7717/LMP7718 have a supply voltage range of 1.8V to 5.5V and can operate from a single supply. The LMP7717/LMP7718 each feature a rail-to-rail output stage. Both amplifiers are part of the LMP™ precision amplifier family and are ideal for a variety of instrumentation applications. 1 23 • • • • (Typical 5V Supply, Unless Otherwise Noted) Input Offset Voltage: ±150 µV (max) Input Referred Voltage Noise: 5.8 nV/√Hz Input Bias Current: 100 fA Gain Bandwidth Product: 88 MHz Supply Voltage Range: 1.8V to 5.5V Supply Current Per Channel – LMP7717: 1.15 mA – LMP7718: 1.30 mA Rail-to-Rail Output Swing – @ 10 kΩ Load: 25 mV from Rail – @ 2 kΩ Load: 45 mV from Rail Ensured 2.5V and 5.0V Performance Total Harmonic Distortion: 0.04% @1 kHz, 600Ω Temperature Range: −40°C to 125°C APPLICATIONS • • • • • • ADC Interface Photodiode Amplifiers Active Filters and Buffers Low Noise Signal Processing Medical Instrumentation Sensor Interface Applications The LMP7717 family provides optimal performance in low voltage and low noise systems. A CMOS input stage, with typical input bias currents in the range of a few femto-Amperes, and an input common mode voltage range, which includes ground, make the LMP7717/LMP7718 ideal for low power sensor applications where high speeds are needed. The LMP7717/LMP7718 are manufactured using TI’s advanced VIP50 process. The LMP7717 is offered in either a 5-Pin SOT-23 or an 8-Pin SOIC package. The LMP7718 is offered in either the 8-Pin SOIC or the 8-Pin VSSOP. 1 2 3 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. LMP is a trademark of Texas Instruments. All other trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2007–2013, Texas Instruments Incorporated LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com Typical Application CF RF IIN CCM + + VOUT CD - VB CIN = CD + CCM VOUT = - RF IIN Figure 1. Photodiode Transimpedance Amplifier VOLTAGE NOISE (nV/ Hz) 1000 5V 100 2.5V 10 1 0.1 1 10 100 1k 10k FREQUENCY (Hz) Figure 2. Input Referred Voltage Noise vs. Frequency These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 2 Submit Documentation Feedback Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 Absolute Maximum Ratings (1) (2) ESD Tolerance (3) Human Body Model 2000V Machine Model 200V Charge-Device Model 1000V VIN Differential ±0.3V Supply Voltage (V+ – V−) 6.0V Input/Output Pin Voltage V+ +0.3V, V− −0.3V −65°C to 150°C Storage Temperature Range Junction Temperature (4) +150°C For soldering specifications: see product folder at www.ti.com/ and http://www.ti.com/lit/SNOA549 (1) (2) (3) (4) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test conditions, see 5V Electrical Characteristics(). If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and specifications. Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of JEDEC)Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC). The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board. Operating Ratings (1) Temperature Range (2) −40°C to 125°C Supply Voltage (V+ – V−) −40°C ≤ TA ≤ 125°C 2.0V to 5.5V 0°C ≤ TA ≤ 125°C 1.8V to 5.5V Package Thermal Resistance (θJA (2)) (1) (2) 5-Pin SOT-23 180°C/W 8-Pin SOIC 190°C/W 8-Pin VSSOP 236°C/W Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and the test conditions, see 5V Electrical Characteristics(). The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 3 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com 2.5V Electrical Characteristics (1) Unless otherwise specified, all limits are ensured for TA = 25°C, V+ = 2.5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbol Parameter Min (2) Conditions Typ (3) Max (2) Units ±20 ±180 ±480 µV ±4 μV/°C VOS Input Offset Voltage TC VOS Input Offset Voltage Temperature Drift (4) (5) LMP7717 −1.0 LMP7718 −1.8 Input Bias Current VCM = 1.0V See (6) and IB (5) −40°C ≤ TA ≤ 85°C 0.05 1 25 −40°C ≤ TA ≤ 125°C 0.05 1 100 .006 0.5 50 IOS Input Offset Current VCM = 1.0V See (5) CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 1.4V 83 80 94 PSRR Power Supply Rejection Ratio 2.0V ≤ V+ ≤ 5.5V, VCM = 0V 85 80 100 1.8V ≤ V+ ≤ 5.5V, VCM = 0V 85 98 Common Mode Voltage Range CMRR ≥ 60 dB CMRR ≥ 55 dB AVOL Open Loop Voltage Gain VOUT = 0.15V to 2.2V, RL = 2 kΩ to V+/2 LMP7717 88 82 98 LMP7718 84 80 92 VOUT = 0.15V to 2.2V, RL = 10 kΩ to V+/2 LMP7717 92 88 110 LMP7718 90 86 95 Output Voltage Swing High Output Voltage Swing Low IOUT IS (1) (2) (3) (4) (5) (6) (7) 4 Output Current Supply Current per Amplifier pA dB dB CMVR VOUT pA −0.3 −0.3 1.5 1.5 dB RL = 2 kΩ to V+/2 25 70 77 RL = 10 kΩ to V+/2 20 60 66 RL = 2 kΩ to V+/2 30 70 73 RL = 10 kΩ to V+/2 15 60 62 Sourcing to V− VIN = 200 mV See (7) 36 30 47 Sinking to V+ VIN = –200 mV See (7) 7.5 5 15 V mV from either rail mA LMP7717 0.95 1.30 1.65 LMP7718 per channel 1.1 1.5 1.85 mA Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of the device such that TJ = TA. No specification of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ > TA. Limits are 100% production tested at 25°C. Limits over the operating temperature range are specified through correlations using the statistical quality control (SQC) method. Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will also depend on the application and configuration. The typical values are not tested and are not specified on shipped production material. Offset voltage average drift is determined by dividing the change in VOS by temperature change. Parameter is specified by design and/or characterization and is not test in production. Positive current corresponds to current flowing into the device. The short circuit test is a momentary test, the short circuit duration is 1.5 ms. Submit Documentation Feedback Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 2.5V Electrical Characteristics(1) (continued) Unless otherwise specified, all limits are ensured for TA = 25°C, V+ = 2.5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbol SR Parameter Conditions Slew Rate Min (2) Typ (3) AV = +10, Rising (10% to 90%) 32 AV = +10, Falling (90% to 10%) 24 Max (2) Units V/μs GBW Gain Bandwidth AV = +10, RL = 10 kΩ 88 MHz en Input Referred Voltage Noise Density f = 1 kHz 6.2 nV/√Hz in Input Referred Current Noise Density f = 1 kHz 0.01 pA/√Hz THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RL = 600Ω 0.01 % Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 5 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com 5V Electrical Characteristics (1) Unless otherwise specified, all limits are ensured for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbo l VOS Parameter Min (2) Conditions Input Offset Voltage Typ (3) Max (2) Units ±10 ±150 ±450 µV ±4 μV/°C TC VOS Input Offset Voltage Temperature Drift (4) (5) LMP7717 −1.0 LMP7718 −1.8 IB VCM = 2.0V See (6)and (5) Input Bias Current −40°C ≤ TA ≤ 85°C 0.1 1 25 −40°C ≤ TA ≤ 125°C 0.1 1 100 .01 0.5 50 IOS Input Offset Current VCM = 2.0V See (5) CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 3.7V 85 80 100 PSRR Power Supply Rejection Ratio 2.0V ≤ V+ ≤ 5.5V, VCM = 0V 85 80 100 1.8V ≤ V+ ≤ 5.5V, VCM = 0V 85 98 dB Common Mode Voltage Range CMRR ≥ 60 dB CMRR ≥ 55 dB AVOL Open Loop Voltage Gain VOUT = 0.3V to 4.7V, RL = 2 kΩ to V+/2 LMP7717 88 82 107 LMP7718 84 80 90 VOUT = 0.3V to 4.7V, RL = 10 kΩ to V+/2 LMP7717 92 88 110 LMP7718 90 86 95 RL = 2 kΩ to V+/2 LMP7717 35 70 77 LMP7718 45 80 83 25 60 66 LMP7717 42 70 73 LMP7718 45 80 83 25 60 66 Output Voltage Swing High −0.3 −0.3 RL = 10 kΩ to V+/2 Output Voltage Swing Low RL = 2 kΩ to V+/2 RL = 10 kΩ to V+/2 IOUT (1) (2) (3) (4) (5) (6) (7) 6 Output Short Circuit Current pA dB CMVR VOUT pA 4 4 Sourcing to V− VIN = 200 mV See (7) 46 38 60 Sinking to V+ VIN = –200 mV See (7) 10.5 6.5 21 V dB mV from either rail mA Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of the device such that TJ = TA. No specification of parametric performance is indicated in the electrical tables under conditions of internal self-heating where TJ > TA. Limits are 100% production tested at 25°C. Limits over the operating temperature range are specified through correlations using the statistical quality control (SQC) method. Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will also depend on the application and configuration. The typical values are not tested and are not specified on shipped production material. Offset voltage average drift is determined by dividing the change in VOS by temperature change. Parameter is specified by design and/or characterization and is not test in production. Positive current corresponds to current flowing into the device. The short circuit test is a momentary test, the short circuit duration is 1.5 ms. Submit Documentation Feedback Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 5V Electrical Characteristics(1) (continued) Unless otherwise specified, all limits are ensured for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbo l IS SR Typ (3) Max (2) LMP7717 1.15 1.40 1.75 LMP7718 per channel 1.30 1.70 2.05 Parameter Min (2) Conditions Supply Current per Amplifier Slew Rate Units mA AV = +10, Rising (10% to 90%) 35 AV = +10, Falling (90% to 10%) 28 88 MHz V/μs GBW Gain Bandwidth AV = +10, RL = 10 kΩ en Input Referred Voltage Noise Density f = 1 kHz 5.8 nV/√Hz in Input Referred Current Noise Density f = 1 kHz 0.01 pA/√Hz f = 1 kHz, AV = 1, RL = 600Ω 0.01 % THD+N Total Harmonic Distortion + Noise CONNECTION DIAGRAMS OUTPUT V - 5 1 2 + +IN + V 4 3 -IN Figure 3. 5-Pin SOT-23 (LMP7717) Top View N/C -IN +IN V - 1 8 2 7 3 + 6 4 5 N/C + V OUTPUT N/C Figure 4. 8-Pin SOIC (LMP7717) Top View -IN A +IN A V - 8 1 2 3 4 - OUT A 7 + +- 6 5 + V OUT B -IN B +IN B Figure 5. 8-Pin SOIC/VSSOP (LMP7718) Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 7 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com TYPICAL PERFORMANCE CHARACTERISTICS Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2. TCVOS Distribution (LMP7717) Offset Voltage Distribution 25 25 VS = 5V -40°C d TA d 125qC VS = 2.5V, 5V PERCENTAGE (%) PERCENTAGE (%) UNITS TESTED: 10,000 15 VCM = VS/2 UNITS TESTED: 10,000 20 20 V CM = VS/2 10 5 15 10 5 0 -4 -3 -2 -1 0 1 0 -200 2 100 Figure 6. Figure 7. 20 VCM = VS/2 UNITS TESTED: 10,000 15 VS = 2.5V VCM = VS/2 UNITS TESTED:10,000 20 10 200 Offset Voltage Distribution 25 -40°C d TA d 125°C VS = 2.5V, 5V PERCENTAGE (%) PERCENTAGE (%) 0 OFFSET VOLTAGE (PV) TCVOS Distribution (LMP7717) 25 -100 TCVOS (PV/°C) 5 15 10 5 0 -4 -3 -2 -1 0 -200 0 -100 0 100 TCVOS (PV/°C) OFFSET VOLTAGE (PV) Figure 8. Figure 9. Supply Current vs. Supply Voltage (LMP7717) Offset Voltage vs. VCM 2 200 200 VS = 1.8V 1.6 125°C 25°C 1.2 0.8 -40°C 0.4 OFFSET VOLTAGE (PV) SUPPLY CURRENT (mA) 150 -40°C 100 50 25°C 0 -50 125°C -100 -150 0 1.5 2.5 3.5 4.5 5.5 6.0 -200 -0.3 + Figure 10. 8 Submit Documentation Feedback 0 0.3 0.6 0.9 1.2 1.5 VCM (V) V (V) Figure 11. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 TYPICAL PERFORMANCE CHARACTERISTICS (continued) Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2. Offset Voltage vs. VCM Offset Voltage vs. VCM 200 200 VS = 2.5V VS = 5V 150 -40°C OFFSET VOLTAGE (PV) OFFSET VOLTAGE (PV) 150 100 25°C 50 0 125°C -50 -100 -150 -40°C 50 25°C 0 125°C -50 -100 -150 -200 -0.3 0 0.3 0.6 0.9 1.2 1.5 1.8 -200 -0.3 2.1 0.7 1.7 2.7 3.7 VCM (V) VCM (V) Figure 12. Figure 13. Offset Voltage vs. Temperature Slew Rate vs. Supply Voltage 150 36 100 34 50 32 4.7 RISING EDGE SLEW RATE (V/Ps) OFFSET VOLTAGE (PV) 100 VS = 2.5V 0 LMP7717 -50 -100 VS = 5V -150 30 28 26 FALLING EDGE 24 LMP7718 -200 -40 -20 0 20 40 60 22 1.5 80 100 120 125 3.5 4.5 5.5 + V (V) Figure 14. Figure 15. Input Bias Current Over Temperature Input Bias Current vs. VCM 50 1000 VS = 5V 25°C 500 + V = 5V 40 30 0 20 -500 IBIAS (pA) INPUT BIAS CURRENT (fA) 2.5 TEMPERATURE (°C) -40°C -1000 -1500 125°C 10 0 85°C -10 -20 -2000 -30 -2500 -40 -50 -3000 0 1 2 3 4 0 1 2 3 4 VCM (V) VCM (V) Figure 16. Figure 17. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 9 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com TYPICAL PERFORMANCE CHARACTERISTICS (continued) Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2. Sourcing Current vs. Supply Voltage 200 80 150 70 100 50 25°C 0 125°C -50 125°C 60 -40°C ISOURCE (mA) OFFSET VOLTAGE (PV) Offset Voltage vs. Supply Voltage -40°C 50 25°C 40 30 -100 20 -150 10 0 -200 1.5 2.5 3.5 4.5 5.5 1 6 2 3 4 5 6 + VS (V) V (V) Figure 18. Figure 19. Sinking Current vs. Supply Voltage Sourcing Current vs. Output Voltage 35 70 30 60 125°C 125°C 50 ISOURCE (mA) ISINK (mA) 25 25°C 20 15 -40°C 10 -40°C 40 25°C 30 20 5 10 0 0 1 2 3 4 5 6 0 1 2 + 3 4 V (V) VOUT (V) Figure 20. Figure 21. Sinking Current vs. Output Voltage Positive Output Swing vs. Supply Voltage 30 5 40 RLOAD = 10 k: 125°C 35 VOUT FROM RAIL (mV) 25 ISINK (mA) 20 25°C 15 10 -40°C 30 125°C 25 25°C 20 -40°C 15 10 5 5 0 0 10 1 2 3 4 5 0 1.8 2.5 3.2 3.9 4.6 V (V) Figure 22. Figure 23. Submit Documentation Feedback 5.3 6 + VOUT (V) Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 TYPICAL PERFORMANCE CHARACTERISTICS (continued) Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2. Negative Output Swing vs. Supply Voltage Positive Output Swing vs. Supply Voltage 25 50 -40°C 45 20 VOUT FROM RAIL (mV) VOUT FROM RAIL (mV) 25°C 15 125°C 10 5 125°C 40 25°C 35 30 25 20 -40°C 15 10 RLOAD = 10 k: 0 1.8 2.5 3.2 5 3.9 4.6 5.3 RLOAD = 2 k: 0 1.8 2.5 3.2 6 3.9 + V (V) Figure 25. Negative Output Swing vs. Supply Voltage Positive Output Swing vs. Supply Voltage -40°C 90 25°C 40 VOUT FROM RAIL (mV) VOUT FROM RAIL (mV) 6 100 50 35 125°C 30 25 20 15 125°C 70 50 40 10 5.3 -40°C 30 20 4.6 25°C 60 5 3.9 RLOAD = 600: 80 10 RLOAD = 2 k: 0 1.8 2.5 3.2 0 1.8 6 2.5 3.2 3.9 + 4.6 5.3 6 + V (V) V (V) Figure 26. Figure 27. Negative Output Swing vs. Supply Voltage Input Referred Voltage Noise vs. Frequency 1000 120 125°C RLOAD = 600: 25°C VOLTAGE NOISE (nV/ Hz) 100 VOUT FROM RAIL (mV) 5.3 Figure 24. 45 80 -40°C 60 40 20 0 1.8 4.6 + V (V) 2.5 3.2 3.9 4.6 5.3 6 5V 100 2.5V 10 1 0.1 1 10 100 + V (V) FREQUENCY (Hz) Figure 28. Figure 29. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 1k 10k Submit Documentation Feedback 11 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com TYPICAL PERFORMANCE CHARACTERISTICS (continued) Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2. Overshoot and Undershoot vs. CLOAD Time Domain Voltage Noise 70 OVERSHOOT AND UNDERSHOOT % VS = ±2.5V 400 nV/DIV VCM = 0.0V US% 60 50 OS% 40 30 20 10 0 1 s/DIV 20 0 40 80 60 100 120 CLOAD (pF) Figure 30. Figure 31. THD+N vs. Frequency THD+N vs. Frequency 0.04 0.05 RL = 600: RL = 600: 0.04 THD+N (%) THD+N (%) 0.03 + 0.03 V = 2.5V 0.02 AV = +10 VO = 1 VPP 0.02 VO = 4 VPP 0.01 0.01 + V = 5V AV = +10 RL = 100 k: 0 10 100 RL = 100 k: 1k 10k 0 10 100k 100 1k 10k 100k FREQUENCY (Hz) FREQUENCY (Hz) Figure 32. Figure 33. THD+N vs. Peak-to-Peak Output Voltage (VOUT) THD+N vs. Peak-to-Peak Output Voltage (VOUT) -40 -40 -50 -60 RL = 600: -70 V+ = 2.5V f = 1 kHz AV = +10 -80 0.01 12 THD+N (dB) THD+N (dB) -50 RL = 100 k: 0.1 1 10 RL = 600: -60 -70 -80 V+ = 5V f = 1 kHz AV = +10 -90 0.01 RL = 100 k: 0.1 1 OUTPUT AMPLITUDE (VPP) OUTPUT AMPLITUDE (VPP) Figure 34. Figure 35. Submit Documentation Feedback 10 Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 TYPICAL PERFORMANCE CHARACTERISTICS (continued) Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2. Closed Loop Output Impedance vs. Frequency Open Loop Gain and Phase 100 1000 100 80 60 60 GAIN 40 40 20 20 + V = 5V 0 CL = 20 pF RL = 2 k:, 10 k: -20 100k 1M OUTPUT IMPEDANCE (:) 80 PHASE (°) GAIN (dB) PHASE 100 10 1 0 -20 100M 10M 0.1 10k 100k 1M 10M 100M FREQUENCY (Hz) FREQUENCY (Hz) Figure 36. Figure 37. Crosstalk Rejection Small Signal Transient Response, AV = +10 140 120 10 mV/DIV CROSSTALK REJECTION RATION (dB) 160 100 80 60 40 VIN = 2 mVPP f = 1 MHz, AV = +10 20 + V = 5V, CL = 10 pF 0 1k 100k 10k 1M 10M 100 ns/DIV 100M FREQUENCY (Hz) Figure 39. Large Signal Transient Response, AV = +10 Small Signal Transient Response, AV = +10 10 mV/DIV 200 mV/DIV Figure 38. VIN = 100 mVPP VIN = 2 mVPP f = 1 MHz, AV = +10 f = 1 MHz, AV = +10 V = 2.5V, CL = 10 pF V = 2.5V, CL = 10 pF + + 100 ns/DIV 100 ns/DIV Figure 40. Figure 41. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 13 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com TYPICAL PERFORMANCE CHARACTERISTICS (continued) Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2. PSRR vs. Frequency Large Signal Transient Response, AV = +10 100 -PSRR, 5V 90 PSRR (dB) 200 mV/DIV 80 -PSRR, 2.5V +PSRR, 5V +PSRR, 2.5V 70 60 VIN = 100 mVPP 50 f = 1 MHz, AV = +10 + V = 5V, CL = 10 pF 40 100 100 ns/DIV 1k 10k 100k 1M FREQUENCY (Hz) Figure 42. Figure 43. CMRR vs. Frequency Input Common Mode Capacitance vs. VCM 120 25 + V = 5V 100 20 + 80 CCM (pF) CMRR (dB) V = 2.5V 60 + V = 5V 15 10 40 5 20 0 10 100 1k 10k 100k 1M 10M 0 0 FREQUENCY (Hz) Submit Documentation Feedback 2 3 4 VCM (V) Figure 44. 14 1 Figure 45. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 APPLICATION INFORMATION ADVANTAGES OF THE LMP7717/LMP7718 Wide Bandwidth at Low Supply Current The LMP7717/LMP7718 are high performance op amps that provide a GBW of 88 MHz with a gain of 10 while drawing a low supply current of 1.15 mA. This makes them ideal for providing wideband amplification in data acquisition applications. With the proper external compensation the LMP7717 can be operated at gains of ±1 and still maintain much faster slew rates than comparable unity gain stable amplifiers. The increase in bandwidth and slew rate is obtained without any additional power consumption over the LMP7715. Low Input Referred Noise and Low Input Bias Current The LMP7717/LMP7718 have a very low input referred voltage noise density (5.8 nV/√Hz at 1 kHz). A CMOS input stage ensures a small input bias current (100 fA) and low input referred current noise (0.01 pA/√Hz). This is very helpful in maintaining signal integrity, and makes the LMP7717/LMP7718 ideal for audio and sensor based applications. Low Supply Voltage The LMP7717 and the LMP7718 have performance ensured at 2.5V and 5V supply. These parts are ensured to be operational at all supply voltages between 2.0V and 5.5V, for ambient temperatures ranging from −40°C to 125°C, thus utilizing the entire battery lifetime. The LMP7717/LMP7718 are also ensured to be operational at 1.8V supply voltage, for temperatures between 0°C and 125°C optimizing their usage in low-voltage applications. RRO and Ground Sensing Rail-to-Rail output swing provides the maximum possible dynamic range. This is particularly important when operating at low supply voltages. An innovative positive feedback scheme is used to boost the current drive capability of the output stage. This allows the LMP7717/LMP7718 to source more than 40 mA of current at 1.8V supply. This also limits the performance of the these parts as comparators, and hence the usage of the LMP7717 and the LMP7718 in an open-loop configuration is not recommended. The input common-mode range includes the negative supply rail which allows direct sensing at ground in single supply operation. Small Size The small footprints of the LMP7717 packages and the LMP7718 packages save space on printed circuit boards, and enable the design of smaller electronic products, such as cellular phones, pagers, or other portable systems. Long traces between the signal source and the op amp make the signal path more susceptible to noise pick up. The physically smaller LMP7717 or LMP7718 packages allow the op amp to be placed closer to the signal source, thus reducing noise pickup and maintaining signal integrity. USING THE DECOMPENSATED LMP7717 Advantages of Decompensated Op Amp A unity gain stable op amp, which is fully compensated, is designed to operate with good stability down to gains of ±1. The large amount of compensation does provide an op amp that is relatively easy to use; however, a decompensated op amp is designed to maximize the bandwidth and slew rate without any additional power consumption. This can be very advantageous. The LMP7717/LMP7718 require a gain of ±10 to be stable. However, with an external compensation network (a simple RC network) these parts can be stable with gains of ±1 and still maintain the higher slew rate. Looking at the Bode plots for the LMP7717 and its closest equivalent unity gain stable op amp, the LMP7715, one can clearly see the increased bandwidth of the LMP7717. Both plots are taken with a parallel combination of 20 pF and 10 kΩ for the output load. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 15 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com 100 100 100 80 80 80 80 60 60 60 60 40 40 20 20 0 0 100 20 20 0 0 -20 1k 10k 100k 1M -20 100M 10M -20 1k 10k 100k 1M 10M PHASE (°) 40 GAIN (dB) GAIN 40 PHASE (°) GAIN (dB) PHASE -20 100M FREQUENCY (Hz) FREQUENCY (Hz) Figure 46. LMP7717 AVOL vs. Frequency Figure 47. LMP7715 AVOL vs. Frequency Figure 46 shows the much larger 88 MHz bandwidth of the LMP7717 as compared to the 17 MHz bandwidth of the LMP7715 shown in Figure 47. The decompensated LMP7717 has five times the bandwidth of the LMP7715. What is a Decompensated Op Amp? The differences between the unity gain stable op amp and the decompensated op amp are shown in Figure 48. This Bode plot assumes an ideal two pole system. The dominant pole of the decompensated op amp is at a higher frequency, f1, as compared to the unity gain stable op amp which is at fd as shown in Figure 48. This is done in order to increase the speed capability of the op amp while maintaining the same power dissipation of the unity gain stable op amp. The LMP7717/LMP7718 have a dominant pole at 1.6 kHz. The unity gain stable LMP7715/LMP7716 have their dominant pole at 300 Hz. DECOMPENSATED OP AMP AOL UNITY-GAIN STABLE OP AMP Gmin fGBWP fd f1 fu f2 fu' Figure 48. Open Loop Gain for Unity Gain Stable Op Amp and Decompensated Op Amp Having a higher frequency for the dominate pole will result in: 1. The DC open loop gain (AVOL) extending to a higher frequency. 2. A wider closed loop bandwidth. 3. Better slew rate due to reduced compensation capacitance within the op amp. The second open loop pole (f2) for the LMP7717/LMP7718 occurs at 45 MHz. The unity gain (fu’) occurs after the second pole at 51 MHz. An ideal two pole system would give a phase margin of 45° at the location of the second pole. The LMP7717/LMP7718 have parasitic poles close to the second pole, giving a phase margin closer to 0°. Therefore it is necessary to operate the LMP7717/LMP7718 at a closed loop gain of 10 or higher, or to add external compensation in order to assure stability. For the LMP7715, the gain bandwidth product occurs at 17 MHz. The curve is constant from fd to fu which occurs before the second pole. For the LMP7717/LMP7718 the GBW = 88 MHz and is constant between f1 and f2. The second pole at f2 occurs before AVOL =1. Therefore fu’ occurs at 51 MHz, well before the GBW frequency of 88 MHz. For decompensated op amps the unity gain frequency and the GBW are no longer equal. Gmin is the minimum gain for stability and for the LMP7717/LMP7718 this is a gain of 18 to 20 dB. 16 Submit Documentation Feedback Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 Input Lead-Lag Compensation The recommended technique which allows the user to compensate the LMP7717/LMP7718 for stable operation at any gain is lead-lag compensation. The compensation components added to the circuit allow the user to shape the feedback function to make sure there is sufficient phase margin when the loop gain is as low as 0 dB and still maintain the advantages over the unity gain op amp. Figure 49 shows the lead-lag configuration. Only RC and C are added for the necessary compensation. RF RIN RC LMP7717 C Figure 49. LMP7717 with Lead-Lag Compensation for Inverting Configuration To cover how to calculate the compensation network values it is necessary to introduce the term called the feedback factor or F. The feedback factor F is the feedback voltage VA-VB across the op amp input terminals relative to the op amp output voltage VOUT. VA - V B F= VOUT (1) From feedback theory the classic form of the feedback equation for op amps is: VOUT VIN = A 1 + AF (2) A is the open loop gain of the amplifier and AF is the loop gain. Both are highly important in analyzing op amps. Normally AF >>1 and so the above equation reduces to: VOUT VIN 1 F = (3) Deriving the equations for the lead-lag compensation is beyond the scope of this datasheet. The derivation is based on the feedback equations that have just been covered. The inverse of feedback factor for the circuit in Figure 49 is: © § ¨ ¨ © F § = ¨¨1 + RF §¨1 + s(Rc + RIN || RF) C 1 + sRcC RIN ¨ © § ¨ ¨ © 1 (4) where 1/F's pole is located at fp = 1 2SRcC (5) 1/F's zero is located at fz = 1 2S Rc + RIN || RF)C (6) 1 F =1+ f=0 RF RIN (7) The circuit gain for Figure 49 at low frequencies is −RF/RIN, but F, the feedback factor is not equal to the circuit gain. The feedback factor is derived from feedback theory and is the same for both inverting and non-inverting configurations. Yes, the feedback factor at low frequencies is equal to the gain for the non-inverting configuration. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 17 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 f=f © § ¨ ¨ © F = ¨¨1 + RIN || RF RF §¨ 1+ RC RIN ¨© § ¨ ¨ © § 1 www.ti.com (8) From this formula, we can see that • 1/F's zero is located at a lower frequency compared with 1/F's pole. • 1/F's value at low frequency is 1 + RF/RIN. • This method creates one additional pole and one additional zero. • This pole-zero pair will serve two purposes: – To raise the 1/F value at higher frequencies prior to its intercept with A, the open loop gain curve, in order to meet the Gmin = 10 requirement. For the LMP7717 some overcompensation will be necessary for good stability. – To achieve the previous purpose above with no additional loop phase delay. Please note the constraint 1/F ≥ Gmin needs to be satisfied only in the vicinity where the open loop gain A and 1/F intersect; 1/F can be shaped elsewhere as needed. The 1/F pole must occur before the intersection with the open loop gain A. In order to have adequate phase margin, it is desirable to follow these two rules: Rule 1 1/F and the open loop gain A should intersect at the frequency where there is a minimum of 45° of phase margin. When over-compensation is required the intersection point of A and 1/F is set at a frequency where the phase margin is above 45°, therefore increasing the stability of the circuit. Rule 2 1/F’s pole should be set at least one decade below the intersection with the open loop gain A in order to take advantage of the full 90° of phase lead brought by 1/F’s pole which is F’s zero. This ensures that the effect of the zero is fully neutralized when the 1/F and A plots intersect each other. Calculating Lead-Lag Compensation for LMP7717 Figure 50 is the same plot as Figure 46, but the AVOL and phase curves have been redrawn as smooth lines to more readily show the concepts covered, and to clearly show the key parameters used in the calculations for lead-lag compensation. 100 PHASE ADDITIONAL COMPENSATION GAIN (dB) and PHASE (°C) 80 AVOL 60 45° PHASE MARGIN 40 20 1 with ADDITIONAL F COMPENSATION 1 F 0 -20 1k 2nd POLE f2 10k 100k 1M GBP 10M 100M FREQUENCY (Hz) Figure 50. LMP7717/LMP7718 Simplified Bode Plot To obtain stable operation with gains under 10 V/V the open loop gain margin must be reduced at high frequencies to where there is a 45° phase margin when the gain margin of the circuit with the external compensation is 0 dB. The pole and zero in F, the feedback factor, control the gain margin at the higher frequencies. The distance between F and AVOL is the gain margin; therefore, the unity gain point (0 dB) is where F crosses the AVOL curve. For the example being used RIN = RF for a gain of −1. Therefore F = 6 dB at low frequencies. At the higher frequencies the minimum value for F is 18 dB for 45° phase margin. From Equation 5 we have the following relationship: 18 Submit Documentation Feedback Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com RF RIN § ¨ ¨ © + § ¨ ¨1 © + RIN || RF RC § ¨ ¨ © § ¨1 ¨ © SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 = 18 dB = 7.9 (9) Now set RF = RIN = R. With these values and solving for RC we have RC = R/5.9. Note that the value of C does not affect the ratio between the resistors. Once the value of the resistors is set, then the position of the pole in F must be set. A 2 kΩ resistor is used for RF and RIN in this design. Therefore the value for RC is set at 330Ω, the closest standard value for 2 kΩ/5.9. Rewriting Equation 2 to solve for the minimum capacitor value gives the following equation: C = 1/(2πfpRC) (10) The feedback factor curve, F, intersects the AVOL curve at about 12 MHz. Therefore the pole of F should not be any larger than 1.2 MHz. Using this value and RC = 330Ω the minimum value for C is 390 pF. Figure 51 shows that there is too much overshoot, but the part is stable. Increasing C to 2.2 nF did not improve the ringing, as shown in Figure 52. Figure 51. First Try at Compensation, Gain = −1 Figure 52. C Increased to 2.2 nF, Gain = −1 Some over-compensation appears to be needed for the desired overshoot characteristics. Instead of intersecting the AVOL curve at 18 dB, 2 dB of over-compensation will be used, and the AVOL curve will be intersected at 20 dB. Using Equation 5 for 20 dB, or 10 V/V, the closest standard value of RC is 240Ω. The following two waveforms show the new resistor value with C = 390 pF and 2.2 nF. Figure 54 shows the final compensation and a very good response for the 1 MHz square wave. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 19 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com Figure 53. RC = 240Ω and C = 390 pF, Gain = −1 Figure 54. RC = 240Ω and C = 2.2 nF, Gain = −1 To summarize, the following steps were taken to compensate the LMP7717 for a gain of −1: 1. Values for Rc and C were calculated from the Bode plot to give an expected phase margin of 45°. The values were based on RIN = RF = 2 kΩ. These calculations gave Rc = 330Ω and C = 390 pF. 2. To reduce the ringing C was increased to 2.2 nF which moved the pole of F, the feedback factor, farther away from the AVOL curve. 3. There was still too much ringing so 2 dB of over-compensation was added to F. This was done by decreasing RC to 240Ω. The LMP7715 is the fully compensated part which is comparable to the LMP7717. Using the LMP7715 in the same setup, but removing the compensation network, provided the response shown in Figure 55 . Figure 55. LMP7715 Response 20 Submit Documentation Feedback Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 For large signal response the rise and fall times are dominated by the slew rate of the op amps. Even though both parts are quite similar the LMP7717 will give rise and fall times about 2.5 times faster than the LMP7715. This is possible because the LMP7717 is a decompensated op amp and even though it is being used at a gain of −1, the speed is preserved by using a good technique for external compensation. Non-Inverting Compensation For the non-inverting amp the same theory applies for establishing the needed compensation. When setting the inverting configuration for a gain of −1, F has a value of 2. For the non-inverting configuration both F and the actual gain are the same, making the non-inverting configuration more difficult to compensate. Using the same circuit as shown in Figure 49, but setting up the circuit for non-inverting operation (gain of +2) results in similar performance as the inverting configuration with the inputs set to half the amplitude to compensate for the additional gain. Figure 56 below shows the results. Figure 56. RC = 240Ω and C = 2.2 nF, Gain = +2 Figure 57. LMP7715 Response Gain = +2 The response shown in Figure 56 is close to the response shown in Figure 54. The part is actually slightly faster in the non-inverting configuration. Decreasing the value of RC to around 200Ω can decrease the negative overshoot but will have slightly longer rise and fall times. The other option is to add a small resistor in series with the input signal. Figure 57 shows the performance of the LMP7715 with no compensation. Again the decompensated parts are almost 2.5 times faster than the fully compensated op amp. The most difficult op amp configuration to stabilize is the gain of +1. With proper compensation the LMP7717/LMP7718 can be used in this configuration and still maintain higher speeds than the fully compensated parts. Figure 58 shows the gain = 1, or the buffer configuration, for these parts. Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 21 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com RF RC RP LMP7717 C + Figure 58. LMP7717 with Lead-Lag Compensation for Non-Inverting Configuration Figure 58 is the result of using Equation 5 and additional experimentation in the lab. RP is not part of Equation 5, but it is necessary to introduce another pole at the input stage for good performance at gain = +1. Equation 5 is shown below with RIN = ∞. + RF Rc § ¨ ¨ © § ¨1 ¨ © = 18 dB = 7.9 (11) Using 2 kΩ for RF and solving for RC gives RC = 2000/6.9 = 290Ω. The closest standard value for RC is 300Ω. After some fine tuning in the lab RC = 330Ω and RP = 1.5 kΩ were chosen as the optimum values. RP together with the input capacitance at the non-inverting pin inserts another pole into the compensation for the LMP7717. Adding this pole and slightly reducing the compensation for 1/F (using a slightly higher resistor value for RC) gives the optimum response for a gain of +1. Figure 59 is the response of the circuit shown in Figure 58. Figure 60 shows the response of the LMP7715 in the buffer configuration with no compensation and RP = RF = 0. Figure 59. RC = 330Ω and C = 10 nF, Gain = +1 22 Submit Documentation Feedback Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 Figure 60. LMP7715 Response Gain = +1 With no increase in power consumption the decompensated op amp offers faster speed than the compensated equivalent part . These examples used RF = 2 kΩ. This value is high enough to be easily driven by the LMP7717/LMP7718, yet small enough to minimize the effects from the parasitic capacitance of both the PCB and the op amp. NOTE When using the LMP7717/LMP7718, proper high frequency PCB layout must be followed. The GBW of these parts is 88 MHz, making the PCB layout significantly more critical than when using the compensated counterparts which have a GBW of 17 MHz. TRANSIMPEDANCE AMPLIFIER An excellent application for either the LMP7717 or the LMP7718 is as a transimpedance amplifier. With a GBW product of 88 MHz these parts are ideal for high speed data transmission by light. The circuit shown on the front page of the datasheet is the circuit used to test the LMP7717/LMP7718 as transimpedance amplifiers. The only change is that VB is tied to the VCC of the part, thus the direction of the diode is reversed from the circuit shown on the front page. Very high speed components were used in testing to check the limits of the LMP7717/LMP7718 in a transimpedance configuration. The photodiode part number is PIN-HR040 from OSI Optoelectronics. The diode capacitance for this part is only about 7 pF for the 2.5V bias used (VCC to virtual ground). The rise time for this diode is 1 nsec. A laser diode was used for the light source. Laser diodes have on and off times under 5 nsec. The speed of the selected optical components allowed an accurate evaluation of the LMP7717 as a transimpedance amplifier. TIs evaluation board for decompensated op amps, PN 551013271-001 A, was used and only minor modifications were necessary and no traces had to be cut. CF 2.5V 2.5V RF DPHOTO CD CCM LMP7717 VOUT + -2.5V Figure 61. Transimpedance Amplifier Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 23 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com Figure 61 is the complete schematic for a transimpedance amplifier. Only the supply bypass capacitors are not shown. CD represents the photodiode capacitance which is given on its datasheet. CCM is the input common mode capacitance of the op amp and, for the LMP7717 it is shown in the last graph of the TYPICAL PERFORMANCE CHARACTERISTICS section of this datasheet. In Figure 61 the inverting input pin of the LMP7717 is kept at virtual ground. Even though the diode is connected to the 2.5V line, a power supply line is AC ground, thus CD is connected to ground. Figure 62 shows the schematic needed to derive F, the feedback factor, for a transimpedance amplifier. In Figure 62, CD + CCM = CIN. Therefore it is critical that the designer knows the diode capacitance and the op amp input capacitance. The photodiode is close to an ideal current source once its capacitance is included in the model. What kind of circuit is this? Without CF there is only an input capacitor and a feedback resistor. This circuit is a differentiator! Remember, differentiator circuits are inherently unstable and must be compensated. In this case CF compensates the circuit. CF RF VA IDIODE CIN VOUT LMP7717 + Figure 62. Transimpedance Feedback Model Using feedback theory, F = VA/VOUT, this becomes a voltage divider giving the following equation: F= 1 + sCFRF 1 + sRF (CF + CIN) (12) The noise gain is 1/F. Because this is a differentiator circuit, a zero must be inserted. The location of the zero is given by: ´ ¶z = 1 1 + sRF (CF + CIN) (13) CF has been added for stability. The addition of this part adds a pole to the circuit. The pole is located at: ´ ¶p = 1 1 + sCFRF (14) To attain maximum bandwidth and still have good stability the pole is to be located on the open loop gain curve which is A. If additional compensation is required one can always increase the value of CF, but this will also reduce the bandwidth of the circuit. Therefore A = 1/F, or AF = 1. For A the equation is: A= ZGBW Z = ´ ¶GBW ´ ¶ (15) The expression fGBW is the gain bandwidth product of the part. For a unity gain stable part this is the frequency where A = 1. For the LMP7717 fGBW = 88 MHz. Multiplying A and F results in the following equation: ´ ¶ GBW ´ ¶ ´ ¶ GBW ´ ¶ 1+ x 1+ 24 x § ¨ ¨ © 1 + sCFRF 1 + sRF (CF + CIN) § CFRF ¨ ¨ © CFRF = 2 RF (CF + CIN) CFRF =1 § ¨ ¨ © P § ¨ ¨ © AF ´ = ¶ 2 Submit Documentation Feedback (16) Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 For the above equation s = jω. To find the actual amplitude of the equation the square root of the square of the real and imaginary parts are calculated. At the intersection of F and A, we have: Z= 1 CFRF (17) After a bit of algebraic manipulation the above equation reduces to: §C + C IN ¨ F ¨ CF © § ¨ ¨ © 1+ 2 2 = 8S2 ´ ¶GBW RF CF 2 2 (18) In the above equation the only unknown is CF. In trying to solve this equation the fourth power of CF must be dealt with. An excel spread sheet with this equation can be used and all the known values entered. Then through iteration, the value of CF when both sides are equal will be found. That is the correct value for CF and of course the closest standard value is used for CF. Before moving to the lab, the transfer function of the transimpedance amplifier must be found and the units must be in Ohms. VOUT = -RF 1 + sCFRF x IDIODE (19) The LMP7717 was evaluated for RF = 10 kΩ and 100 kΩ, representing a somewhat lower gain configuration and with the 100 kΩ feedback resistor a fairly high gain configuration. The RF = 10 kΩ is covered first. Looking at the Input Common Mode Capacitance vs. VCM chart for CCM for the operating point selected CCM = 15 pF. Note that for split supplies VCM = 2.5V, CIN = 22 pF and fGBW = 88 MHz. Solving for CF the calculated value is 1.75 pF, so 1.8 pF is selected for use. Checking the frequency of the pole finds that it is at 8.8 MHz, which is right at the minimum gain recommended for this part. Some over compensation was necessary for stability and the final selected value for CF is 2.7 pF. This moves the pole to 5.9 MHz. Figure 63 and Figure 64 show the rise and fall times obtained in the lab with a 1V output swing. The laser diode was difficult to drive due to thermal effects making the starting and ending point of the pulse quite different, therefore the two separate scope pictures. Figure 63. Fall Time Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 25 LMP7717, LMP7718 SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 www.ti.com Figure 64. Rise Time In Figure 63 the ringing and the hump during the on time is from the laser. The higher drive levels for the laser gave ringing in the light source as well as light changing from the thermal characteristics. The hump is due to the thermal characteristics. Solving for CF using a 100 kΩ feedback resistor, the calculated value is 0.54 pF. One of the problems with more gain is the very small value for CF. A 0.5 pF capacitor was used, its measured value being 0.64 pF. For the 0.64 pF location the pole is at 2.5 MHz. Figure 65 shows the response for a 1V output. Figure 65. High Gain Response A transimpedance amplifier is an excellent application for the LMP7717. Even with the high gain using a 100 kΩ feedback resistor, the bandwidth is still well over 1 MHz. Other than a little over compensation for the 10 kΩ feedback resistor configuration using the LMP7717 was quite easy. Of course a very good board layout was also used for this test. 26 Submit Documentation Feedback Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 LMP7717, LMP7718 www.ti.com SNOSAY7H – MARCH 2007 – REVISED MARCH 2013 REVISION HISTORY Changes from Revision G (March 2013) to Revision H • Page Changed layout of National Data Sheet to TI format .......................................................................................................... 26 Copyright © 2007–2013, Texas Instruments Incorporated Product Folder Links: LMP7717 LMP7718 Submit Documentation Feedback 27 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) LMP7717MA/NOPB ACTIVE SOIC D 8 95 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LMP77 17MA LMP7717MAE/NOPB ACTIVE SOIC D 8 250 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LMP77 17MA LMP7717MF/NOPB ACTIVE SOT-23 DBV 5 1000 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 AT4A LMP7717MFE/NOPB ACTIVE SOT-23 DBV 5 250 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 AT4A LMP7717MFX/NOPB ACTIVE SOT-23 DBV 5 3000 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 AT4A LMP7718MA/NOPB ACTIVE SOIC D 8 95 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LMP77 18MA LMP7718MAE/NOPB ACTIVE SOIC D 8 250 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LMP77 18MA LMP7718MAX/NOPB ACTIVE SOIC D 8 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LMP77 18MA LMP7718MM/NOPB ACTIVE VSSOP DGK 8 1000 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 AP4A LMP7718MME/NOPB ACTIVE VSSOP DGK 8 250 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 AP4A (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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