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LMR14030
SNVSA81A – FEBRUARY 2015 – REVISED APRIL 2015
LMR14030 SIMPLE SWITCHER® 40 V 3.5 A, 2.2 MHz Step-Down Converter with 40 µA IQ
1 Features
3 Description
•
•
•
•
•
•
•
The LMR14030 is a 40 V, 3.5 A step down regulator
with an integrated high-side MOSFET. With a wide
input range from 4 V to 40 V, it’s suitable for various
applications from industrial to automotive for power
conditioning from unregulated sources. The
regulator’s quiescent current is 40 µA in Sleep-mode,
which is suitable for battery powered systems. An
ultra-low 1 μA current in shutdown mode can further
prolong battery life. A wide adjustable switching
frequency range allows either efficiency or external
component size to be optimized. Internal loop
compensation means that the user is free from the
tedious task of loop compensation design. This also
minimizes the external components of the device. A
precision enable input allows simplification of
regulator control and system power sequencing. The
device also has built-in protection features such as
cycle-by-cycle current limit, thermal sensing and
shutdown due to excessive power dissipation, and
output overvoltage protection.
1
•
•
•
•
•
•
•
•
4 V to 40 V Input Range
3.5 A Continuous Output Current
Ultra-low 40 µA Operating Quiescent Current
90 mΩ High-Side MOSFET
Minimum Switch-On Time: 75 ns
Current Mode Control
Adjustable Switching Frequency from 200 kHz to
2.5 MHz
Frequency Synchronization to External Clock
Internal Compensation for Ease of Use
High Duty Cycle Operation Supported
Precision Enable Input
1 µA Shutdown Current
External Soft-start
Thermal, Overvoltage and Short Protection
8-Pin HSOIC with PowerPAD™ Package
Device Information(1)
2 Applications
•
•
•
•
Automotive Battery Regulation
Industrial Power Supplies
Telecom and Datacom Systems
Battery Powered System
PART NUMBER
PACKAGE
BODY SIZE (NOM)
LMR14030SDDA
HSOIC-8
4.89 mm x 3.90 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
4 Simplified Schematic
VIN up to 40 V
Efficiency vs Output Current
CIN
VIN
BOOT
EN
100
CBOOT
90
L
80
SW
RT
D
70
RFBT
COUT
SS
FB
CSS
GND
RFBB
Efficiency (%)
RT/SYNC
VOUT
60
50
40
30
20
10
VOUT = 5 V
VOUT = 3.3 V
VIN = 12 V, gSW = 500 kHz
0
0.001
0.01
0.1
IOUT (A)
1
10
D001
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LMR14030
SNVSA81A – FEBRUARY 2015 – REVISED APRIL 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
8
Features ..................................................................
Applications ...........................................................
Description .............................................................
Simplified Schematic.............................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
1
2
3
4
7.1
7.2
7.3
7.4
7.5
7.6
7.7
4
4
4
4
5
5
6
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Detailed Description .............................................. 8
8.1 Overview ................................................................... 8
8.2 Functional Block Diagram ......................................... 8
8.3 Feature Description................................................... 9
9
Application and Implementation ........................ 15
9.1 Application Information............................................ 15
9.2 Typical Application ................................................. 15
10 Power Supply Recommendations ..................... 21
11 Layout................................................................... 21
11.1 Layout Guidelines ................................................. 21
11.2 Layout Example .................................................... 22
12 Device and Documentation Support ................. 23
12.1 Trademarks ........................................................... 23
12.2 Electrostatic Discharge Caution ............................ 23
12.3 Glossary ................................................................ 23
13 Mechanical, Packaging, and Orderable
Information ........................................................... 23
5 Revision History
Changes from Original (February 2015) to Revision A
•
2
Page
Changed from Product Preview to production Data ............................................................................................................... 1
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SNVSA81A – FEBRUARY 2015 – REVISED APRIL 2015
6 Pin Configuration and Functions
HSOIC
8-Pin
Top View
HSOIC PACKAGE
(TOP VIEW)
BOOT
1
VIN
2
EN
Thermal Pad
(9)
3
RT/SYNC
4
8
SW
7
GND
6
SS
5
FB
Pin Functions
PIN
TYPE
(1)
DESCRIPTION
NAME
NO.
BOOT
1
O
Bootstrap capacitor connection for high-side MOSFET driver. Connect a high quality 0.1 μF
capacitor from BOOT to SW.
VIN
2
I
Connect to power supply and bypass capacitors CIN. Path from VIN pin to high frequency
bypass CIN and GND must be as short as possible.
EN
3
I
Enable pin, with internal pull-up current source. Pull below 1.2 V to disable. Float or connect
to VIN to enable. Adjust the input under voltage lockout with two resistors. See the Enable
and Adjusting Under voltage lockout section.
RT/SYNC
4
I
Resistor Timing or External Clock input. An internal amplifier holds this pin at a fixed voltage
when using an external resistor to ground to set the switching frequency. If the pin is pulled
above the PLL upper threshold, a mode change occurs and the pin becomes a
synchronization input. The internal amplifier is disabled and the pin is a high impedance clock
input to the internal PLL. If clocking edges stop, the internal amplifier is re-enabled and the
operating mode returns to frequency programming by resistor.
FB
5
I
Feedback input pin, connect to the feedback divider to set VOUT. Do not short this pin to
ground during operation.
SS
6
O
Soft-start control pin. Connect to a capacitor to set soft-start time.
GND
7
G
System ground pin.
SW
8
O
Switching output of the regulator. Internally connected to high-side power MOSFET. Connect
to power inductor.
Thermal Pad
9
G
Major heat dissipation path of the die. Must be connected to ground plane on PCB.
(1)
I = Input, O = Output, G = Ground
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SNVSA81A – FEBRUARY 2015 – REVISED APRIL 2015
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7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
Input Voltages
(1)
MIN
MAX
VIN, EN to GND
-0.3
44
BOOT to GND
-0.3
49
SS to GND
-0.3
5
FB to GND
-0.3
7
RT/SYNC to GND
-0.3
3.6
BOOT to SW
Output Voltages
UNIT
V
6.5
V
SW to GND
-3
44
TJ
Junction temperature
-40
150
°C
Tstg
Storage temperature
-65
150
°C
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
7.2 ESD Ratings
PARAMETER
V(ESD)
(1)
(2)
DEFINITION
VALUE
Human body model (HBM) (1)
Electrostatic
discharge
UNIT
2
Charged device model (CDM) (2)
kV
0.5
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VIN
VOUT
Buck Regulator
Control
Temperature
MAX
4
40
0.8
28
BOOT
45
SW
-1
FB
0
5
EN
0
40
RT/SYNC
0
3.3
SS
Frequency
MIN
UNIT
V
40
0
3
Switching frequency range at RT mode
200
2500
Switching frequency range at SYNC mode
250
2300
Operating junction temperature, TJ
-40
125
V
kHz
°C
7.4 Thermal Information
THERMAL METRIC (1)
DDA
8 PINS
RθJA
Junction-to-ambient thermal resistance
42.5
ψJT
Junction-to-top characterization parameter
9.9
ψJB
Junction-to-board characterization parameter
25.4
RθJC(top)
Junction-to-case (top) thermal resistance
56.1
RθJC(bot)
Junction-to-case (bottom) thermal resistance
3.8
RθJB
Junction-to-board thermal resistance
25.5
(1)
4
UNIT
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
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7.5 Electrical Characteristics
Limits apply over the recommended operating junction temperature (TJ) range of -40°C to +125°C, unless otherwise stated.
Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise specified, the following
conditions apply: VIN = 4.0 V to 40 V
PARAMETER
TEST CONDITION
MIN
Rising threshold
3.5
TYP
MAX
UNIT
40
V
3.7
3.9
V
POWER SUPPLY (VIN PIN)
VIN
Operation input voltage
UVLO
Under voltage lockout thresholds
4
Hysteresis
285
ISHDN
Shutdown supply current
VEN = 0 V, TA = 25°C, 4.0 V ≤ VIN ≤ 40 V
1.0
IQ
Operating quiescent current (nonswitching)
VFB = 1.0 V, TA = 25°C
40
mV
3.0
μA
μA
ENABLE (EN PIN)
VEN_TH
EN Threshold Voltage
IEN_PIN
EN PIN current
IEN_HYS
1.05
1.20
Enable threshold +50 mV
-4.6
Enable threshold -50 mV
-1.0
EN hysteresis current
1.38
V
μA
-3.6
μA
3
μA
EXTERNAL SOFT-START
ISS
SS pin current
TA = 25°C
VOLTAGE REFERENCE (FB PIN)
VFB
Feedback voltage
TJ = 25°C
0.744
0.750
0.756
V
TJ = -40°C to 125°C
0.735
0.750
0.765
V
90
180
mΩ
5.5
6.6
A
HIGH-SIDE MOSFET
RDS_ON
On-resistance
VIN = 12 V, BOOT to SW = 5.8 V
High-side MOSFET CURRENT LIMIT
ILIMT
Current limit
VIN = 12 V, TA = 25°C, Open Loop
4.4
THERMAL PERFORMANCE
TSHDN
Thermal shutdown threshold
170
THYS
Hysteresis
12
°C
7.6 Switching Characteristics
over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITION
RT = 49.9 kΩ, 1% accuracy
MIN
TYP
MAX
UNIT
400
500
600
kHz
fSW
Switching frequency
VSYNC_HI
SYNC clock high level threshold
VSYNC_LO
SYNC clock low level threshold
TSYNC_MIN
Minimum SYNC input pulse width
Measured at 500 kHz, VSYNC_HI > 3 V,
VSYNC_LO < 0.3 V
30
ns
TLOCK_IN
PLL lock in time
Measured at 500 kHz
100
µs
TON_MIN
Minimum controllable on time
VIN = 12 V, BOOT to SW = 5.8 V, ILoad =
1A
75
ns
DMAX
Maximum duty cycle
fSW = 200 kHz
97
%
1.7
0.5
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7.7 Typical Characteristics
100
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 500 kHz, L = 5.6 µH, COUT = 47 µF x 2, TA =
25°C.
60
50
40
30
60
50
40
30
20
20
VIN = 36 V
VIN = 24 V
VIN = 12 V
10
0
0.001
0.01
VOUT = 5 V
0.1
IOUT (A)
1
VIN = 36 V
VIN = 24 V
VIN = 12 V
10
0
0.001
10
fSW = 500 kHz
VOUT = 5 V
100
90
90
80
80
70
70
60
50
40
30
1
10
D003
fSW = 1 MHz
60
50
40
30
20
20
VIN = 24 V
VIN = 12 V
VIN = 5 V
10
0
0.001
0.01
VOUT = 3.3 V
0.1
IOUT (A)
1
VIN = 20 V
VIN = 12 V
VIN = 5 V
10
0
0.001
10
0.01
0.1
IOUT (A)
D009
fSW = 1 MHz
VOUT = 3.3 V
Figure 3. Efficiency vs. Load Current
1
10
D010
fSW = 2.2 MHz
Figure 4. Efficiency vs. Load Current
125
0.2
Nominal Switching Frequency (%)
VIN = 36 V
VIN = 24 V
VIN = 12 V
0.15
VOUT Deviation (%)
0.1
IOUT (A)
Figure 2. Efficiency vs. Load Current
100
Efficiency (%)
Efficiency (%)
Figure 1. Efficiency vs. Load Current
0.1
0.05
0
-0.05
0.001
VOUT = 5 V
VFB Falling
VFB Rising
100
75
50
25
0
0.01
0.1
IOUT (A)
1
fSW = 500 kHz
10
0
0.1
D004
VOUT = 5 V
Figure 5. Load Regulation
6
0.01
D002
0.2
0.3
0.4
VFB (V)
0.5
0.6
0.7
D005
fSW = 500 kHz
Figure 6. Frequency vs VFB
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Typical Characteristics (continued)
6
6
5.5
5.5
5
5
4.5
4.5
VOUT (V)
VOUT (V)
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 500 kHz, L = 5.6 µH, COUT = 47 µF x 2, TA =
25°C.
4
3.5
4
3.5
3
3
IOUT = 3.5 A
IOUT = 1.75 A
IOUT = 0.35 A
2.5
IOUT = 3.5 A
IOUT = 1.75 A
IOUT = 0.35 A
2.5
2
2
4
4.5
5
5.5
6
6.5
VIN (V)
VOUT = 5 V
4
4.5
5.5
6
6.5
VIN (V)
fSW = 500 kHz
VOUT = 5V
Figure 7. Dropout Curve
D012
fSW = 1 MHz
Figure 8. Dropout Curve
6
3.5
5.5
3.3
5
3.1
4.5
VOUT (V)
VOUT (V)
5
D011
4
2.9
2.7
3.5
2.5
3
IOUT = 3.5 A
IOUT = 1.75 A
IOUT = 0.35 A
2.5
2
4
4.5
5
5.5
6
VOUT = 5 V
2.1
3.5
6.5
VIN (V)
IOUT = 3.5 A
IOUT = 1.75 A
IOUT = 0.35 A
2.3
3.75
fSW = 2.2 MHz
VOUT = 3.3 V
Figure 9. Dropout Curve
4.25
VIN (V)
4.5
4.75
5
D014
fSW = 2.2 MHz
Figure 10. Dropout Curve
45
3.75
40
3.7
IQ
35
UVLO_H
3.65
30
UVLO (V)
IQ & ISHDN (µA)
4
D013
25
20
15
3.6
3.55
3.5
UVLO_L
10
ISHDN
5
3.45
0
0
5
10
15
20
25
VIN (V)
30
35
40
45
3.4
-50
-25
D006
0
25
50
75
Temperature (°C)
100
125
150
D007
IOUT = 0 A
Figure 11. Shut-down Current and Quiescent Current
Figure 12. UVLO Threshold
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8 Detailed Description
8.1 Overview
The LMR14030 SIMPLE SWITCHER® regulator is an easy to use step-down DC-DC converter that operates
from 4.0 V to 40 V supply voltage. It integrates a 90 mΩ (typical) high-side MOSFET, and is capable of delivering
up to 3.5 A DC load current with exceptional efficiency and thermal performance in a very small solution size.
The operating current is typically 40 μA under no load condition (not switching). When the device is disabled, the
supply current is typically 1 μA. An extended family is available in 2 A and 5 A load options in pin to pin
compatible packages.
The LMR14030 implements constant frequency peak current mode control with Sleep-mode at light load to
achieve high efficiency. The device is internally compensated, which reduces design time, and requires fewer
external components. The switching frequency is programmable from 200 kHz to 2.5 MHz by an external resistor
RT. The LMR14030 is also capable of synchronization to an external clock within the 250 kHz to 2.3 MHz
frequency range, which allows the device to be optimized to fit small board space at higher frequency, or high
efficient power conversion at lower frequency.
Other optional features are included for more comprehensive system requirements, including precision enable,
adjustable soft-start time, and approximate 97% duty cycle by BOOT capacitor recharge circuit. These features
provide a flexible and easy to use platform for a wide range of applications. Protection features include over
temperature shutdown, VOUT over voltage protection (OVP), VIN under-voltage lockout (UVLO), cycle-by-cycle
current limit, and short-circuit protection with frequency fold-back.
8.2 Functional Block Diagram
EN
VIN
Enable
Comparator
Thermal
Shutdown
UVLO
Shutdown
Shutdown
Logic
Voltage
Reference
Enable
Threshold
Boot
Charge
OV
Boot
UVLO
FB
ERROR
AMPLIFIER
Shutdown
PWM
Comparator
BOOT
PWM
Control
Logic
Comp
Components
6
Slope
Compensation
SW
Frequency
Shift
VIN
Oscillator
with PLL
SS
GND
8
Bootstrap
Control
RT/SYNC
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8.3 Feature Description
8.3.1 Fixed Frequency Peak Current Mode Control
The following operation description of the LMR14030 will refer to the Function Block Diagram and to the
waveforms in Figure 13. LMR14030 output voltage is regulated by turning on the high-side N-MOSFET with
controlled ON time. During high-side switch ON time, the SW pin voltage swings up to approximately VIN, and the
inductor current iL increase with linear slope (VIN – VOUT) / L. When high-side switch is off, inductor current
discharges through freewheel diode with a slope of –VOUT / L. The control parameter of Buck converter is defined
as Duty Cycle D = tON /TSW, where tON is the high-side switch ON time and TSW is the switching period. The
regulator control loop maintains a constant output voltage by adjusting the duty cycle D. In an ideal Buck
converter, where losses are ignored, D is proportional to the output voltage and inversely proportional to the input
voltage: D = VOUT / VIN.
VSW
SW Voltage
D = tON/ TSW
VIN
tON
tOFF
t
0
-VD
Inductor Current
iL
TSW
ILPK
IOUT
ûiL
t
0
Figure 13. SW Node and Inductor Current Waveforms in
Continuous Conduction Mode (CCM)
The LMR14030 employs fixed frequency peak current mode control. A voltage feedback loop is used to get
accurate DC voltage regulation by adjusting the peak current command based on voltage offset. The peak
inductor current is sensed from the high-side switch and compared to the peak current to control the ON time of
the high-side switch. The voltage feedback loop is internally compensated, which allows for fewer external
components, makes it easy to design, and provides stable operation with almost any combination of output
capacitors. The regulator operates with fixed switching frequency at normal load condition. At very light load, the
LMR14030 will operate in Sleep-mode to maintain high efficiency and the switching frequency will decrease with
reduced load current.
8.3.2 Slope Compensation
The LMR14030 adds a compensating ramp to the MOSFET switch current sense signal. This slope
compensation prevents sub-harmonic oscillations at duty cycle greater than 50%. The peak current limit of the
high-side switch is not affected by the slope compensation and remains constant over the full duty cycle range.
8.3.3 Sleep-mode
The LMR14030 operates in Sleep-mode at light load currents to improve efficiency by reducing switching and
gate drive losses. If the output voltage is within regulation and the peak switch current at the end of any
switching cycle is below the current threshold of 300 mA, the device enters Sleep-mode. The Sleep-mode current
threshold is the peak switch current level corresponding to a nominal internal COMP voltage of 400 mV.
When in Sleep-mode, the internal COMP voltage is clamped at 400mV and the high-side MOSFET is inhibited,
and the device draws only 40 μA (typical) input quiescent current. Since the device is not switching, the output
voltage begins to decay. The voltage control loop responds to the falling output voltage by increasing the internal
COMP voltage. The high-side MOSFET is enabled and switching resumes when the error amplifier lifts internal
COMP voltage above 400 mV. The output voltage recovers to the regulated value, and internal COMP voltage
eventually falls below the Sleep-mode threshold at which time the device again enters Sleep-mode.
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Feature Description (continued)
8.3.4 Low Dropout Operation and Bootstrap Voltage (BOOT)
The LMR14030 provides an integrated bootstrap voltage regulator. A small capacitor between the BOOT and
SW pins provides the gate drive voltage for the high-side MOSFET. The BOOT capacitor is refreshed when the
high-side MOSFET is off and the external low side diode conducts. The recommended value of the BOOT
capacitor is 0.1 μF. A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 16 V or
higher is recommended for stable performance over temperature and voltage.
When operating with a low voltage difference from input to output, the high-side MOSFET of the LMR14030 will
operate at approximate 97% duty cycle. When the high-side MOSFET is continuously on for 5 or 6 switching
cycles (5 or 6 switching cycles for frequency lower than 1 MHz, and 10 or 11 switching cycles for frequency
higher than 1 MHz) and the voltage from BOOT to SW drops below 3.2 V, the high-side MOSFET is turned off
and an integrated low side MOSFET pulls SW low to recharge the BOOT capacitor.
Since the gate drive current sourced from the BOOT capacitor is small, the high-side MOSFET can remain on for
many switching cycles before the MOSFET is turned off to refresh the capacitor. Thus the effective duty cycle of
the switching regulator can be high, approaching 97%. The effective duty cycle of the converter during dropout is
mainly influenced by the voltage drops across the power MOSFET, the inductor resistance, the low side diode
voltage and the printed circuit board resistance.
8.3.5 Adjustable Output Voltage
The internal voltage reference produces a precise 0.75 V (typical) voltage reference over the operating
temperature. The output voltage is set by a resistor divider from output voltage to the FB pin. It is recommended
to use 1% tolerance or better and temperature coefficient of 100 ppm or lower divider resistors. Select the low
side resistor RFBB for the desired divider current and use Equation 1 to calculate high-side RFBT. Larger value
divider resistors are good for efficiency at light load. However, if the values are too high, the regulator will be
more susceptible to noise and voltage errors from the FB input current may become noticeable. RFBB in the
range from 10 kΩ to 100 kΩ is recommended for most applications.
VOUT
RFBT
FB
RFBB
Figure 14. Output Voltage Setting
RFBT
VOUT 0.75
RFBB
0.75
(1)
8.3.6 Enable and Adjustable Under-voltage Lockout
The LMR14030 is enabled when the VIN pin voltage rises above 3.7 V (typical) and the EN pin voltage exceeds
the enable threshold of 1.2 V (typical). The LMR14030 is disabled when the VIN pin voltage falls below 3.52 V
(typical) or when the EN pin voltage is below 1.2 V. The EN pin has an internal pull-up current source (typically
IEN = 1 μA) that enables operation of the LMR14030 when the EN pin is floating.
Many applications will benefit from the employment of an enable divider RENT and RENB in Figure 15 to establish
a precision system UVLO level for the stage. System UVLO can be used for supplies operating from utility power
as well as battery power. It can be used for sequencing, ensuring reliable operation, or supply protection, such
as a battery. An external logic signal can also be used to drive EN input for system sequencing and protection.
When EN terminal voltage exceeds 1.2 V, an additional hysteresis current (typically IHYS = 3.6 μA) is sourced out
of EN terminal. When the EN terminal is pulled below 1.2 V, IHYS current is removed. This additional current
facilitates adjustable input voltage UVLO hysteresis. Use Equation 2 and Equation 3 to calculate RENT and RENB
for desired UVLO hysteresis voltage.
10
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Feature Description (continued)
IEN_HYS
IEN
VIN
VIN
RENT
VEN
EN
RENB
Figure 15. System UVLO By Enable Dividers
RENT
RENB
VSTART VSTOP
IHYS
(2)
VEN
VSTART VEN
IEN
RENT
(3)
where VSTART is the desired voltage threshold to enable LMR14030, VSTOP is the desired voltage threshold to
disable device.
8.3.7 External Soft-start
The LMR14030 has soft-start pin for programmable output ramp up time. The soft-start feature is used to prevent
inrush current impacting the LMR14030 and its load when power is first applied. The soft-start time can be
programed by connecting an external capacitor CSS from SS pin to GND. An internal current source (typically ISS
= 3 μA) charges CSS and generates a ramp from 0 V to VREF. The soft-start time can be calculated by
Equation 4:
CSS (nF) u VREF (V)
tSS (ms)
ISS (PA)
(4)
The internal soft-start resets while device is disabled or in thermal shutdown.
8.3.8 Switching Frequency and Synchronization (RT/SYNC)
The switching frequency of the LMR14030 can be programmed by the resistor RT from the RT/SYNC pin and
GND pin. The RT/SYNC pin can’t be left floating or shorted to ground. To determine the timing resistance for a
given switching frequency, use Equation 5 or the curve in Figure 16. Table 1 gives typical RT values for a given
fSW.
RT (k:)
32537 u ¦SW N+] 1.045
(5)
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Feature Description (continued)
140
120
RT (k:)
100
80
60
40
20
0
0
500
1000
1500
Frequency (kHz)
2000
2500
D008
Figure 16. RT vs Frequency Curve
Table 1. Typical Frequency Setting RT Resistance
fSW (kHz)
RT (kΩ)
200
127
350
71.5
500
49.9
750
32.4
1000
23.7
1500
15.8
2000
11.5
2200
10.5
The LMR14030 switching action can also be synchronized to an external clock from 250 kHz to 2.3 MHz.
Connect a square wave to the RT/SYNC pin through either circuit network shown in Figure 17. Internal oscillator
is synchronized by the falling edge of external clock. The recommendations for the external clock include: high
level no lower than 1.7 V, low level no higher than 0.5 V and have a pulse width greater than 30 ns. When using
a low impedance signal source, the frequency setting resistor RT is connected in parallel with an AC coupling
capacitor CCOUP to a termination resistor RTERM (e.g., 50 Ω). The two resistors in series provide the default
frequency setting resistance when the signal source is turned off. A 10 pF ceramic capacitor can be used for
CCOUP. Figure 18, Figure 19 and Figure 20 show the device synchronized to an external system clock.
CCOUP
PLL
PLL
Lo-Z
Clock
Source
RT
RT/SYNC
RTERM
Hi-Z
Clock
Source
RT/SYNC
RT
Figure 17. Synchronizing to an External Clock
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SYNC (2 V/DIV)
SYNC (2 V/DIV)
SW (5 V/DIV)
SW (5 V/DIV)
iL (500 mA/DIV)
iL (1 A/DIV)
Time (2 µs/DIV)
Time (2 µs/DIV)
Figure 18. Synchronizing in CCM
Figure 19. Synchronizing in DCM
SYNC (2 V/DIV)
SW (5 V/DIV)
iL (500 mA/DIV)
Time (10 µs/DIV)
Figure 20. Synchronizing in Sleep-mode Mode
Equation 6 calculates the maximum switching frequency limitation set by the minimum controllable on time and
the input to output step down ratio. Setting the switching frequency above this value will cause the regulator to
skip switching pulses to achieve the low duty cycle required at maximum input voltage.
¦SW(max)
§ IOUT u RIND VOUT VD ·
u¨
¸
tON ¨© VIN_MAX IOUT u RDS_ON VD ¸¹
1
(6)
where
• IOUT = Output current
• RIND = Inductor series resistance
• VIN_MAX = Maximum input voltage
• VOUT = Output voltage
• VD = Diode voltage drop
• RDS_ON = High-side MOSFET switch on resistance
• tON = Minimum on time
8.3.9 Over Current and Short Circuit Protection
The LMR14030 is protected from over current condition by cycle-by-cycle current limiting on the peak current of
the high-side MOSFET. High-side MOSFET over-current protection is implemented by the nature of the Peak
Current Mode control. The high-side switch current is compared to the output of the Error Amplifier (EA) minus
slope compensation every switching cycle. Please refer to Functional Block Diagram for more details. The peak
current of high-side switch is limited by a clamped maximum peak current threshold which is constant. So the
peak current limit of the high-side switch is not affected by the slope compensation and remains constant over
the full duty cycle range.
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The LMR14030 also implements a frequency fold-back to protect the converter in severe over-current or short
conditions. The oscillator frequency is divided by 2, 4, and 8 as the FB pin voltage decrease to 75%, 50%, 25%
of VREF. The frequency fold-back increases the off time by increasing the period of the switching cycle, so that it
provides more time for the inductor current to ramp down and leads to a lower average inductor current. Lower
frequency also means lower switching loss. Frequency fold-back reduces power dissipation and prevents
overheating and potential damage to the device.
8.3.10 Overvoltage Protection
The LMR14030 employs an output overvoltage protection (OVP) circuit to minimize voltage overshoot when
recovering from output fault conditions or strong unload transients in designs with low output capacitance. The
OVP feature minimizes output overshoot by turning off high-side switch immediately when FB voltage reaches to
the rising OVP threshold which is nominally 109% of the internal voltage reference VREF. When the FB voltage
drops below the falling OVP threshold which is nominally 107% of VREF, the high-side MOSFET resumes normal
operation.
8.3.11 Thermal Shutdown
The LMR14030 provides an internal thermal shutdown to protect the device when the junction temperature
exceeds 170°C (typical). The high-side MOSFET stops switching when thermal shundown activates. Once the
die temperature falls below 158°C (typical), the device reinitiates the power up sequence controlled by the
internal soft-start circuitry.
14
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The LMR14030 is a step down DC-to-DC regulator. It is typically used to convert a higher DC voltage to a lower
DC voltage with a maximum output current of 3.5 A. The following design procedure can be used to select
components for the LMR14030. This section presents a simplified discussion of the design process.
9.2 Typical Application
The LMR14030 only requires a few external components to convert from wide voltage range supply to a fixed
output voltage. A schematic of 5 V/3.5 A application circuit is shown in Figure 21. The external components have
to fulfill the needs of the application, but also the stability criteria of the device’s control loop.
7 V to 36 V
VIN
CIN
CBOOT
BOOT
L
EN
5 V / 3.5 A
SW
COUT
D
RFBT
RT/SYNC
FB
RFBB
SS
RT
GND
CSS
Figure 21. Application Circuit, 5V Output
9.2.1 Design Requirements
This example details the design of a high frequency switching regulator using ceramic output capacitors. A few
parameters must be known in order to start the design process. These parameters are typically determined at the
system level:
Input Voltage, VIN
7 V to 36 V, Typical 12 V
Output Voltage, VOUT
5.0 V
Maximum Output Current IO_MAX
3.5 A
Transient Response 0.35 A to 3.5 A
5%
Output Voltage Ripple
50 mV
Input Voltage Ripple
400 mV
Switching Frequency fSW
500 kHz
Soft-start time
5 ms
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9.2.2 Detailed Design Procedure
9.2.2.1
Output Voltage Set-Point
The output voltage of LMR14030 is externally adjustable using a resistor divider network. The divider network is
comprised of top feedback resistor RFBT and bottom feedback resistor RFBB. Equation 7 is used to determine the
output voltage:
VOUT 0.75
RFBT
RFBB
0.75
(7)
Choose the value of RFBT to be 100 kΩ. With the desired output voltage set to 5 V and the VFB = 0.75 V, the RFBB
value can then be calculated using Equation 7. The formula yields to a value 17.65 kΩ. Choose the closest
available value of 17.8 kΩ for RFBB.
9.2.2.2
Switching Frequency
For desired frequency, use Equation 8 to calculate the required value for RT.
RT (k:)
32537 u ¦SW N+] 1.045
(8)
For 500 kHz, the calculated RT is 49.2 kΩ and standard value 49.9 kΩ can be used to set the switching
frequency at 500 kHz.
9.2.2.3
Output Inductor Selection
The most critical parameters for the inductor are the inductance, saturation current and the RMS current. The
inductance is based on the desired peak-to-peak ripple current ΔiL. Since the ripple current increases with the
input voltage, the maximum input voltage is always used to calculate the minimum inductance LMIN. Use
Equation 10 to calculate the minimum value of the output inductor. KIND is a coefficient that represents the
amount of inductor ripple current relative to the maximum output current. A reasonable value of KIND should be
20%-40%. During an instantaneous short or over current operation event, the RMS and peak inductor current
can be high. The inductor current rating should be higher than current limit.
VOUT u (VIN_MAX VOUT )
'iL
VIN _ MAX u L u ¦SW
(9)
LMIN
VIN_MAX VOUT
IOUT u KIND
u
VOUT
VIN_MAX u ¦SW
(10)
In general, it is preferable to choose lower inductance in switching power supplies, because it usually
corresponds to faster transient response, smaller DCR, and reduced size for more compact designs. But too low
of an inductance can generate too large of an inductor current ripple such that over current protection at the full
load could be falsely trigged. It also generates more conduction loss since the RMS current is slightly higher.
Larger inductor current ripple also implies larger output voltage ripple with same output capacitors. With peak
current mode control, it is not recommended to have too small of an inductor current ripple. A larger peak current
ripple improves the comparator signal to noise ratio.
For this design example, choose KIND = 0.4, the minimum inductor value is calculated to be 6.12 µH, and a
nearest standard value is chosen: 6.5 µH. A standard 6.5 μH ferrite inductor with a capability of 5 A RMS current
and 7A saturation current can be used.
9.2.2.4
Output Capacitor Selection
The output capacitor(s), COUT, should be chosen with care since it directly affects the steady state output voltage
ripple, loop stability and the voltage over/undershoot during load current transients.
The output ripple is essentially composed of two parts. One is caused by the inductor current ripple going
through the Equivalent Series Resistance (ESR) of the output capacitors:
'VOUT_ESR 'iL u ESR KIND u IOUT u ESR
(11)
The other is caused by the inductor current ripple charging and discharging the output capacitors:
KIND u IOUT
'iL
'VOUT_C
8 u ¦SW u COUT 8 u ¦SW u COUT
16
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The two components in the voltage ripple are not in phase, so the actual peak-to-peak ripple is smaller than the
sum of two peaks.
Output capacitance is usually limited by transient performance specifications if the system requires tight voltage
regulation with presence of large current steps and fast slew rate. When a fast large load increase happens,
output capacitors provide the required charge before the inductor current can slew up to the appropriate level.
The regulator’s control loop usually needs three or more clock cycles to respond to the output voltage droop. The
output capacitance must be large enough to supply the current difference for three clock cycles to maintain the
output voltage within the specified range. Equation 13
Equation 13 shows the minimum output capacitance needed for specified output undershoot. When a sudden
large load decrease happens, the output capacitors absorb energy stored in the inductor. The catch diode can’t
sink current so the energy stored in the inductor results in an output voltage overshoot. Equation 14 calculates
the minimum capacitance required to keep the voltage overshoot within a specified range.
3 u (IOH IOL )
COUT !
¦SW u 9US
(13)
COUT !
2
2
IOH
IOL
2
(VOUT VOS )2 VOUT
uL
(14)
where
• KIND = Ripple ratio of the inductor ripple current (ΔiL / IOUT)
• IOL = Low level output current during load transient
• IOH = High level output current during load transient
• VUS = Target output voltage undershoot
• VOS = Target output voltage overshoot
For this design example, the target output ripple is 50 mV. Presuppose ΔVOUT_ESR = ΔVOUT_C = 50 mV, and
chose KIND = 0.4. Equation 11 yields ESR no larger than 35.7 mΩ and Equation 12 yields COUT no smaller than 7
μF. For the target over/undershoot range of this design, VUS = VOS = 5% × VOUT = 250 mV. The COUT can be
calculated to be no smaller than 75.6 μF and 30.8 μF by Equation 13 and Equation 14 respectively. In summary,
the most stringent criteria for the output capacitor is 75.6 μF. Two 47 μF, 16 V, X7R ceramic capacitors with 5
mΩ ESR are used in parallel.
9.2.2.5
Schottky Diode Selection
The breakdown voltage rating of the diode is preferred to be 25% higher than the maximum input voltage. The
current rating for the diode should be equal to the maximum output current for best reliability in most
applications. In cases where the input voltage is much greater than the output voltage the average diode current
is lower. In this case it is possible to use a diode with a lower average current rating, approximately (1-D) × IOUT
however the peak current rating should be higher than the maximum load current. A 4 A to 5 A rated diode is a
good starting point.
9.2.2.6
Input Capacitor Selection
The LMR14030 device requires high frequency input decoupling capacitor(s) and a bulk input capacitor,
depending on the application. The typical recommended value for the high frequency decoupling capacitor is 4.7
μF to 10 μF. A high-quality ceramic capacitor type X5R or X7R with sufficiency voltage rating is recommended.
To compensate the derating of ceramic capacitors, a voltage rating of twice the maximum input voltage is
recommended. Additionally, some bulk capacitance can be required, especially if the LMR14030 circuit is not
located within approximately 5 cm from the input voltage source. This capacitor is used to provide damping to the
voltage spike due to the lead inductance of the cable or the trace. For this design, two 2.2 μF, X7R ceramic
capacitors rated for 100 V are used. A 0.1 μF for high-frequency filtering and place it as close as possible to the
device pins.
9.2.2.7
Bootstrap Capacitor Selection
Every LMR14030 design requires a bootstrap capacitor (CBOOT). The recommended capacitor is 0.1 μF and rated
16 V or higher. The bootstrap capacitor is located between the SW pin and the BOOT pin. The bootstrap
capacitor must be a high-quality ceramic type with an X7R or X5R grade dielectric for temperature stability.
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9.2.2.8 Soft-start Capacitor Selection
Use Equation 15 in order to calculate the soft-start capacitor value:
t (ms) u ISS (PA)
CSS (nF) SS
VREF (V)
(15)
where
• CSS = Soft-start capacitor value
• ISS = Soft-start charging current (3 μA)
• tSS = Desired soft-start time
For the desired soft-start time of 5 ms and soft-start charging current of 3.0 μA, Equation 15 yields a soft-start
capacitor value of 20 nF, a standard 22 nF ceramic capacitor is used.
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9.2.3 Application Curves
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 500 kHz, L = 5.6 µH, COUT = 47 µF x 2, TA =
25°C.
VIN (5 V/DIV)
VIN (5 V/DIV)
EN (1 V/DIV)
VOUT (1 V/DIV)
VOUT (1 V/DIV)
iL (2 A/DIV)
Time (2 ms/DIV)
VIN = 12 V
VOUT = 5 V
Time (2 ms/DIV)
IOUT = 2 A
VIN = 12 V
Figure 22. Start-up By EN
VOUT = 5 V
IOUT = 2 A
Figure 23. Start-up By VIN
SW (5 V/DIV)
SW (5 V/DIV)
iL (200 mA/DIV)
iL (200 mA/DIV)
VOUT(ac) (10 mV/DIV)
VOUT(ac) (10 mV/DIV)
Time (2 ms/DIV)
VIN = 12 V
VOUT = 5 V
Time (2 µs/DIV)
IOUT = 0 A
VIN = 12 V
VOUT = 5 V
Figure 24. Sleep-mode
IOUT = 100 mA
Figure 25. DCM Mode
SW (5 V/DIV)
VOUT(ac) (200 mV/DIV)
iL (500 mA/DIV)
IOUT (1 A/DIV)
VOUT(ac) (10 mV/DIV)
Time (2 µs/DIV)
VIN = 12 V
VOUT = 5 V
Time (100 µs/DIV)
IOUT = 1.5 A
IOUT: 20% → 80%
of 3.5 A
Slew rate = 100
mA/μs
Figure 26. CCM Mode
Figure 27. Load Transient
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Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 500 kHz, L = 5.6 µH, COUT = 47 µF x 2, TA =
25°C.
VOUT (1 V/DIV)
VOUT (1 V/DIV)
iL (2 A/DIV)
iL (2 A/DIV)
Time (40 µs/DIV)
VIN = 12 V
VOUT = 5 V
Time (2 ms/DIV)
VIN = 12 V
Figure 28. Output Short
20
VOUT = 5 V
Figure 29. Output Short Recovery
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10 Power Supply Recommendations
The LMR14030 is designed to operate from an input voltage supply range between 4 V and 40 V. This input
supply should be able to withstand the maximum input current and maintain a stable voltage. The resistance of
the input supply rail should be low enough that an input current transient does not cause a high enough drop at
the LMR14030 supply voltage that can cause a false UVLO fault triggering and system reset. If the input supply
is located more than a few inches from the LMR14030, additional bulk capacitance may be required in addition to
the ceramic input capacitors. The amount of bulk capacitance is not critical, but a 47 μF or 100 μF electrolytic
capacitor is a typical choice.
11 Layout
11.1 Layout Guidelines
Layout is a critical portion of good power supply design. The following guidelines will help users design a PCB
with the best power conversion performance, thermal performance, and minimized generation of unwanted EMI.
1. The feedback network, resistor RFBT and RFBB, should be kept close to the FB pin. VOUT sense path away
from noisy nodes and preferably through a layer on the other side of a shielding layer.
2. The input bypass capacitor CIN must be placed as close as possible to the VIN pin and ground. Grounding
for both the input and output capacitors should consist of localized top side planes that connect to the GND
pin and PAD.
3. The inductor L should be placed close to the SW pin to reduce magnetic and electrostatic noise.
4. The output capacitor, COUT should be placed close to the junction of L and the diode D. The L, D, and COUT
trace should be as short as possible to reduce conducted and radiated noise and increase overall efficiency.
5. The ground connection for the diode, CIN, and COUT should be as small as possible and tied to the system
ground plane in only one spot (preferably at the COUT ground point) to minimize conducted noise in the
system ground plane
6. For more detail on switching power supply layout considerations see Application Note AN-1149
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11.2 Layout Example
Output Bypass
Capacitor
Output
Inductor
Rectifier Diode
BOOT
Capacitor
Input Bypass
Capacitor
BOOT
UVLO Adjust
Resistor
SW
VIN
GND
EN
SS
RT/SYNC
FB
Soft-Start
Capacitor
Output Voltage
Set Resistor
Frequency
Set Resistor
Thermal VIA
Signal VIA
Figure 30. Layout
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12 Device and Documentation Support
12.1 Trademarks
PowerPAD is a trademark of Texas Instruments.
SIMPLE SWITCHER is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.2 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.3 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LMR14030SDDA
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
DB3SP
LMR14030SDDAR
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
DB3SP
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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