LP3906
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SNVS456M – AUGUST 2006 – REVISED MAY 2013
Dual High-Current Step-Down DC/DC and Dual Linear Regulator with I2C Compatible
Interface
Check for Samples: LP3906
FEATURES
KEY SPECIFICATIONS
•
•
1
2
•
•
•
•
•
•
•
Compatible with Advanced Applications
Processors and FPGAs
2 LDOs for Powering Internal Processor
Functions and I/Os
High Speed Serial Interface for Independent
Control of Device Functions and Settings
Precision Internal Reference
Thermal Overload Protection
Current Overload Protection
24-lead 5 × 4 × 0.8 mm WQFN Package
Software Programmable Regulators
•
APPLICATIONS
•
•
•
FPGA, DSP Core Power
Applications Processors
Peripheral I/O Power
DC/DC Converter (Buck)
– 1.5A Output Current
– Programmable VOUT from:
– Buck1 : 0.8V–2.0V
– Buck2 : 1.0V–3.5V
– Up to 96% Efficiency
– 2 MHz PWM Switching Frequency
– ±3% Output Voltage Accuracy
– Automatic Soft Start
Linear Regulators (LDO)
– Programmable VOUT of 1.0V–3.5V
– ±3% Output Voltage Accuracy
– 300 mA Output Currents
– 25 mW (Typ) Dropout
DESCRIPTION
The LP3906 is a multi-function, programmable Power
Management Unit, optimized for low power FPGAs,
Microprocessors and DSPs.
Typical Application Circuits
VINLDO12
EN_T
1 PF
ENLDO1
SYNC
ENLDO2
VIN1
10 PF
ENSW1
2.2 PH
ENSW2
SW1
LDO1
FB1
10 PF
0.47 PF
GND_SW1
VINLDO1
LP3906
1 PF
VIN2
VINLDO2
1 PF
10 PF
LDO2
2.2 PH
0.47 PF
SW2
SDA
FB2
10 PF
SCL
GND_SW2
GND_L
GND_C
DAP
1
2
AVDD
1 PF
Figure 1. Typical Application Circuit
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2006–2013, Texas Instruments Incorporated
LP3906
SNVS456M – AUGUST 2006 – REVISED MAY 2013
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For information about how schottky diodes can reduce noise in high load, high Vin applications, see Buck Output
Ripple Management.
DESCRIPTION (CONTINUED)
This device integrates two highly efficient 1.5A Step-Down DC/DC converters with dynamic voltage management
(DVM), two 300mA Linear Regulators and a 400kHz I2C compatible interface to allow a host controller access to
the internal control registers of the LP3906. The LP3906 additionally features programmable power-on
sequencing and is offered in a tiny 5 x 4 x 0.8mm WQFN-24 pin package.
2
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DC SOURCE
4.5 ± 5.5V
+
Li-ion/polymer cell 3.6V - 4.2V
FPGA
12
1 PF
13
VIN1
10 PF
VIN2
19
1 PF
VINLDO1
20
1 PF
AVDD
1 PF
VINLDO2
VINLDO12
LP3906 PMIC
Cvdd
4.7 PF
1
10 PF
SYNC
23
7
Clock
divider
OSC
6
BUCK1
AVDD
10
EN_T
Lsw1 2.2 PH 1.2V
VBUCK1
SW1
*FB1
Csw1
10 PF
CPU
CORE
11
15
ENLDO1
16
Power
ON-OFF
Logic
ENLDO2
22
BUCK2
AVDD
ENSW1
Lsw2 2.2 PH 3.3V
2
VBUCK2
Csw2
SW2
10 PF
FB2
I/O
*
21
24
vref
ENSW2
VINLDO1
VinLDO12
LDO1
Thermal
Shutdown
RESET
9
I2C
8
LDO2
18
1.8V
Cldo2
0.47 PF
Power On
Reset
Logic
Control
and
registers
5
Cldo1
0.47 PF
LDO2
SDA
GND_SW1
14
Programmable
Application
Processor
VINLDO2
BIAS
SCL
LDO1
3.3V
3
GND_SW2
4
GND_C
17
GND_L
VBUCK2
3.3V
Flash
Figure 2. Typical Application Circuit
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Connection Diagram
19
18
17
16
15
14
13
20
12
21
11
22
10
23
9
24
8
1
2
3
4
5
6
7
Figure 3. 24-Lead WQFN Package (Top View)
See Package Number NHZ0024B
Table 1. Default Voltage Options (1) (2)
Part Number
Package
Marking
Buck1
Buck2
LDO1
LDO2
LP3906SQ-DJXI/NOPB
06-DJXI
0.9V
1.8V
3.3V
1.8V
LP3906SQ-FXPI/NOPB
06-FXPI
1.0V
3.3V
2.5V
1.8V
LP3906SQ-JXXI/NOPB
06-JXXI
1.2V
3.3V
3.3V
1.8V
LP3906SQ-PPXP/NOPB
06-PPXP
1.5V
2.5V
3.3V
2.5V
LP3906SQ-TKXII/NOPB (3)
06-TKXII (3)
1.25V
3.3V
1.8V
1.8V
LP3906SQ-VPFP/NOPB
06-VPFP
1.8V
2.5V
1.5V
2.5V
LP3906SQE-PPXP/NOPB
06-PPXP
1.5V
2.5V
3.3V
2.5V
LP3906SQE-VPFP/NOPB
06-VPFP
1.8V
2.5V
1.5V
2.5V
LP3906SQX-DJXI/NOPB
06-DJXI
0.9V
1.8V
3.3V
1.8V
LP3906SQX-FXPI/NOPB
06-FXPI
1.0V
3.3V
2.5V
1.8V
LP3906SQX-JXXI/NOPB
06-JXXI
1.2V
3.3V
3.3V
1.8V
LP3906SQX-PPXP/NOPB
06-PPXP
1.5V
2.5V
3.3V
2.5V
LP3906SQX-TKXII/NOPB (3)
06-TKXII (3)
1.25V
3.3V
1.8V
1.8V
LP3906SQX-VPFP/NOPB
06-VPFP
1.8V
2.5V
1.5V
2.5V
(1)
(2)
(3)
4
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
Package drawings, thermal data, and symbolization are available at www.ti.com/packaging.
Option only available with default EN_T of 001; all other options use 010.
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Pin Descriptions (1)
(1)
Pin Name
Pin No.
I/O
Type
VIN2
SW2
Description
1
I
PWR
Power in from either DC source or Battery to Buck 2
2
O
PWR
Buck2 switcher output pin
GND_SW2
3
G
G
Buck2 NMOS Power Ground
GND_C
4
G
G
Non switching core ground pin
GND_SW1
5
G
G
Buck1 NMOS Power Ground
SW1
6
O
PWR
Buck1 switcher output pin
VIN1
7
I
PWR
Power in from either DC source or Battery to buck 1
SDA
8
I/O
D
I2C Data (Bidirectional)
SCL
9
I
D
I2C Clock
EN_T
10
I
D
Enable for preset power on sequence.
Buck1 input feedback terminal
FB1
11
I
A
AVDD
12
I
PWR
Analog Power for Buck converters.
VINLDO1
13
I
PWR
Power in from either DC source or Battery to input terminal of LDO1
LDO1 Output
LDO1
14
O
PWR
ENLDO1
15
I
D
LDO1 enable pin, a logic HIGH enables the LDO1
ENLDO2
16
I
D
LDO2 enable pin, a logic HIGH enables the LDO2
GND_L
17
G
G
LDO Ground
LDO2
18
O
PWR
LDO2 Output
VINLDO2
19
I
PWR
Power in from either DC source or battery to input terminal to LDO2
VinLDO12
20
I
PWR
Analog Power for Internal Functions (VREF, BIAS, I2C, Logic)
FB2
21
I
A
Buck2 input feedback terminal
ENSW1
22
I
D
Enable Pin for Buck1 switcher, a logic HIGH enables Buck1
SYNC
23
I
D
Frequency Synchronization pin which allows the user to connect an external
clock signal PLL to synchronize the PMIC internal oscillator.
Enable Pin for Buck2 switcher, a logic HIGH enables Buck2
ENSW2
24
I
D
DAP
DAP
GND
GND
Connection isn't necessary for electrical performance, but it is recommended
for better thermal dissipation.
A: Analog Pin D: Digital Pin G: Ground Pin PWR: Power Pin I: Input Pin I/O: Input/Output Pin O: Output Pin
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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ABSOLUTE MAXIMUM RATINGS (1) (2) (3)
−0.3V to +6V
VIN, SDA, SCL
GND to GND SLUG
±0.3V
Power Dissipation (PD_MAX)
(TA=85°C, TMAX=125°C, ) (4)
1.43 W
Junction Temperature (TJ-MAX)
150°C
−65°C to +150°C
Storage Temperature Range
Maximum Lead Temperature (Soldering)
ESD Ratings
Human Body Model
(1)
(2)
(3)
(4)
(5)
6
260°C
(5)
2 kV
Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Operating Ratings are conditions under
which operation of the device is specified. Operating Ratings do not imply specified performance limits. For specified performance limits
and associated test conditions, see the Electrical Characteristics.
All voltages are with respect to the potential at the GND pin.
If Military/Aerospace specified devices are required, please contact the TI Sales Office/Distributors for availability and specifications.
In applications where high power dissipation and/or poor package thermal resistance is present, the maximum ambient temperature may
have to be derated. Maximum ambient temperature (TA-MAX) is dependent on the maximum operating junction temperature (TJ-MAX-OP =
125°C), the maximum power dissipation of the device in the application (PD-MAX), and the junction-to-ambient thermal resistance of the
part/package in the application (θJA), as given by the following equation: TA-MAX = TJ-MAX-OP − (θJA × PD-MAX). See APPLICATION
NOTES.
The Human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. (MILSTD - 883 3015.7)
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OPERATING RATINGS (1) (2) (3)
Bucks
VIN
2.7V to 5.5V
VEN
0 to (VIN + 0.3V)
−40°C to +125°C
Junction Temperature (TJ) Range
Ambient Temperature (TA) Range
Thermal Properties (5) (6)
(4)
−40°C to +85°C
(4)
Junction-to-Ambient Thermal Resistance
(θJA)NHZ0024B
(1)
(2)
(3)
(4)
(5)
(6)
28°C/W
Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Operating Ratings are conditions under
which operation of the device is specified. Operating Ratings do not imply specified performance limits. For specified performance limits
and associated test conditions, see the Electrical Characteristics.
All voltages are with respect to the potential at the GND pin.
Min and Max limits are ensured by design, test, or statistical analysis. Typical numbers are not specifications, but do represent the most
likely norm.
Junction-to-ambient thermal resistance is highly application and board-layout dependent. In applications where high maximum power
dissipation exists, special care must be paid to thermal dissipation issues in board design.
Internal thermal shutdown circuitry protects the device from permanent damage. Thermal shutdown engages at TJ = 160°C (typ.) and
disengages at TJ = 140°C (typ.)
In applications where high power dissipation and/or poor package thermal resistance is present, the maximum ambient temperature may
have to be derated. Maximum ambient temperature (TA-MAX) is dependent on the maximum operating junction temperature (TJ-MAX-OP =
125°C), the maximum power dissipation of the device in the application (PD-MAX), and the junction-to-ambient thermal resistance of the
part/package in the application (θJA), as given by the following equation: TA-MAX = TJ-MAX-OP − (θJA × PD-MAX). See APPLICATION
NOTES.
GENERAL ELECTRICAL CHARACTERISTICS (1) (2) (3) (4)
Unless otherwise noted, VIN = 3.6V, Typical values and limits appearing in normal type apply for TJ = 25°C. Limits appearing
in boldface type apply over the entire junction temperature range for operation, −40°C to +125°C.
Parameter
VPOR
Power-On Reset Threshold
TSD
TSDH
(1)
(2)
(3)
(4)
Test Conditions
Min
VDD Falling Edge
Typ
Max
Units
1.9
V
Thermal Shutdown Threshold
160
°C
Themal Shutdown Hysteresis
20
°C
Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Operating Ratings are conditions under
which operation of the device is specified. Operating Ratings do not imply specified performance limits. For specified performance limits
and associated test conditions, see the Electrical Characteristics.
All voltages are with respect to the potential at the GND pin.
Min and Max limits are specified by design, test, or statistical analysis. Typical numbers are not specified, but do represent the most
likely norm.
This specification is ensured by design.
I2C COMPATIBLE INTERFACE ELECTRICAL SPECIFICATIONS (1)
Unless otherwise noted, VIN = 3.6V. Typical values and limits appearing in normal type apply for TJ = 25°C. Limits appearing
in boldface type apply over the entire junction temperature range for operation, −40°C to +125°C
Parameter
FCLK
Clock Frequency
tBF
Bus-Free Time Between Start and Stop
Test Conditions
Min
Typ
Max
Units
400
kHz
See (1)
1.3
µs
(1)
tHOLD
Hold Time Repeated Start Condition
See
0.6
µs
tCLKLP
CLK Low Period
See (1)
1.3
µs
tCLKHP
CLK High Period
See (1)
0.6
µs
(1)
tSU
Set Up Time Repeated Start Condition
See
0.6
µs
tDATAHLD
Data Hold time
See (1)
0
µs
tDATASU
Data Set Up Time
See (1)
100
ns
TSU
Set Up Time for Start Condition
See (1)
0.6
µs
TTRANS
Maximum Pulse Width of Spikes that
Must be Suppressed by the Input Filter of
Both DATA & CLK Signals.
See (1)
(1)
50
ns
This specification is ensured by design.
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LOW DROP OUT REGULATORS, LDO1 AND LDO2
Unless otherwise noted, VIN = 3.6, CIN = 1.0 µF, COUT = 0.47 µF. Typical values and limits appearing in normal type apply for
TJ = 25°C. Limits appearing in boldface type apply over the entire junction temperature range for operation, −40°C to
+125°C. (1) (2) (3) (4) (5) (6) (7)
Parameter
Test Conditions
Min
Typ
Max
Units
1.74
5.5
V
−3
3
%
0.15
%/V
0.011
%/mA
VIN
Operational Voltage Range
VINLDO1 and VINLDO2 PMOS
pins (8)
VOUT Accuracy
Output Voltage Accuracy (Default VOUT)
Load current = 1 mA
ΔVOUT
Line Regulation
VIN = (VOUT + 0.3V) to 5.0V,
(7)
, Load Current = 1 mA
Load Regulation
VIN = 3.6V,
Load Current = 1 mA to IMAX
ISC
Short Circuit Current Limit
LDO1-2, VOUT = 0V
500
VIN – VOUT
Dropout Voltage
Load Current = 50 mA (5)
25
PSRR
Power Supply Ripple Rejection
F = 10 kHz, Load Current = IMAX
45
dB
θn
Supply Output Noise
10 Hz < F < 100 KHz
80
µVrms
Quiescent Current “On”
IOUT = 0 mA
40
µA
Quiescent Current “On”
IOUT = IMAX
60
µA
0.03
µA
300
µs
µF
IQ (6)
(9)
(10)
Quiescent Current “Off”
EN is de-asserted
TON
Turn On Time
Start up from shut-down
COUT
Output Capacitor
Capacitance for stability 0°C ≤ TJ
≤ 125°C
0.33
0.47
−40°C ≤ TJ ≤ 125°C
0.68
1.0
ESR
5
mA
200
mV
µF
500
mΩ
(1)
(2)
All voltages are with respect to the potential at the GND pin.
Min and Max limits are ensured by design, test, or statistical analysis. Typical numbers are not specifications, but do represent the most
likely norm.
(3) CIN, COUT: Low-ESR Surface-Mount Ceramic Capacitors (MLCCs) used in setting electrical characteristics.
(4) The device maintains a stable, regulated output voltage without a load.
(5) Dropout voltage is the voltage difference between the input and the output at which the output voltage drops to 100 mV below its
nominal value.
(6) Quiescent current is defined here as the difference in current between the input voltage source and the load at VOUT.
(7) VIN minimum for line regulation values is 1.8V.
(8) Pins 13, 19 can operate from Vin min of 1.74 to a Vin max of 5.5V this rating is only for the series pass pmos power fet. It allows the
system design to use a lower voltage rating if the input voltage comes from a buck output.
(9) The Iq can be defined as the standing current of the LP3906 when the I2C bus is active and all other power blocks have been disabled
via the I2C bus, or it can be defined as the I2C bus active, and the other power blocks are active under no load condition. These two
values can be used by the system designer when the LP3906 is powered using a battery. If the user plans to use the HW enable pins to
disable each block of the IC please contact the factory applications for IQ details.
(10) The Iq exhibits a higher current draw when the EN pin is de-asserted because the I2C buffer pins draw an additional 2µA
8
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BUCK CONVERTERS SW1, SW2
Unless otherwise noted, VIN = 3.6, CIN = 10 µF, COUT = 10 µF, LOUT = 2.2 µH ceramic. Typical values and limits appearing in
normal type apply for TJ = 25°C. Limits appearing in boldface type apply over the entire junction temperature range for
operation, −40°C to +125°C. (1) (2) (3) (4) (5)
Parameter
VOUT
Test Conditions
Min
Typ
−3
Max
Units
Output Voltage Accuracy
Default VOUT
Line Regulation
2.7< VIN < 5.5
IO =10 mA
0.089
%/V
0.0013
%/mA
Load Regulation
100 mA < IO < IMAX
Eff
Efficiency
Load Current = 250 mA
ISHDN
Shutdown Supply Current
fOSC
Sync Mode Clock Frequency
+3
%
96
%
EN is de-asserted
0.01
µA
Synchronized from 13 MHz system
clock
2.0
MHz
2.0
MHz
Internal Oscillator Frequency
IPEAK
Peak Switching Current Limit
IQ (5)
Quiescent Current “On”
RDSON (P)
Pin-Pin Resistance PFET
RDSON (N)
Pin-Pin Resistance NFET
TON
Turn On Time
Start up from shut-down
CIN
Input Capacitor
Capacitance for stability
10
µF
CO
Output Capacitor
Capacitance for stability
10
µF
(1)
(2)
(3)
(4)
(5)
No load PFM Mode
2.0
A
33
µA
200
mΩ
180
mΩ
500
µs
All voltages are with respect to the potential at the GND pin.
Min and Max limits are ensured by design, test, or statistical analysis. Typical numbers are not specifications, but do represent the most
likely norm.
CIN, COUT: Low-ESR Surface-Mount Ceramic Capacitors (MLCCs) used in setting electrical characteristics.
The device maintains a stable, regulated output voltage without a load.
The Iq can be defined as the standing current of the LP3906 when the I2C bus is active and all other power blocks have been disabled
via the I2C bus, or it can be defined as the I2C bus active, and the other power blocks are active under no load condition. These two
values can be used by the system designer when the LP3906 is powered using a battery. If the user plans to use the HW enable pins to
disable each block of the IC please contact the factory applications for IQ details.
I/O ELECTRICAL CHARACTERISTICS
Unless otherwise noted: Typical values and limits appearing in normal type apply for TJ = 25°C. Limits appearing in boldface
type apply over the entire junction temperature range for operation, TJ = 0°C to +125°C. (1)
Parameter
VIL
Input Low Level
VIH
Input High Level
(1)
Test Conditions
Limit
Min
Max
Units
0.4
1.2
V
V
This specification is ensured by design.
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TYPICAL PERFORMANCE CHARACTERISTICS — LDO
TA = 25°C unless otherwise noted
Output Voltage Change
vs
Temperature (LDO2)
Vin = 4.3V, Vout = 1.8V, 100 mA load
2.00
2.00
1.50
1.50
1.00
1.00
VOUT CHANGE (%)
VOUT CHANGE (%)
Output Voltage Change
vs
Temperature (LDO1)
Vin = 4.3V, Vout = 3.3V, 100 mA load
0.50
0.00
-0.50
0.50
0.00
-0.50
-1.00
-1.00
-1.50
-1.50
-2.00
-50 -35 -20 -5 10 25 40 55 70 85 100
-2.00
-50 -35 -20 -5 10 25 40 55 70 85 100
TEMPERATURE (°C)
10
TEMPERATURE (°C)
Figure 4.
Figure 5.
Load Transient (LDO1)
3.6 Vin, 3.3 Vout, 0 – 100 mA load
Load Transient (LDO2)
3.6 Vin, 1.8 Vout, 0 – 100 mA load
Figure 6.
Figure 7.
Line Transient (LDO1)
3.6 - 4.5 Vin, 3.3 Vout, 150 mA load
Line Transient (LDO2)
3 – 4.2 Vin, 1.8 Vout, 150 mA load
Figure 8.
Figure 9.
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TYPICAL PERFORMANCE CHARACTERISTICS — LDO (continued)
TA = 25°C unless otherwise noted
Enable Start-up time (LDO1) )
0-3.6 Vin, 3.3 Vout, 1mA load
Enable Start-up time (LDO2)
0 – 3.6 Vin, 1.8 Vout, 1 mA load
Figure 10.
Figure 11.
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TYPICAL PERFORMANCE CHARACTERISTICS — BUCK
TA = 25°C unless otherwise noted
Output Voltage
vs.
Supply Voltage
(Vout = 1.0 V)
Shutdown Current
vs.
Temp
0.15
0.12
1.040
1.035
1.030
1.025
1.020
1.015
1.010
1.005
1.000
0.995
0.990
OUTPUT VOLTAGE (V)
SHUTDOWN CURRENT (éA)
IOUT = 20 mA
IOUT = 750 mA
VIN = 5.5V
0.09
VIN = 3.6V
0.06
VIN = 2.7V
IOUT = 1.5A
0.03
0.00
-40
-20
0
20
40
60
80
2.5
3.0
TEMPERATURE (°C)
4.5
5.0
5.5
Figure 13.
Output Voltage
vs.
Supply Voltage
(Vout = 1.8V)
Output Voltage
vs.
Supply Voltage
(Vout = 3.5V)
3.550
IOUT = 20 mA
1.830
3.540
IOUT = 750 mA
1.825
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
4.0
Figure 12.
1.835
IOUT = 20 mA
IOUT = 1.5 A
1.820
1.815
1.810
IOUT = 1.5A
3.530
3.520
3.510
1.805
3.500
1.800
2.5
3.490
4.0
IOUT = 750 mA
3.0
3.5
4.0
4.5
5.0
5.5
4.5
SUPPLY VOLTAGE (V)
5.0
5.5
SUPPLY VOLTAGE (V)
Figure 14.
Figure 15.
Buck 1 Efficiency
vs
Output Current
(Forced PWM Mode, Vout =1.2V, L= 2.2µH)
Buck 1 Efficiency
vs
Output Current
(Forced PWM Mode, Vout =2.0V, L= 2.2µH)
100
100
90
90
VIN = 2.8V
70
60
VIN = 5.5V
50
VIN = 3.6V
40
30
70
50
30
20
10
0
10
1
10
2
10
3
10
4
OUTPUT CURRENT (mA)
VIN = 3.6V
40
10
10
VIN = 5.5V
60
20
0
-3
-2
-1
10 10
10
VIN = 2.8V
80
EFFICIENCY (%)
EFFICIENCY (%)
80
0
-2
10
10
-1
10
0
10
1
10
2
10
3
10
4
OUTPUT CURRENT (mA)
Figure 16.
12
3.5
SUPPLY VOLTAGE (V)
Figure 17.
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TYPICAL PERFORMANCE CHARACTERISTICS — BUCK (continued)
TA = 25°C unless otherwise noted
Buck 1 Efficiency
vs
Output Current
(PFM to PWM mode, Vout =1.2V, L= 2.2µH)
Buck 1 Efficiency
vs
Output Current
(PFM to PWM mode, Vout =2.0V, L= 2.2µH)
100
100
VIN = 2.8V
80
EFFICIENCY (%)
EFFICIENCY (%)
VIN = 2.8V
90
90
VIN = 5.5V
70
VIN = 3.6V
60
80
VIN = 5.5V
70
VIN = 3.6V
60
50
50
40
40
30
10-3 10 -2 10 -1
10 0
10 1 10 2
10 3
30
10-2
10 4
10-1
100
101
102
103
104
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
Figure 18.
Figure 19.
Buck 2 Efficiency
vs
Output Current
(Forced PWM Mode, Vout =1.8V, L= 2.2µH)
Buck 2 Efficiency
vs
Output Current
(Forced PWM Mode, Vout =3.3V, L= 2.2µH)
100
100
90
90
VIN = 4.3V
70
VIN = 5.5V
60
50
40
30
70
50
40
30
20
10
10
10
0
10
1
10
2
10
3
10
0
-2
10
4
VIN = 5.5V
60
20
0
-2
-1
10
10
VIN = 4.3V
80
EFFICIENCY (%)
EFFICIENCY (%)
80
10
-1
OUTPUT CURRENT (mA)
10
0
10
1
10
2
10
3
10
4
OUTPUT CURRENT (mA)
Figure 20.
Figure 21.
Buck 2 Efficiency
vs
Output Current
(PFM to PWM Mode, Vout =1.8V, L= 2.2µH)
Buck 2 Efficiency
vs
Output Current
(PFM to PWM Mode, Vout =3.3V, L= 2.2µH)
100
100
VIN = 4.3V
VIN = 4.3V
90
EFFICIENCY (%)
EFFICIENCY (%)
90
80
VIN = 5.5V
70
60
50
VIN = 5.5V
80
70
60
50
40
-2
-1
10
10
10
0
10
1
10
2
10
3
10
4
OUTPUT CURRENT (mA)
40
-2
10
-1
10
10
0
1
10
10
2
10
3
10
4
OUTPUT CURRENT (mA)
Figure 22.
Figure 23.
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TYPICAL PERFORMANCE CHARACTERISTICS — BUCK (continued)
TA = 25°C unless otherwise noted
14
Load Transient Response
Vout = 1.2 (PWM Mode)
Mode Change by Load Transient
Vout = 1.2V (PWM to PFM)
Figure 24.
Figure 25.
Line Transient Response
Vin = 3 – 3.6 V, Vout = 1.2 V, 250 mA load
Line Transient Response
Vin = 3 – 3.6 V, Vout = 3.3 V, 250 mA load
Figure 26.
Figure 27.
Start up into PWM Mode
Vout = 1.8 V, 1.2 A load
Start up into PWM Mode
Vout = 3.3 V, 1.2 A load
Figure 28.
Figure 29.
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TYPICAL PERFORMANCE CHARACTERISTICS — BUCK (continued)
TA = 25°C unless otherwise noted
Start up into PFM Mode
Vout = 1.8 V, 30 mA load
Start up into PFM Mode
Vout = 3.3 V, 30 mA load
Figure 30.
Figure 31.
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HIGH VIN-HIGH LOAD OPERATION
Additional information is provided when the IC is operated at extremes of Vin and regulator loads. These are
described in terms of the Junction Temperature and Buck Output Ripple Management.
Junction Temperature
The maximum junction temperature TJ-MAX-OP of 125ºC of the IC package.
The following equations demonstrate junction temperature determination, ambient temperature TA-MAX and
Total chip power must be controlled to keep TJ below this maximum:
TJ-MAX-OP = TA-MAX + (θJA) [ °C/ Watt] * (PD-MAX) [Watts]
Total IC power dissipation PD-MAX is the sum of the individual power dissipation of the four regulators plus a
minor amount for chip overhead. Chip overhead is Bias, TSD & LDO analog.
PD-MAX = PLDO1 + PLD02 + PBUCK1 + PBUCK2 + (0.0001A * Vin) [Watts].
Power dissipation of LDO1
PLDO1 = (VinLDO1- VoutLDO1) * IoutLDO1 [V*A]
Power dissipation of LDO2
PLDO2 = (VinLDO2 - VoutLDO2) * IoutLDO2 [V*A]
Power dissipation of Buck1
PBuck1 = PIN – POUT =
VoutBuck1* IoutBuck1 * (1 -η1) / η1 [V*A]
η1 = efficiency of buck 1
Power dissipation of Buck2
PBuck2 = PIN – POUT =
VoutBuck2 * IoutBuck2 * (1 - η2) / η2 [V*A]
η2 = efficiency of buck 2
η is the efficiency for the specific condition taken from efficiency graphs.
Buck Output Ripple Management
If Vin and ILoad increase, the output ripple associated with the Buck Regulators also increases. Figure 32 shows
the safe operating area. To ensure operation in the area of concern it is recommended that the system designer
circumvents the output ripple issues to install schottky diodes on the Bucks(s) that are expected to perform under
these extreme corner conditions.
(Schottky diodes are recommended to reduce the output ripple if the system requirements include this shaded
area of operation. Vin > 5.2 V and Iload > 1.24 A)
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5.5
VIN (V)
5.0
4.5
4.0
3.5
3.0
0
0.5
1.0
1.5
LOAD CURRENT (A)
Figure 32. LP3906 Buck Converter
VIN vs ILOAD Operating Ranges
Thermal Performance of the WQFN Package
The LP3906 is a monolithic device with integrated power FETs. For that reason, it is important to pay special
attention to the thermal impedance of the WQFN package and to the PCB layout rules in order to maximize
power dissipation of the WQFN package.
The WQFN package is designed for enhanced thermal performance and features an exposed die attach pad at
the bottom center of the package that creates a direct path to the PCB for maximum power dissipation.
Compared to the traditional leaded packages where the die attach pad is embedded inside the molding
compound, the WQFN reduces one layer in the thermal path.
The thermal advantage of the WQFN package is fully realized only when the exposed die attach pad is soldered
down to a thermal land on the PCB board with thermal vias planted underneath the thermal land. Based on
thermal analysis of the WQFN package, the junction-to-ambient thermal resistance (θJA) can be improved by a
factor of two when the die attach pad of the WQFN package is soldered directly onto the PCB with thermal land
and thermal vias, as opposed to an alternative with no direct soldering to a thermal land. Typical pitch and outer
diameter for thermal vias are 1.27mm and 0.33mm respectively. Typical copper via barrel plating is 1oz, although
thicker copper may be used to further improve thermal performance. The LP3906 die attach pad is connected to
the substrate of the IC and therefore, the thermal land and vias on the PCB board need to be connected to
ground (GND pin).
For more information on board layout techniques, refer to Application Note AN–1187 (Literature Number
SNOA401) “Leadless Lead frame Package (WQFN).” This application note also discusses package handling,
solder stencil and the assembly process.
DC/DC Converters
OVERVIEW
The LP3906 supplies the various power needs of the application by means of two Linear Low Drop Regulators
LDO1 and LDO2 and two Buck converters SW1 and SW2. The table here under lists the output characteristics of
the various regulators.
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Table 2. SUPPLY SPECIFICATION
Output
Supply
Load
LDO1
LDO2
VOUT Range(V)
Resolution (mV)
IMAX
Maximum Output Current (mA)
analog
1.0 to 3.5
100
300
analog
1.0 to 3.5
100
300
SW1
digital
0.8 to 2.0
50
1500
SW2
digital
1.0 to 3.5
100
1500
LINEAR LOW DROP-OUT REGULATORS (LDOS)
LDO1 and LDO2 are identical linear regulators targeting analog loads characterized by low noise requirements.
LDO1 and LDO2 are enabled through the ENLDO pin or through the corresponding LDO1 or LDO2 control
register. The output voltages of both LDOs are register programmable. The default output voltages are factory
programmed during Final Test, which can be tailored to the specific needs of the system designer.
VLDO
VIN
LDO
Register
controlled
+
ENLDO
-
Vref
GND
NO-LOAD STABILITY
The LDOs will remain stable and in regulation with no external load. This is an important consideration in some
circuits, for example CMOS RAM keep-alive applications.
LDO1 AND LDO2 CONTROL REGISTERS
LDO1 and LDO2 can be configured by means of the LDO1 and LDO2 control registers. The output voltage is
programmable in steps of 100mV from 1.0V to 3.5V by programming bits D4-0 in the LDO Control registers. Both
LDO1 and LDO2 are enabled by applying a logic 1 to the ENLDO1 and ENLDO2 pin. Enable/disable control is
also provided through enable bit of the LDO1 and LDO2 control registers. The value of the enable LDO bit in the
register is logic 1 by default. The output voltage can be altered while the LDO is enabled.
SW1, SW2: Synchronous Step Down Magnetic DC/DC Converters
FUNCTIONAL DESCRIPTION
The LP3906 incorporates two high efficiency synchronous switching buck regulators, SW1 and SW2 that deliver
a constant voltage from a single Li-Ion battery to the portable system processors. Using a voltage mode
architecture with synchronous rectification, both bucks have the ability to deliver up to 1500mA depending on the
input voltage and output voltage (voltage head room), and the inductor chosen (maximum current capability).
There are three modes of operation depending on the current required - PWM, PFM, and shutdown. PWM mode
handles current loads of approximately 70mA or higher, delivering voltage precision of +/-3% with 90% efficiency
or better. Lighter output current loads cause the device to automatically switch into PFM for reduced current
consumption (IQ = 15 µA typ.) and a longer battery life. The Standby operating mode turns off the device, offering
the lowest current consumption. PWM or PFM mode is selected automatically or PWM mode can be forced
through the setting of the buck control register.
Both SW1 and SW2 can operate up to a 100% duty cycle (PMOS switch always on) for low drop out control of
the output voltage. In this way the output voltage will be controlled down to the lowest possible input voltage.
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Additional features include soft-start, under-voltage lock-out, current overload protection, and thermal overload
protection.
CIRCUIT OPERATION DESCRIPTION
A buck converter contains a control block, a switching PFET connected between input and output, a synchronous
rectifying NFET connected between the output and ground (BCKGND pin) and a feedback path. During the first
portion of each switching cycle, the control block turns on the internal PFET switch. This allows current to flow
from the input through the inductor to the output filter capacitor and load. The inductor limits the current to a
ramp with a slope of
VIN - VOUT
L
(1)
by storing energy in a magnetic field. During the second portion of each cycle, the control block turns the PFET
switch off, blocking current flow from the input, and then turns the NFET synchronous rectifier on. The inductor
draws current from ground through the NFET to the output filter capacitor and load, which ramps the inductor
current down with a slope of
-VOUT
L
(2)
The output filter stores charge when the inductor current is high, and releases it when low, smoothing the voltage
across the load.
PWM OPERATION
During PWM operation the converter operates as a voltage-mode controller with input voltage feed forward. This
allows the converter to achieve excellent load and line regulation. The DC gain of the power stage is proportional
to the input voltage. To eliminate this dependence, feed forward voltage inversely proportional to the input
voltage is introduced.
INTERNAL SYNCHRONOUS RECTIFICATION
While in PWM mode, the buck uses an internal NFET as a synchronous rectifier to reduce rectifier forward
voltage drop and associated power loss. Synchronous rectification provides a significant improvement in
efficiency whenever the output voltage is relatively low compared to the voltage drop across an ordinary rectifier
diode.
CURRENT LIMITING
A current limit feature allows the converter to protect itself and external components during overload conditions.
PWM mode implements current limiting using an internal comparator that trips at 2.0 A (typ). If the output is
shorted to ground the device enters a timed current limit mode where the NFET is turned on for a longer duration
until the inductor current falls below a low threshold, ensuring inductor current has more time to decay, thereby
preventing runaway.
A current limit feature allows the buck to protect itself and external components during overload conditions PWM
mode implements cycle-by-cycle current limiting using an internal comparator that trips at 2000mA (typical).
PFM OPERATION
At very light loads, the converter enters PFM mode and operates with reduced switching frequency and supply
current to maintain high efficiency.
The part will automatically transition into PFM mode when either of two conditions occurs for a duration of 32 or
more clock cycles:
1. The inductor current becomes discontinuous
or
2. The peak PMOS switch current drops below the IMODE level
VIN
(Typically IMODE < 66 mA +
)
160:
(3)
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During PFM operation, the converter positions the output voltage slightly higher than the nominal output voltage
during PWM operation, allowing additional headroom for voltage drop during a load transient from light to heavy
load. The PFM comparators sense the output voltage via the feedback pin and control the switching of the output
FETs such that the output voltage ramps between 0.8% and 1.6% (typical) above the nominal PWM output
voltage. If the output voltage is below the ‘high’ PFM comparator threshold, the PMOS power switch is turned on.
It remains on until the output voltage exceeds the ‘high’ PFM threshold or the peak current exceeds the IPFM level
set for PFM mode. The typical peak current in PFM mode is:
VIN
IPFM = 66 mA +
80:
(4)
Once the PMOS power switch is turned off, the NMOS power switch is turned on until the inductor current ramps
to zero. When the NMOS zero-current condition is detected, the NMOS power switch is turned off. If the output
voltage is below the ‘high’ PFM comparator threshold (see Figure 33), the PMOS switch is again turned on and
the cycle is repeated until the output reaches the desired level. Once the output reaches the ‘high’ PFM
threshold, the NMOS switch is turned on briefly to ramp the inductor current to zero and then both output
switches are turned off and the part enters an extremely low power mode. Quiescent supply current during this
‘sleep’ mode is less than 30 µA, which allows the part to achieve high efficiencies under extremely light load
conditions. When the output drops below the ‘low’ PFM threshold, the cycle repeats to restore the output voltage
to ~1.6% above the nominal PWM output voltage.
If the load current should increase during PFM mode (see Figure 33) causing the output voltage to fall below the
‘low2’ PFM threshold, the part will automatically transition into fixed-frequency PWM mode.
SW1, SW2 OPERATION
SW1 and SW2 have selectable output voltages ranging from 0.8V to 3.5V (typ.). Both SW1 and SW2 in the
LP3906 are I2C register controlled and are enabled by default through the internal state machine of the LP3906
following a Power-On event that moves the operating mode to the Active state. (see POWER ON). The SW1 and
SW2 output voltages revert to default values when the power on sequence has been completed. The default
output voltage for each buck converter is factory programmable. (See APPLICATION NOTES).
SW1, SW2 can be enabled/disabled through the corresponding control register.
The Modulation mode PWM/PFM is by default automatic and depends on the load as described above in the
functional description. The modulation mode can be overridden by setting I2C bit to a logic 1 in the corresponding
buck control register, forcing the buck to operate in PWM mode regardless of the load condition.
High PFM Threshold
~1.016 * Vout
PFM Mode at Light Load
Load current
increases
Low1 PFM Threshold
~1.008 * Vout
ZA
xi
s
Nfet on
drains
inductor
current
until
I inductor = 0
Current load
increases,
draws Vout
towards
Low2 PFM
Threshold
Low PFM
Threshold,
turn on
PFET
Low2 PFM Threshold,
switch back to PWMmode
Zs
Axi
Pfet on
until
Ipfm limit
reached
High PFM
Voltage
Threshold
reached,
go into
sleep mode
Low2 PFM Threshold
Vout
PWM Mode at
Moderate to Heavy
Loads
Figure 33.
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SHUTDOWN MODE
During shutdown the PFET switch, reference, control and bias circuitry of the converters are turned off. The
NFET switch will be on in shutdown to discharge the output. When the converter is enabled, soft start is
activated. It is recommended to disable the converter during the system power up and under voltage conditions
when the supply is less than 2.8V.
SOFT START
The soft-start feature allows the power converter to gradually reach the initial steady state operating point, thus
reducing start-up stresses and surges. The two LP3906 buck converters have a soft-start circuit that limits inrush current during start-up. During start-up the switch current limit is increased in steps. Soft start is activated
only if EN goes from logic low to logic high after VIN reaches 2.8V. Soft start is implemented by increasing switch
current limit in steps of 213 mA, 425 mA, 850 mA and 1700 mA (typ. Switch current limit). The start-up time
thereby depends on the output capacitor and load current demanded at start-up.
LOW DROPOUT OPERATION
The LP3906 can operate at 100% duty cycle (noswitching; PMOS switch completely on) for low drop out support
of the output voltage. In this way the output voltage will be controlled down to the lowest possible input voltage.
When the device operates near 100% duty cycle, output voltage ripple is approximately 25 mV. The minimum
input voltage needed to support the output voltage is:
VIN, MIN = ILOAD * (RDSON, PFET + RINDUCTOR) + VOUT
Where:
ILOAD
Load current
RDSON, PFET
Drain to source resistance of
PFET switch in the triode region
RINDUCTOR
Inductor resistance
FLEXIBLE POWER SEQUENCING OF MULTIPLE POWER SUPPLIES
The LP3906 provides several options for power on sequencing. The two bucks can be individually controlled with
ENSW1 and ENSW2. The two LDOs can also be individually controlled with ENLDO1 and ENLDO2.
If the user desires a set power on sequence, all four enables should be tied LOW so that the regulators don’t
automatically enable when power is supplied. The user can then program the chip through I2C and raise EN_T
from LOW to HIGH to activate the power on sequencing.
POWER ON
EN_T assertion causes the LP3906 to emerge from Standby mode to Full Operation mode at a preset timing
sequence. By default, the enables for the LDOs and Bucks are internally pulled up, which causes the part to turn
ON automatically. If the user wishes to have a preset timing sequence to power on the regulators, the external
regulator enables must be tied LOW. Otherwise, simply tie the enables of each specific regulator HIGH.
EN_T is edge triggered with rising edge signaling the chip to power on. The EN_T input is deglitched and the
default is set at 1 ms. As shown in the next 2 diagrams, a rising EN_T edge will start a power on sequence, while
a falling EN_T edge will start a shutdown sequence. If EN_T is high, toggling the external enables of the
regulators will have no effect on the chip.
Table 3. Default Power ON Sequence:
t1 (ms)
t2 (ms)
t3 (ms)
t4 (ms)
1.5
2.0
3
6
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NOTE
LP3906 The default Power on delays can be reprogrammed at final test or by using I2C
registers to 1, 1.5, 2, 3, 6, or 11 ms.
The regulators can also be programmed through I2C to turn on and off. By default, the I2C enables for the
regulators are ON.
The regulators are on following the pattern below:
Regulators on = (I2C enable) AND (External pin enable OR EN_T high).
NOTE
The EN_T power-up sequencing may also be employed immediately after VIN is applied to
the device. However, VIN must be stable for approximately 8ms minimum before EN_T be
asserted high to ensure internal bias, reference, and the Flexible POR timing are
stabilized. This initial EN_T delay is necessary only upon first time device power-on for
power sequencing function to operate properly.
2
I C
Regulator ON
EN_T
Ext Enable
Pins
LP3906 Default Power-Up Sequence
EN_T
t1
Vout Buck1
t2
Vout Buck2
t3
Vout LDO1
t4
Vout LDO2
Table 4. Power-On Timing Specification
Symbol
Description
Min
Typ
Max
Units
t1
Programmable Delay from EN_T assertion to VCC_Buck1 On
1.5
ms
t2
Programmable Delay from EN_T assertion to VCC_Buck2 On
2
ms
t3
Programmable Delay from EN_T assertion to VCC_LDO1 On
3
ms
t4
Programmable Delay from EN_T assertion to VCC_LDO2 On
6
ms
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NOTE
LP3906 The default Power on delays can be reprogrammed at final test or I2C to 1, 1.5, 2,
3, 6, or 11 ms.
LP3906 Default Power-Off Sequence
EN_T
Vout Buck1
t1
Vout Buck2
t2
Vout LDO1
t3
Vout LDO2
t4
Symbol
Description
Min
Typ
Max
Units
t1
Programmable Delay from EN_T deassertion to VCC_Buck1 Off
1.5
ms
t2
Programmable Delay from EN_T deassertion to VCC_Buck2 Off
2
ms
t3
Programmable Delay from EN_T deassertion to VCC_LDO1 Off
3
ms
t4
Programmable Delay from EN_T deassertion to VCC_LDO2 Off
6
ms
NOTE
LP3906 The default Power on delays can be reprogrammed at final test to 0, .5, 1, 2, 5, or
10 ms. Default setting is the same as the on sequence.
Power-On-Reset
The LP3906 is equipped with an internal Power-On-Reset (“POR”) circuit that will reset the logic when VDD <
VPOR. This ensures that the logic is properly initialized when VDD rises above the minimum operating voltage of
the Logic and the internal oscillator that clocks the Sequential Logic in the Control section.
I2C Compatible Serial Interface
I2C SIGNALS
The LP3906 features an I2C compatible serial interface, using two dedicated pins: SCL and SDA for I2C clock
and data respectively. Both signals need a pull-up resistor according to the I2C specification. The LP3906
interface is an I2C slave that is clocked by the incoming SCL clock.
Signal timing specifications are according to the I2C bus specification. The maximum bit rate is 400 kbit/s. See
I2C specification from Philips for further details.
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I2C DATA VALIDITY
The data on the SDA line must be stable during the HIGH period of the clock signal (SCL), e.g.- the state of the
data line can only be changed when CLK is LOW.
SCL
SDA
data
change
allowed
data
valid
data
change
allowed
data
valid
data
change
allowed
Figure 34. I2C Signals: Data Validity
I2C START AND STOP CONDITIONS
START and STOP bits classify the beginning and the end of the I2C session. START condition is defined as the
SDA signal transitioning from HIGH to LOW while the SCL line is HIGH. STOP condition is defined as the SDA
transitioning from LOW to HIGH while the SCL is HIGH. The I2C master always generates START and STOP
bits. The I2C bus is considered to be busy after START condition and free after STOP condition. During data
transmission, I2C master can generate repeated START conditions. First START and repeated START
conditions are equivalent, function-wise.
SDA
SCL
S
P
START condition
STOP
condition
Figure 35. START and STOP Conditions
TRANSFERRING DATA
Every byte put on the SDA line must be eight bits long, with the most significant bit (MSB) being transferred first.
Each byte of data has to be followed by an acknowledge bit. The acknowledged related clock pulse is generated
by the master. The transmitter releases the SDA line (HIGH) during the acknowledge clock pulse. The receiver
must pull down the SDA line during the 9th clock pulse, signifying acknowledgement. A receiver which has been
addressed must generate an acknowledgement (“ACK”) after each byte has been received.
After the START condition, the I2C master sends a chip address. This address is seven bits long followed by an
eighth bit which is a data direction bit (R/W). Please note that according to industry I2C standards for 7-bit
addresses, the MSB of an 8-bit address is removed, and communication actually starts with the 7th most
significant bit. For the eighth bit (LSB), a “0” indicates a WRITE and a “1” indicates a READ. The second byte
selects the register to which the data will be written. The third byte contains data to write to the selected register.
LP3906 has a chip address of 60’h, which is factory programmed.
LSB
MSB
ADR6
bit7
ADR5
bit6
ADR4
bit5
ADR3
bit4
ADR2
bit3
ADR1
bit2
ADR0
bit1
1
1
0
0
0
0
0
R/W
bit0
I2C SLAVE address (chip address)
Figure 36. I2C Chip Address
24
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start
msb Chip Address lsb w ack
ack from slave
ack from slave
ack from slave
msb Register Add lsb
ack
msb
DATA
lsb
ack
stop
ack
stop
SCL
1 2
3 4 5 6
7
8
9 1 2 3 ...
SDA
start
id = K¶60
w ack
addr = K¶02
DGGUHVV K¶$$ GDWD
ack
w = write (SDA = “0”)
r = read (SDA = “1”)
ack = acknowledge (SDA pulled down by either master or slave)
rs = repeated start
id = LP3906 chip address : 0x60
Figure 37. I2C Write Cycle
When a READ function is to be accomplished, a WRITE function must precede the READ function, as shown in
the Read Cycle waveform.
ack from slave
start
msb Chip Address lsb w ack
ack from slave repeated start
msb Register Add lsb
ack
rs
SCL
ack from slave
msb Chip Address lsb
r ack
id = K¶60
r ack
data from slave
msb
DATA
ack from master
lsb
ack stop
.
SDA
start
id = K¶60
w ack
register addr = K¶10
ack
rs
GDWD DGGU K¶6A
ack stop
Figure 38. I2C Read Cycle
LP3906 Control Registers
Register Address
Register Name
Read/Write
0x02
ICRA
R
Interrupt Status Register A
Register Description
0x07
SCR1
R/W
System Control 1 Register
0x10
BKLDOEN
R/W
Buck and LDO Output Voltage Enable Register
0x11
BKLDOSR
R
Buck and LDO Output Voltage Status Register
0x20
VCCR
R/W
Voltage Change Control Register 1
0x23
B1TV1
R/W
Buck 1 Target Voltage 1 Register
0x24
B1TV2
R/W
Buck 1 Target Voltage 2 Register
0x25
B1RC
R/W
Buck 1 Ramp Control
0x29
B2TV1
R/W
Buck 2 Target Voltage 1 Register
0x2A
B2TV2
R/W
Buck 2 Target Voltage 2 Register
0x2B
B2RC
R/W
Buck 2 Ramp Control
0x38
BFCR
R/W
Buck Function Register
0x39
LDO1VCR
R/W
LDO1 Voltage control Registers
0x3A
LDO2VCR
R/W
LDO2 Voltage control Registers
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INTERRUPT STATUS REGISTER (ISRA) 0X02
This register informs the user of the temperature status of the chip.
D7-2
D1
D0
Name
—
Temp 125°C
—
Access
—
R
—
Data
Reserved
Status bit for thermal warning PMIC T>125°C
0 – PMIC Temp. < 125°C
1 – PMIC Temp. > 125°C
Reserved
Reset
0
0
0
CONTROL 1 REGISTER (SCR1) 0X07
This register allows the user to select the preset delay sequence for power-on timing, to switch between PFM
and PWM mode for the bucks, and also to select between an internal and external clock for the bucks.
The external LDO and SW enables should be pulled LOW to allow the blocks to sequence correctly through
assertion of the EN_T pin.
D7
D6-4
D3
D2
D1
D0
Name
—
EN_DLY
—
FPWM2
FPWM1
ECEN
Access
—
R/W
—
R/W
R/W
R/W
Data
Reserved
Selects the preset
delay sequence from
EN_T assertion
(shown below)
Reserved
Buck 2 PWM /PFM
Mode select
0 – Auto Switch PFM PWM operation
1 – PWM Mode Only
Buck 1 PWM /PFM
Mode select
0 – Auto Switch PFM PWM operation
1 – PWM Mode Only
External Buck Clock
Select
0 – Internal 2 MHz
Oscillator clock
1 – External 13 MHz
Oscillator clock
Reset
0
010
1
0
0
0
EN_DLY PRESET DELAY SEQUENCE AFTER EN_T ASSERTION
Delay (ms)
EN_DLY
Buck1
Buck2
LDO1
LDO2
000
1
1
1
1
001
1
1.5
2
2
010
1.5
2
3
6
011
1.5
2
1
1
100
1.5
2
3
6
101
1.5
1.5
2
2
110
3
2
1
1.5
111
2
3
6
11
BUCK AND LDO OUTPUT VOLTAGE ENABLE REGISTER (BKLDOEN) – 0X10
This register controls the enables for the Bucks and LDOs.
D7
D6
D5
D4
D3
D2
D1
D0
Name
—
LDO2EN
—
LDO1EN
—
BK2EN
—
BK1EN
Access
—
R/W
—
R/W
—
R/W
—
R/W
Data
Reserved
0 – Disable
1 – Enable
Reserved
0 – Disable
1 – Enable
Reserved
0 – Disable
1 – Enable
Reserved
0 – Disable
1 – Enable
Reset
0
1
1
1
0
1
0
1
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BUCK AND LDO STATUS REGISTER (BKLDOSR) – 0X11
This register monitors whether the Bucks and LDOs meet the voltage output specifications.
D7
D6
D5
D4
D3
D2
D1
D0
Name
BKS_OK
LDOS_OK
LDO2_OK
LDO1_OK
—
BK2_OK
—
BK1_OK
Access
R
R
R
R
—
R
—
R
Data
0 – Buck 1-2
Not Valid
1 – Bucks Valid
0 – LDO 1-2
Not Valid
1 – LDOs Valid
0 – LDO2 Not
Valid
1 – LDO2 Valid
0 – LDO1 Not
Valid
1 – LDO1 Valid
Reserve
d
0 – Buck2 Not
Valid
1 – Buck2 Valid
Reserve
d
0 – Buck1 Not
Valid
1 – Buck1 Valid
Reset
0
0
0
0
0
0
0
0
BUCK VOLTAGE CHANGE CONTROL REGISTER 1 (VCCR) – 0X20
This register selects and controls the output target voltages for the buck regulators.
D7-6
D5
D4
D3-2
D1
D0
Name
—
B2VS
B2GO
—
B1VS
B1GO
Access
—
R/W
R/W
—
R/W
R/W
Data
Reserved
Buck2 Target Voltage
Select
0 – B2VT1
1 – B2VT2
Buck2 Voltage Ramp
CTRL
0 – Hold
1 – Ramp to B2VS
selection
Reserved
Buck1 Target Voltage
Select
0 – B1VT1
1 – B1VT2
Buck1 Voltage Ramp
CTRL
0 – Hold
1 – Ramp to B1VS
selection
Reset
00
0
0
00
0
0
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BUCK1 TARGET VOLTAGE 1 REGISTER (B1TV1) – 0X23
This register allows the user to program the output target voltage of Buck 1.
D7-5
D4-0
Name
—
BK1_VOUT1
Access
—
R/W
Data
Reserved
Buck1 Output Voltage (V)
5’h00
Ext Ctrl
5’h01
0.80
5’h02
0.85
5’h03
0.90
5’h04
0.95
5’h05
1.00
5’h06
1.05
5’h07
1.10
5’h08
1.15
5’h09
1.20
5’h0A
1.25
5’h0B
1.30
5’h0C
1.35
5’h0D
1.40
5’h0E
1.45
5’h0F
1.50
5’h10
1.55
5’h11
1.60
5’h12
1.65
5’h13
1.70
5’h14
1.75
5’h15
1.80
5’h16
1.85
5’h17
1.90
5’h18
1.95
5’h19
2.00
5’h1A–5’h1F
Reset
28
000
2.00
Factory Programmed Default
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BUCK 1 TARGET VOLTAGE 2 REGISTER (B1TV2) – 0X24
This register allows the user to program the output target voltage of Buck 1.
D7-5
D4-0
Name
—
BK1_VOUT2
Access
—
R/W
Data
Reserved
Buck1 Output Voltage (V)
5’h00
Ext Ctrl
5’h01
0.80
5’h02
0.85
5’h03
0.90
5’h04
0.95
5’h05
1.00
5’h06
1.05
5’h07
1.10
5’h08
1.15
5’h09
1.20
5’h0A
1.25
5’h0B
1.30
5’h0C
1.35
5’h0D
1.40
5’h0E
1.45
5’h0F
1.50
5’h10
1.55
5’h11
1.60
5’h12
1.65
5’h13
1.70
5’h14
1.75
5’h15
1.80
5’h16
1.85
5’h17
1.90
5’h18
1.95
5’h19
2.00
5’h1A–5’h1F
Reset
000
2.00
Factory Programmed Default
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BUCK 1 RAMP CONTROL REGISTER (B1RC) - 0x25
This register allows the user to program the rate of change between the target voltages of Buck 1.
D7
D6-4
D3-0
Name
----
----
B1RS
Access
----
----
R/W
Reserved
Reserved
Data
Data Code
Ramp Rate mV/us
4h'0
Instant
4h'1
1
4h'2
2
4h'3
3
4h'4
4
4h'5
5
4h'6
6
4h'7
7
4h'8
8
4h'9
9
4h'A
10
4h'B - 4h'F
Reset
30
0
010
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BUCK 2 TARGET VOLTAGE 1 REGISTER (B2TV1) – 0X29
This register allows the user to program the output target voltage of Buck 2.
D7-5
D4-0
Name
—
BK2_VOUT1
Access
—
R/W
Data
Reserved
Buck2 Output Voltage (V)
5’h00
Ext Ctrl
5’h01
1.0
5’h02
1.1
5’h03
1.2
5’h04
1.3
5’h05
1.4
5’h06
1.5
5’h07
1.6
5’h08
1.7
5’h09
1.8
5’h0A
1.9
5’h0B
2.0
5’h0C
2.1
5’h0D
2.2
5’h0E
2.4
5’h0F
2.5
5’h10
2.6
5’h11
2.7
5’h12
2.8
5’h13
2.9
5’h14
3.0
5’h15
3.1
5’h16
3.2
5’h17
3.3
5’h18
3.4
5’h19
3.5
5’h1A–5’h1F
Reset
000
3.5
Factory Programmed Default
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BUCK 2 TARGET VOLTAGE 2 REGISTER (B2TV2) – 0X2A
This register allows the user to program the output target voltage of Buck 2.
D7-5
D4-0
Name
—
BK2_VOUT2
Access
—
R/W
Data
Reserved
Buck2 Output Voltage (V)
5’h00
Ext Ctrl
5’h01
1.0
5’h02
1.1
5’h03
1.2
5’h04
1.3
5’h05
1.4
5’h06
1.5
5’h07
1.6
5’h08
1.7
5’h09
1.8
5’h0A
1.9
5’h0B
2.0
5’h0C
2.1
5’h0D
2.2
5’h0E
2.4
5’h0F
2.5
5’h10
2.6
5’h11
2.7
5’h12
2.8
5’h13
2.9
5’h14
3.0
5’h15
3.1
5’h16
3.2
5’h17
3.3
5’h18
3.4
5’h19
3.5
5’h1A–5’h1F
Reset
32
000
3.5
Factory Programmed Default
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BUCK 2 RAMP CONTROL REGISTER (B2RC) - 0x2B
This register allows the user to program the rate of change between the target voltages of Buck 2
D7
D6-4
D3-0
Name
----
----
B2RS
Access
----
----
R/W
Reserved
Reserved
Data
Data Code
Ramp Rate mV/us
4h'0
Instant
4h'1
1
4h'2
2
4h'3
3
4h'4
4
4h'5
5
4h'6
6
4h'7
7
4h'8
8
4h'9
9
4h'A
10
4h'B - 4h'F
Reset
0
010
10
1000
BUCK FUNCTION REGISTER (BFCR) – 0x38
Clock Frequency
This register allows the Buck switcher clock frequency to be spread across a wider range, allowing for less
Electro-magnetic Interference (EMI). The spread spectrum modulation frequency refers to the rate at which the
frequency ramps up and down, centered at 2 MHz.
Spread Spectrum
frequency
Peak frequency deviation
2 kHz triangle
wave
10 kHz triangle
wave
2 MHz
Time
D7-2
Name
Access
D1
D0
—
BK_SLOMOD
BK_SSEN
—
R/W
R/W
Buck Spread Spectrum Modulation
0 – 10 kHz triangular wave
1 – 2 kHz triangular wave
Spread Spectrum Function Output
0 – Disabled
1 – Enabled
1
0
Data
Reserved
Reset
000010
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LDO1 CONTROL REGISTER (LDO1VCR) – 0X39
This register allows the user to program the output target voltage of LDO 1.
D7-5
D4-0
Name
—
LDO1_OUT
Access
—
R/W
Data
Reserved
LDO1 Output voltage (V)
5’h00
1.0
5’h01
1.1
5’h02
1.2
5’h03
1.3
5’h04
1.4
5’h05
1.5
5’h06
1.6
5’h07
1.7
5’h08
1.8
5’h09
1.9
5’h0A
2.0
5’h0B
2.1
5’h0C
2.2
5’h0D
2.3
5’h0E
2.4
5’h0F
2.5
5’h10
2.6
5’h11
2.7
5’h12
2.8
5’h13
2.9
5’h14
3.0
5’h15
3.1
5’h16
3.2
5’h17
3.3
5’h18
3.4
5’h19
3.5
5’h1A–5’h1F
Reset
34
000
3.5
Factory Programmed Default
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LDO2 CONTROL REGISTER (LDO2VCR) – 0X3A
This register allows the user to program the output target voltage of LDO 2.
D7-5
D4-0
Name
—
LDO2_OUT
Access
—
R/W
Data
Reserved
LDO2 Output voltage (V)
5’h00
1.0
5’h01
1.1
5’h02
1.2
5’h03
1.3
5’h04
1.4
5’h05
1.5
5’h06
1.6
5’h07
1.7
5’h08
1.8
5’h09
1.9
5’h0A
2.0
5’h0B
2.1
5’h0C
2.2
5’h0D
2.3
5’h0E
2.4
5’h0F
2.5
5’h10
2.6
5’h11
2.7
5’h12
2.8
5’h13
2.9
5’h14
3.0
5’h15
3.1
5’h16
3.2
5’h17
3.3
5’h18
3.4
5’h19
3.5
5’h1A–5’h1F
Reset
000
3.5
Factory Programmed Default
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APPLICATION NOTES
SYSTEM CLOCK INPUT (SYNC) PIN
Pin 23 of the chip allows for a system clock input in order to synchronize the buck converters in PWM mode. This
is useful if the user wishes to force the bucks to work synchronously with the system. Otherwise, the user should
tie the pin to GND and the bucks will operate on an internal 2 MHz clock.
The signal applied to the SYNC pin must be 13 MHz as per application processor specifications, but we can be
contacted to modify that specification if so desired. Upon inputting the 13 MHz clock signal, the bucks will scale it
down and continue to run at 2 MHz based off the 13 MHz clock.
ANALOG POWER SIGNAL ROUTING
All power inputs should be tied to the main VDD source (i.e. battery), unless the user wishes to power it from
another source. (i.e. external LDO output).
The analog VDD inputs power the internal bias and error amplifiers, so they should be tied to the main VDD. The
analog VDD inputs must have an input voltage between 2.7 and 5.5 V, as specified on pg. 6 of the datasheet.
The other VINs (VINLDO1, VINLDO2, VIN1, VIN2) can actually have inputs lower than 2.7V, as long as it's higher
than the programmed output (+0.3V, to be safe).
The analog and digital grounds should be tied together outside of the chip to reduce noise coupling.
COMPONENT SELECTION
Inductors for SW1 and SW2
There are two main considerations when choosing an inductor; the inductor should not saturate and the inductor
current ripple is small enough to achieve the desired output voltage ripple. Care should be taken when reviewing
the different saturation current ratings that are specified by different manufacturers. Saturation current ratings are
typically specified at 25ºC, so ratings at maximum ambient temperature of the application should be requested
from the manufacturer.
There are two methods to choose the inductor saturation current rating:
Method 1
The saturation current is greater than the sum of the maximum load current and the worst case average to peak
inductor current. This can be written as follows:
Isat > Ioutmax + Iripple
where
Iripple =
§1·
©f¹
x
§VIN - VOUT· x § VOUT·
© 2L ¹ © VIN ¹
(5)
IRIPPLE: Average to peak inductor current
IOUTMAX: Maximum load current
VIN:
Maximum input voltage to the buck
L:
Min inductor value including worse case tolerances (30% drop can be considered for method 1)
f:
Minimum switching frequency (1.6 MHz)
VOUT: Buck Output voltage
Method 2
A more conservative and recommended approach is to choose an inductor that has saturation current rating
greater than the maximum current limit of 2375 mA.
Given a peak-to-peak current ripple (IPP) the inductor needs to be at least:
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§VIN - VOUT· x § VOUT· x § 1 ·
© IPP ¹ © VIN ¹ © f ¹
(6)
Inductor
LSW1,2
Value
Unit
2.2
µH
Description
SW1,2 inductor
Notes
D.C.R. 70 mΩ
External Capacitors
The regulators on the LP3906 require external capacitors for regulator stability. These are specifically designed
for portable applications requiring minimum board space and smallest components. These capacitors must be
correctly selected for good performance.
LDO CAPACITOR SELECTION
Input Capacitor
An input capacitor is required for stability. It is recommended that a 1.0 μF capacitor be connected between the
LDO input pin and ground (this capacitance value may be increased without limit).
This capacitor must be located a distance of not more than 1 cm from the input pin and returned to a clean
analog ground. Any good quality ceramic, tantalum, or film capacitor may be used at the input.
Important: Tantalum capacitors can suffer catastrophic failures due to surge currents when connected to a low
impedance source of power (like a battery or a very large capacitor). If a tantalum capacitor is used at the input,
it must be ensured by the manufacturer to have a surge current rating sufficient for the application.
There are no requirements for the ESR (Equivalent Series Resistance) on the input capacitor, but tolerance and
temperature coefficient must be considered when selecting the capacitor to ensure the capacitance will remain
approximately 1.0 μF over the entire operating temperature range.
Output Capacitor
The LDOs on the LP3906 are designed specifically to work with very small ceramic output capacitors. A 1.0 µF
ceramic capacitor (temperature types Z5U, Y5V or X7R) with ESR between 5 mΩ to 500 mΩ, are suitable in the
application circuit.
It is also possible to use tantalum or film capacitors at the device output, COUT (or VOUT), but these are not as
attractive for reasons of size and cost.
The output capacitor must meet the requirement for the minimum value of capacitance and also have an ESR
value that is within the range 5 mΩ to 500 mΩ for stability.
Capacitor Characteristics
The LDOs are designed to work with ceramic capacitors on the output to take advantage of the benefits they
offer. For capacitance values in the range of 0.47 µF to 4.7 µF, ceramic capacitors are the smallest, least
expensive and have the lowest ESR values, thus making them best for eliminating high frequency noise. The
ESR of a typical 1.0 µF ceramic capacitor is in the range of 20 mΩ to 40 mΩ, which easily meets the ESR
requirement for stability for the LDOs.
For both input and output capacitors, careful interpretation of the capacitor specification is required to ensure
correct device operation. The capacitor value can change greatly, depending on the operating conditions and
capacitor type.
In particular, the output capacitor selection should take account of all the capacitor parameters, to ensure that the
specification is met within the application. The capacitance can vary with DC bias conditions as well as
temperature and frequency of operation. Capacitor values will also show some decrease over time due to aging.
The capacitor parameters are also dependent on the particular case size, with smaller sizes giving poorer
performance figures in general. As an example, the graph below shows a typical graph comparing different
capacitor case sizes in a Capacitance vs. DC Bias plot.
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CAP VALUE (% of NOMINAL 1 PF)
SNVS456M – AUGUST 2006 – REVISED MAY 2013
0603, 10V, X5R
100%
80%
60%
0402, 6.3V, X5R
40%
20%
0
1.0
2.0
3.0
4.0
5.0
DC BIAS (V)
Figure 39. Graph Showing a Typical Variation in Capacitance vs. DC Bias
As shown in the graph, increasing the DC Bias condition can result in the capacitance value that falls below the
minimum value given in the recommended capacitor specifications table. Note that the graph shows the
capacitance out of spec for the 0402 case size capacitor at higher bias voltages. It is therefore recommended
that the capacitor manufacturers' specifications for the nominal value capacitor are consulted for all conditions,
as some capacitor sizes (e.g. 0402) may not be suitable in the actual application.
The ceramic capacitor’s capacitance can vary with temperature. The capacitor type X7R, which operates over a
temperature range of −55°C to +125°C, will only vary the capacitance to within ±15%. The capacitor type X5R
has a similar tolerance over a reduced temperature range of −55°C to +85°C. Many large value ceramic
capacitors, larger than 1 µF are manufactured with Z5U or Y5V temperature characteristics. Their capacitance
can drop by more than 50% as the temperature varies from 25°C to 85°C. Therefore X7R is recommended over
Z5U and Y5V in applications where the ambient temperature will change significantly above or below 25°C.
Tantalum capacitors are less desirable than ceramic for use as output capacitors because they are more
expensive when comparing equivalent capacitance and voltage ratings in the 0.47 µF to 4.7 µF range.
Another important consideration is that tantalum capacitors have higher ESR values than equivalent size
ceramics. This means that while it may be possible to find a tantalum capacitor with an ESR value within the
stable range, it would have to be larger in capacitance (which means bigger and more costly) than a ceramic
capacitor with the same ESR value. It should also be noted that the ESR of a typical tantalum will increase about
2:1 as the temperature goes from 25°C down to −40°C, so some guard band must be allowed.
Input Capacitor Selection for SW1 and SW2
A ceramic input capacitor of 10 µF, 6.3V is sufficient for the magnetic dc/dc converters. Place the input capacitor
as close as possible to the input of the device. A large value may be used for improved input voltage filtering.
The recommended capacitor types are X7R or X5R. Y5V type capacitors should not be used. DC bias
characteristics of ceramic capacitors must be considered when selecting case sizes like 0805 and 0603. The
input filter capacitor supplies current to the PFET switch of the dc/dc converter in the first half of each cycle and
reduces voltage ripple imposed on the input power source. A ceramic capacitor’s low ESR (Equivalent Series
Resistance) provides the best noise filtering of the input voltage spikes due to fast current transients. A capacitor
with sufficient ripple current rating should be selected. The Input current ripple can be calculated as:
Irms = Ioutmax
where
r=
VOUT
VOUT §
r2 ·
1+
VIN ©
VIN
12 ¹
(Vin ± Vout) x Vout
L x f x Ioutmax x Vin
(7)
The worse case is when VIN = 2VOUT
38
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SNVS456M – AUGUST 2006 – REVISED MAY 2013
Output Capacitor Selection for SW1, SW2
A 10 µF, 6.3V ceramic capacitor should be used on the output of the sw1 and sw2 magnetic dc/dc converters.
The output capacitor needs to be mounted as close as possible to the output of the device. A large value may be
used for improved input voltage filtering. The recommended capacitor types are X7R or X5R. Y5V type
capacitors should not be used. DC bias characteristics of ceramic capacitors must be considered when selecting
case sizes like 0805 and 0603. DC bias characteristics vary from manufacturer to manufacturer and DC bias
curves should be requested from them and analyzed as part of the capacitor selection process.
The output filter capacitor of the magnetic dc/dc converter smoothes out current flow from the inductor to the
load, helps maintain a steady output voltage during transient load changes and reduces output voltage ripple.
These capacitors must be selected with sufficient capacitance and sufficiently low ESD to perform these
functions.
The output voltage ripple is caused by the charging and discharging of the output capacitor and also due to its
ESR and can be calculated as follows:
Iripple
Vpp-c =
4xfxC
(8)
Voltage peak-to-peak ripple due to ESR can be expressed as follows:
VPP–ESR = 2 × IRIPPLE × RESR
(9)
Because the VPP-C and VPP-ESR are out of phase, the rms value can be used to get an approximate value of the
peak-to-peak ripple:
Vpp-rms =
Vpp-c2 + Vpp-esr2
(10)
Note that the output voltage ripple is dependent on the inductor current ripple and the equivalent series
resistance of the output capacitor (RESR). The RESR is frequency dependent as well as temperature dependent.
The RESR should be calculated with the applicable switching frequency and ambient temperature.
Min Value
Unit
CLDO1
Capacitor
0.47
µF
LDO1 output capacitor
Description
Ceramic, 6.3V, X5R
Recommended Type
CLDO2
0.47
µF
LDO2 output capacitor
Ceramic, 6.3V, X5R
CSW1
10.0
µF
SW1 output capacitor
Ceramic, 6.3V, X5R
CSW2
10.0
µF
SW2 output capacitor
Ceramic, 6.3V, X5R
I2C Pullup Resistor
Both I2C_SDA and I2C_SCL terminals need to have pullup resistors connected to VINLDO12 or to the power
supply of the I2C master. The values of the pull-up resistors (typ. ∼1.8kΩ) are determined by the capacitance of
the bus. Too large of a resistor combined with a given bus capacitance will result in a rise time that would violate
the max. rise time specification. A too small resistor will result in a contention with the pull-down transistor on
either slave(s) or master.
Operation without I2C Interface
Operation of the LP3906 without the I2C interface is possible if the system can operate with default values for the
LDO and Buck regulators. (See Factory Programmable Options .) The I2C-less system must rely on the correct
default output values of the LDO and Buck converters.
Factory Programmable Options
The following options are EPROM programmed during final test of the LP3906. The system designer that needs
specific options is advised to contact the local Texas Instruments sales office.
Factory programmable options
Enable delay for power on
Current value
code 010 (see BUCK VOLTAGE CHANGE CONTROL REGISTER 1 (VCCR) – 0X20)
SW1 ramp speed
8 mV/µs
SW2 ramp speed
8 mV/µs
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The I2C Chip ID address is offered as a metal mask option. The current value equals 0x60.
MODE BOUNCE
PFM-PWM transition at low load current.
To improve efficiency at lower load currents LP3906 buck converters employ an automatically invoked PFM
mode for the low load operation. The PFM mode operates with a much lower value quiescent current (IQ) than
the PWM mode of operation that is used in the higher load currents.
As shown in the datasheet section about SW operation, there is a DC voltage difference between the two modes
of operation, with Vout PFM being typically 1.2% higher than Vout PWM. So there is a DC voltage level transition
and some associated dynamic perturbation at the mode transition point.
The transition between the two modes of operation has an associated hysterisis in the transition current value,
That is, the transition point for increasing current (PFM to PWM) is at a higher value that the decreasing current
(PWM to PFM). This hysterisis is to ensure that in the event that the load current values equals the PFM PWM
transition value, the device will not make multiple transitions between modes; this reduces the noise at this load
by eliminating multiple transitions between modes (also known as mode bounce).
Under some conditions of high Vin and Low Vout the hystersis value is reduced and some amount of mode
bounce can occur. Under these conditions, the regulator still maintains DC regulation, however the output ripple
is more pronounced. Refer to the attached Vout vs Vin chart below that shows the operational area that may
exhibit this increased output ripple. If the application is expected to be operated in the area of concern AND have
a static load current of the transition current value, the user can avoid the possible noise increase by invoking the
components’ “Force PWM mode”.
3.5
3.0
VOUT (V)
2.5
2.0
1.5
1.0
0
3.5
4.0
4.5
5.0
5.5
VIN (V)
Figure 40. LP3906 Buck Converter
VOUT vs VIN Operating Range
during PFM-PWM-PPM Transition
40
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SNVS456M – AUGUST 2006 – REVISED MAY 2013
REVISION HISTORY
Changes from Revision L (May 2013) to Revision M
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 40
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LP3906SQ-DJXI/NOPB
ACTIVE
WQFN
NHZ
24
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
06-DJXI
LP3906SQ-FXPI/NOPB
ACTIVE
WQFN
NHZ
24
1000
RoHS & Green
SN
Level-1-260C-UNLIM
LP3906SQ-JXXI/NOPB
ACTIVE
WQFN
NHZ
24
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
06-JXXI
LP3906SQ-VPFP/NOPB
ACTIVE
WQFN
NHZ
24
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
06-VPFP
06-FXPI
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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