OPA381
OPA2381
SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
Precision, Low Power, 18MHz
Transimpedance Amplifier
FEATURES
DESCRIPTION
D OVER 250kHz TRANSIMPEDANCE
D
D
D
D
D
D
D
D
D
D
D
D
The OPA381 family of transimpedance amplifiers provides
18MHz of Gain Bandwidth (GBW), with extremely high
precision, excellent long-term stability, and very low 1/f noise.
The OPA381 features an offset voltage of 25µV (max), offset
drift of 0.1µV/°C (max), and bias current of 3pA. The OPA381
far exceeds the offset, drift, and noise performance that
conventional JFET op amps provide.
BANDWIDTH
DYNAMIC RANGE: 5 Decades
EXCELLENT LONG-TERM STABILITY
LOW VOLTAGE NOISE: 10nV/√Hz
BIAS CURRENT: 3pA
OFFSET VOLTAGE: 25µV (max)
OFFSET DRIFT: 0.1µV/°C (max)
GAIN BANDWIDTH: 18MHz
QUIESCENT CURRENT: 800µA
FAST OVERLOAD RECOVERY
SUPPLY RANGE: 2.7V to 5.5V
SINGLE AND DUAL VERSIONS
MicroPACKAGE: DFN-8, MSOP-8
The signal bandwidth of a transimpedance amplifier depends
largely on the GBW of the amplifier and the parasitic
capacitance of the photodiode, as well as the feedback
resistor. The 18MHz GBW of the OPA381 enables a transimpedance bandwidth of > 250kHz in most configurations.
The OPA381 is ideally suited for fast control loops for power
level measurement on an optical fiber.
As a result of the high precision and low-noise characteristics
of the OPA381, a dynamic range of 5 decades can be
achieved. This capability allows the measurement of signal
currents on the order of 10nA, and up to 1mA in a single I/V
conversion stage. In contrast to logarithmic amplifiers, the
OPA381 provides very wide bandwidth throughout the full
dynamic range. By using an external pulldown resistor to
–5V, the output voltage range can be extended to include 0V.
APPLICATIONS
D
D
D
D
D
PRECISION I/V CONVERSION
PHOTODIODE MONITORING
OPTICAL AMPLIFIERS
CAT-SCANNER FRONT-END
PHOTO LAB EQUIPMENT
The OPA381 and OPA2381 are both available in MSOP-8
and DFN-8 (3mm x 3mm) packages. They are specified
from –40°C to +125°C.
RF
+5V
OPA381 RELATED DEVICES
7
OPA381
2
6
VOUT
(0V to 4.4V)
C DIO DE
RP
(Optional
Pulldown
Resistor)
Photodiode
1MΩ
65pF
−5V
100kΩ
3
75pF
PRODUCT
FEATURES
OPA380
90MHz GBW, 2.7V to 5.5V Supply
Transimpedance Amplifier
OPA132
16MHz GBW, Precision FET Op Amp ±15V
OPA300
150MHz GBW, Low-Noise, 2.7V to 5.5V Supply
OPA335
10µV VOS, Zero-Drift, 2.5V to 5V Supply
OPA350
500µV VOS, 38MHz, 2.5V to 5V Supply
OPA354
100MHz GBW CMOS, RRIO, 2.5V to 5V Supply
OPA355
200MHz GBW CMOS, 2.5V to 5V Supply
OPA656/7
230MHz, Precision FET, ±5V
4
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright 2004, Texas Instruments Incorporated
! !
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
ABSOLUTE MAXIMUM RATINGS(1)
ELECTROSTATIC DISCHARGE SENSITIVITY
Voltage Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +7V
Signal Input Terminals(2), Voltage . . . . . (V−) −0.5V to (V+) + 0.5V
Current . . . . . . . . . . . . . . . . . . . . . ±10mA
Short-Circuit Current(3) . . . . . . . . . . . . . . . . . . . . . . . . Continuous
Operating Temperature Range . . . . . . . . . . . . . . . −40°C to +125°C
Storage Temperature Range . . . . . . . . . . . . . . . . . −65°C to +150°C
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
Lead Temperature (soldering, 10s) . . . . . . . . . . . . . . . . . . . . . +300°C
OPA381 ESD Rating (Human Body Model) . . . . . . . . . . . . . . . 2000V
OPA2381 ESD Rating (Human Body Model) . . . . . . . . . . . . . . 1500V
(1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods
may degrade device reliability. These are stress ratings only, and
functional operation of the device at these or any other conditions
beyond those specified is not implied.
(2) Input terminals are diode clamped to the power-supply rails. Input
signals that can swing more than 0.5V beyond the supply rails
should be current limited to 10mA or less.
(3) Short-circuit to ground; one amplifier per package.
This integrated circuit can be damaged by ESD. Texas
Instruments recommends that all integrated circuits be
handled with appropriate precautions. Failure to observe
proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could
cause the device not to meet its published specifications.
PACKAGE/ORDERING INFORMATION(1)
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
OPA381
MSOP-8
DGK
−40°C to +125°C
A64
OPA381
DFN-8
DRB
−40°C to +125°C
A65
OPA2381
MSOP-8
DGK
−40°C to +125°C
A62
OPA2381
DFN-8
DRB
−40°C to +125°C
A63
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
OPA381AIDGKT
OPA381AIDGKR
OPA381AIDRBT
OPA381AIDRBR
OPA2381AIDGKT
OPA2381AIDGKR
OPA2381AIDRBT
OPA2381AIDRBR
Tape and Reel, 250
Tape and Reel, 2500
Tape and Reel, 250
Tape and Reel, 3000
Tape and Reel, 250
Tape and Reel, 2500
Tape and Reel, 250
Tape and Reel, 3000
(1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet.
PIN ASSIGNMENTS
Top View
OPA381
OPA381
NC (1 )
1
8
NC (1 )
−In
2
7
+In
3
V−
4
NC (1 )
1
V+
−In
2
6
Out
+In
3
5
NC (1 )
V−
4
Exposed
Thermal
Die Pad
on
Underside
8 NC (1 )
7 V+
6 Out
5 NC (1 )
DFN−8
MSOP−8
NOTE: (1) NC indicates no internal connection.
OPA2381
Out A
1
8
V+
Out A
1
− In A
2
7
Out B
−In A
2
+In A
3
6
− In B
+In A
3
V−
4
5
+In B
V−
4
MSOP−8
2
OPA2381
Exposed
Thermal
Die Pad
on
Underside
DFN−8
8 V+
7 Out B
6 −In B
5 +In B
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
ELECTRICAL CHARACTERISTICS: VS = +2.7V to +5.5V
Boldface limits apply over the temperature range, TA = −40°C to +125°C.
All specifications at TA = +25°C, RL = 10kΩ connected to VS/2, and VOUT = VS/2, unless otherwise noted.
OPA381
PARAMETER
OFFSET VOLTAGE
Input Offset Voltage
Drift
vs Power Supply
Over Temperature
Long-Term Stability(1)
Channel Separation, dc
INPUT BIAS CURRENT
Input Bias Current
Over Temperature
Input Offset Current
NOISE
Input Voltage Noise, f = 0.1Hz to 10Hz
Input Voltage Noise Density, f = 10kHz
Input Voltage Noise Density, f > 1MHz
Input Current Noise Density, f = 10kHz
INPUT VOLTAGE RANGE
Common-Mode Voltage Range
Common-Mode Rejection Ratio
CONDITION
VOS
dVOS/dT
PSRR
MIN
VS = +5V, VCM = 0V
VS = +2.7V to +5.5V, VCM = 0V
VS = +2.7V to +5.5V, VCM = 0V
IB
VCM = VS/2
IOS
VCM = VS/2
en
en
en
in
OUTPUT
Voltage Output Swing from Positive Rail
Voltage Output Swing from Negative Rail
Voltage Output Swing from Positive Rail
Voltage Output Swing from Negative Rail
Output Current
Short-Circuit Current
Capacitive Load Drive
Open-Loop Output Impedance
POWER SUPPLY
Specified Voltage Range
Quiescent Current (per amplifier)
Over Temperature
TEMPERATURE RANGE
Specified and Operating Range
Storage Range
Thermal Resistance
MSOP-8
DFN-8
UNITS
7
0.03
3.5
25
0.1
20
20
µV
µV/°C
µV/V
µV/V
VCM
CMRR
VS = +5V, (V−) < VCM < (V+) − 1.8V
AOL
0.05V < VO < (V+) − 0.6V, VCM = VS/2, VS = 5V
GBW
SR
V−
95
IOUT
ISC
CLOAD
RO
VS
IQ
110
106
F = 1MHz, IO = 0
pA
110
(V+) − 1.8V
V
dB
1
1013|| 2.5
pF
Ω || pF
135
135
dB
dB
18
12
7
7
200
MHz
V/µs
µs
µs
ns
400
600
30
50
400
600
−20
0
10
20
See Typical Characteristics
250
2.7
IO = 0A
pA
µVPP
nV/√Hz
nV/√Hz
fA/√Hz
3
70
10
20
G = +1
VS = +5V, 4V Step, G = +1, OPA381
VS = +5V, 4V Step, G = +1, OPA2381
VIN • G = > VS
RL = 10kΩ
RL = 10kΩ
RP = 10kΩ to −5V(2)
RP = 10kΩ to −5V(2)
µV/V
3
±50
See Typical Characteristics
6
±100
VS = +5V, VCM = 0V
VS = +5V, VCM = 0V
VS = +5V, VCM = 0V
VS = +5V, VCM = 0V
0V < VO < (V+) − 0.6V, VCM = 0V, RP = 10kΩ to −5V(2), VS = 5V
FREQUENCY RESPONSE
Gain-Bandwidth Product
Slew Rate
Settling Time, 0.0015%(3)
Settling Time, 0.003%(3)
Overload Recovery Time(4), (5)
MAX
See Note (1)
1
INPUT IMPEDANCE
Differential Capacitance
Common-Mode Resistance and Capacitance
OPEN-LOOP GAIN
Open-Loop Voltage Gain
TYP
0.8
−40
−65
mV
mV
mV
mV
mA
mA
Ω
5.5
1
1.1
V
mA
mA
+125
+150
°C
°C
qJA
150
65
°C/W
°C/W
(1) High temperature operating life characterization of zero-drift op amps applying the techniques used in the OPA381 have repeatedly demonstrated randomly
distributed variation approximately equal to measurement repeatability of 1µV. This consistency gives confidence in the stability and repeatability of these zerodrift techniques.
(2) Tested with output connected only to R , a pulldown resistor connected between V
P
OUT and −5V, as shown in Figure 3. See also Applications section, Achieving
Output Swing to Negative Rail.
(3) Transimpedance frequency of 250kHz.
(4) Time required to return to linear operation.
(5) From positive rail.
3
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
TYPICAL CHARACTERISTICS: VS = +2.7V to +5.5V
All specifications at TA = +25°C, RL = 10kΩ connected to VS/2, and VOUT = VS/2, unless otherwise noted.
POWER−SUPPLY REJECTION RATIO AND
COMMON−MODE REJECTION vs FREQUENCY
OPEN−LOOP GAIN AND PHASE vs FREQUENCY
Open−Loop Gain (dB)
120
200
140
150
120
100
100
80
50
60
0
40
−50
Gain
−100
20
0
−150
−20
−200
10
100
1k
10k
100k
1M
10M
100
PSRR, CMRR (dB)
Phase
Phase (_ )
140
PSRR
80
60
40
20
0
CMRR
−20
−40
−60
100M
10
100
1k
Frequency (Hz)
PHASE MARGIN vs LOAD CAPACITANCE
100k
1M
10M
100M
QUIESCENT CURRENT vs TEMPERATURE
90
1.00
RS = 100Ω
70
0.90
Quiescent Current (mA)
80
Phase Margin (_)
10k
Frequency (Hz)
100pF
60
50kΩ
RS
50
CL
RS = 50Ω
40
30
RS = 0Ω
20
0.85
0.80
0.75
5.5V
0.70
0.65
2.7V
0.60
0.55
10
0.50
0
100
200 300 400 500 600 700
800 900 1000
−40 −25
0
CL Load Capacitance (pF)
25
50
75
100
125
Temperature (_ C)
QUIESCENT CURRENT vs SUPPLY VOLTAGE
INPUT BIAS CURRENT vs TEMPERATURE
1.00
1000
0.85
Input Bias Current (pA)
Quiescent Current (mA)
0.90
0.80
0.75
0.70
0.65
0.60
100
10
0.55
1
0.50
2.7
3.1
3.5
3.9
4.3
Supply Voltage (V)
4
4.7
5.1
5.5
−40 −25
0
25
50
Temperature (_ C)
75
100
125
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
TYPICAL CHARACTERISTICS: VS = +2.7V to +5.5V (continued)
All specifications at TA = +25°C, RL = 10kΩ connected to VS/2, and VOUT = VS/2, unless otherwise noted.
OUTPUT VOLTAGE SWING vs OUTPUT CURRENT
(VS = 5.5V)
INPUT BIAS CURRENT
vs COMMON−MODE VOLTAGE
(V+)
50
(V+) − 1
30
20
Output Swing (V)
Input Bias Current (pA)
40
−IB
10
0
−10
+IB
−20
−30
(V+) − 2
+125_ C
+25°C
(V−) + 2
−40_ C
(V−) + 1
−40
−50
(V−)
0
0.5
1.0
1.5
2.0
2.5
3.0
0
3.5
5
15
20
25
OFFSET VOLTAGE DRIFT
PRODUCTION DISTRIBUTION
OUTPUT VOLTAGE SWING vs OUTPUT CURRENT
(VS = 2.7V)
(V+)
10
Output Current (mA)
Common−Mode Voltage (V)
(V+) − 0.35
(V+) −1.05
Population
(V+) −1.40
+125_ C
+25_ C
(V−) + 1.40
−40_C
(V−) + 1.05
(V−) + 0.70
(V−) + 0.35
(V−)
5
10
15
20
25
−0.10
−0.09
−0.08
−0.07
−0.06
−0.05
−0.04
−0.03
−0.02
−0.01
0.00
0.01
0.02
0.03
0.04
0.05
0.06
0.07
0.08
0.09
0.10
0
Output Current (mA)
Offset Voltage Drift (µV/_C)
OFFSET VOLTAGE PRODUCTION DISTRIBUTION
GAIN BANDWIDTH vs POWER SUPPLY VOLTAGE
20
Gain Bandwidth (MHz)
19
Population
18
17
16
15
14
13
25.00
20.00
15.00
10.00
5.00
0.00
−5.00
−10.00
−15.00
−20.00
12
−25.00
Output Swing (V)
(V+) − 0.70
2.5
3.0
3.5
4.0
4.5
5.0
5.5
Power Supply Voltage (V)
Offset Voltage (µV)
5
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
TYPICAL CHARACTERISTICS: VS = +2.7V to +5.5V (continued)
All specifications at TA = +25°C, RL = 10kΩ connected to VS/2, and VOUT = VS/2, unless otherwise noted.
CF
RF
CSTRAY
OPA381
Transimpedance Gain (V/A in dB)
Circuit for Transimpedance Amplifier Characteristic curves on this page.
6
TRANSIMPEDANCE AMP CHARACTERISTIC
150
CDIODE = 10pF
140
130 RF = 10MΩ
120
110 RF = 1MΩ
CF = 0.5pF
100
90 RF = 100kΩ
CF = 2pF
80
70 RF = 10kΩ
60
CF = 4pF
50
40
30
20 CSTRAY (parasitic) = 0.2pF
10
100
1k
10k
100k
1M
10M
100M
Frequency (Hz)
Transimpedance Gain (V/A in dB)
TRANSIMPEDANCE AMP CHARACTERISTIC
150
CDIODE = 50pF
140
130 RF = 10MΩ
120
CF = 1pF
110 RF = 1MΩ
100
CF = 3pF
90 R = 100kΩ
F
80
70 R = 10kΩ
CF = 8pF
F
60
50
40
30
20 CSTRAY (parasitic) = 0.2pF
10
100
1k
10k
100k
1M
10M
100M
Frequency (Hz)
Transimpedance Gain (V/A in dB)
Transimpedance Gain (V/A in dB)
Transimpedance Gain (V/A in dB)
CDIODE
TRANSIMPEDANCE AMP CHARACTERISTIC
150
CDIODE = 100pF
140
130 RF = 10MΩ C = 0.5pF
F
120
CF = 1pF
110 RF = 1MΩ
100
CF = 4pF
90 R = 100kΩ
F
80
70 R = 10kΩ
CF = 12pF
F
60
50
40
30
20 CSTRAY (parasitic) = 0.2pF
10
100
1k
10k
100k
1M
10M
100M
Frequency (Hz)
TRANSIMPEDANCE AMP CHARACTERISTIC
150
CDIODE = 20pF
140
130 RF = 10MΩ
120
110 RF = 1MΩ
CF = 0.5pF
100
90 RF = 100kΩ
CF = 2pF
80
70 RF = 10kΩ
CF = 5pF
60
50
40
30
20 CSTRAY (parasitic) = 0.2pF
10
100
1k
10k
100k
1M
10M
100M
Frequency (Hz)
TRANSIMPEDANCE AMP CHARACTERISTIC
150
CDIODE = 1pF
140
130 RF = 10MΩ
120
110 RF = 1MΩ
100
CF = 0.5pF
90 RF = 100kΩ
80
70 RF = 10kΩ
CF = 2pF
60
50
40
30
20 C
STRAY (parasitic) = 0.2pF
10
100
1k
10k
100k
1M
10M
100M
Frequency (Hz)
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
TYPICAL CHARACTERISTICS: VS = +2.7V to +5.5V (continued)
All specifications at TA = +25°C, and RL = 10kΩ connected to VS/2, unless otherwise noted.
LARGE−SIGNAL STEP RESPONSE
(with pull−down)
SMALL−SIGNAL STEP RESPONSE
(with or without pull−down)
200kHz (CF = 16pF)
1MHz
(CF = 3pF)
1V/div
50mV/div
CF
50kΩ
3pF
50kΩ
OPA381
O PA3 81
10kΩ
10kΩ
VP = 0V or −5V
VP
−5V
Time (100ns/div)
Time (100ns/div)
OVERLOAD RECOVERY
LARGE−SIGNAL STEP RESPONSE
(without pull−down)
6
VOUT (V/div)
200kHz
(CF = 16pF)
1MHz
(CF = 3pF)
20kΩ
4
250µA
2
10kΩ
Nonlinear Linear
Operation Operation
VP
0
50kΩ
IIN (mA/div)
OPA381
OPA381
10kΩ
0.8
VP = 0V or −5V
OPA2381
I IN
0
0
100 200 300 400 500 600 700 800 900 1000
Time (100ns/div)
Time (ns)
CHANNEL SEPARATION vs INPUT FREQUENCY
INPUT VOLTAGE NOISE SPECTRAL DENSITY
160
1000
Channel Separation (dB)
140
Input Voltage Noise (nV/√(Hz)
OPA381
IIN
CF
1V/div
40pF
VOUT
100
10
OPA2381
120
100
80
60
40
20
0
−20
−40
1
10
100
1k
10k
100k
Frequency (Hz)
1M
10M
10
100
1k
10k
100k
1M
10M
100M
Input Frequency (Hz)
7
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
APPLICATIONS INFORMATION
BASIC OPERATION
The OPA381 is a high-precision transimpedance
amplifier with very low 1/f noise. Due to its unique
architecture, the OPA381 has excellent long-term input
voltage offset stability.
The OPA381 performance results from an internal
auto-zero amplifier combined with a high-speed
amplifier. The OPA381 has been designed with circuitry
to improve overload recovery and settling time over that
achieved by a traditional composite approach. It has
been specifically designed and characterized to
accommodate circuit options to allow 0V output
operation (see Figure 3).
The OPA381 is used in inverting configurations, with the
noninverting input used as a fixed biasing point.
Figure 1 shows the OPA381 in a typical configuration.
Power-supply pins should be bypassed with 1µF
ceramic or tantalum capacitors. Electrolytic capacitors
are not recommended.
CF
OPERATING VOLTAGE
OPA381 series op amps are fully specified from 2.7V to
5.5V over a temperature range of −40°C to +125°C.
Parameters that vary significantly with operating
voltages or temperature are shown in the Typical
Characteristics.
INTERNAL OFFSET CORRECTION
The OPA381 series op amps use an auto-zero topology
with a time-continuous 18MHz op amp in the signal
path. This amplifier is zero-corrected every 100µs using
a proprietary technique. Upon power-up, the amplifier
requires approximately 400µs to achieve specified VOS
accuracy, which includes one full auto-zero cycle of
approximately 100µs and the start-up time for the bias
circuitry. Prior to this time, the amplifier will function
properly but with unspecified offset voltage.
This design has virtually no aliasing and low noise. Zero
correction occurs at a 10kHz rate, but there is virtually
no fundamental noise energy present at that frequency
due to internal filtering. For all practical purposes, any
glitches have energy at 20MHz or higher and are easily
filtered, if necessary. Most applications are not sensitive
to such high-frequency noise, and no filtering is
required.
RF
INPUT VOLTAGE
+5V
1µF
λ
OPA381
VOUT(1)
(0.5V to 4.4V)
The input common-mode voltage range of the OPA381
series extends from V− to (V+) –1.8V. With input signals
above this common-mode range, the amplifier will no
longer provide a valid output value, but it will not latch
or invert.
VBIAS = 0.5V
INPUT OVERVOLTAGE PROTECTION
NOTE: (1) VOUT = 0.5V in dark conditions.
Figure 1. OPA381 Typical Configuration
8
Device inputs are protected by ESD diodes that will
conduct if the input voltages exceed the power supplies
by more than approximately 500mV. Momentary
voltages greater than 500mV beyond the power supply
can be tolerated if the current is limited to 10mA. The
OPA381 family features no phase inversion when the
inputs extend beyond supplies if the input is current
limited.
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OUTPUT RANGE
ACHIEVING OUTPUT SWING TO GROUND
The OPA381 is specified to swing within at least 600mV
of the positive rail and 50mV of the negative rail with a
10kΩ load while maintaining good linearity. Swing to the
negative rail while maintaining linearity can be extended
to 0V—see the section, Achieving Output Swing to
Ground. See the Typical Characteristic curve, Output
Voltage Swing vs Output Current.
Some applications require output voltage swing from
0V to a positive full-scale voltage (such as +4.096V)
with excellent accuracy. With most single-supply op
amps, problems arise when the output signal
approaches 0V, near the lower output swing limit of a
single-supply op amp. A good single-supply op amp
may swing close to single-supply ground, but will not
reach 0V.
The OPA381 can swing slightly closer than specified to
the positive rail; however, linearity will decrease and a
high-speed overload recovery clamp limits the amount
of positive output voltage swing available—see
Figure 2.
The output of the OPA381 can be made to swing to 0V,
or slightly below, on a single-supply power source. This
extended output swing requires the use of another
resistor and an additional negative power supply. A
pulldown resistor may be connected between the
output and the negative supply to pull the output down
to 0V; see Figure 3.
25
20
VS = 5.5V
15
VOS (µV)
10
RF
5
λ
0
V+ = +5V
−5
−10
−15
OPA381
R P = 10kΩ to −5V
R L = 10kΩ to VS/2
−20
−25
−1
0
1
2
3
4
5
VOUT
R P = 10kΩ
V− = Gnd
6
VOUT (V)
RP =
−VS
500µA
−VS = −5V
Negative Supply
Figure 2. Effect of High-Speed Overload
Recovery Clamp on Output Voltage
OVERLOAD RECOVERY
The OPA381 has been designed to prevent output
saturation. After being overdriven to the positive rail, it
will typically require only 200ns to return to linear
operation. The time required for negative overload
recovery is greater, unless a pulldown resistor
connected to a more negative supply is used to extend
the output swing all the way to the negative rail—see the
following section, Achieving Output Swing to Ground.
Figure 3. Amplifier with Pull-Down Resistor to
Achieve VOUT = 0V
The OPA381 has an output stage that allows the output
voltage to be pulled to its negative supply rail using this
technique. However, this technique only works with
some types of output stages. The OPA381 has been
designed to perform well with this method. Accuracy is
excellent down to 0V. Reliable operation is assured over
the specified temperature range.
9
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
BIASING PHOTODIODES IN SINGLE-SUPPLY
CIRCUITS
The +IN input can be biased with a positive DC voltage
to offset the output voltage and allow the amplifier
output to indicate a true zero photodiode measurement
when the photodiode is not exposed to any light. It will
also prevent the added delay that results from coming
out of the negative rail. This bias voltage appears
across the photodiode, providing a reverse bias for
faster operation. An RC filter placed at this bias point will
reduce noise. (Refer to Figure 4.) This bias voltage can
also serve as an offset bias point for an ADC with range
that does not include ground.
D the desired transimpedance gain (RF);
D the Gain Bandwidth Product (GBW) for the
OPA381 (18MHz).
With these three variables set, the feedback capacitor
value (CF) can be set to control the frequency response.
CSTRAY is the stray capacitance of RF, which is 0.2pF for
a typical surface-mount resistor.
To achieve a maximally flat 2nd-order Butterworth
frequency response, the feedback pole should be set
to:
1
+
2pR FǒCF ) CSTRAYǓ
Ǹ4pRGBWC
F
(1)
TOT
Bandwidth is calculated by:
CF(1)
< 1pF
f *3dB +
Ǹ2pRGBWC
F
RF
10MΩ
ID
OPA381
VOUT = IDRF + VBIAS
0.1µF
(2)
These
equations
will
result
in
maximum
transimpedance bandwidth. For even higher
transimpedance bandwidth, the high-speed CMOS
OPA380 (90MHz GBW), the OPA300 (150MHz GBW),
or the OPA656 (230MHz GBW) may be used.
V+
λ
Hz
TOT
100kΩ
For additional information, refer to Application Bulletin
AB−050 (SBOA055), Compensate Transimpedance
Amplifiers Intuitively, available for download at
www.ti.com.
+VBIAS
[0V to (V+) − 1.8V]
CF(1)
NOTE: (1) CF is optional to prevent gain peaking.
It includes the stray capacitance of RF.
RF
10MΩ
Figure 4. Photodiode with Filtered Reverse Bias
Voltage
CSTRAY(2)
+5V
TRANSIMPEDANCE AMPLIFIER
Wide bandwidth, low input bias current and low input
voltage and current noise make the OPA381 an ideal
wideband photodiode transimpedance amplifier. Low
voltage noise is important because photodiode
capacitance causes the effective noise gain of the
circuit to increase at high frequency.
The key elements to a transimpedance design are
shown in Figure 5:
D the total input capacitance (CTOT), consisting of the
photodiode capacitance (CDIODE) plus the parasitic
common-mode and differential-mode input
capacitance (2.5pF + 1pF for the OPA381);
10
λ
CTOT(3)
VOUT
OPA381
RP (optional
pulldown resistor)
− 5V
NOTE: (1) CF is optional to prevent gain peaking.
(2) CSTRAY is the stray capacitance of RF
(typically, 0.2pF for a surface−mount resistor).
(3) CTOT is the photodiode capacitance plus OPA381
input capacitance.
Figure 5. Transimpedance Amplifier
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
TRANSIMPEDANCE BANDWIDTH AND
NOISE
RF = 50kΩ
(a)
Limiting the gain set by RF can decrease the noise
occurring at the output of the transimpedance circuit.
However, all required gain should occur in the
transimpedance stage, since adding gain after the
transimpedance amplifier generally produces poorer
noise performance. The noise spectral density
produced by RF increases with the square-root of RF,
whereas the signal increases linearly. Therefore,
signal-to-noise ratio is improved when all the required
gain is placed in the transimpedance stage.
C STRAY = 0.2pF
λ
RF = 50kΩ
(b)
CSTRAY = 0.2pF
CF = 16pF
λ
OPA381
VOUT
VBIAS
RF = 50kΩ
(c)
CSTRAY = 0.2pF
Using RDIODE to represent the equivalent diode
resistance, and CTOT for equivalent diode capacitance
plus OPA381 input capacitance, the noise zero, fZ, is
calculated by:
ǒRDIODE ) RFǓ
fZ +
2pRDIODER FǒC TOT ) C FǓ
VOUT
VBIAS
Total noise increases with increased bandwidth. Limit
the circuit bandwidth to only that required. Use a
capacitor, CF, across the feedback resistor, RF, to limit
bandwidth (even if not required for stability), if total
output noise is a concern.
Figure 6a shows the transimpedance circuit without any
feedback capacitor. The resulting transimpedance gain
of this circuit is shown in Figure 7. The –3dB point is
approximately 3MHz. Adding a 16pF feedback
capacitor (Figure 6b) will limit the bandwidth and result
in a –3dB point at approximately 200kHz (seen in
Figure 7). Output noise will be further reduced by
adding a filter (RFILTER and CFILTER) to create a second
pole (Figure 6c). This second pole is placed within the
feedback loop to maintain the amplifier’s low output
impedance. (If the pole was placed outside the
feedback loop, an additional buffer would be required
and would inadvertently increase noise and dc error).
OPA381
CF = 22pF
RFILTER
= 102kΩ
λ
OPA381
VOUT
CFILTER
= 3.9nF
(3)
VBIAS
Figure 6. Transimpedance Circuit Configurations
with Varying Total and Integrated Noise Gain
11
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
120
500
Integrated Output Noise (µVrms)
Transimpedance Gain (dB)
100
−3dB at 200kHz
80
See Figure 6c
60
40
20
0
−20
100
CDIODE = 10pF
See Figure 6a
CDIODE = 10pF
400
310µVrms
300
See Figure 6a
200
68µVrms
25µVrms
See Figure 6b
100
See Figure 6c
See Figure 6b
0
1k
10k
100k
1M
10M
100M
100
1k
10k
Frequency (Hz)
100k
1M
10M
100M
Frequency (Hz)
Figure 7. Transimpedance Gains for Circuits in
Figure 6
Figure 9. Integrated Output Noise for Circuits in
Figure 6
The effects of these circuit configurations on output
noise are shown in Figure 8 and on integrated output
noise in Figure 9. A 2-pole Butterworth filter (maximally
flat in passband) is created by selecting the filter values
using the equation:
Figure 10 shows the effects of diode capacitance on
integrated output noise, using the circuit in Figure 6c.
(4)
The circuit in Figure 6b rolls off at 20dB/decade. The
circuit with the additional filter shown in Figure 6c rolls
off at 40dB/decade, resulting in improved noise
performance.
400
Output Noise (µV/√Hz)
CDIODE = 10pF
300
CDIODE
= 100pF
50
56µVrms
CDIODE
= 50pF
37µVrms
40
CDIODE
= 20pF
30
28µVrms
20
CDIODE
= 1pF
See Figure 6c
10
CDIODE
= 10pF
See Figure 6a
25µVrms
23µVrms
0
200
1
10
100
1k
10k
100k
1M
10M
100M
Frequency (Hz)
100
0
100
See Figure 6b
Figure 10. Integrated Output Noise for Various
Values of CDIODE for Circuit in Figure 6c
See Figure 6c
1k
10k
100k
1M
10M
100M
Frequency (Hz)
Figure 8. Output Noise for Circuits in Figure 6
12
60
Integrated Output Noise (µVrms)
C FRF + 2C FILTERR FILTER
For additional information, refer to Noise Analysis of
FET Transimpedance Amplifiers (SBOA060), and
Noise Analysis for High Speed Op Amps (SBOA066),
available for download from the TI web site.
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
BOARD LAYOUT
CAPACITIVE LOAD AND STABILITY
Minimize photodiode capacitance and stray
capacitance at the summing junction (inverting input).
This capacitance causes the voltage noise of the op
amp to be amplified (increasing amplification at high
frequency). Using a low-noise voltage source to
reverse-bias a photodiode can significantly reduce its
capacitance. Smaller photodiodes have lower
capacitance. Use optics to concentrate light on a small
photodiode.
The OPA381 series op amps can drive greater than
100pF pure capacitive load. Increasing the gain
enhances the amplifier’s ability to drive greater
capacitive loads. See the Phase Margin vs Load
Capacitance typical characteristic curve.
Circuit board leakage can degrade the performance of
an otherwise well-designed amplifier. Clean the circuit
board carefully. A circuit board guard trace that
encircles the summing junction and is driven at the
same voltage can help control leakage. See Figure 11.
One method of improving capacitive load drive in the
unity-gain configuration is to insert a 10Ω to 20Ω
resistor inside the feedback loop, as shown in
Figure 12. This reduces ringing with large capacitive
loads while maintaining DC accuracy.
RF
CF(3)
RF
V+
λ
OPA381
VOUT
RS
20Ω
λ
VOUT
OPA381
VB(1)
CL
V−
Guard ring
Figure 11. Connection of Input Guard
RL
VPD(2)
NOTES: (1) VB = GND or pedestal voltage to reverse bias the photodiode.
(2) VPD = GND or −5V.
(3) CF x RF ≥ 2CL x RS.
OTHER WAYS TO MEASURE SMALL
CURRENTS
Logarithmic amplifiers are used to compress extremely
wide dynamic range input currents to a much narrower
range. Wide input dynamic ranges of 8 decades, or
100pA to 10mA, can be accommodated for input to a
12-bit ADC. (Suggested products: LOG101, LOG102,
LOG104, LOG112.)
Extremely small currents can be accurately measured
by integrating currents on a capacitor. (Suggested
product: IVC102.)
Low-level currents can be converted to high-resolution
data words. (Suggested product: DDC112.)
For further information on the range of products
available, search www.ti.com using the above specific
model names or by using keywords transimpedance
and logarithmic.
Figure 12. Series Resistor in Unity-Gain Buffer
Configuration Improves Capacitive Load Drive
DRIVING 16-BIT ANALOG-TO-DIGITAL
CONVERTERS (ADC)
The OPA381 series is optimized for driving a 16-bit ADC
such as the ADS8325. The OPA381 op amp buffers the
converter input capacitance and resulting charge
injection while providing signal gain. Figure 13 shows
the OPA381 in a single-ended method of interfacing the
ADS8325 16-bit, 100kSPS ADC. For additional
information, refer to the ADS8325 data sheet.
13
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SBOS313B − AUGUST 2004 − REVISED NOVEMBER 2004
CF
RF
SW1
100Ω
ADS8325
OPA381
C1
1µF
R1
1MΩ
1nF
VIN
V+
1µF
RC values shown are optimized for the
ADS8325 values may vary for other ADCs.
OPA381
Figure 13. Driving 16-Bit ADCs
VOUT
VBIAS
INVERTING AMPLIFIER
Its excellent dc precision characteristics make the
OPA381 also useful as an inverting amplifier. Figure 14
shows it configured for use on a single-supply set to a
gain of 10.
Figure 15. Precision Integrator
CF
DFN (DRB) THERMALLYENHANCED PACKAGE
R1
100kΩ
R2
10kΩ
V+
VIN
OPA381
VBIAS
VOUT =
VBIAS −
R2
R1
x VIN
Figure 14. Inverting Gain
PRECISION INTEGRATOR
With its low offset voltage, the OPA381 is well-suited for
use as an integrator. Some applications require a
means to reset the integration. The circuit shown in
Figure 15 uses a mechanical switch as the reset
mechanism. The switch is opened at the beginning of
the integration period. It is shown in the open position,
which is the integration mode. With the values of R1 and
C1 shown, the output changes −1V/s per volt of input.
14
One of the package options for the OPA381 and
OPA2381 is the DFN-8 package, a thermally-enhanced
package designed to eliminate the use of bulky heat
sinks and slugs traditionally used in thermal packages.
The absence of external leads eliminates bent-lead
concerns and issues.
Although the power dissipation requirements of a given
application might not require a heat sink, for mechanical
reliability, the exposed power pad must be soldered to
the board and connected to V− (pin 4). This package
can be easily mounted using standard PCB assembly
techniques. See Application Note SLUA271, QFN/SON
PCB Attachment, located at www.ti.com. These DFN
packages have reliable solderability with either SnPb or
Pb-free solder paste.
PACKAGE OPTION ADDENDUM
www.ti.com
28-Apr-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
OPA2381AIDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
A62
OPA2381AIDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
A62
OPA2381AIDRBT
ACTIVE
SON
DRB
8
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
A63
OPA381AIDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
A64
OPA381AIDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
A64
OPA381AIDRBT
ACTIVE
SON
DRB
8
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
A65
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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