®
OPA2607
OPA
260
7
OPA
260
7
For most current data sheet and other product
information, visit www.burr-brown.com
Dual, High Output, Current-Feedback
OPERATIONAL AMPLIFIER
TM
FEATURES
DESCRIPTION
● WIDEBAND ±12V OPERATION: 25MHz (G = +8)
● UNITY GAIN STABLE: 35MHz (G = +1)
The OPA2607 provides a high output voltage swing and low
distortion required for low turns ratio ADSL upstream driver
applications. Operating on a ±12V supply, the OPA2607 consumes a low 8.0mA/channel quiescent current to deliver a very
high 250mA peak output current. Guaranteed output current of
180mA supports even the most demanding ADSL CPE requirements with low harmonic distortion. Differential driver applications will deliver < –75dBc distortion at the peak upstream power
levels of full rate ADSL. Using a differential driver design, as
shown below, the OPA2607 can deliver a high 38Vp-p voltage
swing into a 1:0.8 step-down transformer to meet the ADSL CPE
upstream power requirements. This low turns ratio actually provides a step up to the much weaker downstream signal arriving on
the line side of this transformer, extending the DSL modem’s
reach.
●
●
●
●
●
●
●
HIGH OUTPUT CURRENT: 250mA
OUTPUT VOLTAGE SWING: ±10.5V (VS = ±12V)
HIGH SLEW RATE: 600V/µs
LOW SUPPLY CURRENT: 8mA/channel
FLEXIBLE POWER CONTROL (SO-14)
±6V TO ±16V SUPPLY RANGE
POWER PACKAGING
APPLICATIONS
●
●
●
●
xDSL LINE DRIVER
LOW-NOISE ADSL RECEIVER
LOW-COST VIDEO DA
LOW-COST UPGRADE TO LT1207/AD812
+12V
1/2
OPA2607
1.21kΩ
78.7Ω 1:0.8
100nF
4.8Vp-p
38Vp-p
348Ω
1.21kΩ
100Ω
15Vp-p
78.7Ω
Power control features are included in the SO-14 package version
to allow system power to be minimized. Two logic control lines
allow four quiescent power settings. These include full power,
power cutback for short loops, idle state for no signal transmission
but line match maintenance, and shutdown for power off with a
high impedance output. An additional I ADJ pin allows the maximum supply current to be adjusted ±25% from the nominal value.
Connecting this pin to +VCC will increase the full power quiescent
to 20mA, increasing the peak output current available, while
connecting this pin to –VCC will decrease the full power quiescent
to 12mA where a lower peak output current is required. The
digital control lines continue to scale the total quiescent current
from these new maximum levels in the same proportional steps as
before.
The OPA2607 is available in three package styles. For power
package
driver applications, a thermally enhanced
1/2
OPA2607
–12V
Low Turns Ratio ADSL Upstream Driver
with a heat slug is available in both SO-8 and SO-14 pinouts. For
lower power receiver applications, a standard SO-8 package is
available.
OPA2607 RELATED PRODUCTS
SINGLES
DUALS
TRIPLES
NOTES
OPA681
—
OPA2681
OPA2677
OPA3681
—
Single +12V Capable
Single +12V Capable
International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111
Twx: 910-952-1111 • Internet: http://www.burr-brown.com/ • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132
®
©
2000 Burr-Brown Corporation
SBOS128
PDS-1615A
1
Printed in U.S.A. August, 2000
OPA2607
SPECIFICATIONS: VS = ±12V
RF = 1.21kΩ, RL = 100Ω, and G = +8, unless otherwise noted.
OPA2607H, U, N
TYP
PARAMETER
AC PERFORMANCE (see Figure 1)
Small-Signal Bandwidth (VO = 0.5Vp-p)
Bandwidth for 0.1dB Gain Flatness
Large-Signal Bandwidth
Slew Rate
Rise/Fall Time
Spurious Free Dynamic Range(4)
Input Voltage Noise
Non-Inverting Input Current Noise
Inverting Input Current Noise
Differential Gain
Differential Phase
Channel-to-Channel Crosstalk
DC PERFORMANCE(5)
Open-Loop Transimpedance Gain
Input Offset Voltage
Average Offset Voltage Drift
Non-Inverting Input Bias Current
Average Non-Inverting Input Bias Current Drift
Inverting Input Bias Current
Average Inverting Input Bias Current Drift
INPUT
Common-Mode Input Range (CMIR)
Common-Mode Rejection Ratio (CMRR)
Non-Inverting Input Impedance
Inverting Input Resistance
OUTPUT
Voltage Output Swing
Current Output, Sourcing
Current Output, Sinking
Closed-Loop Output Impedance
Power Control (SO-14 only)
Maximum Logic 0
Minimum Logic 1
Logic Input Current
Supply Current at Full Power
Supply Current at Power Cutback
Supply Current at Idle Power
Supply Current at Shutdown
Output Impedance in Idle Power
Output Impedance in Shutdown
Shutdown Isolation
Maximum Adjusted Quiescent Current
Minimum Adjusted Quiescent Current
POWER SUPPLY
Minimum Operating Voltage
Specified Operating Voltage
Maximum Operating Voltage
Maximum Quiescent Current
Minimum Quiescent Current
Power Supply Rejection Ratio (PSRR)
TEMPERATURE RANGE
Specification: H, U, N
Thermal Resistance, θJA
H PSO-8 Power Package(6)
U SO-8
N PSO-14 Power Package(6)
CONDITIONS
+25°C
G = +1, RF = 1.50kΩ
G = +2, RF = 1.43kΩ
G = +4, RF = 1.37kΩ
G = +8, RF = 1.21kΩ
G = +8, VO = 0.5Vp-p
G = +8, VO = 20Vp-p
G = +8, VO = 20V Step
G = +8, VO = 0.5V Step
VO = 2Vp-p, 1MHz, RL = 100Ω
VO = 20Vp-p, 150kHz, RL = 150Ω
35
28
25
25
6
13
600
14
77
75
1.7
11
15
0.01
0.01
–60
GUARANTEED
+25°C(2)
0°C to
70°C(3)
–40°C to
+85°C(3)
UNITS
MIN/ TEST
MAX LEVEL(1)
MHz
MHz
MHz
MHz
MHz
MHz
V/µs
ns
dB
dB
nV/√Hz
pA/√Hz
pA/√Hz
%
degrees
dB
typ
typ
typ
min
typ
min
min
min
min
min
max
max
max
typ
typ
typ
C
C
C
B
C
B
B
B
B
B
B
B
B
C
C
C
19
18
17
10.6
470
18
66
70
2.0
13
17
9.0
400
20
60
58
2.6
13
17
7.9
350
21
57
57
2.7
13
17
950
±1.5
440
±7
±3
±12
±4
±40
390
±8
–20
±15
–70
±58
–380
310
±8.5
–25
±20
–100
±70
–425
kΩ
mV
µV/°C
µA
nA/°C
µA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
±10.0
53
±9.9
52
±9.8
51
Open-Loop
±10.3
64
250 || 4
33
V
dB
kΩ || pF
Ω
min
min
typ
typ
A
A
C
C
No Load, Hard Limit
RL = 100Ω, Hard Limit
RL = 150Ω, SFDR > 67dB, 150kHz
VO = 0
VO = 0
G = +8, f ≤ 10kHz
±11.2
±10.5
±10.2
310
250
0.02
±10.9
±9.9
±10.8
±9.8
±10.7
±9.7
210
180
175
150
140
110
V
V
V
mA
mA
Ω
min
min
typ
min
min
typ
A
A
C
A
A
C
V
V
µA
mA
mA
mA
mA
Ω
kΩ || pF
dB
mA
mA
max
min
max
typ
typ
typ
typ
typ
typ
typ
typ
typ
C
C
C
C
C
C
C
C
C
C
C
C
V
V
V
mA
mA
dB
min
typ
max
max
min
min
B
C
A
A
A
A
–40 to +85
°C
typ
C
50
125
45
°C/W
°C/W
°C/W
typ
typ
typ
C
C
C
NTSC, G = +2, RL = 150Ω
NTSC, G = +2, RL = 150Ω
f = 1MHz
VO = 0V, RL = 100Ω
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
DIG_REF = Gnd
A0, A1
A0, A1
0V to 4.5V
A0 = 1, A1 = 1, IADJ = open
A0 = 0, A1 = 1, IADJ = open
A0 = 1, A1 = 0, IADJ = open
A0 = 0, A1 = 0, IADJ = open
Closed-Loop, f < 1MHz
0.8
2
60
16
13
3.8
1.3
0.7
350 || 17
75
20
12
G = +8, 1MHz
A0 = 1, A1 = 1, IADJ at +VS
A0 = 1, A1 = 1, IADJ at –VS
±12
Total Both Channels, Full Power
Total Both Channels, Full Power
f ≤ 10kHz
16
16
68
Junction-to-Ambient
±6
±6
±6
±16
16.8
15.2
61
±16
17
13.8
59
±16
17.5
13.3
57
NOTES: (1) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.
(C) Typical value only for information. (2) Junction temperature = ambient for 25°C guaranteed specifications. (3) Junction temperature = ambient at low temperature
limit: junction temperature = ambient +40°C at high temperature limit for over temperature guaranteed specifications. (4) Single amplifier SFDR limited by 2nd Harmonic.
Differential SFDR will be limited by 3rd Harmonic and will be > 15dB higher. (5) Current is considered positive out of node. VCM is the input common-mode voltage.
(6) Slug in power package connected to –VS plane at least 2" x 2" (50mm x 50mm) in size. See the Board Layout Guidelines Section.
®
OPA2607
2
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATIONS
Power Supply ............................................................................. ±16.5VDC
Internal Power Dissipation(1) ............................ See Thermal Information
Differential Input Voltage ..................................................................... ±5V
Input Voltage Range ............................................................................ ±VS
Storage Temperature Range: U, N ................................ –40°C to +125°C
Lead Temperature (soldering, 10s) .............................................. +260°C
Junction Temperature (TJ ) ........................................................... +175°C
ESD Rating (Human Body Model) .................................................. 4000V
(Machine Model) ........................................................... 300V
Top View
Out A
1
8
+VS
–In A
2
7
Out B
+In A
3
6
–In B
–VS
4
5
+In B
NOTE:: (1) Packages must be derated based on specified θJA. Maximum TJ
must be observed.
SO-8, PSO-8
ELECTROSTATIC
DISCHARGE SENSITIVITY
Electrostatic discharge can cause damage ranging from performance degradation to complete device failure. Burr-Brown
Corporation recommends that all integrated circuits be handled
and stored using appropriate ESD protection methods.
ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric
changes could cause the device not to meet published specifications.
–In A
1
14 Out A
+In A
2
13 NC
A0
3
12 DIG_REF
–VS
4
A1
5
10 IADJ
+In B
6
9
NC
–In B
7
8
Out B
Power
Control
11 +VS
PSO-14
NC = No Connection
PACKAGE/ORDERING INFORMATION
PRODUCT
PACKAGE
PACKAGE
DRAWING
NUMBER
OPA2607H
"
OPA2607U
"
OPA2607N
"
PSO-8
"
SO-8
"
PSO-14
"
182-1
"
182
"
235-1
"
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER(1)
TRANSPORT
MEDIA
–40°C to +85°C
"
"
"
"
"
OPA2607H
"
OPA2607U
"
OPA2607N
"
OPA2607H
OPA2607H/2K5
OPA2607U
OPA2607U/2K5
Contact Factory
Contact Factory
Rails
Tape and Reel
Rails
Tape and Reel
Rails
Tape and Reel
NOTE: (1) Models with a slash (/) are available only as Tape and Reel in the quantity indicated after the slash (e.g. /2K5 indicates 2500 devices per reel). Ordering 2500
pieces of the OPA2607U/2K5 will get a single 2500-piece Tape and Reel.
The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes
no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change
without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant
any BURR-BROWN product for use in life support devices and/or systems.
®
3
OPA2607
TYPICAL PERFORMANCE CURVES: VS = ±12V
At TA = +25°C, G = +8, RF = 1.21kΩ, and RL = 100Ω, unless otherwise noted. See Figure 1 for AC performance only.
NON-INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
6
0
G = +2
RF = 1.43kΩ
–3
G = +4
RF = 1.37kΩ
–6
–9
G = +8
RF = 1.21kΩ
–12
–15
G = –8
RF = 1.18kΩ
0
G = –1
RF = 1.43kΩ
–3
–6
G = –2
RF = 1.40kΩ
–9
G = –4
RF = 1.30kΩ
–12
–15
–18
–18
1M
10M
100M
1M
10M
100M
Frequency (Hz)
Frequency (Hz)
NON-INVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
6
6
G = +8
VO = 0.5Vp-p
3
G = –8
0
Normalized Gain (dB)
3
VO = 2Vp-p
–3
VO = 8Vp-p
–6
VO = 16Vp-p
–9
–12
–15
VO = 0.5Vp-p
VO = 2Vp-p
0
–3
VO = 16Vp-p
–6
–9
–12
VO = 8Vp-p
–15
–18
–18
10M
100M
1M
10M
Frequency (Hz)
Frequency (Hz)
NON-INVERTING PULSE RESPONSE
INVERTING PULSE RESPONSE
G = –8
Output Voltage (2V/div)
VO = 0.5Vp-p
Small Signal
Output Voltage (100mV/div)
G = +8
VO = 20Vp-p
Large Signal
VO = 20Vp-p
Large Signal
VO = 0.5Vp-p
Small Signal
Time (100ns/div)
Time (100ns/div)
®
OPA2607
100M
4
Output Voltage (100mV/div)
1M
Output Voltage (2V/div)
Normalized Gain (dB)
VO = 0.5Vp-p
3
Normalized Gain (dB)
Normalized Gain (dB)
6
G = +1
RF = 1.50kΩ
VO = 0.5Vp-p
3
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
TYPICAL PERFORMANCE CURVES: VS = ±12V
(Cont.)
At TA = +25°C, G = +8, RF = 1.21kΩ, and RL = 100Ω, unless otherwise noted. See Figure 1 for AC performance only.
HARMONIC DISTORTION vs OUTPUT VOLTAGE
HARMONIC DISTORTION vs FREQUENCY
–50
–50
f = 1MHz
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
VO = 2Vp-p
–60
–70
3rd-Harmonic
2nd-Harmonic
–80
–60
2nd-Harmonic
–70
–80
3rd-Harmonic
–90
–90
100k
1M
0.1
10M
1
Frequency (Hz)
HARMONIC DISTORTION vs NON-INVERTING GAIN
–50
VO = 2Vp-p
f = 1MHz
Harmonic Distortion (dBc)
VO = 2Vp-p
f = 1MHz
Harmonic Distortion (dBc)
20
HARMONIC DISTORTION vs INVERTING GAIN
–50
–60
–70
3rd-Harmonic
–80
2nd-Harmonic
–90
–60
–70
3rd-Harmonic
–80
2nd-Harmonic
–90
0.1
10
0.1
10
Gain Magnitude (V/V)
Gain Magnitude (–V/V)
HARMONIC DISTORTION vs DUAL SUPPLY VOLTAGE
HARMONIC DISTORTION vs LOAD RESISTANCE
–50
–50
VO = 2Vp-p
fO = 1MHz
Harmonic Distortion (dBc)
Harmonic Distortion (dBC)
10
Output Voltage (Vp-p)
–60
3rd-Harmonic
–70
2nd-Harmonic
–80
VO = 2Vp-p
fO = 1MHz
–60
–70
3rd-Harmonic
–80
2nd-Harmonic
–90
–90
10
100
6
1000
8
10
12
14
16
Dual Supply Voltage (±V)
Load Resistance (Ω)
®
5
OPA2607
TYPICAL PERFORMANCE CURVES: VS = ±12V
(Cont.)
At TA = +25°C, G = +8, RF = 1.21kΩ, and RL = 100Ω, unless otherwise noted. See Figure 1 for AC performance only.
TWO-TONE,3RD-ORDER
INTERMODULATION SPURIOUS
–50
3rd-Order Spurious Level (dBc)
Load Power at Matched 50Ω Load
–60
–70
fO = 2MHz
–80
fO = 1MHz
fO ≤ 500kHz
–90
–5
0
5
10
15
20
Single-Tone Load Power (dBm)
APPLICATIONS INFORMATION
WIDEBAND CURRENT FEEDBACK OPERATION
The OPA2607 gives the exceptional AC performance of a
wideband current feedback op amp with a highly linear,
high power output stage. Requiring only 8.0mA/chan quiescent current, the OPA2607 will swing to within 2V of either
supply rail and deliver in excess of 180mA guaranteed at
room temperature. This low output headroom requirement,
along with supply-voltage independent biasing, gives remarkable single (+15V) supply operation. Previous boosted
output stage amplifiers have typically suffered from very
poor crossover distortion as the output current goes through
zero. The OPA2607 achieves a comparable power gain with
much better linearity. The primary advantage of a currentfeedback op amp over a voltage-feedback op amp is that AC
performance (bandwidth and distortion) is relatively independent of signal gain.
0.1µF
2.2µF
+
50Ω Source
VI
49.9Ω
VO 49.9Ω
1/2
OPA2607
50Ω Load
RF
1.21kΩ
RG
169Ω
Figure 1 shows the DC-coupled, gain of +8, dual powersupply circuit configuration used as the basis of the ±12V
Specifications and Typical Performance Curves. For test
purposes, the input impedance is set to 50Ω with a resistor
to ground and the output impedance is set to 50Ω with a
series output resistor. Voltage swings reported in the specifications are taken directly at the input and output pins while
load powers (dBm) are defined at a matched 50Ω load.
For the circuit of Figure 1, the total effective load will be
100Ω || 1379Ω = 93Ω.
+
2.2µF
0.1µF
–VS
–12V
FIGURE 1. DC-Coupled, G = +8, Bipolar Supply, Specification and Test Circuit.
®
OPA2607
+12V
+VS
6
DESIGN-IN TOOLS
DEMONSTRATION BOARDS
VI
Several PC boards are available to assist in the initial
evaluation of circuit performance using the OPA2607 in its
3 package styles. All are available free as an unpopulated
PC board delivered with descriptive documentation. The
summary information for these boards is shown in Table I.
α
VO
RI
IERR
PRODUCT
PACKAGE
DEMO BOARD
NUMBER
OPA2607U
OPA2607N
OPA2607H
SO-8
SO-14 SO-Cool
SO-8 SO-Cool
DEM-OPA268xU
DEM-OPA2607N
DEM-OPA2607H
ORDERING
NUMBER
MKT-352
MKT-367
MKT-366
Z(S) IERR
RF
RG
TABLE I.
FIGURE 2. Current-Feedback Transfer Function Analysis
Circuit.
Contact the Burr-Brown applications support line to request
any of these boards.
The buffer gain is typically very close to 1.00 and is
normally neglected from signal gain considerations. It will,
however set the CMRR for a single op amp differentialamplifier configuration. For a buffer gain α < 1.0, the
CMRR = –20 • log (1– α) dB.
MACROMODELS AND APPLICATIONS SUPPORT
Computer simulation of circuit performance using SPICE is
often useful when analyzing the performance of analog
circuits and systems. This is particularly true for video and
RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. SPICE
models for some op amps are available through the BurrBrown web site (http://www.burr-brown.com). These models do a good job of predicting small-signal AC and transient
performance under a wide variety of operating conditions.
They do not do as well in predicting the harmonic distortion,
dG/dP, or temperature characteristics. These models do not
attempt to distinguish between the package types in their
small-signal AC performance, nor do they attempt to simulate channel-to-channel coupling.
RI, the buffer output impedance, is a critical portion of the
bandwidth control equation. The OPA2607 has an RI typically about 33Ω.
A current-feedback op amp senses an error current in the
inverting node (as opposed to a differential input error
voltage for a voltage-feedback op amp) and passes this on to
the output through an internal frequency dependent
transimpedance gain. The typical performance curves show
this open-loop transimpedance response. This is analogous
to the open-loop voltage gain curve for a voltage-feedback
op amp. Developing the transfer function for the circuit of
Figure 2 gives Equation 1:
OPERATING SUGGESTIONS
SETTING RESISTOR VALUES TO
OPTIMIZE BANDWIDTH
A current-feedback op amp like the OPA2607 can hold an
almost constant bandwidth over signal gain settings with the
proper adjustment of the external resistor values. This is
shown in the Typical Performance Curves; the small-signal
bandwidth decreases only slightly with increasing gain. Those
curves also show that the feedback resistor has been changed
for each gain setting. The resistor “values” on the inverting
side of the circuit for a current-feedback op amp can be
treated as frequency- response compensation elements while
their “ratios” set the signal gain. Figure 2 shows the smallsignal frequency-response analysis circuit for the OPA2607.
VO
=
VI
1+
R
α 1 + F
R
α NG
G
=
R F 1 + R F + R I NG
R F + R I 1 +
Z (S)
RG
Z (S)
(1)
RF
NG ≡ 1 +
R G
This is written in a loop-gain analysis format where the
errors arising from a finite open-loop gain are shown in the
denominator. If Z(s) were infinite over all frequencies, the
denominator of Equation 1 would reduce to 1 and the ideal
desired signal gain shown in the numerator would be achieved.
The fraction in the denominator of Equation 1 determines
the frequency response. Equation 2 shows this as the loopgain equation:
The key elements of this current feedback op amp model are:
α → Buffer Gain from the Non-inverting Input to the Inverting Input
RI → Buffer Output Impedance
Z (S)
iERR → Feedback Error Current Signal
R F + R I NG
Z(s) → Frequency Dependent Open Loop Transimpedance Gain
from iERR to VO
= Loop Gain
(2)
®
7
OPA2607
If 20 • log (RF + NG • RI) were drawn on top of the openloop transimpedance plot, the difference between the two
would be the loop gain at a given frequency. Eventually,
Z(s) rolls off to equal the denominator of Equation 2 at
which point the loop gain has reduced to 1 (and the curves
have intersected). This point of equality is where the
amplifier’s closed-loop frequency response given by Equation 1 will start to roll off, and is exactly analogous to the
frequency at which the noise gain equals the open-loop
voltage gain for a voltage-feedback op amp. The difference
here is that the total impedance in the denominator of
Equation 2 may be controlled somewhat separately from the
desired signal gain (or NG).
The total impedance going into the inverting input may be
used to adjust the closed-loop signal bandwidth. Inserting a
series resistor between the inverting input and the summing
junction will increase the feedback impedance (denominator
of Equation 2), decreasing the bandwidth. The internal
buffer output impedance for the OPA2607 is slightly influenced by the source impedance looking out of the noninverting input terminal. High-source resistors will have the
effect of increasing RI, decreasing the bandwidth. For those
single-supply applications which develop a midpoint bias at
the non-inverting input through high-valued resistors, the
decoupling capacitor is essential for power-supply ripple
rejection, non-inverting input-noise current shunting, and to
minimize the high frequency value for RI in Figure 2.
The OPA2607 is internally compensated to give a maximally flat frequency response for RF = 1.21kΩ at NG = 8 on
±12V supplies. Evaluating the denominator of Equation 2
(which is the feedback transimpedance) gives an optimal
target of 1.44kΩ. As the signal gain changes, the contribution of the NG x RI term in the feedback transimpedance will
change, but the total can be held constant by adjusting RF.
Equation 3 gives an approximate equation for optimum RF
over signal gain:
R F = 1441Ω – NG R I
INVERTING AMPLIFIER OPERATION
Since the OPA2607 is a wideband, current-feedback op
amp, most of the familiar op amp application circuits are
available to the designer. Those dual op amp applications
that require considerable flexibility in the feedback element
(e.g. integrators, transimpedance, some filters) should consider the unity gain stable voltage-feedback OPA2680, since
the feedback resistor is the compensation element for a
current-feedback op amp. Wideband inverting operations
(and especially summing) are particularly suited to the
OPA2607. Figure 4 shows a typical inverting configuration
where the I/O impedances are 50Ω.
(3)
As the desired signal gain increases, this equation will
eventually predict a negative RF. A somewhat subjective
limit to this adjustment can also be set by holding RG to a
minimum value of 20Ω. Lower values will load both the
buffer stage at the input and the output stage if RF gets too
low—actually decreasing the bandwidth. Figure 3 shows the
recommended RF versus NG. The values for RF versus Gain
shown here are approximately equal to the values used to
generate the typical performance curves. They differ in that
the optimized values used in the typical performance curves
are also correcting for board parasitics not considered in the
simplified analysis leading to Equation 3. The values shown
in Figure 3 give a good starting point for design where
bandwidth optimization is desired.
+12V
Power supply
de-coupling
not shown
50Ω Load
1/2
OPA2607
50Ω
Source
VO
49.9Ω
RF
1.21kΩ
RG
150Ω
VI
RM
75.0Ω
FEEDBACK RESISTOR vs NOISE GAIN
–12V
1600
FIGURE 4. Inverting Gain of –8 with Impedance Matching.
Feedback Resistor, RF (Ω)
1400
1200
In the inverting configuration, two key design considerations must be noted. The first is that the gain resistor (RG)
becomes part of the signal-channel input impedance. If input
impedance matching is desired (which is beneficial whenever the signal is coupled through a cable, twisted pair, long
PC board trace, or other transmission line conductor), it is
normally necessary to add an additional matching resistor
(RM) to ground. RG by itself is normally not set to the
required input impedance since its value, along with the
desired gain, will determine an RF which may be nonoptimal from a frequency response standpoint. The total
input impedance for the source becomes the parallel combination of RG and RM.
1000
800
600
400
200
0
0
5
10
15
20
Noise Gain, NG (V/V)
FIGURE 3. Recommended Feedback Resistor vs Noise
Gain.
®
OPA2607
8
NOISE PERFORMANCE
The second major consideration, touched on in the previous
paragraph, is that the signal-source impedance becomes part
of the noise-gain equation and will have slight effect on the
bandwidth through Equation 1. The values shown in Figure
4 have accounted for this by slightly decreasing RF to reoptimize the bandwidth for the noise gain. In the example of
Figure 4, the RM value combines in parallel with the external
50Ω source impedance, yielding an effective driving impedance of 50Ω || 75Ω = 30.0Ω. This impedance is added in
series with RG for calculating the noise gain, which gives
NG = 7.72 (instead of NG = 9.00 with a 0Ω source). This
value, along with the RF of Figure 3 and the inverting input
impedance of 33Ω, are inserted into Equation 3 to get
RF = 1186Ω.
Wideband current-feedback op amps generally have a higher
output noise than comparable voltage-feedback op amps. The
OPA2607 offers an excellent balance between voltage and
current noise terms to achieve low output noise. The inverting
current noise (15pA/√Hz) is significantly lower than competitive solutions, while the input voltage noise (1.7nV/√Hz) is
lower than most unity gain stable, wideband, voltage-feedback
op amps. This low input voltage noise was achieved at the
price of higher non-inverting input current noise (11pA/√Hz).
As long as the AC source impedance looking out of the noninverting node is less than 100Ω, this current noise will not
contribute significantly to the total output noise. The op amp
input voltage noise and the two input current noise terms
combine to give low output noise under a wide variety of
operating conditions. Figure 5 shows the op amp noise analysis model with all the noise terms included. In this model, all
noise terms are taken to be noise voltage or current density
terms in either nV/√Hz or pA/√Hz.
Note that the non-inverting input in this bipolar-supply
inverting application is connected directly to ground. It is
often suggested that an additional resistor be connected to
ground on the non-inverting input to achieve bias-current
error cancellation at the output. The input bias currents for
a current-feedback op amp are not generally matched in
either magnitude or polarity. Connecting a resistor to ground
on the non-inverting input of the OPA2607 in the circuit of
Figure 4 will actually provide additional gain for that input’s
bias and noise currents, but will not decrease the output DC
error since the input bias currents are not matched.
ENI
1/2
OPA2607
RS
EO
IBN
ERS
OUTPUT CURRENT AND VOLTAGE
The OPA2607 provides outstanding output voltage and
current capabilities. Under no-load conditions at 25°C, the
output voltage typically swings within 0.8V of either supply
rail; the guaranteed swing limit is within 1.1V of either rail.
Into a 5Ω load (the minimum tested load), it is guaranteed to
deliver more than ±180mA.
RF
÷ 4kTRS
IBI
RG
4kT
RG
÷ 4kTRF
4kT = 1.6 x 10–20J
at 290°K
FIGURE 5. Op Amp Noise Analysis Model.
DISTORTION PERFORMANCE
The OPA2607 provides good distortion performance into a
100Ω load on ±12V supplies. Relative to alternative solutions,
it provides exceptional performance into lighter loads.
Increasing the load impedance improves distortion directly.
Remember that the total load includes the feedback network—
in the non-inverting configuration (Figure 1) this is the sum of
RF + RG, while in the inverting configuration it is just RF.
Also, providing an additional supply decoupling capacitor
(0.1µF) between the supply pins (for bipolar operation) improves the 2nd-order distortion slightly (3 to 6dB).
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 4 shows the general form for the
output noise voltage using the terms shown in Figure 5.
(4)
EO =
(E
2
NI
)
+ (I BN R S ) + 4 kTR S NG 2 + (I BI R F ) + 4 kTR F NG
2
2
Dividing this expression by the noise gain (NG = (1+RF /RG ))
will give the equivalent input referred spot noise voltage at the
non-inverting input as shown in Equation 5.
In most op amps, increasing the output voltage swing increases harmonic distortion directly. As the typical performance curves show, the spurious intermodulation powers do
not increase as predicted by a traditional intercept model.
As the fundamental power level increases, the dynamic
range does not decrease significantly. For 2 tones centered
at 1MHz, with 10dBm/tone into a matched 50Ω load
(i.e. 2Vp-p for each tone at the load, which requires 8Vp-p
for the overall 2-tone envelope at the output pin), the typical
performance curves show 85dBc difference between the
test-tone power and the 3rd-order intermodulation spurious
levels.
(5)
2
I R
4 kTR F
2
E N = E NI 2 + (I BN R S ) + 4 kTR S + BI F +
NG
NG
Evaluating these two equations for the OPA2607 circuit and
component values shown in Figure 1 will give a total output
spot noise voltage of 27nV/√Hz and a total equivalent input
spot noise voltage of 3.4nV/√Hz. This total input referred
spot noise voltage is higher than the 1.7nV/√Hz specifica®
9
OPA2607
tion for the op amp voltage noise alone. This reflects the
noise added to the output by the inverting current noise times
the feedback resistor. If the feedback resistor is reduced in
high-gain configurations (as suggested previously), the total
input referred voltage noise given by Equation 5 will approach just the 1.7nV/√Hz of the op amp itself. For example,
going to a gain of +20 using RF = 750Ω will give a total
input referred noise of 2.0nV/√Hz .
The ADSL Upstream Driver shown on the front page will be
used as an example of these calculations. The ADSL
(G.DMT) standard uses a spectrally-efficient coding technique, which produces a near-Gaussian output voltage distribution. Under these conditions, IO(AVE) = 0.8 IO(RMS).
The maximum allowed crest factor in this standard is
CF = 5.33Vpk/Vrms. We now calculate for each amplifier
individually:
(9)
348Ω
100Ω / 2
1.21kΩ +
R L = 78.7Ω +
2
2
0.8
DC ACCURACY AND OFFSET CONTROL
A current-feedback op amp like the OPA2607 provides
exceptional bandwidth in high gains, giving fast pulse settling but only moderate DC accuracy. The typical specifications show an input offset voltage comparable to high-speed
voltage-feedback amplifiers. However, the two input bias
currents are somewhat higher and are unmatched. Whereas
bias current cancellation techniques are very effective with
most voltage-feedback op amps, they do not generally reduce the output DC offset for wideband current-feedback op
amps. Since the two input bias currents are unrelated in both
magnitude and polarity, matching the source impedance
looking out of each input to reduce their error contribution
to the output is ineffective. Evaluating the configuration of
Figure 1, using worst case +25°C input offset voltage and
the two input bias currents, gives a worst case output offset
range equal to:
= 141Ω
I O( PK ) =
I O( AVE )
(10)
2 • (12 V • 20.2 mA) – (25.3mA) • 141Ω
2
= 0.69 W
TJ = 85°C + 0.69 W • 50°C / W
= 120°C
The junction temperature of this example is well below
175°C absolute maximum because the PSO-8 power package has such a low thermal impedance when properly
connected to the –VS power plane (see the Board Layout
Guidelines section). To help illustrate this point, the regular
SO-8 package (OPA2607U) gives TJ = 171°C under the
same conditions.
= ±56.0mV ± 2.4mV ± 48.4mV
= ±107mV
THERMAL ANALYSIS
Maximum desired junction temperature will set the maximum allowed internal power dissipating. In no case should
the maximum junction temperature exceed 175°C. The operating junction temperature is given by:
BOARD LAYOUT GUIDELINES
Achieving optimum performance with a high-frequency
amplifier like the OPA2607 requires careful attention to
board layout parasitics and external component types. Recommendations that will optimize performance include:
(6)
a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the output
and inverting input pins can cause instability. On the noninverting input it can react with the source impedance to
cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be
opened in all of the ground and power planes around those
pins. Otherwise, ground and power planes should be unbroken elsewhere on the board.
where TA is the ambient temperature, PD is the average
power dissipation as calculated below, and θJA is the package thermal resistance shown in the specifications.
The total internal power dissipation of a single amplifier,
assuming bipolar supplies (±VS), is:
(7)
where IQ is the quiescent supply current, IO(AVE) is the
average output current, IO(RMS) is the root-mean-square output current, and RL is the load seen by the output. Under
absolute worst case conditions, with VO = VS/2, Equation 7
becomes:
b) Minimize the distance (< 0.25") from the power supply
pins to high frequency 0.1µF decoupling capacitors. At the
device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power supply
(8)
®
OPA2607
=
PD = 2 • 12 V • 16 mA +
= ± (8 • 7mV) ± (12µA • 25Ω • 8) ± (40µA • 1.21kΩ)
PD = 2 VS I Q + VS2 / ( 4 R L )
I O( PK )
38Vp − p / 2
= 135mA
141Ω
Now calculate the typical junction temperature of both
channels of the OPA2607H (PSO-8
Package)
based on Equations 6 and 7:
where NG = non-inverting signal gain
PD = 2 VS I Q + VS I O( AVE ) – I O2 ( RMS) R L
RL
=
135mA
= 25.3mA
CF
5.33
= 0.8 I O( RMS) = 20.2 mA
I O( RMS) =
± (NG • VOS(MAX)) ± (IBN • RS /2 • NG) ± (IBI • RF)
TJ = TA + PD θ JA
VO( PK )
10
the destination device. Remember also that the terminating
impedance will be the parallel combination of the shunt
resistor and the input impedance of the destination device:
this total effective impedance should be set to match the
trace impedance. The high output voltage and current capability of the OPA2607 allows multiple destination devices to
be handled as separate transmission lines, each with their
own series and shunt terminations. If the 6dB attenuation of
a doubly-terminated transmission line is unacceptable, a
long trace can be series-terminated at the source end only.
Treat the trace as a capacitive load in this case and set the
series resistor value as shown in the plot of “RS vs Capacitive Load”. This will not preserve signal integrity as well as
a doubly-terminated line. If the input impedance of the
destination device is low, there will be some signal attenuation due to the voltage divider formed by the series output
into the terminating impedance.
connections (on pins 4 and 7) should always be decoupled
with these capacitors. An optional supply decoupling capacitor across the two power supplies (for bipolar operation)
will improve 2nd harmonic distortion performance. Larger
(2.2µF to 6.8µF) decoupling capacitors, effective at lower
frequency, should also be used on the main supply pins.
These may be placed somewhat farther from the device and
may be shared among several devices in the same area of the
PC board.
c) Careful selection and placement of external components will preserve the high-frequency performance of
the OPA2607. Resistors should be a very low reactance
type. Surface-mount resistors work best and allow a tighter
overall layout. Metal film and carbon composition axially
leaded resistors can also provide good high-frequency performance. Again, keep their leads and PC board trace length
as short as possible. Never use wirewound-type resistors in
a high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. Other
network components, such as non-inverting input termination resistors, should also be placed close to the package.
Where double-side component mounting is allowed, place
the feedback resistor directly under the package on the other
side of the board between the output and inverting input
pins. The frequency response is primarily determined by the
feedback resistor value as described previously. Increasing
its value will reduce the bandwidth, while decreasing it will
give a more peaked frequency response. The 1.21kΩ feedback resistor used in the typical performance specifications
at a gain of +8 on ±12V supplies is a good starting point for
design. Note that a 1.50kΩ feedback resistor, rather than a
direct short, is recommended for the unity-gain follower
application. A current-feedback op amp requires a feedback
resistor even in the unity-gain follower configuration to
control stability.
e) Do not socket a high speed part like the OPA2607. The
additional lead length and pin-to-pin capacitance introduced
by the socket can create an extremely troublesome parasitic
network which can make it almost impossible to achieve a
smooth, stable frequency response. Best results are obtained
by soldering the OPA2607 onto the board.
f) Use the –VS plane to conduct heat out of the PSO-8 and
PSO-14 Power Packages (OPA2607H and OPA2607N).
These packages attach the die directly to a metal slug in its
bottom, which you should solder to the board. This slug
needs to be connected electrically to the negative supply
plane, which must have a minimum area of 2" x 2" (50mm
x 50mm) to produce the θJA values in the specifications
table. More details will be found in the data sheets that
accompany the demo boards described in the Demonstration
Boards section of this data sheet.
INPUT AND ESD PROTECTION
The OPA2607 is built using a very high-speed complementary bipolar process. All device pins have ESD protection
using internal diodes to the power supplies as shown in
Figure 6. The OPA2607 has an ESD rating of 4000V human
body model, and 300V machine model.
d) Connections to other wideband devices on the board
may be made with short direct traces or through on-board
transmission lines. For short connections, consider the trace
and the input to the next device as a lumped capacitive load.
Relatively wide traces (50 to 100mils) should be used,
preferably with ground and power planes opened up around
them. Estimate the total capacitive load and set RS from the
plot of recommended “RS vs Capacitive Load”. Low parasitic capacitive loads (< 5pF) may not need an RS since the
OPA2607 is nominally compensated to operate with a 2pF
parasitic load. If a long trace is required, and the 6dB signal
loss intrinsic to a doubly-terminated transmission line is
acceptable, implement a matched-impedance transmission
line using microstrip or stripline techniques (consult an ECL
design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary on
board, and in fact a higher-impedance environment will
improve distortion as shown in the “Distortion vs Load”
plots. With a characteristic board-trace impedance defined
based on board material and trace dimensions, a matching
series resistor into the trace from the output of the OPA2607
is used as well as a terminating shunt resistor at the input of
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (e.g. in systems with ±30V supply parts
driving into the OPA2607), current-limiting series resistors
should be added into the two inputs. Keep these resistor
values as low as possible since high values degrade both
noise performance and frequency response.
+V CC
External
Pin
Internal
Circuitry
–V CC
FIGURE 6. Internal ESD Protection.
®
11
OPA2607
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