OPA197-Q1, OPA2197-Q1, OPA4197-Q1
OPA197-Q1,
OPA2197-Q1,
SBOS918A
– MARCH
2018 – REVISEDOPA4197-Q1
JANUARY 2021
SBOS918A – MARCH 2018 – REVISED JANUARY 2021
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OPAx197-Q1 36-V, Precision, Rail-to-Rail Input/Output, Low-Offset Voltage,
Low-Input Bias Current e-trim™ Operational Amplifiers
1 Features
3 Description
•
The OPAx197-Q1 family (OPA197-Q1, OPA2197-Q1,
and OPA4197-Q1) is part of a new generation of 36-V,
e-trim™ operational amplifiers. The OPAx197-Q1
family of e-trim operational amplifiers uses a
proprietary method of package-level trim for offset and
offset temperature drift implemented during the final
steps of manufacturing after the plastic molding
process. This method minimizes the influence of
inherent input transistor mismatch, as well as errors
induced during package molding.
•
•
•
•
•
•
•
•
•
•
•
•
•
•
AEC-Q100 qualified for automotive applications:
– Temperature grade 1: –40°C to +125°C, TA
Low offset voltage: ±250 µV (maximum)
Low offset voltage drift: ±0.2 µV/°C
Low noise: 5.5 nV/√Hz at 1 kHz
High common-mode rejection: 140 dB
Low bias current: ±5 pA
Rail-to-rail input and output
Wide bandwidth: 10 MHz GBW
High slew rate: 20 V/µs
Low quiescent current: 1 mA per amplifier
Wide supply: ±2.25 V to ±18 V, 4.5 V to 36 V
EMI/RFI filtered inputs
Differential input-voltage range to supply rail
High capacitive load drive capability: 1 nF
Industry-standard packages:
– Single and dual channel in very-small, 8-pin
VSSOP
– Quad channel in 14-pin TSSOP
These devices offer outstanding dc precision and ac
performance, including rail-to-rail input/output, low
offset (±5 µV, typical), low offset drift (±0.2 µV/°C,
typical), and a 10-MHz bandwidth.
Unique features such as differential input-voltage
range to the supply rail, high output current (±65 mA),
high capacitive load drive of up to 1 nF, and high slew
rate (20 V/µs) make the OPAx197-Q1 robust, highperformance op amps for high-voltage industrial
applications.
Device Information
2 Applications
•
•
•
•
PACKAGE(1)
PART NUMBER
Inverter and motor control
DC/DC converter
On-board (OBC) and wireless charger
Battery management system (BMS)
OPA197-Q1
OPA2197-Q1
OPA4197-Q1
(1)
BODY SIZE (NOM)
VSSOP (8)
3.00 mm × 3.00 mm
TSSOP (14)
5.00 mm × 4.40 mm
For all available packages, see the package option
addendum at the end of the data sheet.
100
66 Typical Units Shown
75
VOS ( V)
50
25
0
±25
±50
±75
±100
±75
±50
±25
0
25
50
75
Temperature (ƒC)
100
125
150
C001
The OPAx197-Q1 Maintains Ultra-Low Input Offset Voltage Over Temperature
An©IMPORTANT
NOTICEIncorporated
at the end of this data sheet addresses availability, warranty, changes, use in
safety-critical
applications,
Copyright
2021 Texas Instruments
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SBOS918A – MARCH 2018 – REVISED JANUARY 2021
Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Pin Configuration and Functions...................................3
6 Specifications.................................................................. 6
6.1 Absolute Maximum Ratings ....................................... 6
6.2 ESD Ratings .............................................................. 6
6.3 Recommended Operating Conditions ........................6
6.4 Thermal Information: OPA197-Q1 ............................. 7
6.5 Thermal Information: OPA2197-Q1 ........................... 7
6.6 Thermal Information: OPA4197-Q1 ........................... 7
6.7 Electrical Characteristics: VS = ±4 V to ±18 V
(VS = 8 V to 36 V) .........................................................8
6.8 Electrical Characteristics: VS = ±2.25 V to ±4 V
(VS = 4.5 V to 8 V) ......................................................10
6.9 Typical Characteristics.............................................. 12
7 Detailed Description......................................................21
7.1 Overview................................................................... 21
7.2 Functional Block Diagram......................................... 21
7.3 Feature Description...................................................22
7.4 Device Functional Modes..........................................28
8 Application and Implementation.................................. 29
8.1 Application Information............................................. 29
8.2 Typical Applications.................................................. 29
9 Power Supply Recommendations................................33
10 Layout...........................................................................33
10.1 Layout Guidelines................................................... 33
10.2 Layout Examples.................................................... 34
11 Device and Documentation Support..........................35
11.1 Device Support........................................................35
11.2 Documentation Support.......................................... 35
11.3 Receiving Notification of Documentation Updates.. 35
11.4 Support Resources................................................. 35
11.5 Trademarks............................................................. 36
11.6 Electrostatic Discharge Caution.............................. 36
11.7 Glossary.................................................................. 36
12 Mechanical, Packaging, and Orderable
Information.................................................................... 36
4 Revision History
Changes from Revision * (March 2018) to Revision A (January 2021)
Page
• Added OPA4197-Q1 and associated content..................................................................................................... 1
2
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5 Pin Configuration and Functions
NC
1
–IN
2
+IN
3
V–
4
8
NC
–
7
V+
+
6
OUT
5
NC
Not to scale
Figure 5-1. OPA197-Q1 DGK Package, 8-Pin VSSOP, Top View
Pin Functions: OPA197-Q1
PIN
NAME
+IN
I/O
NO.
3
DESCRIPTION
I
Noninverting input
Inverting input
–IN
2
I
NC
1, 5, 8
—
No internal connection (can be left floating)
OUT
6
O
Output
V+
7
—
Positive (highest) power supply
V–
4
—
Negative (lowest) power supply
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OUT A
1
8
V+
–IN A
2
7
OUT B
+IN A
3
6
–IN B
V–
4
5
+IN B
Not to scale
Figure 5-2. OPA2197-Q1 DGK Package, 8-Pin VSSOP, Top View
Pin Functions: OPA2197-Q1
PIN
4
I/O
DESCRIPTION
NAME
DGK (VSSOP)
+IN A
3
I
Noninverting input, channel A
+IN B
5
I
Noninverting input, channel B
–IN A
2
I
Inverting input, channel A
–IN B
6
I
Inverting input, channel B
OUT A
1
O
Output, channel A
OUT B
7
O
Output, channel B
V+
8
—
Positive (highest) power supply
V–
4
—
Negative (lowest) power supply
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OUT A
1
14
OUT D
–IN A
2
13
–IN D
+IN A
3
12
+IN D
V+
4
11
V–
+IN B
5
10
+IN C
–IN B
6
9
–IN C
OUT B
7
8
OUT C
Not to scale
Figure 5-3. OPA4197-Q1 PW Package, 14-Pin TSSOP, Top View
Pin Functions: OPA4197-Q1
PIN
I/O
DESCRIPTION
NAME
NO.
+IN A
3
I
Noninverting input, channel A
+IN B
5
I
Noninverting input, channel B
+IN C
10
I
Noninverting input, channel C
+IN D
12
I
Noninverting input, channel D
–IN A
2
I
Inverting input, channel A
–IN B
6
I
Inverting input, channel B
–IN C
9
I
Inverting input, channel C
–IN D
13
I
Inverting input, channel D
OUT A
1
O
Output, channel A
OUT B
7
O
Output, channel B
OUT C
8
O
Output, channel C
OUT D
14
O
Output, channel D
V+
4
—
Positive (highest) power supply
V–
11
—
Negative (lowest) power supply
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)(1)
MIN
VS
Supply voltage
Single supply, VS = (V+)
40
Dual supply, VS = (V+) – (V–)
Common-mode
Signal input voltage
MAX
±20
(V–) – 0.5
Signal input current
Output short
circuit(2)
Class IIA
TA
Operating temperature
–55
TJ
Junction temperature
Tstg
Storage temperature
(2)
V
±10
mA
150
°C
150
°C
150
°C
Continuous
Latch-up per JESD78D
(1)
V
(V+) + 0.5
(V+) – (V–) +
0.2
Differential
UNIT
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under
Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device
reliability.
Short-circuit to ground, one amplifier per package.
6.2 ESD Ratings
VALUE
UNIT
OPA197-Q1, OPA2197-Q1
V(ESD)
Electrostatic discharge
Human-body model (HBM), per AEC Q100-002(1)
HBM ESD classification level 3A
±4000
Charge device model (CDM), per AEC Q100-011
CDM ESD classification level C4A
±500
Human-body model (HBM), per AEC Q100-002(1)
HBM ESD classification level 2
±2000
Charge device model (CDM), per AEC Q100-011
CDM ESD classification level C5
±750
V
OPA4197-Q1
V(ESD)
(1)
Electrostatic discharge
V
AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
6
VS
Supply voltage
TA
Operating temperature
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Single supply, VS = (V+)
Dual supply, VS = (V+) – (V–)
NOM
MAX
4.5
36
±2.25
±18
–40
125
UNIT
V
°C
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6.4 Thermal Information: OPA197-Q1
OPA197-Q1
THERMAL
METRIC(1)
DGK (VSSOP)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
180.4
°C/W
RθJC(top)
Junction-to-case(top) thermal resistance
67.9
°C/W
RθJB
Junction-to-board thermal resistance
102.1
°C/W
ψJT
Junction-to-top characterization parameter
10.4
°C/W
ψJB
Junction-to-board characterization parameter
100.3
°C/W
RθJC(bot)
Junction-to-case(bottom) thermal resistance
N/A
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
6.5 Thermal Information: OPA2197-Q1
OPA2197-Q1
THERMAL
METRIC(1)
DGK (VSSOP)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
158
°C/W
RθJC(top)
Junction-to-case(top) thermal resistance
48.6
°C/W
RθJB
Junction-to-board thermal resistance
78.7
°C/W
ψJT
Junction-to-top characterization parameter
3.9
°C/W
ψJB
Junction-to-board characterization parameter
77.3
°C/W
RθJC(bot)
Junction-to-case(bottom) thermal resistance
N/A
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
6.6 Thermal Information: OPA4197-Q1
OPA4197-Q1
THERMAL
METRIC(1)
PW (TSSOP)
UNIT
14 PINS
RθJA
Junction-to-ambient thermal resistance
108.1
°C/W
RθJC(top)
Junction-to-case(top) thermal resistance
26.3
°C/W
RθJB
Junction-to-board thermal resistance
54.4
°C/W
ψJT
Junction-to-top characterization parameter
1.4
°C/W
ψJB
Junction-to-board characterization parameter
53.3
°C/W
RθJC(bot)
Junction-to-case(bottom) thermal resistance
N/A
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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6.7 Electrical Characteristics: VS = ±4 V to ±18 V (VS = 8 V to 36 V)
at TA = +25°C, VCM = VOUT = VS / 2, and RL = 10 kΩ connected to VS / 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OFFSET VOLTAGE
VOS
Input offset voltage
±25
±250
TA = 0°C to 85°C
±30
±350
TA = –40°C to +125°C
±50
±400
±10
±250
TA = 0°C to 85°C
±25
±350
TA = –40°C to +125°C
±50
±500
TA = 0°C to 85°C
±0.5
±2.5
TA = –40°C to +125°C
±0.8
±4.5
TA = –40°C to +125°C
±0.3
±1.0
µV/V
±5
±20
pA
±5
nA
±2
±20
pA
±2
nA
VCM = (V+) – 1.5 V
dVOS/dT
Input offset voltage
drift
PSRR
Power-supply
rejection ratio
µV
µV/°C
INPUT BIAS CURRENT
IB
Input bias current
IOS
Input offset current
TA = –40°C to +125°C
TA = –40°C to +125°C
NOISE
En
Input voltage noise
Input voltage noise
density
en
f = 0.1 Hz to 10 Hz
1.3
(V+) – 1.5 V < VCM < (V+) + 0.1 V f = 0.1 Hz to 10 Hz
(V–) – 0.1 V < VCM < (V+) – 3 V
4
(V–) – 0.1 V < VCM < (V+) – 3 V
(V+) – 1.5 V < VCM < (V+) + 0.1 V
Input current noise
density
in
f = 100 Hz
µVPP
10.5
f = 1 kHz
5.5
f = 100 Hz
32
f = 1 kHz
nV/√Hz
12.5
f = 1 kHz
1.5
fA/√Hz
INPUT VOLTAGE
Common-mode
voltage
VCM
CMRR
Common-mode
rejection ratio
(V–) – 0.1
(V+) + 0.1
(V–) – 0.1 V < VCM < (V+) – 3 V
120
140
(V–) < VCM < (V+) – 3 V, TA = –40°C to +125°C
114
126
100
120
86
100
(V+) – 1.5 V < VCM < (V+)
TA = –40°C to +125°C
V
dB
INPUT IMPEDANCE
8
ZID
Differential
ZIC
Common-mode
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100 || 1.6
MΩ || pF
1 || 6.4
1013Ω ||
pF
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6.7 Electrical Characteristics: VS = ±4 V to ±18 V (VS = 8 V to 36 V) (continued)
at TA = +25°C, VCM = VOUT = VS / 2, and RL = 10 kΩ connected to VS / 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
120
134
114
126
126
140
120
134
MAX
UNIT
OPEN-LOOP GAIN
AOL
Open-loop voltage
gain
(V–) + 0.6 V < VO < (V+) – 0.6 V,
RL = 2 kΩ
(V–) + 0.3 V < VO < (V+) – 0.3 V,
RL = 10 kΩ
TA = –40°C to +125°C
TA = –40°C to +125°C
dB
FREQUENCY RESPONSE
GBW
Unity gain bandwidth
SR
Slew rate
G = 1, 10-V step
To 0.01%
ts
Settling time
To 0.001%
tOR
Overload recovery
time
VIN × G = VS
THD+N
Total harmonic
distortion + noise
G = 1, f = 1 kHz, VO = 3.5 VRMS
Crosstalk
10
MHz
20
V/µs
G = 1, 10-V step
1.4
G = 1, 5-V step
0.9
G = 1, 10-V step
2.1
G = 1, 5-V step
1.8
µs
200
ns
µs
0.00008%
OPA4197-Q1 at dc
150
dB
OPA4197-Q1, f = 100 kHz
130
dB
OUTPUT
No load
Positive rail
VO
Voltage output swing
from rail
ISC
95
110
RL = 2 kΩ
430
500
5
15
RL = 10 kΩ
95
110
RL = 2 kΩ
430
500
Short-circuit current
CLOAD
Capacitive load drive
ZO
Open-loop output
impedance
15
RL = 10 kΩ
No load
Negative rail
5
±65
mV
mA
See Section 6.9
f = 1 MHz, IO = 0 A; see Figure 6-29
375
Ω
POWER SUPPLY
IQ
Quiescent current per
IO = 0 A
amplifier
1
TA = –40°C to +125°C
1.2
1.5
mA
TEMPERATURE
Thermal protection
140
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6.8 Electrical Characteristics: VS = ±2.25 V to ±4 V (VS = 4.5 V to 8 V)
at TA = +25°C, VCM = VOUT = VS / 2, and RL = 10 kΩ connected to VS / 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OFFSET VOLTAGE
VCM = (V+) – 3 V
TA = 0°C to 85°C
TA = –40°C to +125°C
VOS
Input offset voltage
(V+) – 3.5 V < VCM < (V+) – 1.5 V
VCM = (V+) – 1.5 V
Input offset voltage
drift
VCM = (V+) – 3 V
dVOS/dT
Input offset voltage
drift
VCM = (V+) – 1.5 V
PSRR
Power-supply
rejection ratio
TA = –40°C to +125°C
dVOS/dT
±5
±250
±8
±350
±10
±400
µV
See Section 7.3.6
±10
±250
TA = 0°C to 85°C
±25
±350
TA = –40°C to +125°C
±50
±500
±0.5
±2.5
±0.8
±4.5
TA = –40°C to +125°C
µV/°C
±2
µV/V
INPUT BIAS CURRENT
IB
Input bias current
IOS
Input offset current
±5
±20
±5
nA
±2
±20
pA
±2
nA
TA = –40°C to +125°C
TA = –40°C to +125°C
pA
NOISE
En
Input voltage noise
Input voltage noise
density
en
(V–) – 0.1 V < VCM < (V+) – 3 V, f = 0.1 Hz to 10 Hz
(V–) – 0.1 V < VCM < (V+) – 3 V
(V+) – 1.5 V < VCM < (V+) + 0.1 V
Input current noise
density
in
1.3
(V+) – 1.5 V < VCM < (V+) + 0.1 V, f = 0.1 Hz to 10 Hz
µVPP
4
f = 100 Hz
10.5
f = 1 kHz
5.5
f = 100 Hz
32
f = 1 kHz
nV/√Hz
12.5
f = 1 kHz
1.5
fA/√Hz
INPUT VOLTAGE
Common-mode
voltage range
VCM
(V–) – 0.1
(V–) – 0.1 V < VCM < (V+) – 3 V
CMRR
Common-mode
rejection ratio
(V+) – 1.5 V < VCM < (V+)
TA = –40°C to +125°C
TA = –40°C to +125°C
(V+) – 3 V < VCM < (V+) – 1.5 V
(V+) + 0.1
94
110
90
104
100
120
84
100
V
dB
See Section 6.9
INPUT IMPEDANCE
ZID
Differential
ZIC
Common-mode
10
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100 || 1.6
MΩ || pF
1 || 6.4
1013Ω ||
pF
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6.8 Electrical Characteristics: VS = ±2.25 V to ±4 V (VS = 4.5 V to 8 V) (continued)
at TA = +25°C, VCM = VOUT = VS / 2, and RL = 10 kΩ connected to VS / 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OPEN-LOOP GAIN
AOL
Open-loop voltage
gain
(V–) + 0.6 V < VO < (V+) – 0.6 V,
RL = 2 kΩ
TA = –40°C to +125°C
(V–) + 0.3 V < VO < (V+) – 0.3 V,
RL = 10 kΩ
TA = –40°C to +125°C
110
120
100
114
110
126
110
120
dB
10
MHz
20
V/µs
1
µs
ns
dB
FREQUENCY RESPONSE
GBW
Unity gain bandwidth
SR
Slew rate
G = 1, 5-V step
ts
Settling time
To 0.01%, VS = ±3 V, G = 1, 5-V step
tOR
Overload recovery
time
VIN × G = VS
200
OPA4197-Q1 at dc
150
dB
OPA4197-Q1, f = 100 kHz
130
dB
Crosstalk
OUTPUT
No load
Positive rail
VO
Voltage output swing
from rail
ISC
95
110
RL = 2 kΩ
430
500
5
15
RL = 10 kΩ
95
110
RL = 2 kΩ
430
500
Short-circuit current
CLOAD
Capacitive load drive
ZO
Open-loop output
impedance
15
RL = 10 kΩ
No load
Negative rail
5
±65
mV
mA
See Section 6.9
f = 1 MHz, IO = 0 A; see Figure 6-29
375
Ω
POWER SUPPLY
IQ
Quiescent current per
IO = 0 A
amplifier
1
TA = –40°C to +125°C
1.2
1.5
mA
TEMPERATURE
Thermal protection
140
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6.9 Typical Characteristics
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
Table 6-1. Table of Graphs
DESCRIPTION
FIGURE
Offset Voltage Production Distribution
Figure 6-1 to Figure 6-6
Offset Voltage Drift Distribution
Figure 6-7 to Figure 6-8
Offset Voltage vs Temperature
Figure 6-9
Offset Voltage vs Common-Mode Voltage
Figure 6-10 to Figure 6-12
Offset Voltage vs Power Supply
Figure 6-13
Open-Loop Gain and Phase vs Frequency
Figure 6-14
Closed-Loop Gain and Phase vs Frequency
Figure 6-15
Input Bias Current vs Common-Mode Voltage
Figure 6-16
Input Bias Current vs Temperature
Figure 6-17
Output Voltage Swing vs Output Current (maximum supply)
Figure 6-18
CMRR and PSRR vs Frequency
Figure 6-19
CMRR vs Temperature
Figure 6-20
PSRR vs Temperature
Figure 6-21
0.1-Hz to 10-Hz Noise
Figure 6-22
Input Voltage Noise Spectral Density vs Frequency
Figure 6-23
THD+N Ratio vs Frequency
Figure 6-24
THD+N vs Output Amplitude
Figure 6-25
Quiescent Current vs Supply Voltage
Figure 6-26
Quiescent Current vs Temperature
Figure 6-27
Open Loop Gain vs Temperature
Figure 6-28
Open Loop Output Impedance vs Frequency
Small Signal Overshoot vs Capacitive Load (100-mV Output Step)
Figure 6-29
Figure 6-30, Figure 6-31
No Phase Reversal
Figure 6-32
Positive Overload Recovery
Figure 6-33
Negative Overload Recovery
Figure 6-34
Small-Signal Step Response (100 mV)
Figure 6-35, Figure 6-36
Large-Signal Step Response
Figure 6-37
Settling Time
Figure 6-38 to Figure 6-41
Short-Circuit Current vs Temperature
Figure 6-42
Maximum Output Voltage vs Frequency
Figure 6-43
Propagation Delay Rising Edge
Figure 6-44
Propagation Delay Falling Edge
Figure 6-45
Crosstalk vs Frequency
Figure 6-46
12
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SBOS918A – MARCH 2018 – REVISED JANUARY 2021
6.9 Typical Characteristics (continued)
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
50
22
Distribution Taken From 190 Amplifiers
40
18
16
Amplifiers (%)
14
12
10
8
6
30
20
10
-10
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
1
2
3
4
5
6
7
8
9
10
75
50
25
0
0
0
-25
2
-50
4
-75
Percentage of Amplifiers (%)
20
Offset Voltage (µV)
Offset Voltage (V)
C013
TA = 125°C
Figure 6-1. Offset Voltage Production Distribution at 25°C
Distribution Taken From 190 Amplifiers
Distribution Taken From 190 Amplifiers
70
60
50
50
Amplifiers (%)
60
40
30
40
30
20
10
10
0
0
-50
-45
-40
-35
-30
-25
-20
-15
-10
-5
0
5
10
15
20
25
30
35
40
45
50
20
-50
-45
-40
-35
-30
-25
-20
-15
-10
-5
0
5
10
15
20
25
30
35
40
45
50
70
Offset Voltage (µV)
Offset Voltage (µV)
TA = 85°C
TA = 0°C
Figure 6-3. Offset Voltage Production Distribution at 85°C
Figure 6-4. Offset Voltage Production Distribution at 0°C
50
40
35
35
25
20
75
50
0
25
5
0
0
10
5
-25
15
10
-50
15
Offset Voltage (µV)
TA = –25°C
75
20
50
25
30
25
30
-75
Amplifiers (%)
40
-75
Amplifiers (%)
Distribution Taken From 190 Amplifiers
45
0
Distribution Taken From 190 Amplifiers
45
-50
50
-25
Amplifiers (%)
Figure 6-2. Offset Voltage Production Distribution at 125°C
Offset Voltage (µV)
TA = –40° C
Figure 6-5. Offset Voltage Production Distribution at –25°C
Figure 6-6. Offset Voltage Production Distribution at –40°C
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6.9 Typical Characteristics (continued)
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
50
30
Distribution Taken From 75 Amplifiers
Distribution Taken From 75 Amplifiers
25
Amplifiers (%)
30
20
20
15
10
10
Offset Voltage Drift (µV/ƒC)
0.8
0.6
0.7
0.4
0.5
0.2
0.3
0
0.1
-0.2
-0.1
-0.4
-0.3
-0.6
-0.5
-0.8
0
1.1
0.9
0.7
0.5
0.3
0.1
-0.1
-0.3
-0.5
-0.7
-0.9
-1.1
0
5
-0.7
Amplifiers (%)
40
Offset Voltage Drift (µV/ƒC)
TA = –40°C to +125°C
TA = 0°C to 85°C
Figure 6-7. Offset Voltage Drift Distribution from
–40°C to +125°C
Figure 6-8. Offset Voltage Drift Distribution from 0°C to 85°C
50
100
75
25
Offset Voltage (V)
50
VOS (V)
25
0
–25
0
VCM = –18.1 V
–25
–50
–75
–50
–100
–75
–50
–25
0
25
50
75
100
125
–20
150
–15
–10
–5
0
5
10
15
20
Common-Mode Voltage (V)
Temperature ( °C)
C001
Figure 6-10. Offset Voltage vs Common-Mode Voltage
Figure 6-9. Offset Voltage vs Temperature
100
200
75
150
5 Typical Units Shown
VCM = +18.1 V
25
Offset Voltage (µV)
Offset Voltage (V)
50
VCM = –18.1 V
0
–25
P-Channel
N-Channel
–50
50
0
–50
–100
VCM = +2.35 V
VCM = –2.35 V
–150
–75
Transition
–100
12.5
100
13.5
14.5
15.5
16.5
17.5
18.5
Common-Mode Voltage (V)
Transition
P-Channel
–200
–2.5 –2.0 –1.5 –1.0 –0.5 0.0 0.5
N-Channel
1.0
1.5
2.0
2.5
Common-Mode Voltage (V)
VS = ±2.25 V
Figure 6-11. Offset Voltage vs Common-Mode Voltage
14
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Figure 6-12. Offset Voltage vs Common-Mode Voltage
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6.9 Typical Characteristics (continued)
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
50
40
120.0
Open-Loop Gain
30
135
100.0
20
Gain (dB)
10
0
–10
80.0
Phase
60.0
90
40.0
Phase (°)
Offset Voltage (µV)
180
140.0
10 Typical Units Shown
–20
20.0
–30
45
0.0
–40
–50
0.0
2.0
4.0
6.0
8.0
10.0 12.0 14.0 16.0 18.0 20.0
1
10
100
1k
10k 100k
Frequency (Hz)
Power Supply Voltage (V)
1M
CLOAD = 15 pF
VS = ±2.25 V to ±18 V
Figure 6-14. Open-Loop Gain and Phase vs Frequency
Figure 6-13. Offset Voltage vs Power Supply
60.0
20
G = -100
G = +1
G = -1
G = -10
15
Input Bias Current (pA)
40.0
Gain (dB)
0
10M 100M
–20.0
20.0
0.0
IB–
10
5
0
IB+
–5
–10
–15
–20
±20.0
1000
10k
100k
1M
18.0
10M
Frequency (Hz)
Figure 6-15. Closed-Loop Gain and Phase vs Frequency
0.0
9.0
18.0
Common-Mode Voltage (V)
C001
Figure 6-16. Input Bias Current vs Common-Mode Voltage
(V–) + 5
6000
Out pu Voltage Swing (V)
IB+
IB Ios
5000
Input Bias Current (pA)
9.0
C003
4000
3000
2000
1000
+125°C
(V–) + 3
(V–) + 2
– 40°C
(V–) + 1
(V–)
Ios
0
(V–) + 4
(V–) – 1
±1000
±75
±50
±25
0
25
50
75
100
125
150
Temperature (ƒC)
Figure 6-17. Input Bias Current vs Temperature
175
0
10
20
30
40
50
Output Current (mA)
C001
60
70
80
C001
Figure 6-18. Output Voltage Swing vs Output Current (Maximum
Supply)
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6.9 Typical Characteristics (continued)
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
Common-Mode Rejection Ratio (µV/V)
Common-Mode Rejection Ratio (dB),
Power-Supply Rejection Ratio (dB)
160.0
140.0
120.0
100.0
80.0
60.0
+PSRR
40.0
CMRR
20.0
-PSRR
0.0
10
8
6
4
VS = ±2.25 V, VCM = V+ - 3 V
2
0
±2
VS = ±18 V, VCM = 0 V
±4
±6
±8
±10
1
10
100
1k
10k
100k
Frequency (Hz)
1M
±75
±50
±25
0
25
50
75
100
125
150
Temperature (ƒC)
C012
Figure 6-19. CMRR and PSRR vs Frequency
C001
Figure 6-20. CMRR vs Temperature
0.6
Noise (400 nV/div)
Power-Supply Rejection Ratio (µV/V)
1
0.8
0.4
0.2
0
–0.2
–0.4
–0.6
–0.8
Peak-to-Peak Noise = VRMS × 6.6 = 1.30 µVpp
–1
–75
–50 –25
0
25
50
75
100
125
Temperature (°C)
Time (1 s/div)
150
C001
C001
Figure 6-22. 0.1-Hz to 10-Hz Noise
Figure 6-21. PSRR vs Temperature
Total Harmonic Distortion + Noise (%)
Voltage Noise Density (nV/√Hz)
0.1
VCM = V+ – 100 mV
N-Channel Input
100
10
VCM = 0 V
P-Channel Input
1
0.1
1
10
100
Frequency (Hz)
1k
10k
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G = +1 V/V, RL = 2 kΩ
G = –1 V/V, RL = 10 kΩ
0.01
–80
G = –1 V/V, RL = 2 kΩ
0.001
–100
0.0001
–120
0.00001
100k
C002
Figure 6-23. Input Voltage Noise Spectral Density vs Frequency
16
–60
G = +1 V/V, RL = 10 kΩ
Total Harmonic Distortion + Noise (dB)
1000
–140
10
100
1k
10k
Frequency (Hz)
VOUT = 3.5 VRMS
BW = 80 kHz
Figure 6-24. THD+N Ratio vs Frequency
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6.9 Typical Characteristics (continued)
1.2
–60
0.01
–80
0.001
–100
0.0001
G = +1
G = +1
G = –1
G = –1
0.00001
0.01
–120
V/V, RL = 10 kΩ
V/V, RL = 2 kΩ
V/V, RL = 10 kΩ
V/V, RL = 2 kΩ
0.1
Quiescent Current (mA)
0.1
Total Harmonic Distortion + Noise (dB)
Total Harmonic Distortion + Noise (%)
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
1.0
0.9
0.8
–140
1
1.1
0
10
4
8
12
16
20
24
28
32
36
Supply Voltage (V)
Output Amplitude (VRMS)
f = 1 kHz, BW = 80 kHz
Figure 6-26. Quiescent Current vs Supply Voltage
Figure 6-25. THD+N vs Output Amplitude
1.2
3.0
VS = 4.5 V
VS = 36 V
Open-Loop Gain (µV/V)
Quiescent Current (mA)
2.0
1.1
VS = ±18 V
1
VS = ±2.25 V
0.9
1.0
0.0
–1.0
–2.0
0.8
–3.0
75
50
25
0
25
50
75
100
125
Temperature (°C)
150
–75
–50
–25
0
25
50
75
100
125
150
Temperature (°C)
C001
RL = 10 kΩ
Figure 6-27. Quiescent Current vs Temperature
Figure 6-28. Open-Loop Gain vs Temperature
10k
50
45
+ 18 V
1k
Overshoot (%)
Output Impedance ( )
40
100
35
+
VIN
-
30
-
RISO
OPA197-Q1
+
CL
-18 V
25
20
RISO = 0 0
Ω
15
25
RISO = 25 Ω
10
RISO = 50 Ω
5
0
10
0
1
10
100
1k
10k
Frequency (Hz)
100k
1M
10M
10p
RI = 1 kΩ
Figure 6-29. Open-Loop Output Impedance vs Frequency
100p
1n
Capacitive Load (F)
C016
RF = 1 kΩ
G = –1
Figure 6-30. Small-Signal Overshoot vs Capacitive Load (100mV Output Step)
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6.9 Typical Characteristics (continued)
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
50
-
35
-
RISO
VIN
+
RL
CL
+
+
-
37 VPP -18 V
Sine Wave
(±18.5 V)
-18 V
-
30
VOUT
OPA197-Q1
OPA197-Q1
+
5 V/div
Overshoot (%)
40
VIN
+ 18 V
+ 18 V
45
25
20
15
VOUT
RISO = 0 Ω0
RISO = 25
25 Ω
RISO = 50
50 Ω
10
5
0
10p
100p
Time (200 μs/div)
1n
Capacitive Load (F)
G=1
Figure 6-32. No Phase Reversal
Figure 6-31. Small-Signal Overshoot vs Capacitive Load (100mV Output Step)
+ 18 V
VOUT
+
VIN
+ 18 V
OPA197-Q1
-
VOUT
+
-18 V
VOUT
+
VIN
VOUT
OPA197-Q1
5 V/div
+
5 V/div
-
-
- 18 V
VIN
VIN
Time (200 ns/div)
RI = 1 kΩ
Time (200 ns/div)
RF = 10 kΩ
G = –10
RI = 1 kΩ
Figure 6-33. Positive Overload Recovery
G = –10
RF = 10 kΩ
Figure 6-34. Negative Overload Recovery
+ 18 V
-
+
OPA197-Q1
+
20 mV/div
20 mV/div
VIN
+ 18 V
-
-
CL
- 18 V
OPA197-Q1
+
VIN
+
-18 V
RL
CL
-
Time (120 ns/div)
Time (100 ns/div)
CL = 10 pF
RL = 1 kΩ
G=1
Figure 6-35. Small-Signal Step Response (100 mV)
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CL = 10 pF
G = –1
Figure 6-36. Small-Signal Step Response (100 mV)
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6.9 Typical Characteristics (continued)
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
2 V/div
Output Delta from Final Value (mV)
4
+ 18 V
-
+
OPA197-Q1
+
VIN
-
CL
-18 V
3
2
1
0
–1
0.01% Settling = ±1 mV
–2
–3
Step Applied at t = 0
–4
0
Time (300 ns/div)
RL = 1 kΩ
0.5
0.75
CL = 10 pF
G = –1
1.25
1.5
1.75
2
G=1
Figure 6-38. Settling Time (10-V Positive Step)
4
Output Delta from Final Value (mV)
4
3
2
1
0
0.01% Settling = ±500 μV
–1
–2
–3
Step Applied at t = 0
–4
3
2
1
0
–1
0.01% Settling = ±1 mV
–2
–3
Step Applied at t = 0
–4
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
1.8
0
0.2
0.4
0.6
0.8
Time (μs)
1
1.2
1.4
1.6
1.8
2
Time (μs)
G=1
G=1
Figure 6-39. Settling Time (5-V Positive Step)
Figure 6-40. Settling Time (10-V Negative Step)
80
4
ISC, Source
3
Short-Circuit Current (mA)
Output Delta from Final Value (mV)
1
Time (μs)
Figure 6-37. Large-Signal Step Response
Output Delta from Final Value (mV)
0.25
2
1
0
0.01% Settling = ±500 μV
–1
–2
ISC, Sink
60
40
20
–3
Step Applied at t = 0
0
–4
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
Time (μs)
1.8
–75
–50 –25
0
25
50
75
100
125
Temperature (°C)
150
C001
G=1
Figure 6-41. Settling Time (5-V Negative Step)
Figure 6-42. Short-Circuit Current vs Temperature
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6.9 Typical Characteristics (continued)
at TA = 25°C, VS = ±18 V, VCM = VS / 2, RLOAD = 10 kΩ connected to VS / 2, and CL = 100 pF, (unless otherwise noted)
30
Maximum output voltage without
slew-rate induced distortion.
VS = ±15 V
Overdrive = 100 mV
Output Voltage (5 V/div)
Output Voltage (VPP)
25
20
15
VS = ±5 V
10
VS = ±2.25 V
5
tpLH = 0.97 s
VOUT Voltage
0
10k
100k
1M
Time (200 ns/div)
10M
Frequency (Hz)
C025
C033
Figure 6-44. Propagation Delay Rising Edge
Figure 6-43. Maximum Output Voltage vs Frequency
-100
VOUT Voltage
Crosstalk (db)
Output Voltage (1 V/div)
-80
tpLH = 1.1 s
Overdrive = 100 mV
-120
-140
-160
Time (200 ns/div)
-180
1k
10k
Figure 6-45. Propagation Delay Falling Edge
20
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100k
1M
Frequency (Hz)
C026
Figure 6-46. Crosstalk vs Frequency
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SBOS918A – MARCH 2018 – REVISED JANUARY 2021
7 Detailed Description
7.1 Overview
The OPAx197-Q1 family of e-trim operational amplifiers use a proprietary method of package-level trim for offset
and offset temperature drift implemented during the final steps of manufacturing after the plastic molding
process. This method minimizes the influence of inherent input transistor mismatch, as well as errors induced
during package molding. The trim communication occurs on the output pin of the standard pinout, and after the
trim points are set, further communication to the trim structure is permanently disabled. Section 7.2 shows the
simplified diagram of the OPAx197-Q1.
Unlike previous e-trim op amps, the OPAx197-Q1 uses a patented two-temperature trim architecture to achieve
a very-low offset voltage of 25 µV (maximum) and low voltage offset drift of 0.5 µV/°C (maximum) over the full
specified temperature range. This level of precision performance at wide supply voltages makes these amplifiers
especially useful for high-impedance industrial sensors, filters, and high-voltage data acquisition.
7.2 Functional Block Diagram
+
NCH Input
Stage
±
IN+
36-V
Differential
Front End
Slew
Boost
High Capacitive
Load
Compensation
+
Output
Stage
VOUT
±
IN±
+
PCH Input
Stage
±
e-WULPŒ
Package Level Trim
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7.3 Feature Description
7.3.1 Input Protection Circuitry
The OPAx197-Q1 use a unique input architecture to eliminate the need for input protection diodes but still
provide robust input protection under transient conditions. Conventional input diode protection schemes shown
in Figure 7-1 can be activated by fast transient step responses, and can introduce signal distortion and settlingtime delays because of alternate current paths, as shown in Figure 7-2. For low-gain circuits, these fast-ramping
input signals forward-bias back-to-back diodes, causing an increase in input current, and resulting in extended
settling time, as shown in Figure 7-3.
V+
V+
+
VIN+
+
VIN+
VOUT
VOUT
OPAx197-Q1
36 V
~0.7 V
VIN
VIN
V
OPA197-Q1 Provides Full 36-V
Differential Input Range
V
Conventional Input Protection
Limits Differential Input Range
Copyright © 2018, Texas Instruments Incorporated
Figure 7-1. OPAx197-Q1 Input Protection Does Not Limit Differential Input Capability
Vn = +10 V
RFILT
+10 V
1
Ron_mux
Sn
1
D
+10 V
CFILT
2
~±9.3 V
CS
CD
Vn+1 = ±10 V RFILT
±10 V
Vin±
2
Ron_mux
Sn+1
~0.7 V
CS
CFILT
Vout
Idiode_transient
±10 V
Input Low Pass Filter
Vin+
Buffer Amplifier
Simplified Mux Model
Figure 7-2. Back-to-Back Diodes Create Settling Issues
Output Delta From Final Value (mV)
100
Standard Input Diode Structure
Extends Settling Time
80
60
40
0.1% Settling = ±10 mV
20
0
–20
OPA197-Q1 Input Structure
Offers Fast Settling
–40
–60
–80
–100
0
5
10
15
20
25
30
35
Time (µs)
40
45
50
55
60
C040
Figure 7-3. OPAx197-Q1 Protection Circuit Maintains Fast-Settling Transient Response
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The OPAx197-Q1 family of operational amplifiers provides a true high-impedance differential input capability for
high-voltage applications. This patented input protection architecture does not introduce additional signal
distortion or delayed settling time, making these devices the optimal op amps for multichannel, high-switched,
input applications. The OPAx197-Q1 tolerate a maximum differential swing (voltage between inverting and
noninverting pins of the op amp) of up to 36 V, making these devices an excellent choice for use as comparators
or in applications with fast-ramping input signals, such as multiplexed data-acquisition systems; see Figure 8-1.
7.3.2 EMI Rejection
The OPAx197-Q1 use integrated electromagnetic interference (EMI) filtering to reduce the effects of EMI from
sources such as wireless communications and densely-populated boards with a mix of analog signal chain and
digital components. EMI immunity can be improved with circuit design techniques; the OPAx197-Q1 benefit from
these design improvements. Texas Instruments has developed the ability to accurately measure and quantify the
immunity of an operational amplifier over a broad frequency spectrum extending from 10 MHz
to 6 GHz. Figure 7-4 shows the results of this testing on the OPAx197-Q1. Table 7-1 shows the EMIRR IN+
values for the OPAx197-Q1 at particular frequencies commonly encountered in real-world applications.
Applications listed in Table 7-1 may be centered on or operated near the particular frequency shown. Detailed
information can also be found in the TI application report EMI Rejection Ratio of Operational Amplifiers available
for download from www.ti.com.
160.0
140.0
PRF = -10 dBm
VSUPPLY = ±18 V
VCM = 0 V
EMIRR IN+ (dB)
120.0
100.0
80.0
60.0
40.0
20.0
0.0
10M
100M
1G
Frequency (Hz)
10G
C017
Figure 7-4. EMIRR Testing
Table 7-1. OPAx197-Q1 EMIRR IN+ For Frequencies of Interest
FREQUENCY
APPLICATION OR ALLOCATION
EMIRR IN+
400 MHz
Mobile radio, mobile satellite, space operation, weather, radar, ultra-high frequency (UHF) applications
44.1 dB
900 MHz
Global system for mobile communications (GSM) applications, radio communication, navigation, GPS (to
1.6 GHz), GSM, aeronautical mobile, UHF applications
52.8 dB
1.8 GHz
GSM applications, mobile personal communications, broadband, satellite, L-band (1 GHz to 2 GHz)
61.0 dB
Bluetooth®,
2.4 GHz
802.11b, 802.11g, 802.11n,
mobile personal communications, industrial, scientific and
medical (ISM) radio band, amateur radio and satellite, S-band (2 GHz to 4 GHz)
69.5 dB
3.6 GHz
Radiolocation, aero communication and navigation, satellite, mobile, S-band
88.7 dB
802.11a, 802.11n, aero communication and navigation, mobile communication, space and satellite
operation, C-band (4 GHz to 8 GHz)
105.5 dB
5 GHz
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7.3.3 Phase Reversal Protection
The OPAx197-Q1 family has internal phase-reversal protection. Many op amps exhibit a phase reversal when
the input is driven beyond its linear common-mode range. This condition is most often encountered in
noninverting circuits when the input is driven beyond the specified common-mode voltage range, causing the
output to reverse into the opposite rail. The OPAx197-Q1 is a rail-to-rail input op amp; therefore, the commonmode range can extend up to the rails. Input signals beyond the rails do not cause phase reversal; instead, the
output limits into the appropriate rail. This performance is shown in Figure 7-5.
VIN
+ 18 V
VOUT
5 V/div
OPA197-Q1
+
+
-
37 VPP -18 V
Sine Wave
(±18.5 V)
VOUT
Time (200 μs/div)
Figure 7-5. No Phase Reversal
7.3.4 Thermal Protection
+30 V
VOUT
The internal power dissipation of any amplifier causes the internal (junction) temperature to rise. This
phenomenon is called self heating. The absolute maximum junction temperature of the OPAx197-Q1 is 150°C
and exceeding this maximum temperature causes damage to the device. The OPAx197-Q1 have a thermal
protection feature that prevents damage from self heating. The protection works by monitoring the temperature
of the device and turning off the op amp output drive for temperatures above 140°C. Figure 7-6 shows an
application example for the OPAx197-Q1 that has significant self heating (159°C) because of the power
dissipation (0.81 W). Thermal calculations indicate that for an ambient temperature of 65°C, the device junction
temperature must reach 187°C. The actual device, however, turns off the output drive to maintain a safe junction
temperature. Figure 7-6 shows how the circuit behaves during thermal protection. During normal operation, the
device acts as a buffer so the output is 3 V. When self heating causes the device junction temperature to
increase above 140°C, the thermal protection forces the output to a high-impedance state and the output is
pulled to ground through resistor RL.
TA = 65°C
PD = 0.81W
R JA = 116°C/W
TJ = 116°C/W × 0.81W + 65°C
TJ = 159°C (expected)
3V
Normal
Operation
0V
Output
High-Z
150°C
OPAx197-Q1
+
±
VIN
3V
+
RL
3V
100 Ÿ ±
140ºC
Temperature
IOUT = 30 mA
Copyright © 2017, Texas Instruments Incorporated
Figure 7-6. Thermal Protection
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7.3.5 Capacitive Load and Stability
The OPAx197-Q1 feature a patented output stage capable of driving large capacitive loads, and in a unity-gain
configuration, directly drive up to 1 nF of pure capacitive load. Increasing the gain enhances the ability of these
amplifiers to drive greater capacitive loads; see Figure 7-7 and Figure 7-8. The particular op-amp circuit
configuration, layout, gain, and output loading are some of the factors to consider when establishing whether an
amplifier is stable in operation.
50
50
45
+ 18 V
45
-
+ 18 V
35
+
VIN
-
30
-
40
RISO
OPA197-Q1
+
Overshoot (%)
Overshoot (%)
40
CL
-18 V
25
35
+
VIN
+
RL
25
RISO = 0 0
Ω
15
25
RISO = 25 Ω
15
10
RISO = 50 Ω
10
20
RISO = 0 Ω0
RISO = 25
25 Ω
RISO = 50
50 Ω
5
0
0
10p
100p
1n
CL
-18 V
-
30
20
5
RISO
OPA197-Q1
10p
100p
Capacitive Load (F)
1n
Capacitive Load (F)
Figure 7-7. Small-Signal Overshoot vs Capacitive
Load (100-mV Output Step)
Figure 7-8. Small-Signal Overshoot vs Capacitive
Load (100-mV Output Step)
For additional drive capability in unity-gain configurations, improve capacitive load drive by inserting a small (10Ω to 20-Ω) resistor, RISO, in series with the output, as shown in Figure 7-9. This resistor significantly reduces
ringing and maintains dc performance for purely capacitive loads. However, if a resistive load is in parallel with
the capacitive load, then a voltage divider is created, thus introducing a gain error at the output and slightly
reducing the output swing. The error introduced is proportional to the ratio RISO / RL, and is generally negligible
at low output levels. A high capacitive load drive makes the OPAx197-Q1 a great choice for applications such as
reference buffers, MOSFET gate drives, and cable-shield drives. The circuit shown in Figure 7-9 uses an
isolation resistor, RISO, to stabilize the output of an op amp. RISO modifies the open-loop gain of the system for
increased phase margin, and results using the OPAx197-Q1 are summarized in Table 7-2. For additional
information on techniques to optimize and design using this circuit, TI Precision Design TIPD128 details
complete design goals, simulation, and test results.
+Vs
Vout
Riso
+
Vin
+
±
Cload
-Vs
Figure 7-9. Extending Capacitive Load Drive With the OPAx197-Q1
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Table 7-2. OPAx197-Q1 Capacitive Load Drive Solution Using Isolation Resistor Comparison of
Calculated and Measured Results
PARAMETER
VALUE
Capacitive Load
100 pF
1000 pF
0.01 µF
0.1 µF
1 µF
Phase Margin
45°
60°
45°
60°
45°
60°
45°
60°
45°
60°
RISO (Ω)
47
360
24
100
20
51
6.2
15.8
2
4.7
Measured
Overshoot (%)
23.2 8.6
10.4
22.5
9
22.1
8.7
23.1
8.6
21
8.6
Calculated PM
45.1°
58.1°
45.8°
59.7°
46.1°
60.1°
45.2°
60.2°
47.2°
60.2°
For step-by-step design procedure, circuit schematics, bill of materials, printed circuit board (PCB) files, simulation results, and test
results, see TI Precision Design TIPD128Capacitive Load Drive Solution using an Isolation Resistor.
7.3.6 Common-Mode Voltage Range
The OPAx197-Q1 are 36-V, true rail-to-rail input operational amplifiers with an input common-mode range that
extends 100 mV beyond either supply rail. This wide range is achieved with paralleled complementary N-channel
and P-channel differential input pairs, as shown in Figure 7-10. The N-channel pair is active for input voltages
close to the positive rail, typically (V+) – 3 V to 100 mV above the positive supply. The P-channel pair is active
for inputs from 100 mV below the negative supply to approximately (V+) – 1.5 V. There is a small transition
region, typically ( V+) –3 V to (V+) – 1.5 V, in which both input pairs are on. This transition region can vary
modestly with process variation, and within this region, PSRR, CMRR, offset voltage, offset drift, noise, and THD
performance may be degraded compared to operation outside this region.
+Vsupply
IS1
VIN±
PCH1
PCH2
NCH4
NCH3
VIN+
e-trimTM
FUSE BANK
VOS TRIM
VOS DRIFT TRIM
±Vsupply
Figure 7-10. Rail-to-Rail Input Stage
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To achieve the best performance for two-stage rail-to-rail input amplifiers, avoid the transition region when
possible. The OPAx197-Q1 use a precision trim for both the N-channel and P-channel regions. This technique
enables significantly lower levels of offset than previous-generation devices, causing variance in the transition
region of the input stages to appear exaggerated relative to offset over the full common-mode range, as shown
in Figure 7-11.
Transition
Region
N-Channel
Region
P-Channel
Region
200
200
100
100
Input Offset Voltage ( V)
Input Offset Voltage ( V)
P-Channel
Region
0
±100
OPA197 e-Trim
Input Offset Voltage vs Vcm
±200
Transition
Region
N-Channel
Region
0
±100
±200
Input Offset Voltage vs Vcm
without e-Trim Input
±300
±15.0
±14.0
«
11.0
12.0
13.0
Common-Mode Voltage (V)
14.0
15.0
±300
±15.0
±14.0
«
11.0
12.0
13.0
Common-Mode Voltage (V)
14.0
15.0
Figure 7-11. Common-Mode Transition vs Standard Rail-to-Rail Amplifiers
7.3.7 Electrical Overstress
Designers often ask questions about the capability of an operational amplifier to withstand electrical overstress
(EOS). These questions tend to focus on the device inputs, but may involve the supply voltage pins or even the
output pin. Each of these different pin functions have electrical stress limits determined by the voltage
breakdown characteristics of the particular semiconductor fabrication process and specific circuits connected to
the pin. Additionally, internal electrostatic discharge (ESD) protection is built into these circuits to protect them
from accidental ESD events both before and during product assembly.
Having a good understanding of this basic ESD circuitry and its relevance to an electrical overstress event is
helpful. Figure 7-12 shows an illustration of the ESD circuits contained in the OPAx197-Q1 (indicated by the
dashed line area). The ESD protection circuitry involves several current-steering diodes connected from the
input and output pins and routed back to the internal power-supply lines, where the diodes meet at an absorption
device or the power-supply ESD cell, internal to the operational amplifier. This protection circuitry is intended to
remain inactive during normal circuit operation.
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TVS
+
±
RF
+VS
OPAx197-Q1
VDD
R1
IN±
100 Ÿ
IN+
100 Ÿ
RS
±
+
Power Supply
ESD Cell
VIN
RL
+
±
VSS
+
±
±VS
TVS
Copyright © 2017, Texas Instruments Incorporated
Figure 7-12. Equivalent Internal ESD Circuitry Relative to a Typical Circuit Application
An ESD event is very short in duration and very high voltage (for example, 1 kV, 100 ns), whereas an EOS event
is long duration and lower voltage (for example, 50 V, 100 ms). The ESD diodes are designed for out-of-circuit
ESD protection (that is, during assembly, test, and storage of the device before being soldered to the PCB).
During an ESD event, the ESD signal is passed through the ESD steering diodes to an absorption circuit
(labeled ESD power-supply circuit). The ESD absorption circuit clamps the supplies to a safe level.
Although this behavior is necessary for out-of-circuit protection, excessive current and damage is caused if
activated in-circuit. A transient voltage suppressors (TVS) can be used to prevent against damage caused by
turning on the ESD absorption circuit during an in-circuit ESD event. Using the appropriate current limiting
resistors and TVS diodes allows for the use of device ESD diodes to protect against EOS events.
7.3.8 Overload Recovery
Overload recovery is defined as the time required for the op amp output to recover from a saturated state to a
linear state. The output devices of the op amp enter a saturation region when the output voltage exceeds the
rated operating voltage, either due to the high input voltage or the high gain. After the device enters the
saturation region, the charge carriers in the output devices require time to return back to the linear state. After
the charge carriers return back to the linear state, the device begins to slew at the specified slew rate. Thus, the
propagation delay in case of an overload condition is the sum of the overload recovery time and the slew time.
The overload recovery time for the OPAx197-Q1 is approximately 200 ns.
7.4 Device Functional Modes
The OPAx197-Q1 have a single functional mode and is operational when the power-supply voltage is greater
than 4.5 V (±2.25 V). The maximum power supply voltage for the OPAx197-Q1 is 36 V (±18 V).
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8 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification, and TI
does not warrant its accuracy or completeness. TI’s customers are responsible for determining
suitability of components for their purposes, as well as validating and testing their design
implementation to confirm system functionality.
8.1 Application Information
The OPAx197-Q1 family offers outstanding dc precision and ac performance. These devices operate up to 36-V
supply rails and offer true rail-to-rail input and output, ultra-low offset voltage and offset voltage drift, as well as
10-MHz bandwidth and high capacitive load drive. These features make the OPAx197-Q1 a robust, highperformance operational amplifier for high-voltage industrial applications.
8.2 Typical Applications
8.2.1 16-Bit Precision Multiplexed Data-Acquisition System
Figure 8-1 shows a 16-bit, differential, 4-channel, multiplexed data-acquisition system. This example is typical in
industrial applications that require low distortion and a high-voltage differential input. The circuit uses the
ADS8864, a 16-bit, 400-kSPS successive-approximation-resistor (SAR) analog-to-digital converter (ADC), along
with a precision, high-voltage, signal-conditioning front end, and a 4-channel differential multiplexer (mux). This
TI Precision Design details the process for optimizing the precision, high-voltage, front-end drive circuit using the
OPAx197-Q1 and OPA140 to achieve excellent dynamic performance and linearity with the ADS8864.
1
2
Very Low Output Impedance
Input-Filter Bandwidth
High-Impedance Inputs
No Differential Input Clamps
Fast Settling-Time Requirements
Attenuate High-Voltage Input Signal
Fast-Settling Time Requirements
Stability of the Input Driver
4
Attenuate ADC Kickback Noise
VREF Output: Value and Accuracy
Low Temp and Long-Term Drift
Voltage
Reference
CH0+
OPAx197-Q1
±20-V,
10-kHz
Sine Wave
3
+
RC Filter
Buffer
RC Filter
Reference Driver
+
CH0-
OPAx197-Q1
Gain
Network
OPAx197-Q1
Gain
Network
+
4:2
Mux
REFP
+
OPAx197-Q1
CH3+
OPAx197-Q1
+
+
Antialiasing
Filter
SAR
ADC
+
VINM
OPAx197-Q1
CH3-
n
16 Bits
400 kSPS
High-Voltage Level Translation
VCM
High-Voltage Multiplexed Input
CONV
Gain
Network
±20-V,
10-kHz
Sine Wave
VINP
OPAx197-Q1
Gain
Network
REF3240
Voltage
Divider
OPA350
VCM Generation Circuit
Counter
n
5
Shmidtt
Trigger
Delay
Digital Counter For Multiplexer
Fast logic transition
Copyright © 2017, Texas Instruments Incorporated
Figure 8-1. OPAx197-Q1 in 16-Bit, 400-kSPS, 4-Channel, Multiplexed Data Acquisition System for HighVoltage Inputs With Lowest Distortion
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8.2.1.1 Design Requirements
The primary objective is to design a ±20-V, differential, 4-channel, multiplexed data acquisition system with
lowest distortion using the 16-bit ADS8864 at a throughput of 400 kSPS for a 10-kHz, full-scale, pure sine-wave
input. The design requirements for this block design are:
•
•
•
•
•
System supply voltage: ±15 V
ADC supply voltage: 3.3 V
ADC sampling rate: 400 kSPS
ADC reference voltage (REFP): 4.096 V
System input signal: A high-voltage differential input signal with a peak amplitude of 10 V and frequency (fIN)
of 10 kHz are applied to each differential input of the multiplexer.
8.2.1.2 Detailed Design Procedure
The purpose of this precision design is to design an optimal high voltage multiplexed data acquisition system for
highest system linearity and fast settling. The overall system block diagram is illustrated in Figure 8-1. The circuit
is a multichannel data acquisition signal chain consisting of an input low-pass filter, mux, mux output buffer,
attenuating SAR ADC driver, digital counter for mux and the reference driver. The architecture allows fast
sampling of multiple channels using a single ADC, providing a low-cost solution. The two primary design
considerations to maximize the performance of a precision multiplexed data acquisition system are the mux input
analog front-end and the high-voltage level translation SAR ADC driver design. However, carefully design each
analog circuit block based on the ADC performance specifications in order to achieve the fastest settling at 16-bit
resolution and lowest distortion system. Figure 8-1 includes the most important specifications for each individual
analog block.
This design systematically approaches each analog circuit block to achieve a 16-bit settling for a full-scale input
stage voltage and linearity for a 10-kHz sinusoidal input signal at each input channel. The first step in the design
is to understand the requirement for extremely low impedance input-filter design for the mux. This understanding
helps in the decision of an appropriate input filter and selection of a mux to meet the system settling
requirements. The next important step is the design of the attenuating analog front-end (AFE) used to level
translate the high-voltage input signal to a low-voltage ADC input when maintaining amplifier stability. The next
step is to design a digital interface to switch the mux input channels with minimum delay. The final design
challenge is to design a high-precision, reference-driver circuit that provides the required REFP reference
voltage with low offset, drift, and noise contributions.
For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test results, refer to TI
Precision Design TIPD151, 16-bit, 400-kSPS, 4-Channel, Multiplexed Data Acquisition System for High Voltage Inputs with Lowest
Distortion.
8.2.1.3 Application Curve
Integral Nonlinearity Error (LSB)
2.0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
–20
–15
–10
–5
0
5
10
15
20
ADC Differential Input (V)
Figure 8-2. ADC 16-Bit Linearity Error for the Multiplexed Data Acquisition Block
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8.2.2 Slew-Rate Limit for Input Protection
In control systems for valves or motors, abrupt changes in voltages or currents can cause mechanical damages.
By controlling the slew rate of the command voltages into the drive circuits, the load voltages ramps up and
down at a safe rate. For symmetrical slew-rate applications (positive slew rate equals negative slew rate), one
additional op amp provides slew-rate control for a given analog gain stage. The unique input protection and high
output current and slew rate of the OPAx197-Q1 make these devices the optimal amplifiers to achieve slew-rate
control for both dual- and single-supply systems.Figure 8-3 shows the OPAx197-Q1 in a slew-rate limit design.
Op Amp Gain Stage
Slew Rate Limiter
C1
470 nF
R1
1.69 NŸ
VEE
VEE
+
R2
1.6 0Ÿ
VIN
OPAx197-Q1
V+
VOUT
OPAx197-Q1
V+
VCC
RL
10 NŸ
VCC
Copyright © 2017, Texas Instruments Incorporated
Figure 8-3. Slew-Rate Limiter Uses One Op Amp
For step-by-step design procedure, circuit schematics, bill of materials, PCB files, simulation results, and test results, see TI
Precision Design TIPD140, Slew Rate Limiter Uses One Op Amp.
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8.2.3 Precision Reference Buffer
The OPAx197-Q1 feature high output-current-drive capability and low input offset voltage, making these devices
an excellent reference buffer to provide an accurate buffered output with ample drive current for transients. For
the 10-µF ceramic capacitor shown in Figure 8-4, a 37.4-Ω isolation resistor (RISO), provides separation of two
feedback paths for optimal stability. Feedback path number one is through RF and is directly at the output
(VOUT). Feedback path number two is through RFx and CF and is connected at the output of the op amp. The
optimized stability components shown for the 10-µF load give a closed-loop signal bandwidth at VOUT of 4 kHz
and still provide a loop gain phase margin of 89°. Any other load capacitances require recalculation of the
stability components: RF, RFx , CF , and RISO.
RF
1 kŸ
RFx
10 kŸ
CF
39 nF
RISO
37.4 Ÿ
VOUT
OPAx197-Q1
V+
CL
10 µF
VREF
2.5 V
VCC
Copyright © 2017, Texas Instruments Incorporated
Figure 8-4. Precision Reference Buffer
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9 Power Supply Recommendations
The OPAx197-Q1 are specified for operation from 4.5 V to 36 V (±2.25 V to ±18 V); many specifications apply
from –40°C to +125°C. Parameters that can exhibit significant variance with regard to operating voltage or
temperature are presented in Section 6.9.
CAUTION
Supply voltages larger than 40 V can permanently damage the device; see Absolute Maximum
Ratings.
Place 0.1-μF bypass capacitors close to the power-supply pins to reduce errors coupling in from noisy or highimpedance power supplies. For more detailed information on bypass capacitor placement, see Section 10.
10 Layout
10.1 Layout Guidelines
For best operational performance of the device, use good PCB layout practices, including:
•
•
•
•
•
•
•
•
Noise can propagate into analog circuitry through the power pins of the circuit as a whole and op amp itself.
Bypass capacitors are used to reduce the coupled noise by providing low-impedance power sources local to
the analog circuitry.
– Connect low-ESR, 0.1-µF ceramic bypass capacitors between each supply pin and ground, placed as
close as possible to the device. A single bypass capacitor from V+ to ground is applicable for singlesupply applications.
Separate grounding for analog and digital portions of circuitry is one of the simplest and most-effective
methods of noise suppression. One or more layers on multilayer PCBs are usually devoted to ground planes.
A ground plane helps distribute heat and reduces EMI noise pickup. Make sure to physically separate digital
and analog grounds paying attention to the flow of the ground current.
To reduce parasitic coupling, run the input traces as far away as possible from the supply or output traces. If
these traces cannot be kept separate, crossing the sensitive trace perpendicular is much better than in
parallel with the noisy trace.
Place the external components as close as possible to the device. As illustrated in Figure 10-2, keep RF and
RG close to the inverting input to minimize parasitic capacitance.
Keep the length of input traces as short as possible. Always remember that the input traces are the most
sensitive part of the circuit.
Consider a driven, low-impedance guard ring around the critical traces. A guard ring can significantly reduce
leakage currents from nearby traces that are at different potentials.
For best performance, clean the PCB following board assembly.
Any precision integrated circuit may experience performance shifts due to moisture ingress into the plastic
package. Following any aqueous PCB cleaning process, bake the PCB assembly to remove moisture
introduced into the device packaging during the cleaning process. A low-temperature, post-cleaning bake at
85°C for 30 minutes is sufficient for most circumstances.
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10.2 Layout Examples
+
VIN
VOUT
RG
RF
Figure 10-1. Schematic Representation
Run the input traces
as far away from
the supply lines
as possible
Place components close
to device and to each
other to reduce parasitic
errors
RF
VS+
N/C
N/C
GND
±IN
V+
VIN
+IN
OUTPUT
V±
N/C
RG
GND
GND
VOUT
Ground (GND) plane on another layer
Use low-ESR,
ceramic bypass
capacitors
Figure 10-2. Operational Amplifier Board Layout for Noninverting Configuration
34
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SBOS918A – MARCH 2018 – REVISED JANUARY 2021
11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 TINA-TI™ Simulation Software (Free Download)
TINA™ is a simple, powerful, and easy-to-use circuit simulation program based on a SPICE engine. TINA-TI™
simulation software is a free, fully functional version of the TINA software, preloaded with a library of macro
models in addition to a range of both passive and active models. TINA-TI simulation software provides all the
conventional dc, transient, and frequency domain analysis of SPICE, as well as additional design capabilities.
Available as a free download from the Analog eLab Design Center, TINA-TI simulation software offers extensive
post-processing capability that allows users to format results in a variety of ways. Virtual instruments offer the
ability to select input waveforms and probe circuit nodes, voltages, and waveforms, creating a dynamic quickstart tool.
Note
These files require that either the TINA software (from DesignSoft™) or TINA-TI software be installed.
Download the free TINA-TI software from the TINA-TI folder.
11.1.1.2 TI Precision Designs
The OPA197 is featured in several Texas Instruments (TI) Precision Designs, available online at http://
www.ti.com/ww/en/analog/precision-designs/. TI Precision Designs are analog solutions created by TI’s precision
analog applications experts and offer the theory of operation, component selection, simulation, complete PCB
schematic and layout, bill of materials, and measured performance of many useful circuits.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
• Texas Instruments, EMI Rejection Ratio of Operational Amplifiers application report
• Texas Instruments, Capacitive Load Drive Solution using an Isolation Resistor reference design
11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
11.4 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
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11.5 Trademarks
e-trim™, TINA-TI™, and TI E2E™ are trademarks of Texas Instruments.
TINA™ and DesignSoft™ are trademarks of DesignSoft, Inc.
Bluetooth® is a registered trademark of Bluetooth SIG, Inc.
All trademarks are the property of their respective owners.
11.6 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
11.7 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
www.ti.com
7-Feb-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
OPA197QDGKRQ1
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
197
OPA2197QDGKRQ1
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
2197
OPA4197QPWRQ1
ACTIVE
TSSOP
PW
14
2000
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 125
O4197Q
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of