OPA4820IPWR

OPA4820 SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 Quad, Unity-Gain Stable, Low-Noise, Voltage-Feedback Operational Amplifier FEATURES D D D D D D DESCRIPTION HIGH BANDWIDTH: 220MHz (G = +2) HIGH OUTPUT CURRENT: ±85mA LOW INPUT NOISE: 2.5nV/√Hz LOW SUPPLY CURRENT: 5.7mA/ch FLEXIBLE SUPPLY VOLTAGE: ±2V to ±6.3V Dual Supply +4V to +12.6V Single Supply EXCELLENT DC ACCURACY: Maximum 25°C Input Offset Voltage = 0.8mV Maximum 25°C Input Offset Current = 500nA The OPA4820 provides a wideband, unity-gain stable, voltage-feedback amplifier with a very low input noise voltage and high output current using a low 5.7mA/ch supply current. At unity-gain, the OPA4820 gives > 600MHz bandwidth with < 1 dB peaking. The OPA4820 complements this high-speed operation with excellent DC precision in a low-power device. A worst-case input offset voltage of ±0.8mV and an offset current of ±500nA give excellent absolute DC precision for pulse amplifier applications. Minimal input and output voltage swing headroom allow the OPA4820 to operate on a single +5V supply with > 2VPP output swing. While not a rail-to-rail (RR) output, this swing will support most emerging analog-to-digital converter (ADC) input ranges with lower power and noise than typical RR output op amps. APPLICATIONS D D D D D D D D D LOW-COST VIDEO LINE DRIVERS ADC PREAMPS ACTIVE FILTERS LOW-NOISE INTEGRATORS PORTABLE TEST EQUIPMENT OPTICAL CHANNEL AMPLIFIERS LOW-POWER, BASEBAND AMPLIFIERS CCD IMAGING CHANNEL AMPLIFIERS OPA4650 UPGRADE Exceptionally low dG/dP (0.01%/0.03°) supports low-cost composite video line driver applications. Existing designs can use the industry-standard quad pinout SO-14 package while emerging high-density portable applications can use the TSSOP-14. RELATED PRODUCTS +12V 1/4 OPA4820 10kΩ 402Ω Transmit Filter 50Ω VCM 133Ω 100Ω Line 14VPP 402Ω DUALS TRIPLES QUADS OPA354 OPA690 — — OPA820 OPA2354 OPA2690 OPA2652 OPA2822 — — OPA3690 — — — OPA4354 — — — — FEATURES CMOS RR Output High Slew Rate SOT23-8 Low-Noise Single Channel 1:1 +6V 2VPP SINGLES 50Ω 10dBm 3.5 Crest Factor 10kΩ 1/4 OPA4820 800Ω AFE 800Ω 402Ω 402Ω VCM 1/4 OPA4820 402Ω Receiver Filter 402Ω 1/4 OPA4820 VCM Low-Noise Transceiver Interface Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. Copyright  2004−2008, Texas Instruments Incorporated
          
             
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"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ABSOLUTE MAXIMUM RATINGS(1) Power Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±6.5V Internal Power Dissipation . . . . . . . . . See Thermal Characteristics Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±1.2V Input Common-Mode Voltage Range . . . . . . . . . . . . . . . . . . . . ±VS Storage Temperature Range . . . . . . . . . . . . . . . . . −65°C to +125°C Lead Temperature (soldering, 10s) . . . . . . . . . . . . . . . . . . . . +300°C Junction Temperature (TJ) . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C ESD Rating: Human Body Model (HBM) . . . . . . . . . . . . . . . . . . . . . . . 2000V Charge Device Model (CDM) . . . . . . . . . . . . . . . . . . . . . 1000V Machine Model (MM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 200V (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not supported. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. PACKAGE/ORDERING INFORMATION(1) PRODUCT PACKAGE-LEAD PACKAGE DESIGNATOR SPECIFIED TEMPERATURE RANGE PACKAGE MARKING OPA4820 ″ SO-14 D −45°C to +85°C OPA4820 OPA4820ID Rails, 58 ″ ″ ″ ″ OPA4820IDR Tape and Reel, 2500 OPA4820 ″ TSSOP-14 PW −45°C to +85°C OPA4820 OPA4820IPWT Tape and Reel, 250 ″ ″ ″ ″ OPA4820IPWR Tape and Reel, 2500 ORDERING NUMBER TRANSPORT MEDIA, QUANTITY (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet or see the TI website at www.ti.com. PIN CONFIGURATION TOP VIEW SO, TSSOP Output A 1 −Input A 2 A Output D 13 −Input D D +Input A 3 12 +Input D +V 4 11 −V +Input B 5 10 +Input C B C −Input B 6 9 −Input C Output B 7 8 Output C OPA4820 2 14 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 ELECTRICAL CHARACTERISTICS: VS = ±5V Boldface limits are tested at +25°C. At RF = 402Ω, RL = 100Ω, and GND = +2, unless otherwise noted. OPA4820ID, IPW TYP PARAMETER AC PERFORMANCE Small-Signal Bandwidth Gain-Bandwidth Product Peaking at a Gain of 1 Bandwidth for 0.1dB Gain Flatness Large-Signal Bandwidth Slew Rate Rise Time and Fall TIme Settling Time to 0.02% Settling Time to 0.1% Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Noninverting Input Current Noise Differential Gain Differential Phase All Hostile Crosstalk, Input-Referred DC PERFORMANCE(4) Open-Loop Voltage Gain (AOL) Input Offset Voltage Average Input Offset Voltage Drift Input Bias Current Average Input Bias Current Drift Input Offset Current Inverting Input Bias Current Drift INPUT Common-Mode Input Range (CMIR)(5) Common-Mode Rejection Ratio (CMRR) Input Impedance, Differential Mode Input Impedance, Common-Mode OUTPUT Output Voltage Swing Output Current Short-Circuit Output Current Closed-Loop Output Impedance POWER SUPPLY Specified Operating Voltage Maximum Operating Voltage Minimum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (−PSRR) THERMAL CHARACTERISTICS Specification: ID, IPW Thermal Resistance, qJA D SO-14 PW TSSOP-14 CONDITIONS +25°C G = +1, VO = 0.1VPP, RF = 25Ω G = +2, VO = 0.1VPP G = +10, VO = 0.1VPP G ≥ 20 VO = 0.1VPP, RF = 25Ω G = +2, VO = 0.1VPP G = +2, 2VPP G = +2, 2V Step G = +2, VO = 2V Step G = +2, VO = 2V Step G = +2, VO = 2V Step G = +2, f = 1MHz, VO = 2VPP RL = 200Ω RL ≥ 500Ω RL = 200Ω RL ≥ 500Ω f > 100kHz f > 100kHz G = +2, NTSC, VO = 1.4VPP, RL = 150Ω G = +2, NTSC, VO = 1.4VPP, RL = 150Ω 3 Channels Driven at 5MHz, 1VPP 4th Channel Measured 650 220 27 250 1 33 80 240 1.5 22 18 VO = 0V, RL = 100Ω VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V −84 −90 −92 −105 2.5 1.7 0.003 0.06 MIN/MAX OVER TEMPERATURE +25°C(1) 0°C to 70°C(2) −40°C to +85°C(2) 150 21 200 140 19 185 135 18 180 190 184 178 −80 −85 −87 −100 2.7 2.6 −79 −83 −86 −97 2.8 2.8 −78 −81 −85 −95 2.9 3.0 −61 66 ±0.25 62 ±0.8 −9 −20 ±100 ±500 VCM = 0V, Input-Referred VCM = 0V VCM = 0V ±4.0 85 18 0.8 6 1.0 RL ≥ 500Ω RL = 100Ω VO = 0 Output Shorted to Ground G = +2, f ≤ 100kHz ±3.7 ±3.6 ±85 ±110 0.04 Junction-to-Ambient Junction-to-Ambient MIN/ MAX TEST LEVEL(3) MHz MHz MHz MHz dB MHz MHz V/µs ns ns ns typ min min min typ typ typ min typ typ typ C B B B C C C B C C C dBc dBc dBc dBc nV/√Hz pA/√Hz % ° max max max max max max typ typ B B B B B B C C dB typ C 61 ±1.2 8 −22 30 ±800 5 60 ±1.5 10 −26 50 ±900 5 dB mV µV/°C µA nA/°C nA nA/°C min max max max max max max A A B A B A B ±3.8 76 ±3.7 75 ±3.6 73 V dB kΩ  pF MΩ  pF min min typ typ A A C C ±3.5 ±3.5 ±70 ±3.45 ±3.45 ±65 ±3.4 ±3.4 ±60 V V mA mA Ω min min min typ typ A A A C C ±6.3 ±6.3 ±6.3 23.4 21.8 64 25 20.2 63 25.8 19.4 62 V V V mA mA dB typ max typ max min min C A C A A A −40 to +85 °C typ C 100 110 °C/W °C/W typ typ C C ±5 VS = ±5V VS = ±5V Input-Referred UNITS ±2 22.6 22.6 72 (1) Junction temperature = ambient for +25°C specifications. Junction temperature = ambient at low temperature limits; junction temperature = ambient +28°C at high temperature limit for over temperature specifications. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of pin. (5) Tested < 3dB below minimum specified CMRR at ± CMIR limits. (2) 3 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 ELECTRICAL CHARACTERISTICS: VS = +5V Boldface limits are tested at +25°C. At RF = 402Ω, RL = 100Ω to 2.5V, and G = +2, unless otherwise noted. OPA4820ID, IPW TYP PARAMETER AC PERFORMANCE Small-Signal Bandwidth Gain-Bandwidth Product Peaking at a Gain of 1 Large-Signal Bandwidth Slew Rate Rise Time and Fall Time Settling Time to 0.02% Settling Time to 0.1% Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Noninverting Input Current Noise All Hostile Crosstalk, Input-Referred DC PERFORMANCE(4) Open-Loop Voltage Gain (AOL) Input Offset Voltage Average Input Offset Voltage Drift Input Bias Current Average Input Bias Current Drift Input Offset Current Inverting Input Bias Current Drift INPUT Least Positive Input Voltage Most Positive Input Voltage Common-Mode Rejection Ratio (CMRR)(5) Input Impedance, Differential-Mode Input Impedance, Common-Mode OUTPUT Least Positive Output Voltage Most Negative Output Voltage Output Current Short-Circuit Output Current Closed-Loop Output Impedance POWER SUPPLY Specified Operating Voltage Maximum Operating Voltage Minimum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (+PSRR) MIN/MAX OVER TEMPERATURE +25°C(1) 0°C to 70°C(2) −40°C to +85°C(2) 148 18 180 135 17 170 130 16 160 135 130 125 −75 −79 −87 −95 2.8 2.5 −74 −77 −86 −93 2.9 2.7 −73 −75 −85 −92 3.0 2.9 CONDITIONS +25°C G = +1, VO = 0.1VPP, RF = 25Ω G = +2, VO = 0.1VPP G = +10, VO = 0.1VPP G ≥ 20 VO = 0.1VPP, RF = 25Ω G = +2, VO = 2VPP G = +2, VO = 2V Step G = +2, VO = 2V Step G = +2, VO = 2V Step G = +2, VO = 2V Step G = +2, f = 1MHz, VO = 2VPP RL = 200Ω RL ≥ 500Ω RL = 200Ω RL ≥ 500Ω f > 100kHz f > 100kHz 3 Channels Driven at 5MHz, 1VPP 4th Channel Measured 520 210 25 230 2 67 190 1.8 25 22 VO = 2.5V, RL = 100Ω VCM = 2.5V VCM = 2.5V VCM = 2.5V VCM = 2.5V VCM = 2.5V VCM = 2.5V 65 ±0.35 60 ±1.3 −8 −18 ±100 ±500 VCM = 2.5V, Input-Referred VCM = 2.5V VCM = 2.5V 0.9 4.4 83 15 1 5 1.3 RL ≥ 500Ω to 2.5V RL = 100Ω to 2.5V RL ≥ 500Ω to 2.5V RL = 100Ω to 2.5V VO = 2.5V Output Shorted to Ground G = +2, f ≤ 100kHz 3.9 3.8 1.2 1.2 ±75 ±105 0.04 −79 −83 −94 −98 2.5 1.6 −61 +4 20.4 20.4 68 UNITS MHz MHz MHz MHz dB MHz V/µs ns ns ns typ min min min typ typ min typ typ typ C B B B C C B C C C dBc dBc dBc dBc nV/√Hz pA/√Hz max max max max max max B B B B B B dB typ C (3) 59 ±1.7 8 −20 30 ±800 5 58 ±2.0 10 −24 50 ±900 5 dB mV µV/°C µA nA/°C µA nA/°C min max max max max max max A A B A B A B 1.1 4.1 74 1.2 4.0 73 1.3 3.9 72 V V dB kΩ  pF MΩ  pF min max min typ typ A A A C C 3.8 3.7 1.3 1.3 ±60 3.75 3.65 1.35 1.35 ±55 3.7 3.6 1.4 1.4 ±50 V V V V mA mA Ω min min min min min typ typ A A A A A C C +12.6 +12.6 +12.6 22.2 18.0 22.6 17.4 23.0 16.8 V V V mA mA dB typ max typ max min typ C A C A A C +5 VS = +5V, 4 Channels VS = +5V, 4 Channels Input-Referred TEST LEVEL MIN/ MAX THERMAL CHARACTERISTICS Specification: ID, IPW −40 to +85 °C typ C Thermal Resistance, qJA D SO-14 Junction-to-Ambient 100 °C/W typ C PW TSSOP-14 Junction-to-Ambient 110 °C/W typ C (1) Junction temperature = ambient for +25°C specifications. (2) Junction temperature = ambient at low temperature limits; junction temperature = ambient +13°C at high temperature limit for over temperature. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current considered positive out of pin. (5) Tested < 3dB below minimum specified CMRR at ± CMIR limits. 4 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. NONINVERTING SMALL−SIGNAL FREQUENCY RESPONSE 3 INVERTING SMALL−SIGNAL FREQUENCY RESPONSE 3 G = +1, RF = 25Ω G = −1 0 Normalized Gain (dB) −3 G = +5 G = +2 −6 G = +10 −9 −12 VO = 0.1VPP RL = 100Ω See Figure 1 −15 −18 1M −3 −6 G = −10 −9 −12 VO = 0.1VPP RL = 100Ω See Figure 2 −15 −18 10M 100M 1G 1 10 Frequency (Hz) 100 NONINVERTING LARGE−SIGNAL FREQUENCY RESPONSE INVERTING LARGE−SIGNAL FREQUENCY RESPONSE 3 VO = 0.5VPP 6 0 3 −3 VO = 1VPP 0 Gain (dB) Gain (dB) VO = 0.5VPP VO = 2VPP −3 VO = 4VPP −6 −12 1 VO = 1VPP VO = 2VPP −6 VO = 4VPP −9 −12 G = +2V/V RL = 100Ω See Figure 1 −9 G = −1V/V RL = 100Ω See Figure 2 −15 −18 10 100 500 1 10 Frequency (MHz) 0 1.5 1.0 Small−Signal ± 100mV Left Scale 0.5 0 −0.1 −0.5 −0.2 −1.0 −0.3 −1.5 −0.4 −2.0 Time (10ns/div) Small−Signal Output Voltage (100mV/div) 0.2 Large−Signal Output Voltage (500mV/div) Small−Signal Output Voltage (100mV/div) Large−Signal ± 1V Right Scale 500 INVERTING PULSE RESPONSE 2.0 G = +2V/V See Figure 1 100 Frequency (MHz) NONINVERTING PULSE RESPONSE 0.4 0.1 500 Frequency (MHz) 9 0.3 G = −2 G = −5 0.4 0.3 2.0 G = −1V/V See Figure 2 1.5 0.2 1.0 0.1 0 0.5 Small−Signal ± 100mV Left Scale −0.1 −0.2 −0.3 0 −0.5 Large−Signal ± 1V Right Scale −0.4 −1.0 −1.5 −2.0 Large−Signal Output Voltage (500mV/div) Normalized Gain (dB) 0 Time (10ns/div) 5 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V (continued) RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. 1MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE −60 −65 −65 Harmonic Distortion (dBc) Harmonic Distortion (dBc) HARMONIC DISTORTION vs LOAD RESISTANCE −60 −70 −75 2nd−Harmonic −80 −85 −90 G = +2V/V f = 1MHz VO = 2VPP −95 −100 3rd−Harmonic G = +2V/V RL = 200Ω VO = 2VPP See Figure 1 −70 −75 −80 2nd−Harmonic −85 −90 3rd−Harmonic −95 −100 −105 −105 100 2.5 1000 3.0 3.5 −70 2nd−Harmonic −75 −80 −85 −90 3rd−Harmonic −95 6.0 −70 −75 −80 2nd−Harmonic −85 −90 3rd−Harmonic −95 −105 −105 0.1 1 0.1 10 1 HARMONIC DISTORTION vs NONINVERTING GAIN f = 1MHz RL = 200Ω VO = 2VPP See Figure 1 −75 −80 HARMONIC DISTORTION vs INVERTING GAIN −70 2nd−Harmonic −75 Harmonic Distortion (dBc) −70 10 Output Voltage Swing (VPP) Frequency (MHz) Harmonic Distortion (dBc) 5.5 −100 −100 −85 −90 3rd−Harmonic −95 −100 −105 2nd−Harmonic −80 −85 3rd−Harmonic −90 −95 f = 1MHz RL = 200Ω VO = 2VPP See Figure 2 −100 −105 −110 −110 1 10 Gain (V/V) 6 5.0 G = +2V/V f = 1MHz RL = 200Ω See Figure 1 −65 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −60 G = +2V/V VO = 2VPP RL = 200Ω See Figure 1 −65 4.5 HARMONIC DISTORTION vs OUTPUT VOLTAGE HARMONIC DISTORTION vs FREQUENCY −60 4.0 Supply Voltage (±VS) Resistance (Ω) 1 10 Gain ( V/V ) 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V (continued) RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. TWO−TONE, 3RD−ORDER INTERMODULATION INTERCEPT INPUT VOLTAGE AND CURRENT NOISE 100 45 Intercept Point (+dBm) Voltage Noise (nV/√Hz) Current Noise (pA/√Hz) PI 1 /4 50Ω 40 10 Voltage Noise (2.5nV/√Hz) PO OP A4820 200Ω 402 Ω 35 402 Ω 30 25 20 Current Noise (1.7pA/√Hz) 1 100 1k 10k 100k 1M 15 10M 0 5 10 Frequency (Hz) Normalized Gain to Capacitive Load (dB) RS (Ω ) 0.1dB Peaking Targeted 10 1 10 100 8 5 CL = 47pF 4 CL = 100pF 3 2 RS VI 1 50Ω VO OPA4820 CL 0 1kΩ(1) 40 2Ω −1 402Ω −2 NOTE: (1 ) 1kΩ is optional. −3 1 1000 10 100 400 Frequency (MHz) OPEN−LOOP GAIN AND PHASE 80 CMRR 80 70 70 60 Open−Loop Gain (dB) Common−Mode Rejection Ratio (dB) Power−Supply Rejection Ratio (dB) 30 CL = 22pF 6 CMRR AND PSRR vs FREQUENCY 60 25 CL = 10pF 7 Capacitive Load (pF) 90 20 FREQUENCY RESPONSE vs CAPACITIVE LOAD RECOMMENDED RS vs CAPACITIVE LOAD 100 1 15 Frequency (MHz) +PSRR 50 40 0 −20 20 log (AOL) −40 −60 50 −80 40 ∠AOL 30 −100 20 −120 20 10 −140 10 0 −160 0 −10 100 30 −PSRR 1k 10k 100k 1M Frequency (Hz) 10M 100M 1k 10k 100k 1M Frequency (Hz) 10M 100M 1G Open−Loop Phase (_) 10 −180 7 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V (continued) RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. OUTPUT VOLTAGE AND CURRENT LIMITATIONS Output Current Power Limit Limit 3 RL = 100Ω 2 VO (V) 10 1 RL = 25Ω 0 RL = 50Ω −1 −2 −3 Output Current −4 Single Channel −100 0.1 1W Internal Limit −5 −150 1 Power Limit 0.01 −50 0 50 100 150 1k 10k 100k IO (mA) 4 5 6 3 4 4 2 1 Output Left Scale 0 0 −2 −1 −2 RL = 100Ω G = +2V/V See Figure 1 −3 −8 Input/Output Voltage (1V/div) 2 Input Voltage (1V/div) Output Voltage (2V/div) Input Right Scale −6 3 Input 1 0 −1 −2 −3 Output RL = 100Ω G = −1V/V See Figure 2 −5 Time (40ns/div) Time (40ns/div) TYPICAL DC DRIFT OVER TEMPERATURE COMPOSITE VIDEO dG/dP 0.14 0.28 0.12 0.24 0.10 0.20 0.08 0.16 dP Negative Video 0.12 0.04 0.08 0.02 Input Offset Voltage (VOS) Left Scale 0.5 0 10x Input Offset Current (IOS) Right Scale −0.5 1 2 3 Video Loads 0 4 10 0 −10 Input Bias Current (IB) Right Scale 0.04 dG Positive Video 0 8 Input Offset Voltage (mV) 0.32 dP Positive Video 20 0.36 Differential Phase (_) Differential Gain (%) dG Negative Video 0.16 0.06 1.0 0.40 G = +2V/V 0.18 100M 2 −4 −4 0.20 10M INVERTING OVERDRIVE RECOVERY NONINVERTING OVERDRIVE RECOVERY 8 −4 1M Frequency (Hz) −1.0 −50 −25 0 25 50 75 Ambient Temperature (_C) 100 −20 125 Input Bias and Offset Current (µA) 4 CLOSED−LOOP OUTPUT IMPEDANCE vs FREQUENCY 1W Internal Output Impedance (Ω ) 5 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V (continued) RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. COMMON−MODE INPUT RANGE AND OUTPUT SWING vs SUPPLY VOLTAGE SUPPLY AND OUTPUT CURRENT vs TEMPERATURE 26 24 Total Supply Current Right Scale Sourcing Output Current Left Scale 90 22 80 20 Sinking Output Current Left Scale 70 −50 −25 0 25 +VIN 4 −VIN 3 +VOUT 2 −VOUT 1 50 75 100 18 125 0 2.5 Ambient Temperature (_ C) 3.5 4.0 4.5 5.0 5.5 6.0 CROSSTALK vs FREQUENCY −10 Common−Mode Input Impedance −20 1M Crosstalk (dB) Input Impedance (Ω ) 3.0 Supply Voltage (±VS) COMMON−MODE AND DIFFERENTIAL INPUT IMPEDANCE 10M RL ≥ 500Ω 5 Voltage Range (V) 100 6 Supply Current (2mA/div) Output Current (10mA/div) 110 100k Differential Input Impedance All Hostile Crosstalk 1VPP Output, 3 Channels, 100Ω Load −30 −40 −50 −60 10k Channel−to−Channel Crosstalk 1VPP Output, 1 Channel, 100ΩLoad −70 −80 1k 100 1k 10k 100k Frequency (Hz) 1M 10M 100M 1 10 100 Frequency (MHz) 9 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V RF = 402Ω, RL = 100Ω to VS/2, and G = +2, unless otherwise noted. NONINVERTING SMALL−SIGNAL FREQUENCY RESPONSE 3 INVERTING SMALL−SIGNAL FREQUENCY RESPONSE 3 G = +1, RF = 25Ω G = −1 0 −3 G = +5 −6 Normalized Gain (dB) G = +2 G = +10 −9 −12 VO = 0.1VPP RL = 100Ω See Figure 3 −15 −18 1 −3 G = −10 −9 −12 VO = 0.1VPP RL = 100Ω See Figure 4 −15 −18 10 100 600 1 10 Frequency (MHz) NONINVERTING LARGE−SIGNAL FREQUENCY RESPONSE VO = 1VPP VO = 1VPP −3 VO = 2VPP Gain (dB) Gain (dB) VO = 0.5VPP 0 3 0 VO = 3VPP −6 VO = 2VPP −6 VO = 3VPP −9 −12 G = +2V/V RL = 100Ω See Figure 3 −9 −12 1 G = −1V/V RL = 100Ω See Figure 4 −15 −18 10 100 500 1 10 Frequency (MHz) 0 1.0 Small−Signal ± 100mV Left Scale 0.5 0 −0.1 −0.5 −0.2 −1.0 −0.3 −1.5 −0.4 −2.0 Time (10ns/div) 10 1.5 Small−Signal Output Voltage (100mV/div) 0.2 Large−Signal Output Voltage (500mV/div) Small−Signal Output Voltage (100mV/div) Large−Signal ± 1V Right Scale 500 INVERTING PULSE RESPONSE 2.0 G = +2V/V See Figure 3 100 Frequency (MHz) NONINVERTING PULSE RESPONSE 0.4 0.1 500 INVERTING LARGE−SIGNAL FREQUENCY RESPONSE 3 VO = 0.5VPP 6 0.3 100 Frequency (MHz) 9 −3 G = −2 G = −5 −6 0.4 0.3 2.0 G = −1V/V See Figure 4 1.5 0.2 1.0 0.1 0.5 0 −0.1 −0.2 −0.3 Small−Signal ± 100mV Left Scale Large−Signal ± 1V Right Scale −0.4 0 −0.5 −1.0 −1.5 −2.0 Time (10ns/div) Large−Signal Output Voltage (500mV/div) Normalized Gain (dB) 0 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V (continued) RF = 402Ω, RL = 100Ω to VS/2, and G = +2, unless otherwise noted. HARMONIC DISTORTION vs FREQUENCY HARMONIC DISTORTION vs LOAD RESISTANCE −60 −60 G = +2V/V f = 1MHz VO = 2VPP −70 −75 2nd−Harmonic −80 −85 −90 3rd−Harmonic −95 G = +2V/V RL = 200Ω VO = 2VPP −65 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −65 −70 2nd−Harmonic −75 −80 −85 −90 3rd−Harmonic −95 −100 −100 −105 −105 100 0.1 1000 1 HARMONIC DISTORTION vs OUTPUT VOLTAGE −60 −70 −75 −80 2nd−Harmonic −85 −90 −95 3rd−Harmonic −100 1 −70 −75 2nd−Harmonic −80 −85 −90 3rd−Harmonic −95 −100 −105 0.1 f = 1MHz RL = 200Ω VO = 2VPP See Figure 3 −65 Harmonic Distortion (dBc) Harmonic Distortion (dBc) HARMONIC DISTORTION vs NONINVERTING GAIN −60 G = +2V/V f = 1MHz RL = 200Ω See Figure 3 −65 10 −105 1 10 Output Voltage Swing (VPP) Gain (V/V) TWO−TONE, 3RD−ORDER INTERMODULATION INTERCEPT HARMONIC DISTORTION vs INVERTING GAIN −65 Harmonic Distortion (dBc) 40 f = 1MHz RL = 200Ω VO = 2VPP See Figure 4 −70 +5V 2nd−Harmonic 0.01µF Intercept Point (+dBm) −60 10 Frequency (MHz) Resistance (Ω) −75 −80 −85 3rd−Harmonic −90 −95 35 806Ω PI 57.6Ω 1/4 O PA4820 806Ω PO 200Ω 402Ω 0.1µF 30 402Ω 0.1µF 25 20 −100 −105 1 10 Gain ( V/V ) 15 0 5 10 15 20 25 30 Frequency (MHz) 11 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V (continued) RF = 402Ω, RL = 100Ω to VS/2, and G = +2, unless otherwise noted. RECOMMENDED RS vs CAPACITIVE LOAD FREQUENCY RESPONSE vs CAPACITIVE LOAD Normalized Gain to Capacitive Load (dB) 100 10 1 1 10 100 1000 C L = 10pF 7 6 5 CL = 22pF 4 CL = 47pF 3 CL = 100pF 2 +5V 0.01µF 1 806Ω RS VI 57.6Ω 0 806Ω CL −1 402Ω −2 −3 1 10 Input Offset Voltage (VOS) Left Scale 0 −0.5 −5 −10 Input Bias Current (IB) Right Scale −25 0 25 50 75 100 −15 125 Output Current (5mA/div) 5 Imput Bias and Offset Current (µA) Input Offset Voltage (mV) 10 10x Input Offset Current (IOS) Right Scale Ambient Temperature (_C) 12 90 15 1.0 −1.5 −50 100 300 SUPPLY AND OUTPUT CURRENT vs TEMPERATURE TYPICAL DC DRIFT OVER TEMPERATURE −1.0 NOTE: (1) 1kΩ is optional. 0.01µF Frequency (MHz) 1.5 0 1kΩ(1) 402Ω Capacitive Load (pF) 0.5 VO OPA820 23 85 Total Supply Current Right Scale 80 75 70 21 20 Sinking Output Current Left Scale 65 60 −50 22 19 Sourcing Output Current Left Scale −25 0 25 50 75 Ambient Temperature (_ C) 100 18 17 125 Supply Current (1mA/div) RS (Ω) 0.1dB Peaking Targeted 8 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 APPLICATIONS INFORMATION +5V WIDEBAND VOLTAGE-FEEDBACK OPERATION Figure 1 shows the gain of +2 configuration used as the basis for most of the Typical Characteristics. Most of the curves were characterized using signal sources with 50Ω driving impedance and with measurement equipment presenting 50Ω load impedance. In Figure 1, the 50Ω shunt resistor at the VI terminal matches the source impedance of the test generator while the 50Ω series resistor at the VO terminal provides a matching resistor for the measurement equipment load. Generally, data sheet specifications refer to the voltage swings at the output pin (VO in Figure 1). The 100Ω load, combined with the 804Ω total feedback network load, presents the OPA4820 with an effective load of approximately 90Ω in Figure 1. +5V +VS + 0.1µF 2.2µF 50Ω Source VIN 50Ω RS 50Ω Load 50Ω VO 1/4 OPA4820 RF 402Ω 0.1µF + RG 402Ω 2.2µF −VS −5V Figure 1. Gain of +2, High-Frequency Application and Characterization Circuit WIDEBAND INVERTING OPERATION Operating the OPA4820 as an inverting amplifier has several benefits and is particularly useful when a matched 50Ω source and input impedance is required. Figure 2 shows the inverting gain of −1 circuit used as the basis of the inverting mode Typical Characteristics. 0.1µF RT 205Ω 0.01µF 50Ω Source 1/4 OPA4820 RG 402Ω 2.2µF 50Ω VO 50Ω Load RF 402Ω VI RM 57.6Ω 0.1µF + Proper printed circuit board (PCB) layout and careful component selection will maximize the performance of the OPA4820 in all applications, as discussed in the following sections of this data sheet. + The combination of speed and dynamic range offered by the OPA4820 is easily achieved in a wide variety of application circuits, providing that simple principles of good design are observed. For example, good power-supply decoupling, as shown in Figure 1, is essential to achieve the lowest possible harmonic distortion and smooth frequency response. 2.2µF −5V Figure 2. Inverting G = −1 Specifications and Test Circuit In the inverting case, just the feedback resistor appears as part of the total output load in parallel with the actual load. For the 100Ω load used in the Typical Characteristics, this gives a total load of 80Ω in this inverting configuration. The gain resistor is set to get the desired gain (in this case 402Ω for a gain of −1) while an additional input matching resistor (RM) can be used to set the total input impedance equal to the source if desired. In this case, RM = 57.6Ω in parallel with the 402Ω gain setting resistor gives a matched input impedance of 50Ω. This matching is only needed when the input needs to be matched to a source impedance, as in the characterization testing done using the circuit of Figure 2. The OPA4820 offers extremely good DC accuracy as well as low noise and distortion. To take full advantage of that DC precision, the total DC impedance looking out of each of the input nodes must be matched to get bias current cancellation. For the circuit of Figure 2, this requires the 205Ω resistor shown to ground on the noninverting input. The calculation for this resistor includes a DC-coupled 50Ω source impedance along with RG and RM. Although this resistor will provide cancellation for the bias current, it must be well decoupled (0.01µF in Figure 2) to filter the noise contribution of the resistor and the input current noise. As the required RG resistor approaches 50Ω at higher gains, the bandwidth for the circuit in Figure 2 will far exceed the bandwidth at that same gain magnitude for the noninverting circuit of Figure 1. This occurs due to the lower noise gain for the circuit of Figure 2 when the 50Ω source impedance is included in the analysis. For instance, at a signal gain of −10 (RG = 50Ω, RM = open, RF = 499Ω) the noise gain for the circuit of Figure 2 will be 1 + 499Ω/(50Ω + 50Ω) = 6 as a result of adding the 50Ω source in the noise gain equation. This gives considerable higher bandwidth than the noninverting gain of +10. Using the 240MHz gain bandwidth product for the OPA4820, an inverting gain of −10 from a 50Ω source to a 50Ω RG gives 42MHz bandwidth, whereas the noninverting gain of +10 gives 27MHz. 13 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 WIDEBAND SINGLE-SUPPLY OPERATION while delivering more than 60mA output current giving 2.4V output swing into 100Ω (5.6dBm maximum at a matched 50Ω load). Figure 3 shows the AC-coupled, single +5V supply, gain of +2V/V circuit configuration used as a basis for the +5V only Electrical and Typical Characteristics. The key requirement for single-supply operation is to maintain input and output signal swings within the useable voltage ranges at both the input and the output. The circuit of Figure 3 establishes an input midpoint bias using a simple resistive divider from the +5V supply (two 806Ω resistors) to the noninverting input. The input signal is then AC-coupled into this midpoint voltage bias. The input voltage can swing to within 0.9V of the negative supply and 0.6V of the positive supply, giving a 3.5VPP input signal range. The input impedance matching resistor (57.6Ω) used in Figure 3 is adjusted to give a 50Ω input match when the parallel combination of the biasing divider network is included. The gain resistor (RG) is AC-coupled, giving the circuit a DC gain of +1. This puts the input DC bias voltage (2.5V) on the output as well. On a single +5V supply, the output voltage can swing to within 1.3V of either supply pin Figure 4 shows the AC-coupled, single +5V supply, gain of −1V/V circuit configuration used as a basis for the +5V only Typical Characteristic curves. In this case, the midpoint DC bias on the noninverting input is also decoupled with an additional 0.01µF decoupling capacitor. This reduces the source impedance at higher frequencies for the noninverting input bias current noise. This 2.5V bias on the noninverting input pin appears on the inverting input pin and, since RG is DC blocked by the input capacitor, will also appear at the output pin. The single-supply test circuits of Figure 3 and Figure 4 show +5V operation. These same circuits can be used over a single-supply range of +4V to +12.6V. Operating on a single +12V supply, with the Absolute Maximum Supply voltage specification of +13V, gives adequate design margin for the typical ±5% supply tolerance. +5V +VS + 0.1µF 50Ω Source 0.01µF 6.8µF 806Ω DIS VI 57.6Ω VO 1/4 OPA4820 806Ω 100Ω VS/2 RF 402Ω RG 402Ω 0.01µF Figure 3. AC-Coupled, G = +2V/V, Single-Supply Specifications and Test Circuit +5V +VS + 0.1µF 6.8µF 806Ω DIS 0.01µF 806Ω RG 0.01µF 402Ω 1/4 OPA4820 VO 100Ω VS/2 RF 402Ω VI Figure 4. AC-Coupled, G = −1V/V, Single-Supply Specifications and Test Circuit 14 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 DIFFERENTIAL INTERFACE APPLICATIONS Dual and quad op amps are particularly suitable to differential input to differential output applications. Typically, these fall into either ADC input interface or line driver applications. Two basic approaches to differential I/O are noninverting or inverting configurations. Since the output is differential, the signal polarity is somewhat meaningless—the noninverting and inverting terminology applies here to where the input is brought into the OPA4820. Each has its advantages and disadvantages. Figure 5 shows a basic starting point for noninverting differential I/O applications. Figure 6 shows a differential I/O stage configured as an inverting amplifier. In this case, the gain resistors (RG) become the input resistance for the source. This provides a better noise performance than the noninverting configuration, but does limit the flexibility in setting the input impedance separately from the gain. +VCC VCM 1/4 OPA4820 +VCC 1/4 OPA4820 RG RF 402Ω RF 402Ω VI RG RF 402Ω VI VO VO RG RF 402Ω 1/4 OPA4820 1/4 OPA4820 −VCC Figure 5. Noninverting Differential I/O Amplifier This approach provides for a source termination impedance that is independent of the signal gain. For instance, simple differential filters may be included in the signal path right up to the noninverting inputs without interacting with the amplifier gain. The differential signal gain for the circuit of Figure 5 is: VO + AD + 1 ) 2 VI RF RG (1) Figure 5 shows the recommended value of 402Ω. However, the gain may be adjusted using just the RG resistor. Various combinations of single-supply or AC-coupled gains can also be delivered using the basic circuit of Figure 5. Common-mode bias voltages on the two noninverting inputs pass on to the output with a gain of 1 since an equal DC voltage at each inverting node creates no current through RG, giving that voltage a commonmode gain of 1 to the output. VCM −VCC Figure 6. Inverting Differential I/O Amplifier The two noninverting inputs provide an easy common-mode control input. This is particularly useful if the source is AC-coupled through either blocking caps or a transformer. In either case, the common-mode input voltages on the two noninverting inputs again have a gain of 1 to the output pins, giving an easy common-mode control for single-supply operation. The input resistors may be adjusted to the desired gain but will also be changing the input impedance as well. The differential gain for this circuit is: VO R +* F VI RG (2) 15 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 DC-COUPLED SINGLE-TO-DIFFERENTIAL CONVERSION stage, creating no current flow and placing the desired VCM at the output of this stage as well. Both the positive and negative output are shown in Figure 8. The previous differential output circuits were set up to receive a differential input as well as provide a differential output. Figure 7 shows one way to provide a single to differential conversion, with DC coupling, and independent output common-mode control using a quad op amp. The circuit of Figure 7 provides several useful features for isolating the input signal from the final outputs. Using the first amplifier as a simple noninverting stage gives an independent adjustment on RI (to set the source loading) while the gain can be easily adjusting in this stage using the RG resistor. The next stage allows a separate output common-mode level to be set up. The desired output common-mode voltage, VCM, is cut in half and applied to the noninverting input of the 2nd stage. The signal path in this stage sees a gain of −1 while this (1/2 × VCM) voltage sees a gain of +2. The output of this 2nd stage is then the original common-mode voltage plus the inverted signal from the output of the first stage. The 2nd stage output appears directly at the output of the noninverting final stage. The inverting node of the inverting output stage is also biased to the common-mode voltage, equal to the common-mode voltage appearing at the output of the 2nd LOW-POWER, DIFFERENTIAL I/O, 4th-ORDER ACTIVE FILTER The OPA4820 can give a very capable gain block for active filters. The quad design lends itself very well to differential active filters. Where the filter topology is looking for a simple gain function to implement the filter, the noninverting configuration is preferred to isolate the filter elements from the gain elements in the design. Figure 9 shows an example of a 10MHz, 4th-order Butterworth, low-pass Sallen-Key filter. The design places the higher Q stage first to allow the lower Q 2nd stage to roll off the peaked noise of the first stage. The resistor values have been adjusted slightly to account for the amplifier group delay. While this circuit is bipolar, using ±5V supplies, it can easily be adapted to single-supply operation. This will add two real zeroes in the response, transforming this circuit into a bandpass. The frequency response for the filter of Figure 9 is shown in Figure 10. +5V 1.5V VCM 200Ω 1/4 OPA4820 750Ω VCM 2 VI RI 1/4 OPA4820 402Ω RG 0.1µF 750Ω 750Ω 1/4 OPA4820 750Ω 402Ω 402Ω +VOUT = VCM + VI (1 + 50Ω 200Ω 1/4 OPA4820 −VOUT = VCM − VI (1 + 250Ω −5V Figure 7. Precision, DC-Coupled, Single-to-Differential Conversion 16 402Ω ) RG 402Ω ) RG 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 3.0 VIN = 0V to 0.5V RG = 402Ω +VOUT 2.5 Voltage (V) 2.0 1.5 1.0 −VOUT 0.5 0 Time (10ns/div) Figure 8. Pulse Response for Figure 7 Schematic 100pF 100pF +5V 161Ω 77Ω 121Ω 294Ω VO/VI = 4V/V 1/4 OPA4820 1/4 OPA4820 250Ω VI 50pF 500Ω 250Ω 161Ω VO 250Ω 1/4 OPA4820 121Ω PD = 225mW 250Ω 500Ω 50pF f−3dB = 10MHz 77Ω 100pF 294Ω 1/4 OPA4820 100pF GD = 2, ωO = 2π 10MHz, Q = 1.31 −5V GD = 2, ωO = 2π 10MHz, Q = 0.54 Figure 9. Low-Power, Differential I/O, 4th-Order Butterworth Active Filter LOW-POWER xDSL TRANSCEIVER INTERFACE 15 Differential Gain (dB) 12 9 6 3 0 −3 −6 −9 1 10 Frequency (MHz) Figure 10. Differential 4th-Order, 10MHz Butterworth Filter 100 With four amplifiers available, the quad OPA4820 can meet the needs for both differential driver and receiver in a low-power xDSL line interface design. A simplified design example is shown on the front page. Two amplifiers are used as a noninverting differential driver while the other two implement the driver echo cancellation and receiver amplifier function. This example shows a single +12V design where the drive side is taking a 2VPP maximum input from the transmit filter and providing a differential gain of 7, giving a maximum 14VPP differential output swing. This is coupled through 50Ω matching resistors and a 1:1 transformer to give a maximum 7VPP on a 100Ω line. This 7VPP corresponds to a 10dBm line power with a 3.5 crest factor. 17 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 The differential receiver is configured as an inverting summing stage where the outputs of the driver are cancelled prior to appearing at the output of the receive amplifiers. This is done by summing the output voltages for the drive amplifiers and their attenuated and inverted levels (at the transformer input) into the inverting inputs of each receiver amplifier. The resistor values are set (see the circuit on the front page) to give perfect drive signal cancellation if the drive signal is attenuated by 1/2, going from the drive amplifier outputs to the transformer input. The signal received through the transformer has a gain of 1 through the receive amplifiers. Higher gain could easily be provided by scaling the resistors summing into the inverting inputs of the receiver amplifiers down while keeping the same ratio between them. DUAL-CHANNEL, DIFFERENTIAL ADC DRIVER Where a low-noise, single-supply, interface to a differential input +5V ADC is required, the circuit of Figure 11 can provide a high dynamic range, medium gain interface for dual high-performance ADCs. The circuit of Figure 11 uses two amplifiers in the differential inverting configuration. The common-mode voltage is set on the noninverting inputs to the supply midscale. In this example, the input signal is coupled in through a 1:2 transformer. This provides both signal gain, single to differential conversion, and a reduction in noise figure. To show a 50Ω input impedance at the input to the transformer, two 200Ω resistors are required on the transformer secondary. These two resistors are also the amplifier gain elements. Since the same DC voltage appears on both inverting nodes in the circuit of Figure 11, no DC current will flow through the transformer, giving a DC gain of 1 to the output for this common-mode voltage, VCM. The circuit of Figure 11 is particularly suitable for a moderate resolution dual ADC used as I/Q samplers. The optional 500Ω resistors to ground on each amplifier output can be added to improve the 2nd- and 3rd-harmonic distortion by > 15dB if higher dynamic range is required. 18 The 5mA added output stage current significantly improves linearity if that is required. The measured 2nd-harmonic distortion is consistently lower than the 3rd-harmonics for this balanced differential design. It is particularly helpful for this low-power design if there are no grounds in the signal path after the low-level signal at the transformer input. The two pull-down resistors do show a signal path ground and should be connected at the same physical point to ground, in order to eliminate imbalanced ground return currents from degrading 2nd-harmonic distortion. VIDEO LINE DRIVING Most video distribution systems are designed with 75Ω series resistors to drive a matched 75Ω cable. In order to deliver a net gain of 1 to the 75Ω matched load, the amplifier is typically set up for a voltage gain of +2, compensating for the 6dB attenuation of the voltage divider formed by the series and shunt 75Ω resistors at either end of the cable. The circuit of Figure 1 applies to this requirement if all references to 50Ω resistors are replaced by 75Ω values. Often, the amplifier gain is further increased to 2.2, which recovers the additional DC loss of a typical long cable run. This change would require the gain resistor (RG) in Figure 1 to be reduced from 402Ω to 335Ω. In either case, both the gain flatness and the differential gain/phase performance of the OPA4820 will provide exceptional results in video distribution applications. Differential gain and phase measure the change in overall small-signal gain and phase for the color sub-carrier frequency (3.58MHz in NTSC systems) versus changes in the large-signal output level (which represents luminance information in a composite video signal). The OPA4820, with the typical 150Ω load of a single matched video cable, shows less than 0.003%/0.06° differential gain/phase errors over the standard luminance range for a positive video (negative sync) signal. Similar performance would be observed for multiple video signals (see Figure 12). 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 +5V 1kΩ VCM 0.1µF 1kΩ 1/4 OPA4820 Dual ADC 500Ω 1:2 50Ω Source 200Ω 800Ω RS 200Ω 800Ω RS 16.7dB Noise Figure CL 1 of 2 Channels 1/4 OPA4820 Gain = 8V/V 18dB 500Ω VCM Figure 11. Single-Supply Differential ADC Driver (1 of 2 channels) 335Ω 402Ω 75Ω Transmission Line 75Ω 1/4 OPA4820 VOUT Video Input 75Ω 75Ω 75Ω VOUT 75Ω 75Ω High output current drive capability allows three back−terminated 75Ω transmission lines to be simultaneously driven. VOUT 75Ω Figure 12. Video Distribution Amplifier SINGLE OP AMP DIFFERENTIAL AMPLIFIER The voltage-feedback architecture of the OPA4820, with its high common-mode rejection ratio (CMRR), will provide exceptional performance in differential amplifier configurations. Figure 13 shows a typical configuration. The starting point for this design is the selection of the RF value in the range of 200Ω to 2kΩ. Lower values reduce the required RG, increasing the load on the V2 source and on the OPA4820 output. Higher values increase output noise as well as the effects of parasitic board and device capacitances. Following the selection of RF, RG must be set to achieve the desired inverting gain for V2. Remember that the bandwidth will be set approximately by the gain bandwidth product (GBP) divided by the noise gain (1 + RF/RG). For accurate differential operation (that is, good CMRR), the ratio R2/R1 must be set equal to RF/RG. +5V Power−supply decoupling not shown. R1 V1 R2 1/4 OPA4820 RG RF V2 50Ω VO = when RF (V − V2) RG 1 R2 R F = R1 R G −5V Figure 13. High-Speed, Single Differential Amplifier 19 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 Usually, it is best to set the absolute values of R2 and R1 equal to RF and RG, respectively; this equalizes the divider resistances and cancels the effect of input bias currents. However, it is sometimes useful to scale the values of R2 and R1 in order to adjust the loading on the driving source, V1. In most cases, the achievable low-frequency CMRR will be limited by the accuracy of the resistor values. The 85dB CMRR of the OPA4820 itself will not determine the overall circuit CMRR unless the resistor ratios are matched to better than 0.003%. If it is necessary to trim the CMRR, then R2 is the suggested adjustment point. 4-CHANNEL DAC TRANSIMPEDANCE AMPLIFIER High-frequency Digital-to-Analog Converters (DACs) require a low-distortion output amplifier to retain their SFDR performance into real-world loads. See Figure 14 for a single-ended output drive implementation. In this circuit, only one side of the complementary output drive signal is used. The diagram shows the signal output current connected into the virtual ground-summing junction of the OPA4820, which is set up as a transimpedance stage or I-V converter. The unused current output of the DAC is connected to ground. If the DAC requires its outputs to be terminated to a compliance voltage other than ground for operation, then the appropriate voltage level may be applied to the noninverting input of the OPA4820. 1/4 OPA4820 High−Speed DAC VO = ID RF RF peaking. To achieve a flat transimpedance frequency response, this pole in the feedback network should be set to: 1 + 2pR FCF GBP Ǹ4pR C F (3) D which will give a corner frequency f−3dB of approximately: f *3dB + GBP Ǹ2pR C F (4) D ACTIVE FILTERS Most active filter topologies will have exceptional performance using the broad bandwidth and unity-gain stability of the OPA4820. Topologies employing capacitive feedback require a unity-gain stable, voltage-feedback op amp. Sallen-Key filters simply use the op amp as a noninverting gain stage inside an RC network. Either current- or voltage-feedback op amps may be used in Sallen-Key implementations. Figure 15 shows an example Sallen-Key low-pass filter, in which the OPA4820 is set up to deliver a low-frequency gain of +2. The filter component values have been selected to achieve a maximally-flat Butterworth response with a 5MHz, −3dB bandwidth. The resistor values have been slightly adjusted to compensate for the effects of the 240MHz bandwidth provided by the OPA4820 in this configuration. This filter may be combined with the ADC driver suggestions to provide moderate (2-pole) Nyquist filtering, limiting noise, and out-of-band harmonics into the input of an ADC. This filter will deliver the exceptionally low harmonic distortion required by high SFDR ADCs such as the ADS850 (14-bit, 10MSPS, 82dB SFDR). CF ID CD C1 150pF GBP →Gain Bandwidth Product (Hz) for the OPA4820. +5V R1 124Ω ID R2 505Ω V1 C2 100pF Figure 14. Wideband, Low-Distortion DAC Transimpedance Amplifier The DC gain for this circuit is equal to RF. At high frequencies, the DAC output capacitance (CD) will produce a zero in the noise gain for the OPA4820 that may cause peaking in the closed-loop frequency response. CF is added across RF to compensate for this noise-gain 20 1/4 OPA4820 VO RF 402Ω Power−supply decoupling not shown. −5V RG 402Ω Figure 15. 5MHz Butterworth Low-Pass Active Filter 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 Another type of filter, a high-Q bandpass filter, is shown in Figure 16. The transfer function for this filter is: R )R s R 3R C4 V OUT 1 4 1 + 1 V IN 2 s ) sR C ) R 1 1 with and R3 (5) 2R 4R 5C 1C 2 R3 w O2 + R 2R 4R 5C 1C 2 wO + 1 Q R 1C 1 (6) and DEMONSTRATION FIXTURES Two printed circuit boards (PCBs) are available to assist in the initial evaluation of circuit performance using the OPA4820 in its two package options. Both of these are offered free of charge as unpopulated PCBs, delivered with a user’s guide. The summary information for these fixtures is shown in the table below. (7) PRODUCT For the values chosen in Figure 16: wO ] 1MHz 2p fO + DESIGN-IN TOOLS (8) PACKAGE ORDERING NUMBER LITERATURE NUMBER OPA4820ID SO-14 DEM-OPA-SO-4A SBOU016 OPA4820IPW TSSOP-14 DEM-OPA-TSSOP-4A SBOU017 Q = 100 See Figure 17 for the frequency response of the filter shown in Figure 16. MACROMODELS AND APPLICATIONS SUPPORT R3 500Ω 1/4 OPA4820 C2 1000pF R1 15.8kΩ R2 158Ω R4 500Ω R5 158Ω 1/4 OPA4820 The demonstration fixtures can be requested at the Texas Instruments web site (www.ti.com) through the OPA4820 product folder. VOUT VIN C1 1000pF Figure 16. High-Q 1MHz Bandpass Filter Computer simulation of circuit performance using SPICE is often a quick way to analyze the performance of the OPA4820 and its circuit designs. This is particularly true for video and RF amplifier circuits where parasitic capacitance and inductance can play a major role on circuit performance. A SPICE model for the OPA4820 is available through the TI web page (www.ti.com). The applications department is also available for design assistance. These models predict typical small-signal AC, transient steps, DC performance, and noise under a wide variety of operating conditions. The models include the noise terms found in the electrical specifications of the data sheet. These models do not attempt to distinguish between the package types in their small-signal AC performance. Gain (dB) OPERATING SUGGESTIONS OPTIMIZING RESISTOR VALUES 6 0 −6 −12 −18 −24 −30 −36 −42 −48 −54 −60 −66 −72 100k 1M 10M Frequency (Hz) Figure 17. High-Q 1MHz Bandpass Filter Frequency Response 100M Since the OPA4820 is a unity-gain stable, voltage-feedback op amp, a wide range of resistor values may be used for the feedback and gain-setting resistors. The primary limits on these values are set by dynamic range (noise and distortion) and parasitic capacitance considerations. Usually, the feedback resistor value should be between 200Ω and 1kΩ. Below 200Ω, the feedback network will present additional output loading which can degrade the harmonic distortion performance of the OPA4820. Above 1kΩ, the typical parasitic capacitance (approximately 0.2pF) across the feedback resistor may cause unintentional band limiting in the amplifier response. A 25Ω feedback resistor is suggested for AV = +1V/V. A good rule of thumb is to target the parallel combination of RF and RG (see Figure 1) to be less than about 200Ω. The combined impedance RF || RG interacts with the inverting input capacitance, placing an additional pole in the feedback network, and thus a zero in the forward 21 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 response. Assuming a 2pF total parasitic on the inverting node, holding RF || RG < 200Ω will keep this pole above 400MHz. By itself, this constraint implies that the feedback resistor RF can increase to several kΩ at high gains. This is acceptable as long as the pole formed by RF and any parasitic capacitance appearing in parallel is kept out of the frequency range of interest. In the inverting configuration, an additional design consideration must be noted. RG becomes the input resistor and therefore the load impedance to the driving source. If impedance matching is desired, RG may be set equal to the required termination value. However, at low inverting gains, the resulting feedback resistor value can present a significant load to the amplifier output. For example, an inverting gain of 2 with a 50Ω input matching resistor (= RG) would require a 100Ω feedback resistor, which would contribute to output loading in parallel with the external load. In such a case, it would be preferable to increase both the RF and RG values, and then achieve the input matching impedance with a third resistor to ground (see Figure 2). The total input impedance becomes the parallel combination of RG and the additional shunt resistor. BANDWIDTH vs GAIN Voltage-feedback op amps exhibit decreasing closed-loop bandwidth as the signal gain is increased. In theory, this relationship is described by the GBP shown in the specifications. Ideally, dividing GBP by the noninverting signal gain (also called the noise gain, or NG) will predict the closed-loop bandwidth. In practice, this only holds true when the phase margin approaches 90°, as it does in high-gain configurations. At low signal gains, most amplifiers will exhibit a more complex response with lower phase margin. The OPA4820 is optimized to give a maximally-flat, 2nd-order Butterworth response in a gain of 2. In this configuration, the OPA4820 has approximately 64° of phase margin and will show a typical −3dB bandwidth of 240MHz. When the phase margin is 64°, the closed-loop bandwidth is approximately √2 greater than the value predicted by dividing GBP by the noise gain. Increasing the gain will cause the phase margin to approach 90° and the bandwidth to more closely approach the predicted value of (GBP/NG). At a gain of +10, the 27MHz bandwidth shown in the Electrical Characteristics agrees with that predicted using the simple formula and the typical GBP of 250MHz. OUTPUT DRIVE CAPABILITY The OPA4820 has been optimized to drive the demanding load of a doubly-terminated transmission line. When a 50Ω line is driven, a series 50Ω into the cable and a terminating 50Ω load at the end of the cable are used. Under these conditions, the cable impedance will appear resistive over a wide frequency range, and the total effective load on the OPA4820 is 100Ω in parallel with the resistance of the feedback network. The electrical characteristics show a 22 ±3.6V swing into this load—which will then be reduced to a ±1.8V swing at the termination resistor. The ±75mA output drive over temperature provides adequate current drive margin for this load. Higher voltage swings (and lower distortion) are achievable when driving higher impedance loads. A single video load typically appears as a 150Ω load (using standard 75Ω cables) to the driving amplifier. The OPA4820 provides adequate voltage and current drive to support up to three parallel video loads (50Ω total load) for an NTSC signal. With only one load, the OPA4820 achieves an exceptionally low 0.01%/0.03° dG/dP error. DRIVING CAPACITIVE LOADS One of the most demanding, and yet very common, load conditions for an op amp is capacitive loading. A high-speed, high open-loop gain amplifier like the OPA4820 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. In simple terms, the capacitive load reacts with the open-loop output resistance of the amplifier to introduce an additional pole into the loop and thereby decrease the phase margin. This issue has become a popular topic of application notes and articles, and several external solutions to this problem have been suggested. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. This does not eliminate the pole from the loop response, but rather shifts it and adds a zero at a higher frequency. The additional zero acts to cancel the phase lag from the capacitive load pole, thus increasing the phase margin and improving stability. The Typical Characteristics show the recommended RS vs Capacitive Load and the resulting frequency response at the load. The criterion for setting the recommended resistor is maximum bandwidth, flat frequency response at the load. Since there is now a passive low-pass filter between the output pin and the load capacitance, the response at the output pin itself is typically somewhat peaked, and becomes flat after the roll-off action of the RC network. This is not a concern in most applications, but can cause clipping if the desired signal swing at the load is very close to the amplifier’s swing limit. Such clipping would be most likely to occur in pulse response applications where the frequency peaking is manifested as an overshoot in the step response. Parasitic capacitive loads greater than 2pF can begin to degrade the performance of the OPA4820. Long PCB traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always consider this effect carefully, and add the recommended series resistor as close as possible to the OPA4820 output pin (see the Board Layout section). 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 DISTORTION PERFORMANCE The OPA4820 is capable of delivering an exceptionally low distortion signal at high frequencies and low gains. The distortion plots in the Typical Characteristics show the typical distortion under a wide variety of conditions. Most of these plots are limited to 100dB dynamic range. The OPA4820 distortion does not rise above −90dBc until either the signal level exceeds 0.9V and/or the fundamental frequency exceeds 500kHz. Distortion in the audio band is ≤ −100dBc. Generally, until the fundamental signal reaches very high frequencies or powers, the 2nd-harmonic will dominate the distortion with a negligible 3rd-harmonic component. Focusing then on the 2nd-harmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network—in the noninverting configuration this is the sum of RF + RG, whereas in the inverting configuration this is just RF (see Figure 1). Increasing the output voltage swing increases harmonic distortion directly. Increasing the signal gain will also increase the 2nd-harmonic distortion. Again, a 6dB increase in gain will increase the 2nd- and 3rd-harmonic by 6dB even with a constant output power and frequency. Finally, the distortion increases as the fundamental frequency increases because of the roll-off in the loop gain with frequency. Conversely, the distortion will improve going to lower frequencies down to the dominant open-loop pole at approximately 100kHz. Starting from the −85dBc 2nd-harmonic for 2VPP into 200Ω, G = +2 distortion at 1MHz (from the Typical Characteristics), the 2nd-harmonic distortion will not show any improvement below 100kHz and will then be: −85dB − 20log (1MHz/100kHz) = −105dBc NOISE PERFORMANCE The OPA4820 complements its low harmonic distortion with low input noise terms. Both the input-referred voltage noise and the two input-referred current noise terms combine to give a low output noise under a wide variety of operating conditions. Figure 18 shows the op amp noise analysis model with all the noise terms included. In this model, all the noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. The total output spot noise voltage is computed as the square root of the squared contributing terms to the output noise voltage. This computation is adding all the contributing noise powers at the output by superposition, then taking the square root to get back to a spot noise voltage. Equation 9 shows the general form for this output noise voltage using the terms presented in Figure 18. EO + Ǹƪ ƫ E2NI ) ǒI BNR SǓ ) 4kTR S NG 2 ) ǒI BIR FǓ ) 4kTR FNG 2 2 (9) ENI 1/4 OPA4820 RS EO IBN ERS RF √4kTRS RG 4kT RG √4kTRF I BI 4kT = 1.6E − 20J at 290_ K Figure 18. Op Amp Noise Analysis Model Dividing this expression by the noise gain (NG = 1 + RF/RG) will give the equivalent input referred spot noise voltage at the noninverting input, as shown in Equation 10. EN + Ǹ E2NI ) ǒIBNRSǓ ) 4kTR S ) 2 ǒ Ǔ IBIRF NG 2 ) 4kTR F NG (10) Evaluating these two equations for the OPA4820 circuit presented in Figure 1 will give a total output spot noise voltage of 6.44nV/√Hz and an equivalent input spot noise voltage of 3.22nV/√Hz. DC OFFSET CONTROL The OPA4820 can provide excellent DC signal accuracy because of its high open-loop gain, high common-mode rejection, high power-supply rejection, and low input offset voltage and bias current offset errors. To take full advantage of this low input offset voltage, careful attention to input bias current cancellation is also required. The high-speed input stage for the OPA4820 has a moderately high input bias current (9µA typ into the pins) but with a very close match between the two input currents—typically 100nA input offset current. The total output offset voltage may be considerably reduced by matching the source impedances looking out of the two inputs. For example, one way to add bias current cancellation to the circuit of Figure 1 would be to insert a 175Ω series resistor into the noninverting input from the 50Ω terminating resistor. When the 50Ω source resistor is DC-coupled, this will increase the source impedance for the noninverting input bias current to 200Ω. Since this is now equal to the impedance looking out of the inverting input (RF || RG), the circuit will cancel the gains for the bias currents to the output leaving only the offset current times the feedback resistor as a residual DC error term at the output. Using a 402Ω feedback resistor, this output error will now be less than ±0.5µA × 402Ω = ±208µV at 25°C. 23 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 THERMAL ANALYSIS The OPA4820 will not require heatsinking or airflow in most applications. Maximum desired junction temperature would set the maximum allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed +150°C. Operating junction temperature (TJ) is given by TA + PD × qJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for a grounded resistive load, be at a maximum when the output is fixed at a voltage equal to 1/2 of either supply voltage (for equal bipolar supplies). Under this worst-case condition, PDL = VS2/(4 × RL), where RL includes feedback network loading. Note that it is the power in the output stage and not in the load that determines internal power dissipation. As a worst-case example, compute the maximum TJ using all channels of an OPA4820IPW (TSSOP-14 package) in the circuit of Figure 1 operating at the maximum specified ambient temperature of +85°C. PD = 10V(25.8mA) + 4 × [52/(4 × (100Ω || 800Ω))] = 539mW Maximum TJ = +85°C + (539mW × 110°C/W) = 144°C This maximum operating junction temperature is below the absolute maximum junction temperature. Most junction temperatures in applications will be lower since an absolute worst-case output stage power was assumed in this calculation. BOARD LAYOUT Achieving optimum performance with a high-frequency amplifier such as the OPA4820 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability: on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. 24 b) Minimize the distance (< 0.25”) from the power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and power-plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power-supply connections should always be decoupled with these capacitors. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PCB. c) Careful selection and placement of external components will preserve the high-frequency performance of the OPA4820. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal-film and carbon composition, axially leaded resistors can also provide good high-frequency performance. Again, keep their leads and PCB trace length as short as possible. Never use wire-wound type resistors in a high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. Other network components, such as noninverting input termination resistors, should also be placed close to the package. Where double-side component mounting is allowed, place the feedback resistor directly under the package on the other side of the board between the output and inverting input pins. Even with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant time constants that can degrade performance. Good axial metal-film or surface-mount resistors have approximately 0.2pF in shunt with the resistor. For resistor values > 1.5kΩ, this parasitic capacitance can add a pole and/or a zero below 500MHz that can effect circuit operation. Keep resistor values as low as possible consistent with load-driving considerations. It has been suggested here that a good starting point for design would be to set RG || RF = 200Ω. Using this setting will automatically keep the resistor noise terms low, and minimize the effect of their parasitic capacitance. 
"#$% www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 d) Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the plot of Recommended RS vs Capacitive Load. Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA4820 is nominally compensated to operate with a 2pF parasitic load. Higher parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin). If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary onboard, and in fact, a higher impedance environment will improve distortion as shown in the distortion versus load plots. With a characteristic board trace impedance defined based on board material and trace dimensions, a matching series resistor into the trace from the output of the OPA4820 is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and input impedance of the destination device; this total effective impedance should be set to match the trace impedance. If the 6dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the plot of RS vs Capacitive Load. This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. e) Socketing a high-speed part like the OPA4820 is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network, which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA4820 onto the board. INPUT AND ESD PROTECTION The OPA4820 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power supplies, as shown in Figure 19. +VCC External Pin −VCC Figure 19. Internal ESD Protection These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (for example, in systems with ±15V supply parts driving into the OPA4820), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible since high values degrade both noise performance and frequency response. Figure 20 shows an example protection circuit for I/O voltages that may exceed the supplies. +5V 50Ω Source Power−supply decoupling not shown. 174Ω V1 50Ω 50Ω D1 D2 1/4 OPA4820 RF 301Ω 50Ω RG 301Ω VO −5V D1 = D2 IN5911 (or equivalent) Figure 20. Gain of +2 with Input Protection 25 www.ti.com SBOS317D − SEPTEMBER 2004 − REVISED AUGUST 2008 Revision History DATE REV PAGE SECTION 8/08 D 2 Absolute Maximum Ratings 21 Design-In Tools 6/06 C 26 Application Information DESCRIPTION Changed Storage Temperature minimum value from −40°C to −65°C. Demonstration fixture numbers changed. Added Revision History table. NOTE: Page numbers for previous revisions may differ from page numbers in the current version. 26 PACKAGE OPTION ADDENDUM www.ti.com 13-Aug-2021 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) OPA4820ID ACTIVE SOIC D 14 50 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA4820 OPA4820IDG4 ACTIVE SOIC D 14 50 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA4820 OPA4820IDR ACTIVE SOIC D 14 2500 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA4820 OPA4820IPWR ACTIVE TSSOP PW 14 2500 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA 4820 OPA4820IPWT ACTIVE TSSOP PW 14 250 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA 4820 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of