OPA604AP

® OPA604 FPO FET-Input, Low Distortion OPERATIONAL AMPLIFIER FEATURES q q q q q q LOW DISTORTION: 0.0003% at 1kHz LOW NOISE: 10nV/√Hz HIGH SLEW RATE: 25V/µs WIDE GAIN-BANDWIDTH: 20MHz UNITY-GAIN STABLE WIDE SUPPLY RANGE: VS = ±4.5 to ±24V APPLICATIONS q q q q q q PROFESSIONAL AUDIO EQUIPMENT PCM DAC I/V CONVERTER SPECTRAL ANALYSIS EQUIPMENT ACTIVE FILTERS TRANSDUCER AMPLIFIER DATA ACQUISITION q DRIVES 600Ω LOAD q DUAL VERSION AVAILABLE (OPA2604) DESCRIPTION The OPA604 is a FET-input operational amplifier designed for enhanced AC performance. Very low distortion, low noise and wide bandwidth provide superior performance in high quality audio and other applications requiring excellent dynamic performance. New circuit techniques and special laser trimming of dynamic circuit performance yield very low harmonic distortion. The result is an op amp with exceptional sound quality. The low-noise FET input of the OPA604 provides wide dynamic range, even with high source impedance. Offset voltage is laser-trimmed to minimize the need for interstage coupling capacitors. The OPA604 is available in 8-pin plastic mini-DIP and SO-8 surface-mount packages, specified for the –25°C to +85°C temperature range. (7) V+ (+) (3) (–) (2) Distortion Rejection Circuitry(1) Output Stage(1) (6) VO (5) (1) (4) V– NOTE: (1) Patents Granted: #5053718, 5019789 International Airport Industrial Park • Mailing Address: PO Box 11400 Tel: (520) 746-1111 • Twx: 910-952-1111 • Cable: BBRCORP • • Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd. • Tucson, AZ 85706 Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 © 1992 Burr-Brown Corporation PDS-1161C Printed in U.S.A. May, 1995 SPECIFICATIONS ELECTRICAL TA = +25°C, VS = ±15V unless otherwise noted. OPA604AP, AU PARAMETER OFFSET VOLTAGE Input Offset Voltage Average Drift Power Supply Rejection INPUT BIAS CURRENT(1) Input Bias Current Input Offset Current NOISE Input Voltage Noise Noise Density: f = 10Hz f = 100Hz f = 1kHz f = 10kHz Voltage Noise, BW = 20Hz to 20kHz Input Bias Current Noise Current Noise Density, f = 0.1Hz to 20kHz INPUT VOLTAGE RANGE Common-Mode Input Range Common-Mode Rejection INPUT IMPEDANCE Differential Common-Mode OPEN-LOOP GAIN Open-Loop Voltage Gain FREQUENCY RESPONSE Gain-Bandwidth Product Slew Rate Settling Time: 0.01% 0.1% Total Harmonic Distortion + Noise (THD+N) OUTPUT Voltage Output Current Output Short Circuit Current Output Resistance, Open-Loop POWER SUPPLY Specified Operating Voltage Operating Voltage Range Current TEMPERATURE RANGE Specification Storage Thermal Resistance(2), θJA VO = ±10V, RL = 1kΩ G = 100 20Vp-p, RL = 1kΩ G = –1, 10V Step G = 1, f = 1kHz VO = 3.5Vrms, RL = 1kΩ RL = 600Ω VO = ±12V ±11 80 ±12 80 CONDITION MIN TYP ±1 ±8 100 50 ±3 MAX ±5 UNITS mV µV/°C dB pA pA VS = ±5 to ±24V VCM = 0V VCM = 0V 80 25 15 11 10 1.5 4 ±13 100 1012 || 8 1012 || 10 100 20 25 1.5 1 0.0003 nV/√Hz nV/√Hz nV/√Hz nV/√Hz µVp-p fA/√Hz V dB Ω || pF Ω || pF dB MHz V/µs µs µs % VCM = ±12V 15 ±12 ±35 ±40 25 ±15 ±5.3 V mA mA Ω V V mA °C °C °C/W ±4.5 ±24 ±6 +85 +125 –25 –40 90 NOTES: (1) Typical performance, measured fully warmed-up. (2) Soldered to circuit board—see text. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® OPA604 2 PIN CONFIGURATION Top View Offset Trim –In +In –VS 1 2 3 4 8 7 6 5 ABSOLUTE MAXIMUM RATINGS DIP, SOIC No Internal Connection +VS Output Offset Trim Power Supply Voltage ........................................................................ ±25V Input Voltage ............................................................... (V–)–1V to (V+)+1V Output Short Circuit to Ground ................................................ Continuous Operating Temperature ................................................... –40°C to +100°C Storage Temperature ...................................................... –40°C to +125°C Junction Temperature .................................................................... +150°C Lead Temperature (soldering, 10s) AP .......................................... +300°C Lead Temperature (soldering, 3s) AU ............................................ +260°C ORDERING INFORMATION MODEL OPA604AP OPA604AU PACKAGE 8-Pin Plastic DIP SO-8 Surface-Mount TEMP. RANGE –25°C to +85°C –25°C to +85°C ELECTROSTATIC DISCHARGE SENSITIVITY Any integrated circuit can be damaged by ESD. Burr-Brown recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet published specifications. PACKAGE INFORMATION MODEL OPA604AP OPA604AU PACKAGE 8-Pin Plastic DIP SO-8 Surface-Mount PACKAGE DRAWING NUMBER(1) 006 182 NOTE: (1) For detailed drawing and dimension table, please see end of data sheet, or Appendix D of Burr-Brown IC Data Book. ® 3 OPA604 TYPICAL PERFORMANCE CURVES TA = +25°C, VS = ±15V unless otherwise noted. TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 1 VO = 3.5Vrms 1kΩ TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT VOLTAGE 0.1 See “Distortion Measurements” for description of test method. VO 1kΩ 0.1 THD + N (%) Measurement BW = 80kHz See “Distortion Measurements” for description of test method. THD + N (%) 0.01 f = 1kHz Measurement BW = 80kHz 0.001 0.01 G = 100V/V G = 10V/V 0.001 G = 1V/V 0.0001 20 100 1k Frequency (Hz) 10k 20k 0.0001 0.1 1 10 100 Output Voltage (Vp-p) OPEN-LOOP GAIN/PHASE vs FREQUENCY 120 100 80 60 40 G 20 0 –20 1 10 100 1k 10k 100k 1M 10M Frequency (Hz) –180 1 1 –135 φ –90 Phase Shift (Degrees) 100 0 1k INPUT VOLTAGE AND CURRENT NOISE SPECTRAL DENSITY vs FREQUENCY 1k Voltage Noise 100 10 10 Current Noise 10 100 1k Frequency (Hz) 10k 100k 1 1M INPUT BIAS AND INPUT OFFSET CURRENT vs TEMPERATURE 100nA 10nA 10nA INPUT BIAS AND INPUT OFFSET CURRENT vs INPUT COMMON-MODE VOLTAGE 1nA Input Offset Current (pA) Input Bias Current (pA) Input Bias Current (pA) Input Bias Current 1nA 100 1nA Input Bias Current 100 100 Input Offset Current 10 100 Input Offset Current 10 –15 10 10 1 1 –75 –50 –25 0 25 50 75 100 0.1 125 –10 –5 0 5 10 1 15 Ambient Temperature (°C) Common-Mode Voltage (V) ® OPA604 4 Input Offset Current (pA) 10nA 1nA Current Noise (fA/ Hz) –45 Voltage Gain (dB) Voltage Noise (nV/ Hz) TYPICAL PERFORMANCE CURVES TA = +25°C, VS = ±15V unless otherwise noted. (CONT) INPUT BIAS CURRENT vs TIME FROM POWER TURN-ON 1nA Common-Mode Rejection (dB) 120 COMMON-MODE REJECTION vs COMMON-MODE VOLTAGE VS = ±24VDC Input Bias Current (pA) 110 100 VS = ±15VDC 100 10 VS = ±5VDC 90 1 0 1 2 3 4 5 Time After Power Turn-On (min) 80 –15 –10 –5 0 5 10 15 Common-Mode Voltage (V) POWER SUPPLY AND COMMON-MODE REJECTION vs FREQUENCY 120 Power Supply Rejection (dB) AOL, PSR, AND CMR vs SUPPLY VOLTAGE 120 100 80 60 40 20 0 10M Common-Mode Rejection (dB) +PSR 120 100 80 –PSR 60 40 20 0 10 110 AOL, PSR, CMR (dB) CMR 100 PSR AOL CMR 90 80 70 5 10 15 Supply Voltage (±VS) 20 25 100 1k 10k 100k 1M Frequency (Hz) GAIN-BANDWIDTH AND SLEW RATE vs SUPPLY VOLTAGE 28 33 28 GAIN-BANDWIDTH AND SLEW RATE vs TEMPERATURE 30 Slew Rate Gain-Bandwidth (MHz) Gain-Bandwidth (MHz) 24 Slew Rate (V/µs) 20 25 20 Gain-Bandwidth G = +100 20 16 21 16 15 12 5 10 15 Supply Voltage (±VS) 20 17 25 12 –75 –50 –25 0 25 50 75 100 10 125 Temperature (°C) Slew Rate (V/µs) ® Gain-Bandwidth G = +100 Slew Rate 29 24 25 5 OPA604 TYPICAL PERFORMANCE CURVES TA = +25°C, VS = ±15V unless otherwise noted. (CONT) SETTLING TIME vs CLOSED-LOOP GAIN 5 VO = 10V Step RL = 1kΩ CL = 50pF 30 MAXIMUM OUTPUT VOLTAGE SWING vs FREQUENCY 4 VS = ±15V Output Voltage (Vp-p) –1000 20 Settling Time (µs) 3 0.01% 2 0.1% 1 10 0 –1 –10 –100 Closed-Loop Gain (V/V) 0 10k 100k Frequency (Hz) 1M 10M SUPPLY CURRENT vs TEMPERATURE 7 LARGE-SIGNAL TRANSIENT RESPONSE Supply Current (mA) 6 VS = ±24VDC 5 Output Voltage (V) VS = ±15VDC +10 FPO Bleed to edge –10 VS = ±5VDC 4 3 –75 –50 –25 0 25 50 75 100 125 Ambient Temperature (°C) 0 5 Time (µs) 10 SMALL-SIGNAL TRANSIENT RESPONSE 60 SHORT-CIRCUIT CURRENT vs TEMPERATURE Output Voltage (mV) +100 Short-Circuit Current (mA) ISC+ and ISC– 50 40 –100 30 20 0 1µs Time (µs) 2µs –75 –50 –25 0 25 50 75 100 125 Ambient Temperature (°C) ® OPA604 6 TYPICAL PERFORMANCE CURVES TA = +25°C, VS = ±15V unless otherwise noted. (CONT) POWER DISSIPATION vs SUPPLY VOLTAGE 0.5 Total Power Dissipation (W) 1.4 MAXIMUM POWER DISSIPATION vs TEMPERATURE 0.45 Power Dissipation (W) 0.40 0.35 0.30 0.25 0.20 0.15 0.10 0.05 6 8 10 12 Typical high-level music RL = 600Ω Worst case sine wave RL = 600Ω 1.2 1.0 0.8 0.6 0.4 0.2 0 Maximum Specified Operating Temperature 85°C θJ-A = 90°C/W Soldered to Circuit Board (see text) No signal or no load 14 16 18 20 22 24 0 25 50 75 100 125 150 Supply Voltage, ±VS (V) Ambient Temperature (°C) APPLICATIONS INFORMATION OFFSET VOLTAGE ADJUSTMENT The OPA604 offset voltage is laser-trimmed and will require no further trim for most applications. As with most amplifiers, externally trimming the remaining offset can change drift performance by about 0.3µV/°C for each 100µV of adjusted offset. The OPA604 can replace many other amplifiers by leaving the external null circuit unconnected. The OPA604 is unity-gain stable, making it easy to use in a wide range of circuitry. Applications with noisy or high impedance power supply lines may require decoupling capacitors close to the device pins. In most cases, a 1µF tantalum capacitor at each power supply pin is adequate. +VCC Op amp distortion can be considered an internal error source which can be referred to the input. Figure 2 shows a circuit which causes the op amp distortion to be 101 times greater than normally produced by the op amp. The addition of R3 to the otherwise standard noninverting amplifier configuration alters the feedback factor or noise gain of the circuit. The closed-loop gain is unchanged, but the feedback available for error correction is reduced by a factor of 101. This extends the measurement limit, including the effects of the signal-source purity, by a factor of 101. Note that the input signal and load applied to the op amp are the same as with conventional feedback without R3. Validity of this technique can be verified by duplicating measurements at high gain and/or high frequency where the distortion is within the measurement capability of the test equipment. Measurements for this data sheet were made with the Audio Precision System One which greatly simplifies such repetitive measurements. The measurement technique can, however, be performed with manual distortion measurement instruments. CAPACITIVE LOADS The dynamic characteristics of the OPA604 have been optimized for commonly encountered gains, loads and operating conditions. The combination of low closed-loop gain and capacitive load will decrease the phase margin and may lead to gain peaking or oscillations. Load capacitance reacts with the op amp’s open-loop output resistance to form an additional pole in the feedback loop. Figure 3 shows various circuits which preserve phase margin with capacitive load. Request Application Bulletin AB-028 for details of analysis techniques and applications circuits. For the unity-gain buffer, Figure 3a, stability is preserved by adding a phase-lead network, RC and CC. Voltage drop 7 2 3 4 (1) OPA604 5 6 1 ±50mV Typical Trim Range –VCC NOTE: (1) 50kΩ to 1MΩ Trim Potentiometer (100kΩ Recommended) FIGURE 1. Offset Voltage Trim. DISTORTION MEASUREMENTS The distortion produced by the OPA604 is below the measurement limit of virtually all commercially available equipment. A special test circuit, however, can be used to extend the measurement capabilities. ® 7 OPA604 across RC will reduce output voltage swing with heavy loads. An alternate circuit, Figure 3b, does not limit the output with low load impedance. It provides a small amount of positive feedback to reduce the net feedback factor. Input impedance of this circuit falls at high frequency as op amp gain rolloff reduces the bootstrap action on the compensation network. Figures 3c and 3d show compensation techniques for noninverting amplifiers. Like the follower circuits, the circuit in Figure 3d eliminates voltage drop due to load current, but at the penalty of somewhat reduced input impedance at high frequency. Figures 3e and 3f show input lead compensation networks for inverting and difference amplifier configurations. NOISE PERFORMANCE Op amp noise is described by two parameters—noise voltage and noise current. The voltage noise determines the noise performance with low source impedance. Low noise bipolar-input op amps such as the OPA27 and OPA37 provide very low voltage noise. But if source impedance is greater than a few thousand ohms, the current noise of bipolar-input op amps react with the source impedance and will dominate. At a few thousand ohms source impedance and above, the OPA604 will generally provide lower noise. POWER DISSIPATION The OPA604 is capable of driving a 600Ω load with power supply voltages up to ±24V. Internal power dissipation is increased when operating at high power supply voltage. The typical performance curve, Power Dissipation vs Power Supply Voltage, shows quiescent dissipation (no signal or no load) as well as dissipation with a worst case continuous sine wave. Continuous high-level music signals typically produce dissipation significantly less than worst case sine waves. Copper leadframe construction used in the OPA604 improves heat dissipation compared to conventional plastic packages. To achieve best heat dissipation, solder the device directly to the circuit board and use wide circuit board traces. OUTPUT CURRENT LIMIT Output current is limited by internal circuitry to approximately ±40mA at 25°C. The limit current decreases with increasing temperature as shown in the typical curves. R1 R2 SIG. DIST. GAIN GAIN 1 R3 OPA604 VO = 10Vp-p (3.5Vrms) 10 100 101 101 101 R1 ∞ 500Ω 50Ω R2 5kΩ 5kΩ 5kΩ R3 50Ω 500Ω ∞ Generator Output Analyzer Input Audio Precision System One Analyzer(1) RL 1kΩ IBM PC or Compatible NOTE: (1) Measurement BW = 80kHz FIGURE 2. Distortion Test Circuit. ® OPA604 8 (a) (b) CC 820pF RC OPA604 ei 750Ω CL 5000pF CC = 120 X 10–12 CL ei 2kΩ RC = CC = 10Ω R2 4CL X 1010 – 1 CL X 103 RC R2 eo CC 0.47µF RC CL 5000pF OPA604 eo (c) R1 10kΩ R2 10kΩ CC 24pF RC OPA604 ei 50 CL R2 25Ω CL 5000pF eo ei R1 2kΩ RC 20Ω CC 0.22µF (d) R2 2kΩ OPA604 eo CL 5000pF CC = RC = R2 2CL X 1010 – (1 + R2/R1) CL X 103 RC CC = (e) R2 e1 2kΩ R1 ei 2kΩ RC 20Ω CC 0.22µF RC = R2 2CL X 1010 – (1 + R2/R1) RC = CC = CL X RC 103 CC = OPA604 eo CL 5000pF e2 2kΩ CC 0.22µF R3 2kΩ RC 20Ω R1 (f) R2 2kΩ OPA604 eo CL 5000pF R4 2kΩ R2 2CL X 1010 – (1 + R2/R1) CL X 103 RC NOTE: Design equations and component values are approximate. User adjustment is required for optimum performance. FIGURE 3. Driving Large Capacitive Loads. ® 9 OPA604 R4 22kΩ C3 R1 VIN 2.7kΩ 22kΩ C1 3000pF 10kΩ C2 2000pF fp = 20kHz OPA604 VO R2 R3 100pF FIGURE 4. Three-Pole Low-Pass Filter. R1 VIN 6.04kΩ R2 4.02kΩ R5 2kΩ OPA604 VO C3 1000pF R2 4.02kΩ 1 2 1 2 OPA2604 OPA2604 Low-pass 3-pole Butterworth f–3dB = 40kHz C1 1000pF R4 5.36kΩ See Application Bulletin AB-026 for information on GIC filters. C2 1000pF FIGURE 5. Three-Pole Generalized Immittance Converter (GIC) Low-Pass Filter. 1 7.87kΩ – VIN + 100pF 2 10kΩ 10kΩ OPA2604 OPA604 VO G=1 1 7.87kΩ 100kHz Input Filter 2 OPA2604 10kΩ 10kΩ FIGURE 6. Differential Amplifier with Low-Pass Filter. ® OPA604 10 100Ω 10kΩ NOTE: (1) C1 ≈ COUT 2π R f f c G = 101 (40dB) OPA604 Piezoelectric Transducer 1MΩ(1) NOTE: (1) Provides input bias current return path. RF = Internal feedback resistance = 1.5kΩ fC = Crossover frequency = 8MHz 10 5 PCM63 20-bit 6 D/A 9 Converter C1(1) OPA604 VO = ±3Vp To low-pass filter. FIGURE 7. High Impedance Amplifier. FIGURE 8. Digital Audio DAC I-V Amplifier. OPA604 A2 I2 R4 OPA604 A1 VIN R2 I1 R3 51Ω IL = I1 + I2 51Ω VOUT R1 VOUT = VIN (1+R2/R1) Load FIGURE 9. Using Two OPA604 Op Amps to Double the Output Current to a Load. ® 11 OPA604 SOUND QUALITY The following discussion is provided, recognizing that not all measured performance behavior explains or correlates with listening tests by audio experts. The design of the OPA604 included consideration of both objective performance measurements, as well as an awareness of widely held theory on the success and failure of previous op amp designs. SOUND QUALITY The sound quality of an op amp is often the crucial selection criteria—even when a data sheet claims exceptional distortion performance. By its nature, sound quality is subjective. Furthermore, results of listening tests can vary depending on application and circuit configuration. Even experienced listeners in controlled tests often reach different conclusions. Many audio experts believe that the sound quality of a high performance FET op amp is superior to that of bipolar op amps. A possible reason for this is that bipolar designs generate greater odd-order harmonics than FETs. To the human ear, odd-order harmonics have long been identified as sounding more unpleasant than even-order harmonics. FETs, like vacuum tubes, have a square-law I-V transfer function which is more linear than the exponential transfer function of a bipolar transistor. As a direct result of this square-law characteristic, FETs produce predominantly even-order harmonics. Figure 10 shows the transfer function of a bipolar transistor and FET. Fourier transformation of both transfer functions reveals the lower odd-order harmonics of the FET amplifier stage. FFT VO I1 800µA R1 75Ω (+) J1 (–) J2 J3 R2 75Ω R5 500Ω R6 500Ω R7 4kΩ J4 Distortion Rejection Circuitry I2 200µA J5 Output Stage R10 10kΩ Q1 Q3 Q2 Q4 R11 10kΩ R3 1kΩ R4 1kΩ R8 3kΩ R9 3kΩ THE OPA604 DESIGN The OPA604 uses FETs throughout the signal path, including the input stage, input-stage load, and the important phase-splitting section of the output stage. Bipolar transistors are used where their attributes, such as current capability are important, and where their transfer characteristics have minimal impact. The topology consists of a single folded-cascode gain stage followed by a unity-gain output stage. Differential input transistors J1 and J2 are special large-geometry, P-channel JFETs. Input stage current is a relatively high 800µA, providing high transconductance and reducing voltage noise. Laser trimming of stage currents and careful attention to symmetry yields a nearly symmetrical slew rate of ±25V/µs. The JFET input stage holds input bias current to approximately 50pA or roughly 3000 times lower than common bipolar-input audio op amps. This dramatically reduces noise with high-impedance circuitry. The drains of J1 and J2 are cascoded by Q1 and Q2, driving the input stage loads, FETs J3 and J4. Distortion reduction circuitry (patented) linearizes the openloop response and increases voltage gain. The 20MHz bandwidth of the OPA604 further reduces distortion through the user-connected feedback loop. The output stage consists of a JFET phase-splitter loaded into high speed all-NPN output drivers. Output transistors are biased by a special circuit to prevent cutoff, even with full output swing into 600Ω loads. 1 VBE = 1kHz + DC Bias IC IC V (mA) BE log (VO) fO 2fO 3fO 4fO 5fO 0 0 0.65 VBE (V) 1 0 1234 5 Frequency (kHz) VGS = 1kHz + DC Bias 1 VGS FFT –ID (mA) ID VO log (VO) fO 2fO 3fO 4fO 5fO 0 1 VGS (V) 0 0 1234 5 Frequency (kHz) FIGURE 10. I-V and Spectral Response of NPN and JFET. ® OPA604 12