OPA653
OP
A6
53
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SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
Wideband, Fixed Gain, JFET-Input AMPLIFIER
Check for Samples: OPA653
FEATURES
DESCRIPTION
•
•
•
•
•
•
•
•
•
•
The OPA653 combines a very wideband
voltage-feedback operational amplifier with a
JFET-input stage with internal gain setting resistors to
achieve an ultra-high, dynamic-range amplifier for
fixed gain of +2-V/V or –1-V/V applications.
1
23
•
HIGH BANDWIDTH: 500 MHz (G = +2 V/V)
HIGH SLEW RATE: 2675 V/μs (4-V Step)
EXCELLENT THD: –71dBc at 10 MHz
LOW INPUT VOLTAGE NOISE: 6.1 nV/√Hz
FAST OVERDRIVE RECOVERY: 8 ns
FAST SETTLING TIME (1% 4-V Step): 7.9 ns
LOW INPUT OFFSET VOLTAGE: ±1 mV
LOW INPUT BIAS CURRENT: ±10 pA
HIGH INPUT IMPEDANCE: 1012 Ω||2.5 pF
INTERNAL GAIN SETTING RESISTORS:
G = +2 V/V or G = –1 V/V
HIGH OUTPUT CURRENT: 70 mA
APPLICATIONS
•
•
•
•
•
TEST AND MEASUREMENT FRONT-END
HIGH-INPUT IMPEDANCE PROBES
DATA ACQUISITION CARDS
OSCILLOSCOPE INPUT
ADC INPUT AMPLIFIER
The 500-MHz wide gain of +2-V/V bandwidth is
complemented by a very high 2675-V/μs slew rate
and fast settling time that make it ideal for
time-domain and pulse-oriented applications.
Excellent –72-dBc THD distortion performance at
10 MHz makes the OPA653 an excellent choice for
frequency-domain and FFT analysis applications.
Additionally, with the low 6.1-nV/√Hz voltage noise,
low bias current, and high impedance JFET input, it
supports very low noise, wideband, high input
impedance
applications.
Examples
include
high-impedance probes, data acquisition cards, and
oscilloscope front-ends.
RELATED
OPERATIONAL AMPLIFIER
PRODUCTS
NONINVERTING GAIN OF +2 V/V
50-W TEST ENVIRONMENT
+6 V
0.1 mF
10 mF
SLEW
RATE
(V/μs)
VOLTAGE
NOISE
(nV/√Hz)
DEVICE
VS (V)
BW
(MHz)
OPA356
+5
200
300
5.80
Unity-Gain
Stable CMOS
OPA656
±5
500
290
7
Unity-Gain
Stable
JFET-Input
OPA657
±5
350
700
4.8
Gain of +7
Stable
JFET-Input
OPA659
±6
650
2550
8.9
Unity-Gain
Stable
JFET-Input
THS4631
±15
105
900
7
Unity-Gain
Stable
JFET-Input
VS+
50-W
Source
VIN+
VOUT
OPA653
49.9 W
VIN-
49.9 W
50-W
Load
160 W
160 W
VS0.1 mF
10 mF
AMPLIFIER
DESCRIPTION
-6 V
SMALL-SIGNAL FREQUENCY RESPONSE
6
3
Signal Gain (dB)
0
-3
-6
-9
-12
-15
VO = 200 mVPP
-18
100 k
1M
10 M
100 M
1G
Frequency (Hz)
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008–2009, Texas Instruments Incorporated
OPA653
SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION (1)
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
OPA653
SOT23-5
DBV
–40°C to +85°C
BZW
OPA653
VSON-8
DRB
–40°C to +85°C
OBEI
(1)
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
OPA653IDBVT
Tape and Reel, 250
OPA653IDBVR
Tape and Reel, 3000
OPA653IDRBT
Tape and Reel, 250
OPA653IDRBR
Tape and Reel, 3000
For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
ABSOLUTE MAXIMUM RATINGS
Over operating free-air temperature range (unless otherwise noted).
OPA653
UNIT
Power Supply Voltage VS+ to VS–
±6.5
V
Input Voltage
±VS
V
Input Current
100
mA
100
mA
Output Current
Continuous Power Dissipation
See Thermal Characteristics
Operating Free Air Temperature Range, TA
–40 to +85
°C
Storage Temperature Range
–65 to +150
°C
Maximum Junction Temperature, TJ
+150
°C
Maximum Junction Temperature, TJ (continuous operation for long term reliability)
+125
°C
Human Body Model (HBM)
4000
V
Charge Device Model (CDM)
1000
V
Machine Model
200
V
ESD
Rating:
DRB PACKAGE
VSON-8
(TOP VIEW)
Note:
2
DRV PACKAGE
SOT23-5
(TOP VIEW)
NC
1
8
NC
VIN-
2
7
VS+
VIN+
3
6
VS-
4
5
VOUT
1
-VS
2
VOUT
NC
VIN+
3
5
VS+
4
VIN-
NC: No connection.
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SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
ELECTRICAL CHARACTERISTICS: VS = ±6 V
At G = +2 V/V, RL = 100 Ω, and TA = +25°C, unless otherwise noted.
OPA653
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL (1)
AC PERFORMANCE
Small-Signal Bandwidth
VO = 200 mVPP
500
MHz
C
Bandwidth for 0.1-dB Flatness
VO = 2 VPP
75
MHz
C
Large-Signal Bandwidth
VO = 2 VPP
475
MHz
B
Slew Rate
VO = 4-V Step
2675
V/μs
B
Rise and Fall Time
VO = 4-V Step
1.3
ns
C
Settling Time to 1%
VO = 4-V Step
7.9
ns
C
Pulse Response Overshoot
VO = 4-V Step
14
%
C
2nd harmonic
–72
dBc
C
3rd harmonic
–90
dBc
C
Second-order Intermodulation
–75
dBc
C
Third-order Intermodulation
–96
dBc
C
Harmonic Distortion
Intermodulation Distortion
f = 10 MHz, VO = 2 VPP
VO = 2-VPP envelope (each tone 1 VPP),
f1 = 10 MHz, f2 = 11 MHz
Input Voltage Noise
f > 100 kHz
6.1
nV/√Hz
C
Input Current Noise
f > 100 kHz
1.8
fA/√Hz
C
TA = +25°C
±0.5
±1.25
%
A
TA = –40° C to +85°C
±0.6
±1.35
%
B
160
180
Ω
A
dB
C
DC PERFORMANCE
Gain Error
Internal RF and RG
140
Open-Loop Voltage Gain (AOL) (2)
Input Offset Voltage
Average input offset voltage drift
Input Bias Current, Noninverting input
Average input bias current drift,
noninverting input
62
TA = +25°C, VCM = 0 V
±1
±5
mV
A
TA = –40°C to +85°C, VCM = 0 V
DRB package
±1.5
±7
mV
B
TA = –40°C to +85°C, VCM = 0 V
DBV package
±1.5
±8.9
mV
B
TA = –40°C to +85°C, VCM = 0 V
DRB package
±10
±30
μV/°C
B
TA = –40°C to +85°C, VCM = 0 V
DBV package
±10
±60
μV/°C
B
TA = +25°C, VCM = 0 V
±10
±50
pA
A
TA = 0°C to + 70°C, VCM = 0 V
±240
±1200
pA
B
TA = –40°C to +85°C, VCM = 0 V
±640
±3200
pA
B
TA = 0°C to +70°C, VCM = 0 V
±5
±26
pA/°C
B
TA = –40°C to +85°C, VCM = 0 V
±7
±34
pA/°C
B
V
C
INPUT
Input Voltage Range
VIN+ = VIN–
±3.5
12
Noninverting Input Impedance
10
2.5
Ω
pF
C
OUTPUT
Output Voltage Swing
Output Voltage Swing
Output Current, Sourcing, Sinking
Closed-Loop Output Impedance
(1)
(2)
TA = +25°C, No load
±4.35
±4.45
V
A
TA = +25°C, RL = 100 Ω
±3.2
±3.6
V
A
TA = –40°C to +85°C, No load
±4.2
±4.3
V
B
TA = –40°C to +85°C, RL = 100 Ω
±3.07
±3.2
V
B
TA = +25°C
±60
±70
mA
A
TA = –40°C to +85°C
±56
±66
mA
B
0.16
Ω
C
f = 100 kHz
Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Open loop gain is for informational use only. Open loop gain is from simulation and not measured in a closed-loop amplifier.
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OPA653
SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
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ELECTRICAL CHARACTERISTICS: VS = ±6 V (continued)
At G = +2 V/V, RL = 100 Ω, and TA = +25°C, unless otherwise noted.
OPA653
PARAMETER
UNIT
TEST
LEVEL (1)
±6.5
V
B
33.5
mA
A
35.7
mA
B
66
dB
A
64
dB
B
°C
C
CONDITIONS
MIN
TYP
MAX
TA = –40°C to +85°C
±3.5
±6
TA = +25°C
30.5
32
TA = –40°C to +85°C
28.3
TA = +25°C, ±VS = 5.5 V to 6.5 V
62
TA = –40°C to +85°C, ±VS = 5.5 V to 6.5 V
60
POWER SUPPLY
Specified Operating Voltage
Quiescent Current
Power-Supply Rejection Ratio
THERMAL CHARACTERISTICS
Specified Operating Range
DBV and DRB Packages
–40
Thermal Resistance, θ JA
+85
Junction-to-ambient
DBV
SOT23-5
105
°C/W
C
DRB
VSON-8
55
°C/W
C
space
space
TYPICAL CHARACTERISTICS
Table of Graphs
TITLE
FIGURE
Small-Signal Frequency Response
VO = 200 mVPP
Figure 1
Noninverting Large-Signal Frequency Response
Gain = +2 V/V
Figure 2
Inverting Large-Signal Frequency Response
Gain = –1 V/V
Figure 3
Input-Referred Voltage and Current Noise Spectral Density
Figure 4
Noninverting Transient Response
0.5-V Step
Figure 5
Inverting Transient Response
0.5-V Step
Figure 6
Harmonic Distortion vs Frequency
Figure 7, Figure 8
Harmonic Distortion vs Load
Figure 9
Harmonic Distortion vs Output Voltage
Figure 10
Harmonic Distortion vs ±Supply Voltage
Figure 11
Two-Tone, Second- and Third-Order Intermodulation Distortion vs Frequency
Figure 12
Noninverting Overdrive Recovery
Gain = +2 V/V
Figure 13
Inverting Overdrive Recovery
Gain = –1 V/V
Figure 14
Power-Supply Rejection Ratio vs Frequency
Figure 15
Frequency Response vs Capacitive Load
Figure 16
Recommended RISO vs Capacitive Load
Figure 17
Closed-Loop Output Impedance vs Frequency
Figure 18
Slew Rate vs VOUT Step
Figure 19
Output Voltage Swing vs RLOAD
Figure 20
4
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SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
TYPICAL CHARACTERISTICS: VS = ±6 V
At G = +2 V/V, RL = 100 Ω, and TA = +25°C, unless otherwise noted.
SMALL-SIGNAL FREQUENCY RESPONSE
(VO = 200 mVPP)
NONINVERTING LARGE-SIGNAL FREQUENCY RESPONSE
8
G = -1 V/V
3
VO = 1.0 VPP
7
0
6
VO = 0.5 VPP
Signal Gain (dB)
Normalized Signal Gain (dB)
6
G = +2 V/V
-3
-6
-9
5
VO = 4.0 VPP
4
3
VO = 2.0 VPP
-12
2
-15
1
-18
100 k
G = +2 V/V
0
100 k
1M
10 M
1M
100 M
1G
VO = 6.0 VPP
10 M
Frequency (Hz)
INVERTING LARGE-SIGNAL FREQUENCY RESPONSE
INPUT-REFERRED VOLTAGE AND CURRENT NOISE
SPECTRAL DENSITY
VO = 2.0 VPP
Input-Referred Voltage Noise (nV/ÖHz)
Input-Referred Current Noise (fA/ÖHz)
Figure 2.
2
VO = 1.0 VPP
0
VO = 6.0 VPP
-2
VO = 4.0 VPP
-4
VO = 0.5 VPP
-6
-8
G = -1 V/V
-10
100 k
1M
1k
Input-Referred
Voltage Noise
100
10
Input-Referred
Current Noise
1
10 M
100 M
1G
10
100
10 M
1M
Figure 4.
NONINVERTING TRANSIENT RESPONSE
INVERTING TRANSIENT RESPONSE
3
G = +2 V/V
100 k
Frequency (Hz)
Figure 3.
3
10 k
1k
Frequency (Hz)
G = -1 V/V
4-V Step
2
4-V Step
2
2-V Step
2-V Step
1
0
VOUT (V)
1
VOUT (V)
1G
Figure 1.
4
Signal Gain (dB)
100 M
Frequency (Hz)
0.5-V Step
0
-1
-1
-2
-2
-3
0.5-V Step
-3
0
5
10
15
20
25
30
35
40
45
50
0
Time (ns)
5
10
15
20
25
30
35
40
45
50
Time (ns)
Figure 5.
Figure 6.
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OPA653
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TYPICAL CHARACTERISTICS: VS = ±6 V (continued)
At G = +2 V/V, RL = 100 Ω, and TA = +25°C, unless otherwise noted.
NONINVERTING HARMONIC DISTORTION
vs FREQUENCY
-40
VS = ±6 V
RL = 100 W
VO = 2 VPP
Gain = +2 V/V
-50
-60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
-40
INVERTING HARMONIC DISTORTION
vs FREQUENCY
Second Harmonic
-70
-80
Third
Harmonic
-90
-100
VS = ±6 V
RL = 100 W
VO = 2 VPP
Gain = -1 V/V
-50
-60
-70
Third
Harmonic
-80
-90
-100
1M
10 M
100 M
1M
10 M
Frequency (Hz)
-50
Figure 7.
Figure 8.
HARMONIC DISTORTION
vs LOAD
HARMONIC DISTORTION
vs OUTPUT VOLTAGE
-50
-65
-70
-75
VS = ±6 V
RL = 100 W
f = 10 MHz
Gain = +2 V/V
-55
Harmonic Distortion (dBc)
-60
Second Harmonic
-80
-85
-90
-95
-60
-65
Second Harmonic
-70
-75
-80
-85
Third Harmonic
-90
-95
Third Harmonic
-100
-100
100
200
300
400
500
600
700
800
900
1k
1
3
2
Load (W)
Figure 10.
HARMONIC DISTORTION
vs ±VSUPPLY VOLTAGE
TWO-TONE INTERMODULATION DISTORTION
vs FREQUENCY
-40
Intermodulation Distortion (dBc)
Harmonic Distortion (dBc)
Second Harmonic
-80
-85
Third Harmonic
-90
-100
f = 10 MHz
RL = 100 W
VOUT = 2 VPP
Gain = +2 V/V
4.0
5
Figure 9.
-75
-95
4
4.5
6
VOUT (VPP)
-70
VS = ±6 V
RL = 100 W
VOUT = 2-VPP Envelope
Gain = +2 V/V
-50
-60
Second Intermodulation
-70
-80
Third Intermodulation
-90
-100
5.0
5.5
6.0
0
±Supply Voltage (V)
Figure 11.
6
100 M
Frequency (Hz)
VS = ±6 V
f = 10 MHz
VO = 2 VPP
Gain = +2 V/V
-55
Harmonic Distortion (dBc)
Second Harmonic
50 M
100 M
Frequency (Hz)
150 M
Figure 12.
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SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
TYPICAL CHARACTERISTICS: VS = ±6 V (continued)
At G = +2 V/V, RL = 100 Ω, and TA = +25°C, unless otherwise noted.
INVERTING OVERDRIVE RECOVERY
(GAIN = –1V/V)
2
4
1
2
0
0
VOUT
Right Scale
VIN
Left Scale
-1
-2
-2
-4
-3
-6
0
20
40
60
4
3
VOUT
Right Scale
VIN
Left Scale
2
2
1
1
0
0
-1
-1
-2
-2
-3
-3
-4
-4
0
80 100 120 140 160 180 200 220 240
20
40
60
80
100 120 140 160 180 200
Time (ns)
Time (ns)
Figure 13.
Figure 14.
POWER-SUPPLY REJECTION RATIO
vs FREQUENCY
FREQUENCY RESPONSE
vs CLOAD (RLOAD = 1 kΩ)
5
80
70
CL = 10 pF
RISO = 46.4 W
0
Normalized Gain (dB)
60
PSRR (dB)
4
3
Input Voltage (V)
6
50
+PSRR
-PSRR
40
30
20
-50
CL = 100 pF
RISO = 10.7 W
-10
CL = 47 pF
RISO = 15.0 W
-15
CL = 22 pF
RISO = 23.7 W
-20
10
0
100 k
10 M
1M
-25
10 M
100 M
Output Voltage (V)
3
Output Voltage (V)
Input Voltage (V)
NONINVERTING OVERDRIVE RECOVERY
(GAIN = +2V/V)
100 M
Frequency (Hz)
1G
Frequency (Hz)
Figure 15.
Figure 16.
RISO vs CLOAD
(RLOAD = 1 kΩ)
CLOSED-LOOP OUTPUT IMPEDANCE
vs FREQUENCY
100
RISO (W)
Closed-Loop Output Impedance (W)
100
10
10
100
10
1
0.1
100 k
1M
10 M
100 M
1G
Frequency (Hz)
CLOAD (pF)
Figure 17.
Figure 18.
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TYPICAL CHARACTERISTICS: VS = ±6 V (continued)
At G = +2 V/V, RL = 100 Ω, and TA = +25°C, unless otherwise noted.
SLEW RATE
vs VOUT STEP
OUTPUT VOLTAGE SWING
vs RLOAD
4000
5
4
Falling
3
2
Rising
VOUT (V)
Slew Rate (V/ms)
3000
2000
1
VOUT High
0
VOUT Low
-1
-2
1000
-3
-4
0
-5
1
2
3
4
5
6
0
VOUT Step (V)
1k
RLOAD (W)
Figure 19.
8
100 200 300 400 500 600 700 800 900
Figure 20.
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SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
APPLICATION INFORMATION
Wideband, Noninverting, and Inverting
Operation
The OPA653 is a very broadband, voltage-feedback
amplifier with internal gain-setting resistors that set a
fixed gain of +2 V/V or -1 V/V and a high-impedance
JFET-input stage. Its very high bandwidth of 500 MHz
can be used to either deliver high signal bandwidths
at a gain of +2 V/V or, if driven from a low-impedance
source, a gain of –1 V/V . The OPA653 is designed to
provide very low noise and accurate pulse response
with low overshoot and ringing. To achieve the full
performance of the OPA653, careful attention to
printed circuit board (PCB) layout and component
selection is required as discussed in the remaining
sections of this data sheet.
Figure 21 shows the noninverting gain of +2-V/V
circuit that is used as the basis for the Typical
Characteristics.
Most
of
the
curves
were
characterized using signal sources with 50-Ω driving
impedance, and with measurement equipment that
presents a 50-Ω load impedance. In Figure 21, the
49.9-Ω shunt resistor to ground at the VIN+ input is
used to match the source impedance of the test
generator and cable, while the 49.9-Ω series output
resistor VOUT provides matching impedance for the
measurement equipment load and cable. Data sheet
voltage swing specifications are taken at the
noninverting input pin, VIN+, or the output pin, VOUT,
unless otherwise noted.
+6 V
0.1 mF
50-W
Source
10 mF
VS+
VIN+
VOUT
49.9 W
50-W
Load
OPA653
49.9 W
VIN-
160 W
160 W
VS0.1 mF
10 mF
-6 V
Figure 22 shows the OPA653 in an inverting gain of
–1 V/V configuration in a 50-Ω test environment as
was used for testing the Typical Characteristics. The
circuit operation is essentially the same as Figure 21
except that a 72.3-Ω termination resistor is now used
between the VIN– input and ground, so that together
with the gain-setting resistor (RG = 160 Ω), the input
impedance is approximately 50 Ω. The VIN+ input is
terminated to ground using a 49.9-Ω resistor as a
precaution to avoid single transistor oscillations at the
input; the value is not critical, but attention should be
paid to avoid large values because of the noise
contribution as noted below.
+6 V
0.1 mF
10 mF
VS+
49.9 W
VIN+
VOUT
50-W
Source
49.9 W
50-W
Load
OPA653
VIN-
160 W
160 W
72.3 W
VS-
0.1 mF
10 mF
-6 V
Figure 22. Inverting Gain of –1 V/V in 50-Ω Test
Environment
Note that the 72.3-Ω input termination resistor and
the 50-Ω source impedance of the test equipment
modify the noise gain to +1.84 V/V and the amplifier
is compensated for optimal performance with a noise
gain of +2 V/V. This compensation reduces the phase
margin and results in more peaking in the frequency
response and more overshoot/ringing in the pulse
response. This effect can be seen by comparing the
inverting and noninverting frequency and pulse
response graphs in the characteristic data. The
amplifier phase margin can be restored in an
application that uses an inverting configuration if it is
driven from a very low impedance source such as an
op amp.
Figure 21. Noninverting Gain of +2 V/V in 50-Ω
Test Environment
space
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OPA653
SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
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OPERATING SUGGESTIONS
Setting Resistor Values to Minimize Noise
The OPA653 provides a low input noise voltage.
Figure 23 shows the op amp noise analysis model
with all the noise terms included. In this model, all the
noise terms are taken to be noise voltage or current
density terms in either nV/√Hz or pA/√Hz.
eN
OPA653
However, attention should be paid to the value of RT
or other source impedance on the noninverting input.
High-value resistive impedance on the noninverting
input can add significant noise; for example, 2.4 kΩ
adds a Johnson voltage noise term equal to the
amplifier itself (6.2 nV/√Hz). So while the JFET input
of the OPA653 is ideal for high source impedance
applications in the noninverting configuration of
Figure 21, the overall bandwidth and noise are limited
by high source impedances.
VIN+
VOUT
RT
IBN
Driving Capacitive Loads
eO
4kTRT
RF = 160 W
IBI
4kTRF
4kT
RG
RG = 160 W
VIN-
Figure 23. Noise Analysis Circuit
The total output spot noise voltage can be computed
as the square root of the squared contributing terms
to the output noise voltage. This calculation adds all
the contributing noise powers at the output by
superposition, then takes the square root to return to
a spot noise voltage. Equation 1 shows the general
form for this output noise voltage using the terms
shown in Figure 23.
2
eO =
[4kTR
]
2
2
1+
T + (IBNRT) + eN
RF
RF
+ (IBIRF)2 + 4kTRF 1 +
RG
RG
(1)
Dividing this expression by the noise gain = 1 +
RF/RG gives the equivalent input-referred spot noise
voltage at the noninverting input as shown in
Equation 2
One of the most demanding and yet very common
load conditions for an op amp is capacitive loading.
The OPA653 is very robust, but care should be taken
with light loading scenarios so output capacitance
does not lead to decreased stability, increased
frequency response peaking, overshoot, and ringing.
When the amplifier output resistance is considered,
capacitive loading introduces an additional pole in the
signal path that reduces the phase margin. Several
external solutions to this problem have been
suggested for standard op amps. Because the
OPA653 has internal gain-setting resistors, the only
real option is to use a series output resistor. This
option is a good solution because when the primary
considerations are frequency response flatness,
pulse response fidelity, and/or distortion, a series
output resistor is the simplest and most effective
technique. The idea is to isolate the capacitive load
from the feedback loop by inserting a series isolation
resistor, RISO, between the amplifier output and the
capacitive load as shown in Figure 24 below. In
effect, this configuration isolates the phase shift from
the loop gain of the amplifier, thus restoring the
phase margin and improving stability.
OPA653
VIN+
VOUT RISO
160 W
VLOAD
CLOAD
VIN-
RLOAD
160 W
2
eNI =
2
4kTRT + (IBNRT) + eN
2
I R
+ BI F
2
+
4kTRF
2
Figure 24. Adding Series Ouput Resistance to
Isolate Capacitive Loads
(2)
Putting high resistor values into Equation 2 can
quickly dominate the total equivalent input-referred
noise. Because the gain-setting resistors, RF and RG,
are internal to the device, the user cannot change this
noise contribution, and the noise gain is equal to +2
V/V.
10
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The Typical Characteristics show the recommended
RISO versus Capacitive Load performance (see
Figure 17) and the resulting frequency response with
a 1-kΩ load. Note that larger RISO values are required
for lower capacitive loading. In this case, a design
target of a maximally-flat frequency response was
used. Lower values of RISO may be used if some
peaking can be tolerated. Long PCB traces,
unmatched cables, and connections to multiple
devices can easily degrade the performance of the
OPA653. Always consider this effect carefully, and
add the recommended series resistor as close as
possible to the OPA653 output pin (see the Board
Layout section). With heavier loads (for example, the
100-Ω load presented in the test circuits used for
testing the Typical Characteristics), the OPA653 is
very robust; RISO can be as low as 10 Ω with
capacitive loads less than 5 pF and continue to show
a flat frequency response.
Board Layout
Distortion Performance
b) Minimize the distance (less than 0.25 in, or 6,35
mm) from the power-supply pins to the
high-frequency, 0.1-μF decoupling capacitors. At the
device pins, the ground and power plane layout
should not be in close proximity to the signal I/O pins.
Use a single point ground, located away from the
input pins, for the positive and negative supply
high-frequency, 0.1-μF decoupling capacitors. Avoid
narrow power and ground traces to minimize
inductance between the pins and the decoupling
capacitors. The power-supply connections should
always be decoupled with these capacitors. Larger
(2.2-μF to 10-μF) decoupling capacitors, effective at
lower frequencies, should also be used on the supply
pins. These larger capacitors may be placed
somewhat farther from the device and may be shared
among several devices in the same area of the PCB.
The OPA653 is capable of delivering low distortion at
high frequencies. The distortion plots in the Typical
Characteristics show the typical distortion under a
wide variety of conditions. Generally, the best
distortion performance can be achieved using higher
power-supply voltage (±6 V is recommended), lower
output voltage swings, and lower loads.
The total load includes the feedback network—in the
noninverting configuration, this value is the sum of
RF + RG = 320 Ω, while in the inverting configuration
the total load is only RF = 160 Ω (see Figure 22).
Power-supply decoupling is critical for harmonic
distortion performance. In particular, for optimal
second-harmonic performance, the high-frequency,
0.1-μF, power-supply decoupling capacitors should
be as close as posible to the positive and negative
supply pins and should be brought to a single point
ground away from the input pins.
Pulse and Transient Response
To achieve optimum pulse and transient response,
the OPA653 should be used in a noise gain of +2 V/V
configuration, with minimal capacitance at the output,
and high-frequency, 0.1-μF, power-supply decoupling
capacitors as close the power pins as possible.
Note: Noise gain of +2 V/V is achieved by tying VIN–
to a 0-Ω point. In noninverting gain of +2 V/V
applications, VIN– should be grounded, and in
inverting gain of –1 V/V applications, VIN– should be
driven from a near-0-Ω source such as an op amp.
Achieving
optimum
performance
with
a
high-frequency amplifier such as the OPA653
requires careful attention to PCB layout parasitics and
external component types. Recommendations that
can optimize device performance include the
following.
a) Minimize parasitic capacitance to any ac ground
for all of the signal input/output (I/O) pins. Parasitic
capacitance on the output and inverting input pins
can cause instability: on the noninverting input, it can
react with the source impedance to cause
unintentional band-limiting. To reduce unwanted
capacitance, a window around the signal I/O pins
should be opened in all of the ground and power
planes around those pins. Otherwise, ground and
power planes should be unbroken elsewhere on the
board.
c) Careful selection and placement of external
components
preserves
the
high-frequency
performance of the OPA653. Resistors should be a
very low reactance type. Surface-mount resistors
work best and allow a tighter overall layout. Metal film
and carbon composition, axially-leaded resistors can
also provide good high-frequency performance.
Again, keep the leads and PCB trace length as short
as possible. Never use wirewound-type resistors in a
high-frequency application. The inverting input pin is
the most sensitive to parasitic capacitance;
consequently, always position the feedback resistor
as close to the negative input as possible. The output
is also sensitive to parasitic capacitance; therefore,
position a series output resistor (in this case, RISO) as
close to the output pin as possible.
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Other network components, such as noninverting
input termination resistors, should also be placed
close to the package. Even with a low parasitic
capacitance, excessively high resistor values can
create significant time constants that can degrade
device performance. Good axial metal film or
surface-mount resistors have approximately 0.2 pF in
shunt with the resistor. For resistor values greater
than 1.5 kΩ, this parasitic capacitance can add a pole
and/or zero below 500 MHz that can affect circuit
operation. Keep resistor values as low as possible.
Using values less than 500 Ω automatically holds the
resistor noise terms low, and minimizes the effects of
parasitic capacitance.
well as a doubly-terminated line. If the input
impedance of the destination device is low, there will
be some signal attenuation as a result of the voltage
divider formed by the series output into the
terminating impedance.
d) Connections to other wideband devices on the
board may be made with short direct traces or
through onboard transmission lines. For short
connections, consider the trace and the input to the
next device as a lumped capacitive load. Relatively
wide traces (50 mils to 100 mils, or 1,27 cm to 2,54
cm) should be used. Estimate the total capacitive
load and set RISO from the plot of Recommended
RISO vs Capacitive Load (Figure 17). Low parasitic
capacitive loads (less than 5 pF) may not need an
RISO because the OPA653 is nominally compensated
to operate with a 2-pF parasitic load.
Input and ESD Protection
Higher parasitic capacitive loads without an RISO are
allowed as the signal gain increases (increasing the
unloaded phase margin). If a long trace is required,
and the 6-dB signal loss intrinsic to a
doubly-terminated transmission line is acceptable,
implement a matched impedance transmission line
using microstrip or stripline techniques (consult an
ECL design handbook for microstrip and stripline
layout techniques). A 50-Ω environment is normally
not necessary onboard, and in fact a higher
impedance environment improves distortion as shown
in the distortion versus load plots. With a
characteristic board trace impedance defined based
on board material and trace dimensions, a matching
series resistor into the trace from the output of the
OPA653 is used as well as a terminating shunt
resistor at the input of the destination device.
Remember also that the terminating impedance is the
parallel combination of the shunt resistor and the
input impedance of the destination device: this total
effective impedance should be set to match the trace
impedance. If the 6-dB attenuation of a
doubly-terminated transmission line is unacceptable,
a long trace can be series-terminated at the source
end only. Treat the trace as a capacitive load in this
case, and set the series resistor value as shown in
the plot of RISO vs Capacitive Load (Figure 17). This
configuration does not preserve signal integrity as
12
e) Socketing a high-speed part such as the
OPA653 is not recommended. The additional lead
length and pin-to-pin capacitance introduced by the
socket can create an extremely troublesome parasitic
network that can make it almost impossible to
achieve a smooth, stable frequency response. Best
results are obtained by soldering the OPA653 directly
onto the board.
The OPA653 is built using a very high-speed
complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very
small geometry devices. These breakdowns are
reflected in the Absolute Maximum Ratings table. All
device pins are protected with internal ESD protection
diodes to the power supplies, as Figure 25 shows.
+VCC
External
Pin
Internal
Circuitry
-VCC
Figure 25. Internal ESD Protection
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30-mA
continuous current. Where higher currents are
possible (for example, in systems with ±12-V supply
parts driving into the OPA653), current limiting series
resistors should be added into the two inputs. Keep
these resistor values as low as possible because high
values degrade both noise performance and
frequency response.
PowerPAD™ Information
The DRB package option is a PowerPAD™ package
that includes a thermal pad for increased thermal
performance. When using thls package, it is
recommended to distribute the negative supply as a
power plane, and tie the PowerPAD to this supply
wlth multiple vias for proper power dissipation.
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OPA653
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SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
For proper operation, the PowerPAD must be tied to
the most negative supply voltage. It is recommended
to use five evenly-spaced vias under the device as
shown in the EVM layer views (see Figure 27).
For more general data and detailed information about
the PowerPAD package, refer to the PowerPAD™
Thermally Enhanced Package application note
(SLMA002).
space
EVALUATION MODULE
Schematic and PCB Layout
Figure 26 is the OPA653EVM schematic. Layers 1 through 4 of the PCB are shown in Figure 27. It is
recommended to follow the layout of the external components near to the amplifier, ground plane construction,
and power routing as closely as possible.
N/C
2
3
+
7
6
4
+
+
Figure 26. OPA653EVM Schematic
Figure 27. OPA653EVM PCB Layers 1 through 4
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OPA653
SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
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Bill of Materials
Table 1 lists the bill of materials for the OPA653EVM as supplied from TI.
Table 1. OPA653EVM Parts List
14
SMD SIZE
REFERENCE
DESIGNATOR
QUANTITY
D
C1, C2
2
(AVX) TAJ106K035R
0603
C3, C4
2
(AVX) 0603YC104KAT2A
R1, R2
2
(ROHM) MCR03EZPJ000
0603
R4
1
0603
R3, R5
2
(ROHM) MCR03EZPFX49R9
Jack, Banana Receptance, 0.25-in
dia. hole
J4, J5, J8
3
(SPC) 813
7
Connector, Edge, SMA PCB jack
J1, J2, J3
3
(JOHNSON) 142-0701-801
8
Test Point, Black
TP1
1
(KEYSTONE) 5001
9
IC, OPA653
U1
1
(TI) OPA653DRB
10
Standoff, 4-40 Hex, 0.625-in length
4
(KEYSTONE) 1808
11
Screw, Phillips, 4-40, .250 in
4
SHR-0440-016-SN
12
Board, Printed Circuit
1
(TI) EDGE# 6506174
13
Bead, Ferrite, 3 A, 80 Ω
2
(STEWARD) HI1206N800R-00
ITEM
DESCRIPTION
1
Cap, 10.0 μF, Tantalum, 10%, 35 V
2
Cap, 0.1 μF, Ceramic, X7R, 16 V
3
Resistor, 0 Ω
0603
4
Open
5
Resistor, 49.9 Ω, 1/10W, 1%
6
1206
FB1, FB2
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MANUFACTURER
PART NUMBER
Copyright © 2008–2009, Texas Instruments Incorporated
Product Folder Link(s): OPA653
OPA653
www.ti.com
SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
EVALUATION BOARD/KIT IMPORTANT NOTICE
Texas Instruments (TI) provides the enclosed product(s) under the following conditions:
This evaluation board/kit is intended for use for ENGINEERING DEVELOPMENT, DEMONSTRATION, OR EVALUATION PURPOSES
ONLY and is not considered by TI to be a finished end-product fit for general consumer use. Persons handling the product(s) must have
electronics training and observe good engineering practice standards. As such, the goods being provided are not intended to be complete
in terms of required design-, marketing-, and/or manufacturing-related protective considerations, including product safety and environmental
measures typically found in end products that incorporate such semiconductor components or circuit boards. This evaluation board/kit does
not fall within the scope of the European Union directives regarding electromagnetic compatibility, restricted substances (RoHS), recycling
(WEEE), FCC, CE or UL, and therefore may not meet the technical requirements of these directives or other related directives.
Should this evaluation board/kit not meet the specifications indicated in the User’s Guide, the board/kit may be returned within 30 days from
the date of delivery for a full refund. THE FOREGOING WARRANTY IS THE EXCLUSIVE WARRANTY MADE BY SELLER TO BUYER
AND IS IN LIEU OF ALL OTHER WARRANTIES, EXPRESSED, IMPLIED, OR STATUTORY, INCLUDING ANY WARRANTY OF
MERCHANTABILITY OR FITNESS FOR ANY PARTICULAR PURPOSE.
The user assumes all responsibility and liability for proper and safe handling of the goods. Further, the user indemnifies TI from all claims
arising from the handling or use of the goods. Due to the open construction of the product, it is the user’s responsibility to take any and all
appropriate precautions with regard to electrostatic discharge.
EXCEPT TO THE EXTENT OF THE INDEMNITY SET FORTH ABOVE, NEITHER PARTY SHALL BE LIABLE TO THE OTHER FOR ANY
INDIRECT, SPECIAL, INCIDENTAL, OR CONSEQUENTIAL DAMAGES.
TI currently deals with a variety of customers for products, and therefore our arrangement with the user is not exclusive.
TI assumes no liability for applications assistance, customer product design, software performance, or infringement of patents or
services described herein.
Please read the User’s Guide and, specifically, the Warnings and Restrictions notice in the User’s Guide prior to handling the product. This
notice contains important safety information about temperatures and voltages. For additional information on TI’s environmental and/or
safety programs, please contact the TI application engineer or visit www.ti.com/esh.
No license is granted under any patent right or other intellectual property right of TI covering or relating to any machine, process, or
combination in which such TI products or services might be or are used.
FCC Warning
This evaluation board/kit is intended for use for ENGINEERING DEVELOPMENT, DEMONSTRATION, OR EVALUATION PURPOSES
ONLY and is not considered by TI to be a finished end-product fit for general consumer use. It generates, uses, and can radiate radio
frequency energy and has not been tested for compliance with the limits of computing devices pursuant to part 15 of FCC rules, which are
designed to provide reasonable protection against radio frequency interference. Operation of this equipment in other environments may
cause interference with radio communications, in which case the user at his own expense will be required to take whatever measures may
be required to correct this interference.
EVM WARNINGS AND RESTRICTIONS
It is important to operate this EVM within the input voltage range of ±3.5 V to ±6.5 V split-supply and the output voltage range of ±3.5 V to
±6.5 V power-supply voltage; do not exceed ±6.5 V power-supply voltage.
Exceeding the specified input range may cause unexpected operation and/or irreversible damage to the EVM. If there are questions
concerning the input range, please contact a TI field representative prior to connecting the input power.
Applying loads outside of the specified output range may result in unintended operation and/or possible permanent damage to the EVM.
Please consult the EVM User's Guide prior to connecting any load to the EVM output. If there is uncertainty as to the load specification,
please contact a TI field representative.
During normal operation, some circuit components may have case temperatures greater than +85°C. The EVM is designed to operate
properly with certain components above +85°C as long as the input and output ranges are maintained. These components include but are
not limited to linear regulators, switching transistors, pass transistors, and current sense resistors. These types of devices can be identified
using the EVM schematic located in the EVM User's Guide. When placing measurement probes near these devices during operation,
please be aware that these devices may be very warm to the touch.
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2009, Texas Instruments Incorporated
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OPA653
SBOS348A – DECEMBER 2008 – REVISED NOVEMBER 2009
www.ti.com
REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Original (December, 2008) to Revision A
Page
•
Deleted lead temperature specification from Absolute Maximum Ratings table .................................................................. 2
•
Added Input offset voltage specifications for DBV package; noted that existing specifications apply to DRB package ...... 3
•
Changed Input offset voltage specifications for DRB package to ±1.5 mV from ±3 mV ...................................................... 3
•
Changed Average input offset voltage drift typical specification for DRB package from ±15μV/°C to ±10μV/°C ................ 3
•
Added PowerPAD™ Information section ............................................................................................................................ 12
•
Corrected Edge number in Figure 27 ................................................................................................................................. 13
16
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Product Folder Link(s): OPA653
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
OPA653IDBVR
ACTIVE
SOT-23
DBV
5
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
BZW
OPA653IDBVT
ACTIVE
SOT-23
DBV
5
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
BZW
OPA653IDRBT
ACTIVE
SON
DRB
8
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OBEI
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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