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OPA656
SBOS196H – DECEMBER 2001 – REVISED SEPTEMBER 2015
OPA656 Wideband, Unity-Gain Stable, FET-Input Operational Amplifier
1 Features
3 Description
•
•
•
•
•
•
The OPA656 device combines a very wideband,
unity-gain stable, voltage-feedback operational
amplifier with a FET-input stage to offer an ultra high
dynamic-range
amplifier
for
Analog-to-Digital
Converter (ADC) buffering and transimpedance
applications. Extremely low DC errors give good
precision in optical applications.
1
500 MHz Unity-gain Bandwidth
Low Input Bias Current: 2 pA
Low Offset And Drift: ±250 µV, ±2 μV/°C
Low Distortion: 74-dB SFDR at 5 MHz
High-Output Current: 70 mA
Low Input Voltage Noise: 7 nV/√Hz
The high unity-gain stable bandwidth and JFET input
allows exceptional performance in high-speed, lownoise integrators.
2 Applications
•
•
•
•
•
•
Wideband Photodiode Amplifiers
Sample-and-Hold Buffers
CCD Output Buffers
ADC Input Buffers
Wideband Precision Amplifiers
Test and Measurement Front Ends
The high input impedance and low bias current
provided by the FET input is supported by the ultralow 7-nV/√Hz input voltage noise to achieve a very
low integrated noise in wideband photodiode
transimpedance applications.
Broad transimpedance bandwidths are achievable
given the OPA656 device’s high 230-MHz gain
bandwidth product. As shown below, a –3-dB
bandwidth of 1 MHz is provided even for a high 1-MΩ
transimpedance gain from a 47-pF source
capacitance.
Device Information(1)
PART NUMBER
OPA656
PACKAGE
BODY SIZE (NOM)
SOIC (8)
2.90 mm × 1.60 mm
SOT-23 (5)
4.90 mm × 3.91 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
1-MΩ Transimpedance Bandwidth
Transimpedance Gain (dB)
130
Wideband Photodiode Transimpedance Amplifier
1MHz Bandwidth
1 pF
120
499 kΩ
110
499 kΩ
100
90
80
10kHz
OPA656
λ
100kHz
1MHz
VO
5MHz
(47 pF)
Frequency
–Vb
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
OPA656
SBOS196H – DECEMBER 2001 – REVISED SEPTEMBER 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Related Operational Amplifier Products..............
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
3
4
7.1
7.2
7.3
7.4
7.5
7.6
Absolute Maximum Ratings ...................................... 4
ESD Ratings ............................................................ 4
Recommended Operating Conditions....................... 4
Thermal Information ................................................. 4
Electrical Characteristics........................................... 5
Electrical Characteristics: VS = ±5 V: High Grade DC
Specifications ............................................................. 7
7.7 Typical Characteristics: VS = ±5 V ............................ 8
8
Detailed Description ............................................ 13
8.1 Overview ................................................................. 13
8.2 Feature Description................................................. 13
8.3 Device Functional Modes........................................ 13
9
Application and Implementation ........................ 14
9.1 Application Information............................................ 14
9.2 Typical Application ................................................. 19
10 Power Supply Recommendations ..................... 21
11 Layout................................................................... 22
11.1 Layout Guidelines ................................................. 22
11.2 Layout Example .................................................... 23
11.3 Thermal Considerations ........................................ 24
12 Device and Documentation Support ................. 24
12.1
12.2
12.3
12.4
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
24
24
24
24
13 Mechanical, Packaging, and Orderable
Information ........................................................... 24
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision G (November 2008) to Revision H
•
Page
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section. ................................................................................................. 1
Changes from Revision F (March 2006) to Revision G
Page
•
Changed Storage Temperature Range from –40°C to 125°C to –65°C to 12°C ................................................................... 4
•
Deleted in the DC Performance section: Drift from Input Offset Current specifications......................................................... 6
Changes from Revision E (March 2006) to Revision F
•
2
Page
Added Design-In Tools paragraph and table ....................................................................................................................... 23
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OPA656
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SBOS196H – DECEMBER 2001 – REVISED SEPTEMBER 2015
5 Related Operational Amplifier Products
DEVICE
VS (V)
BW (MHz)
SLEW RATE
(V/μs)
VOLTAGE NOISE
(nV/√Hz)
OPA656
±5
230
290
7
Unity-Gain Stable FET-Input
OPA657
±5
1600
700
4.8
Gain of +7 stable FET Input
OPA659
±6
350
2550
8.9
Unity-Gain Stable FET-Input
LMH6629
5
4000
1600
0.69
Gain of +10 stable Bipolar Input
THS4631
±15
210
1000
7
Unity-Gain Stable FET-Input
OPA857
5
4750
220
—
Programmable Gain (5 kΩ / 20 kΩ)
Transimpedance Amplifier
AMPLIFIER DESCRIPTION
6 Pin Configuration and Functions
D Package
8-Pin SOIC Surface-Mount
Top View
DBV Package
5-Pin SOT-23
Top View
1
8
NC
VIN–
2
7
+VS
VIN+
3
6
VOUT
-VS
4
5
NC
VOUT
1
-VS
2
VIN+
3
5
+VS
4
VIN–
4
5
NC
3
2
1
A57
Pin Orientation/Package Marking
Pin Functions
PIN
NAME
SOIC
SOT-23
I/O
DESCRIPTION
1
NC
5
—
—
No Connection
I
Inverting Input
Noninverting Input
8
VIN–
2
4
VIN+
3
3
I
–VS
4
2
POW
VOUT
6
1
O
+VS
7
5
POW
Negative Power Supply
Output of amplifier
Positive Power Supply
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OPA656
SBOS196H – DECEMBER 2001 – REVISED SEPTEMBER 2015
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7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
(1)
MIN
Supply voltage (Total Bipolar Supplies)
Internal power dissipation
MAX
UNIT
±6.5
V
See Thermal Information
Differential input voltage
–VS
+VS
Input voltage
–VS
+VS
Junction temperature, TJ
Storage temperature, Tstg
(1)
–65
150
°C
125
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±500
Machine Model
±200
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VS
Total supply voltage
TA
Ambient temperature
MIN
NOM
MAX
8
10
12
UNIT
V
–40
25
85
°C
7.4 Thermal Information
OPA656
THERMAL METRIC (1)
D (SOIC)
DBV (SOT-23)
8 PINS
5 PINS
UNIT
RθJA
Junction-to-ambient thermal resistance
125
150
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
85.2
140.8
°C/W
RθJB
Junction-to-board thermal resistance
75.9
62.8
°C/W
ψJT
Junction-to-top characterization parameter
26.2
24.4
°C/W
ψJB
Junction-to-board characterization parameter
75.4
61.8
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
—
—
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
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SBOS196H – DECEMBER 2001 – REVISED SEPTEMBER 2015
7.5 Electrical Characteristics
RF = 250 Ω, RL = 100 Ω, and G = 2 V/V, unless otherwise noted. See Figure 1 for AC performance.
PARAMETER
TEST
LEVEL (1)
TEST CONDITIONS
MIN
TYP
MAX
UNIT
AC PERFORMANCE (Figure 29)
G = +1 V/V,
VO = 200 mVPP,
RF = 0 Ω
TJ = 25°C (2)
C
500
MHz
G = +2 V/V,
VO = 200 mVPP
TJ = 25°C (2)
C
200
MHz
G = +5 V/V,
VO = 200 mVPP
TJ = 25°C (2)
C
59
MHz
G = +10 V/V,
VO = 200 mVPP
TJ = 25°C (2)
C
23
MHz
Gain-Bandwidth Product
G > +10 V/V
TJ = 25°C (2)
C
230
MHz
Bandwidth for 0.1-dB flatness
G = +2 V/V,
VO = 200 mVPP
TJ = 25°C (2)
C
30
MHz
Peaking at a Gain of +1
VO < 200 mVPP,
RF = 0 Ω
TJ = 25°C (2)
C
1.5
dB
Large-Signal Bandwidth
G = +2 V/V,
VO = 2 VPP
TJ = 25°C (2)
C
75
MHz
Slew Rate
G = +2 V/V,
1-V Step
TJ = 25°C (2)
C
290
V/µs
Rise-and-Fall Time
0.2-V Step
TJ = 25°C (2)
C
1.5
ns
Settling Time to 0.02%
G = +2 V/V,
VO = 2-V Step
(2)
C
21
ns
Harmonic Distortion
G = +2 V/V, f = 5 MHz, VO = 2 VPP
Small-Signal Bandwidth
TJ = 25°C
RL = 200 Ω
TJ = 25°C (2)
RL > 500 Ω
TJ = 25°C (2)
RL = 200 Ω
TJ = 25°C (2)
RL > 500 Ω
TJ = 25°C (2)
Input Voltage Noise
f > 100 kHz
TJ = 25°C (2)
7
nV/√Hz
Input Current Noise
f > 100 kHz
TJ = 25°C (2)
C
1.3
fA/√Hz
Differential Gain
G = +2 V/V, PAL,
TJ = 25°C (2)
RL = 150 Ω
C
0.02%
Differential Phase
G = +2 V/V, PAL,
TJ = 25°C (2)
RL = 150 Ω
C
0.05
2nd-Harmonic
3rd-Harmonic
DC PERFORMANCE
–71
C
–81
C
Open-Loop Voltage Gain (AOL)
VO = 0 V,
RL = 100 Ω
60
TJ = 0°C to +70°C (4)
A
TJ = –40°C to +85°C (4)
VCM = 0 V
65
59
dB
58
TJ = 25°C (2)
Input Offset Voltage
±0.25
TJ = 0°C to +70°C (4)
A
VCM = 0 V
±2
TJ = 0°C to +70°C (4)
TJ = –40°C to +85°C
A
TJ = 0°C to +70°C (4)
A
TJ = –40°C to +85°C (4)
(1)
(2)
(3)
(4)
±20
±1800
pA
±5000
TJ = 25°C (2)
TJ = 0°C to +70°C (4)
µV/°C
±12
±2
TJ = –40°C to +85°C (4)
VCM = 0 V
±12
±12
(4)
TJ = 25°C (2)
VCM = 0 V
mV
±2.6
TJ = 25°C (2)
Average Offset Voltage Drift
±1.8
±2.2
TJ = –40°C to +85°C (4)
Input Bias Current
dBc
–100
(3)
TJ = 25°C (2)
Input Bias Current
dBc
–74
±2
A
±20
±1800
pA
±5000
Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
Junction temperature = ambient for 25°C min/max specifications.
Current is considered positive out-of-node. VCM is the input common-mode voltage.
Junction temperature = ambient at low temperature limit: junction temperature = ambient +20°C at high temperature limit for over
temperature minimum and maximum specifications.
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SBOS196H – DECEMBER 2001 – REVISED SEPTEMBER 2015
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Electrical Characteristics (continued)
RF = 250 Ω, RL = 100 Ω, and G = 2 V/V, unless otherwise noted. See Figure 1 for AC performance.
PARAMETER
TJ = 25°C
Input Offset Current
TEST
LEVEL (1)
TEST CONDITIONS
VCM = 0 V
MIN
TYP
(2)
±1
TJ = 0°C to +70°C (4)
B
MAX
±10
±900
TJ = –40°C to +85°C (4)
UNIT
pA
±2500
INPUT
TJ = 25°C (2)
Most Positive Input Voltage (5)
2.1
TJ = 0°C to +70°C (4)
A
TJ = –40°C to +85°C (4)
Most Negative Input Voltage
TJ = 0°C to +70°C
–4
(4)
A
TJ = –40°C to +85°C (4)
Most Positive Input Voltage
TJ = 0°C to +70°C (4)
TJ = 0°C to +70°C (4)
TJ = 0°C to +70°C (4)
Differential
TJ = 25°C (2)
Common-Mode
–4.5
–3.9
V
80
A
TJ = –40°C to +85°C (4)
Input Impedance
V
–3.8
TJ = 25°C (2)
VCM = ±0.5 V
3.25
2.5
–4
A
TJ = –40°C to +85°C (4)
Common-Mode Rejection Ratio (CMRR)
V
2.4
TJ = 25°C (2)
Most Negative Input Voltage
–3.9
2.6
A
TJ = –40°C to +85°C (4)
(6)
–4.5
–3.8
TJ = 25°C (2)
(6)
V
2
TJ = 25°C (2)
(5)
2.75
2.05
86
78
dB
76
1012 || 0.7
C
12
C
10
Ω || pF
|| 2.8
Ω || pF
±3.9
±3.7
V
OUTPUT
No Load
TJ = 25°C (2)
TJ = 25°C
Voltage Output Swing
RL = 100 Ω
A
(2)
TJ = 0°C to +70°C (4)
±3.3
A
TJ = –40°C to +85°C (4)
TJ = 0°C to +70°C (4)
50
A
TJ = –40°C to +85°C (4)
TJ = 0°C to +70°C (4)
–50
A
TJ = –40°C to +85°C (4)
Closed-Loop Output Impedance
G = +1 V/V,
f = 0.1 MHz
70
48
mA
46
TJ = 25°C (2)
Current Output, Sinking
V
±3.1
TJ = 25°C (2)
Current Output, Sourcing
±3.5
±3.2
–70
–48
mA
–46
TJ = 25°C (2)
C
0.01
TJ = 25°C (2)
C
±5
Ω
POWER SUPPLY
Specified Operating Voltage
TJ = 25°C (2)
Maximum Operating Voltage Range
TJ = 0°C to +70°C (4)
A
±6
TJ = –40°C to +85°C (4)
TJ = 0°C to +70°C (4)
14
A
TJ = –40°C to +85°C (4)
(5)
(6)
6
mA
16.3
TJ = 25°C (2)
TJ = 0°C to +70°C (4)
16
16.2
TJ = –40°C to +85°C (4)
Minimum Quiescent Current
V
±6
TJ = 25°C (2)
Maximum Quiescent Current
V
±6
11.7
A
11.4
14
dB
11.1
Tested 53-dB CMRR.
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Electrical Characteristics (continued)
RF = 250 Ω, RL = 100 Ω, and G = 2 V/V, unless otherwise noted. See Figure 1 for AC performance.
PARAMETER
TEST
LEVEL (1)
TEST CONDITIONS
+VS = 4.50 V
to
5.50 V
Power-Supply Rejection Ratio (+PSRR)
–VS = 4.50 V
to
–5.50 V
Power-Supply Rejection Ratio (–PSRR)
TJ = 25°C
(2)
TJ = 0°C to +70°C (4)
A
TJ = –40°C to +85°C (4)
MIN
TYP
72
76
MAX
UNIT
70
mA
68
TJ = 25°C (2)
56
TJ = 0°C to +70°C (4)
A
TJ = –40°C to +85°C (4)
62
54
dB
52
TEMPERATURE RANGE
TJ = 25°C (2)
Specified Operating Range: U,N Package
Thermal Resistance, θJA
Junction-toAmbient
U: SO-8
N: SOT23-5
–40
85
°C
TJ = 25°C (2)
125
°C/W
TJ = 25°C (2)
150
°C/W
7.6 Electrical Characteristics: VS = ±5 V: High Grade DC Specifications
RF = 250 Ω, RL = 100 Ω, and G = +2 V/V, unless otherwise noted. (1)
PARAMETER
TJ = 25°C
Input Offset Voltage
TEST LEVEL (2)
TEST CONDITIONS
VCM = 0 V
(3)
TJ = 0°C to +70°C (4)
A
TJ = –40°C to +85°C (4)
MIN
TYP
±0.6
±0.1
VCM = 0 V
mV
±0.9
±2
TJ = 0°C to +70°C (4)
A
±1
TJ = 0°C to +70°C (4)
A
TJ = –40°C to +85°C
Common-Mode
Rejection Ratio
(CMRR)
Power-Supply
Rejection Ratio
(+PSRR)
Power-Supply
Rejection Ratio
(–PSRR)
(1)
(2)
(3)
(4)
±0.5
TJ = 0°C to +70°C (4)
A
TJ = 0°C to +70°C (4)
A
TJ = –40°C to +85°C (4)
dB
78
72
dB
70
TJ = 25°C (3)
TJ = 0°C to +70°C (4)
86
74
A
TJ = –40°C to +85°C (4)
–VS = –4.5 V to –5.5 V
95
84
TJ = 25°C (3)
TJ = 0°C to +70°C (4)
pA
±1250
88
TJ = –40°C to +85°C (4)
+VS = 4.5 V to 5.5 V
±5
±450
(4)
TJ = 25°C (3)
VCM = ±0.5 V
pA
±1250
TJ = 25°C (3)
VCM = 0 V
±5
±450
TJ = –40°C to +85°C (4)
Input Offset Current
µV/°C
±6
TJ = 25°C (3)
VCM = 0 V
±6
±6
TJ = –40°C to +85°C (4)
Input Bias Current
UNIT
±0.85
TJ = 25°C (3)
Input Offset Voltage
Drift
MAX
62
A
68
60
dB
58
All other specifications are the same as the standard-grade.
Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation.
Junction temperature = ambient for 25°C min/max specifications.M
Junction temperature = ambient at low temperature limit: junction temperature = ambient +20°C at high temperature limit for over
temperature min/max specifications.
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OPA656
SBOS196H – DECEMBER 2001 – REVISED SEPTEMBER 2015
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7.7 Typical Characteristics: VS = ±5 V
TA = 25°C, G = +2 V/V, RF = 250 Ω, RL = 100 Ω, unless otherwise noted.
6
3
Normalized Gain (dB)
0
G = +2
–3
–6
G = +5
–9
–12
G = +10
See Figure 1
0
–6
–12
1
10
Frequency (MHz)
100
500
9
100
500
VO = 0.5 Vp-p
G = –1
0
VO = 0.5 Vp-p
–3
VO = 1 Vp-p
0
VO = 2 Vp-p
–3
VO = 1 Vp-p
Gain (dB)
3
Gain (dB)
10
Frequency (MHz)
Figure 2. Inverting Small-Signal Frequency Response
6
–6
VO = 2 Vp-p
–9
–12
See Figure 1
See Figure 2
–6
–15
–9
–18
0.5
1
10
Frequency (MHz)
100
500
0.5
1.6
G = +2
0.6
1.2
0.8
Large-Signal Right Scale
0.2
0.4
Small-Signal Left Scale
0
–0.2
–0.4
–0.4
–0.8
See Figure 1
–0.6
–1.2
–0.8
–1.6
1
10
Frequency (MHz)
100
500
Figure 4. Inverting Large-Signal Frequency Response
Small-Signal Output Voltage (200 mV/div)
0.8
Large-Signal Output Voltage (400 mV/div)
Figure 3. Noninverting Large-Signal Frequency Response
1.6
0.8
G = –1
1.2
0.6
0.4
Large-Signal Right Scale
0
0.8
0.4
0.2
Small-Signal Left Scale
0
–0.4
–0.2
–0.8
–0.4
See Figure 2
–1.2
–0.6
–1.6
–0.8
Time (10 ns/div)
Time (10 ns/div)
Figure 5. Noninverting Pulse Response
8
1
3
VO = 0.2 Vp-p
G = +2
Small-Signal Output Voltage (200 mV/div)
See Figure 2
0.5
Figure 1. Noninverting Small-Signal Frequency Response
0
G = –10
–15
–24
0.5
0.4
G = –5
–9
–21
–18
G = –2
–3
–18
–15
G = –1
VO = 200 mVp-p
RF = 402 Ω
6
Large-Signal Output Voltage (400 mV/div)
Normalized Gain (dB)
3
9
G = +1
RF = 0 Ω
VO = 200mVp-p
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Figure 6. Inverting Pulse Response
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Typical Characteristics: VS = ±5 V (continued)
TA = 25°C, G = +2 V/V, RF = 250 Ω, RL = 100 Ω, unless otherwise noted.
–60
–60
VO = 2 Vp-p
f = 5 MHz
–70
–75
2nd Harmonic
–80
–85
–90
3rd Harmonic
–95
–100
See Figure 1
–105
f = 5 MHz
RL = 200 Ω
–65
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–65
–70
2nd Harmonic
–75
–80
–85
3rd Harmonic
–90
–95
–100
–105
–110
100
0.5
1k
1
Output Voltage Swing (Vp-p)
Resistance (Ω)
5
5 MHz
Figure 7. Harmonic Distortion vs Load Resistance
VO = 2 Vp-p
RL = 200 Ω
–60
f = 1 MHz
RL = 200 Ω
–75
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
Figure 8. Harmonic Distortion vs Output Voltage
–70
–50
2nd Harmonic
–70
–80
–90
3rd Harmonic
–100
2nd Harmonic
–80
–85
–90
–95
3rd Harmonic
–100
See Figure 1
–105
See Figure 1
–110
–110
0.1
1
Frequency (MHz)
10
0.5
20
1
Output Voltage Swing (Vp-p)
5
1 MHz
Figure 9. Harmonic Distortion vs Frequency
Figure 10. Harmonic Distortion vs Output Voltage
–60
VO = 2 Vp-p
f = 5 MHz
RL = 200 Ω
–70
2nd Harmonic
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–60
–80
3rd Harmonic
–90
–100
VO = 2 Vp-p
RF = 604 Ω
F = 5 MHz
RL = 200 Ω
–65
–70
2nd Harmonic
–75
3rd Harmonic
–80
–85
See Figure 2, RG and RM Adjusted
See Figure 1, RG Adjusted
–110
–90
1
10
–1
–10
Gain (V/V)
Gain (V/V)
Figure 11. Harmonic Distortion vs Noninverting Gain
Figure 12. Harmonic Distortion vs Inverting Gain
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Typical Characteristics: VS = ±5 V (continued)
TA = 25°C, G = +2 V/V, RF = 250 Ω, RL = 100 Ω, unless otherwise noted.
–30
10
Input Voltage Noise 7nV/√Hz
Input Current Noise 1.3fA/√Hz
3rd-Order Spurious Level (dBc)
1
100
1k
10k
f (Hz)
100k
1M
–50
–60
250 Ω
10 MHz
–70
5 MHz
–80
–90
2 MHz
–8
–6
–4
–2
0
2
4
Single-Tone Load Power (dBm)
80
Open Loop Gain - Magnitude (dB)
+PSRR
80
70
–PSRR
60
50
40
30
20
10k
100k
1M
10M
45
60
0
40
–45
ÐAOL
20
–90
0
–135
–20
–180
Aol Magnitude
Aol Phase
–225
100
100M
1k
10k
Frequency (Hz)
Figure 15. Common-Mode Rejection Ratio and
Power-Supply Rejection Ratio vs Frequency
10
For Maximally Flat Frequency Response
1
100
1k
100k
1M
Frequency (Hz)
10M
100M
1G
D001
Figure 16. Open-Loop Gain and Phase
Normalized Gain to Capacitive Load (dB)
100
8
20log10(AOL)
–40
1k
10
6
Figure 14. 2-Tone, 3rd-Order Intermodulation Spurious
CMRR
90
RS (Ω)
15 MHz
250 Ω
–10
110
100
10
PO
50 Ω
10M
Figure 13. Input Current and Voltage Noise Density
PSRR (dB)
50 Ω
50 Ω OPA656
–100
10
CMRR (dB)
PI
–40
Open Loop Gain - Phase (°)
in (fA/√Hz)
en (nV/√Hz)
100
9
C L = 10 pF
6
C L = 22 pF
3
C L = 100 pF
0
VI
–3
RS
VO
50 Ω OPA656
CL
–6
1 kΩ
250 Ω
–9
250 Ω
–12
1
10
100
500
Capacitive Load (pF)
Frequency (MHz)
Figure 17. Recommended RS vs Capacitive Load
Figure 18. Frequency Response vs Capacitive Load
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Typical Characteristics: VS = ±5 V (continued)
TA = 25°C, G = +2 V/V, RF = 250 Ω, RL = 100 Ω, unless otherwise noted.
1.0
2.0
Input Bias Current (pA)
Input Offset Voltage (mV)
1.5
0.5
0
–0.5
1.0
0.5
0
–0.5
–1.0
–1.5
–1.0
–2.0
–25
0
25
50
75
100
125
–3
–2
Ambient Temperature (°C)
–1
Figure 19. Typical Input Offset Voltage
Over Temperature
1
2
3
Figure 20. Typical Input Bias Current vs
Common-Mode Input Voltage
1000
150
18
Supply Current
Output Current (25 mA/div)
900
800
Input Bias Current (pA)
0
Common-Mode Input Voltage (V)
700
600
500
400
300
200
Right Scale
125
15
Left Scale
100
Sourcing Current
12
75
9
50
6
Left Scale
Sinking Current
25
3
Supply Current (3 mA/div)
–50
100
0
0
–50
–25
0
25
50
75
100
125
0
–50
Ambient Temperature (°C)
5
3.2
4
2.4
3.2
1.6
1.6
0
0
–1.6
–3.2
–4.8
–6.4
0.8
–0.8
RL = 100 Ω
G = +2
–1.6
–2.4
See Figure 1
–8.0
Input and Output Voltage (V)
Output Voltage (V)
4.0
4.8
Output Voltage
Left Scale
25
50
75
100
125
Figure 22. Supply and Output
Current vs Temperature
Input Voltage (V)
Input Voltage
Right Scale
6.4
0
Ambient Temperature (°C)
Figure 21. Typical Input Bias Current
Over Temperature
8.0
–25
3
2
Input
RL = 100 Ω
RF = 402 Ω
G = –1
1
0
–1
–2
Output
–3
–3.2
–4
–4.0
–5
Time (20 ns/div)
See Figure 2
Time (20 ns/div)
Figure 23. Noninverting Input Overdrive Recovery
Figure 24. Inverting Overdrive Recovery
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Typical Characteristics: VS = ±5 V (continued)
TA = 25°C, G = +2 V/V, RF = 250 Ω, RL = 100 Ω, unless otherwise noted.
10
5
1 W Internal Power
3
RL = 100 Ω
2
VO (V)
Output Impedance (Ω)
4
RL = 50 Ω
1
0
RL = 25 Ω
–1
–2
1
0.1
–3
–4
–5
–100 –80
1 W Internal Power
–60 –40 –20
0
20
40
60
80
0.01
1k
100
10k
100k
1M
10M
100M
IO (mA)
Frequency (Hz)
Figure 25. Output Voltage and Current Limitations
Figure 26. Closed-Loop Output Impedance vs Frequency
CMRR (dB)
110
90
70
50
–5
–4
–3
–2
–1
0
1
2
3
4
5
Common-Mode Input Voltage (V)
Figure 27. Common-Mode Rejection Ratio vs Common-Mode Input Voltage
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8 Detailed Description
8.1 Overview
The OPA656 is high gain-bandwidth, voltage feedback operational amplifier featuring a low noise JFET input
stage. The OPA656 is compensated to be unity gain stable. The OPA656 finds wide use in optical front-end
applications and in test and measurement systems that require high input impedance.
8.2 Feature Description
8.2.1 Input and ESD Protection
The OPA656 is built using a very high speed complementary bipolar process. The internal junction breakdown
voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the
Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power
supplies as shown in Figure 28.
These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection
diodes can typically support 30-mA continuous current. Where higher currents are possible (for example, in
systems with ±12-V supply parts driving into the OPA656), current limiting series resistors should be added into
the two inputs. Keep these resistor values as low as possible because high values degrade both noise
performance and frequency response.
+V CC
External
Pin
Internal
Circuitry
–V CC
Figure 28. Internal ESD Protection
8.3 Device Functional Modes
8.3.1 Split-Supply Operation (±4 V to ±6 V)
To facilitate testing with common lab equipment, the OPA656 may be configured to allow for split-supply
operation. This configuration eases lab testing because the mid-point between the power rails is ground, and
most signal generators, network analyzers, oscilloscopes, spectrum analyzers and other lab equipment reference
their inputs and outputs to ground. Figure 29 and Figure 30 show the OPA656 configured in a simple
noninverting and inverting configuration respectively with ±5-V supplies. The input and output will swing
symmetrically around ground. Due to its ease of use, split-supply operation is preferred in systems where signals
swing around ground, but it requires generation of two supply rails.
8.3.2 Single-Supply Operation (8 V to 12 V)
Many newer systems use single power supply to improve efficiency and reduce the cost of the extra power
supply. The OPA656 is designed for use with split-supply configuration; however, it can be used with a singlesupply with no change in performance, as long as the input and output are biased within the linear operation of
the device. To change the circuit from split supply to single supply, level shift all the voltages by 1/2 the
difference between the power supply rails. An additional advantage of configuring an amplifier for single-supply
operation is that the effects of –PSRR will be minimized because the low supply rail has been grounded.
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
9.1.1 Wideband, Noninverting Operation
The OPA656 provides a unique combination of a broadband, unity gain stable, voltage-feedback amplifier with
the DC precision of a trimmed JFET-input stage. Its very high Gain Bandwidth Product (GBP) of 230 MHz can be
used to either deliver high signal bandwidths for low-gain buffers, or to deliver broadband, low-noise
transimpedance bandwidth to photodiode-detector applications. To achieve the full performance of the OPA656,
careful attention to printed-circuit-board (PCB) layout and component selection is required as discussed in the
remaining sections of this data sheet.
Figure 29 shows the noninverting gain of +2 V/V circuit used as the basis for most of the Typical Characteristics.
Most of the curves were characterized using signal sources with 50-Ω driving impedance, and with measurement
equipment presenting a 50-Ω load impedance. In Figure 29, the 50-Ω shunt resistor at the VI terminal matches
the source impedance of the test generator, while the 50-Ω series resistor at the VO terminal provides a matching
resistor for the measurement equipment load. Generally, data sheet voltage swing specifications are at the
output pin (VO in Figure 29) while output power specifications are at the matched 50-Ω load. The total 100-Ω load
at the output combined with the 500-Ω total feedback network load, presents the OPA656 with an effective output
load of 83 Ω for the circuit of Figure 29.
+5 V
+VS
0.1 μF
6.8 μF
+
50 Ω Source
50 Ω Load
VI
VO
50 Ω
50 Ω
OPA656
RF
250 Ω
RG
250 Ω
+
6.8 μF
0.1 μF
–VS
–5 V
Figure 29. Noninverting G = +2 V/V Specifications and Test Circuit
Voltage-feedback operational amplifiers, unlike current feedback products, can use a wide range of resistor
values to set their gain. To retain a controlled frequency response for the noninverting voltage amplifier of
Figure 29, the parallel combination of RF || RG should always < 200 Ω. In the noninverting configuration, the
parallel combination of RF || RG will form a pole with the parasitic input capacitance at the inverting node of the
OPA656 (including layout parasitics). For best performance, this pole should be at a frequency greater than the
closed loop bandwidth for the OPA656. For this reason, TI recommends a direct short from output to inverting
input for the unity gain follower application.
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Application Information (continued)
9.1.2 Wideband, Inverting Gain Operation
The circuit of Figure 30 shows the inverting gain of –1 V/V test circuit used for most of the inverting Typical
Characteristics. In this case, an additional resistor RM is used to achieve the 50-Ω input impedance required by
the test equipment using in characterization. This input impedance matching is optional in a circuit board
environment where the OPA656 is used as an inverting amplifier at the output of a prior stage.
In this configuration, the output sees the feedback resistor as an additional load in parallel with the 100-Ω load
used for test. It is often useful to increase the RF value to decrease the loading on the output (improving
harmonic distortion) with the constraint that the parallel combination of RF || RG < 200 Ω. For higher inverting
gains with the DC precision provided by the FET input OPA656, consider the higher gain bandwidth product
OPA657.
+5 V
+VS
+
0.1 μF
6.8 μF
50 Ω Load
VO
50 Ω
OPA656
50 Ω Source
RG
402 Ω
RF
402 Ω
VI
RM
57.6 Ω
0.1 μF
+
6.8 μF
–VS
–5 V
Figure 30. Inverting G = –1 V/V Specifications and Test Circuit
Figure 30 also shows the noninverting input tied directly to ground. Often, a bias current canceling resistor to
ground is included here to null out the DC errors caused by input bias current effects. This is only useful when
the input bias currents are matched. For a JFET part like the OPA656, the input bias currents do not match but
are so low to begin with (< 5 pA) that DC errors due to input bias currents are negligible. Hence, no resistor is
recommended at the noninverting inputs for the inverting signal path condition.
9.1.3 Operating Suggestions
9.1.3.1 Setting Resistor Values to Minimize Noise
The OPA656 provides a very low input noise voltage while requiring a low 14-mA quiescent supply current. To
take full advantage of this low input noise, careful attention to the other possible noise contributors is required.
Figure 31 shows the operational amplifier noise analysis model with all the noise terms included. In this model,
all the noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz.
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Application Information (continued)
ENI
*
ERS
EO
OPA656
RS
IBN
*
RF
√4kTRS
*
RG
4 kT
RG
IBI
√4kTRF
4 kT = 1.6E –20J
at 290°K
Figure 31. Operational Amplifier Noise Analysis Model
The total output spot noise voltage can be computed as the square root of the squared contributing terms to the
output noise voltage. This computation is adding all the contributing noise powers at the output by superposition,
then taking the square root to get back to a spot noise voltage. Equation 1 shows the general form for this output
noise voltage using the terms shown in Figure 31.
EO =
(E
2
NI
2
)
(
+ (IBNRS ) + 4kTRS NG2 + IBIRF
2
) + 4kTRFNG
(1)
Dividing this expression by the noise gain (GN = 1+RF/RG) will give the equivalent input referred spot noise
voltage at the noninverting input as shown in Equation 2.
2
EN =
ENI2
+ (IBNRS )
2
4kTRF
æI R ö
+ 4kTRS + ç BI F ÷ +
NG
è NG ø
(2)
Putting high resistor values into Equation 2 can quickly dominate the total equivalent input referred noise. A
source impedance on the noninverting input of 3 kΩ will add a Johnson voltage noise term equal to just that for
the amplifier itself (7 nV/√Hz). While the JFET input of the OPA656 is ideal for high source impedance
applications, both the overall bandwidth and noise will be limited by higher source impedances in the
noninverting configuration of Figure 29.
9.1.3.2 Frequency Response Control
Voltage-feedback op amps like the OPA656 exhibit decreasing signal bandwidth as the signal gain is increased.
In theory, this relationship is described by the GBP shown in the Electrical Characteristics. Ideally, dividing GBP
by the noninverting signal gain (also called the Noise Gain, or NG) will predict the closed-loop bandwidth. In
practice, this only holds true when the phase margin approaches 90°, as it does in high-gain configurations. At
low gains (increased feedback factors), most high-speed amplifiers will exhibit a more complex response with
lower phase margin. The OPA656 is compensated to give a maximally flat 2nd-order Butterworth closed loop
response at a noninverting gain of +2 V/V (Figure 29). This results in a typical gain of +2 V/V bandwidth of 200
MHz, far exceeding that predicted by dividing the 230-MHz GBP by 2. Increasing the gain will cause the phase
margin to approach 90° and the bandwidth to more closely approach the predicted value of (GBP/NG). At a gain
of +10 V/V the OPA656 will show the 23-MHz bandwidth predicted using the simple formula and the typical GBP
of 230 MHz.
Unity-gain stable operational amplifiers like the OPA656 can also be bandlimited using a capacitor across the
feedback resistor. For the noninverting configuration of Figure 29, a capacitor across the feedback resistor will
decrease the gain with frequency down to a gain of +1 V/V. For instance, to bandlimit the gain of +2 V/V design
to 20 MHz, a 32-pF capacitor can be placed in parallel with the 250-Ω feedback resistor. This will, however, only
decrease the gain from +2 V/V to +1 V/V. Using a feedback capacitor to limit the signal bandwidth is more
effective in the inverting configuration of Figure 30. Adding that same capacitor to the feedback of Figure 30 will
set a pole in the signal frequency response at 20 MHz, but in this case it will continue to attenuate the signal gain
to below 1. However, the output noise contribution due the input voltage noise of the OPA656 will still only be
reduced to a gain of +1 V/V with the addition of the feedback capacitor.
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Application Information (continued)
9.1.3.3 Driving Capacitive Loads
One of the most demanding and yet very common load conditions for an operational amplifier is capacitive
loading. Often, the capacitive load is the input of an ADC—including additional external capacitance which may
be recommended to improve ADC linearity. A high-speed, high open-loop gain amplifier like the OPA656 can be
very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed
directly on the output pin. When the amplifier’s open loop output resistance is considered, this capacitive load
introduces an additional pole in the signal path that can decrease the phase margin. Several external solutions to
this problem have been suggested. When the primary considerations are frequency response flatness, pulse
response fidelity and/or distortion, the simplest and most effective solution is to isolate the capacitive load from
the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load.
This does not eliminate the pole from the loop response, but rather shifts it and adds a zero at a higher
frequency. The additional zero acts to cancel the phase lag from the capacitive load pole, thus increasing the
phase margin and improving stability.
The Typical Characteristics: VS = ±5 V show the recommended RS versus Capacitive Load and the resulting
frequency response at the load. In this case, a design target of a maximally flat frequency response was used.
Lower values of RS may be used if some peaking can be tolerated. Also, operating at higher gains (than the +2
used in Typical Characteristics: VS = ±5 V) will require lower values of RS for a minimally peaked frequency
response. Parasitic capacitive loads greater than 2 pF can begin to degrade the performance of the OPA656.
Long PC board traces, unmatched cables, and connections to multiple devices can easily cause this value to be
exceeded. Always consider this effect carefully, and add the recommended series resistor as close as possible to
the OPA656 output pin (see Layout section).
9.1.3.4 Distortion Performance
The OPA656 is capable of delivering a low distortion signal at high frequencies over a wide range of gains. The
distortion plots in the Typical Characteristics: VS = ±5 V show the typical distortion under a wide variety of
conditions.
Generally, until the fundamental signal reaches very high frequencies or powers, the 2nd-harmonic will dominate
the distortion with negligible 3rd-harmonic component. Focusing then on the 2nd-harmonic, increasing the load
impedance improves distortion directly. Remember that the total load includes the feedback network—in the
noninverting configuration this is sum of RF + RG, while in the inverting configuration this is just RF (see
Figure 29). Increasing output voltage swing increases harmonic distortion directly. A 6-dB increase in output
swing will generally increase the 2nd-harmonic 12 dB and the 3rd-harmonic 18 dB. Increasing the signal gain will
also increase the 2nd-harmonic distortion. Again a 6-dB increase in gain will increase the 2nd- and 3rd-harmonic
by about 6 dB even with a constant output power and frequency. And finally, the distortion increases as the
fundamental frequency increases due to the rolloff in the loop gain with frequency. Conversely, the distortion will
improve going to lower frequencies down to the dominant open loop pole at approximately 100 kHz. Starting from
the –70 dBc 2nd-harmonic for a 5 MHz, 2VPP fundamental into a 200-Ω load at G = +2 V/V (from the Typical
Characteristics: VS = ±5 V), the 2nd-harmonic distortion for frequencies lower than 100 kHz will be < –105 dBc.
The OPA656 has an extremely low 3rd-order harmonic distortion. This also shows up in the 2-tone 3rd-order
intermodulation spurious (IM3) response curves. The 3rd-order spurious levels are extremely low < –80 dBc) at
low output power levels. The output stage continues to hold them low even as the fundamental power reaches
higher levels. As the Typical Characteristics: VS = ±5 V show, the spurious intermodulation powers do not
increase as predicted by a traditional intercept model. As the fundamental power level increases, the dynamic
range does not decrease significantly. For 2 tones centered at 10 MHz, with 4 dBm/tone into a matched 50-Ω
load (that is, 1VPP for each tone at the load, which requires 4 VPP for the overall 2-tone envelope at the output
pin), the Typical Characteristics: VS = ±5 V show a 78-dBc difference between the test tone and the 3rd-order
intermodulation spurious levels. This exceptional performance improves further when operating at lower
frequencies and/or higher load impedances.
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Application Information (continued)
9.1.3.5 DC Accuracy and Offset Control
The OPA656 can provide excellent DC accuracy due to its high open-loop gain, high common-mode rejection,
high power-supply rejection, and its trimmed input offset voltage (and drift) along with the negligible errors
introduced by the low input bias current. For the best DC precision, a highgrade version (OPA656UB or
OPA656NB) screens the key DC parameters to an even tighter limits. Both standard- and high-grade versions
take advantage of a new final test technique to 100% test input offset voltage drift over temperature. This
discussion will use the high-grade typical and minimum and maximum electrical characteristics for illustration;
however, an identical analysis applies to the standard-grade version.
The total output DC offset voltage in any configuration and temperature will be the combination of a number of
possible error terms. In a JFET part like the OPA656, the input bias current terms are typically quite low but are
unmatched. Using bias current cancellation techniques, more typical in bipolar input amplifiers, does not improve
output DC offset errors. Errors due to the input bias current will only become dominant at elevated temperatures.
The OPA656 shows the typical 2× increase in every 10°C common to JFET-input stage amplifiers. Using the 5pA maximum tested value at 25°C, and a 20°C internal self heating (see Thermal Considerations), the maximum
input bias current at 85°C ambient will be 5 pA × 2(105 – 25)/10 = 1280 pA. For noninverting configurations, this
term only begins to be a significant term versus the input offset voltage for source impedances > 750 kΩ. This
would also be the feedback-resistor value for transimpedance applications (see Figure 32) where the output DC
error due to inverting input bias current is on the order of that contributed by the input offset voltage. In general,
except for these extremely high impedance values, the output DC errors due to the input bias current may be
neglected.
After the input offset voltage itself, the most significant term contributing to output offset voltage is the PSRR for
the negative supply. This term is modeled as an input offset voltage shift due to changes in the negative powersupply voltage (and similarly for the +PSRR). The high-grade test limit for –PSRR is 62 dB. This translates into
1.59-mV/V input offset voltage shift = 10(–62/20). In the worst case, a ±0.38 V (±7.6%) shift in the negative supply
voltage will produce a ±0.6 mV apparent input offset voltage shift. Because this is comparable to the tested limit
of ±0.6 mV input offset voltage, a careful control of the negative supply voltage is required. The +PSRR is tested
to a minimum value of 74 dB. This translates into 10(–74/20) = 0.2 mV/V sensitivity for the input offset voltage to
positive power supply changes.
As an example, compute the worst-case output DC error for the transimpedance circuit of Figure 32 at 25°C and
then the shift over the 0°C to 70°C range given the following assumptions.
Negative Power Supply = –5 V ±0.2V with a ±5mV/°C worst-case shift
Positive Power Supply = +5 V ±0.2V with a ±5mV/°C worst-case shift
Initial 25°C Output DC Error Band
= ±0.3 mV (due to the –PSRR = 1.59 mV/V × ±0.2 V)
±0.04 mV (due to the +PSRR = 0.2 mV/V × ±0.2 V)
±0.6 mV Input Offset Voltage
Total = ±0.94 mV
This would be the worst-case error band in volume production at 25°C acceptance testing given the conditions
stated.
Over the temperature range of 0°C to 70°C, we can expect the following worst-case shifting from initial value. A
20°C internal junction self heating is assumed here.
±0.36 mV (OPA656 high-grade input offset drift) = ±6 μV/°C × (70°C + 20°C – 25°C))
±0.23 mV (–PSRR of 60 dB with 5 mV × (70°C – 25°C) supply shift)
±0.06 mV (+PSRR of 72 dB with 5 mV × (70°C – 25°C) supply shift)
Total = ±0.65 mV
This would be the worst-case shift from initial offset over a 0°C to 70°C ambient for the conditions stated. Typical
initial output DC error bands and shifts over temperature will be much lower than these worst-case estimates.
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Application Information (continued)
In the transimpedance configuration, the CMRR errors can be neglected because the input common mode
voltage is held at ground. For noninverting gain configurations (see Figure 29), the CMRR term must be
considered but will typically be far lower than the input offset voltage term. With a tested minimum of 80 dB (100
μV/V), the added apparent DC error will be no more than ±0.2 mV for a ±2-V input swing to the circuit of
Figure 29.
9.2 Typical Application
The high GBP and low input voltage and current noise for the OPA656 make it an ideal wideband
transimpedance amplifier for moderate to high transimpedance gains.
VB
Supply Decoupling not
shown
+5 V
RS
50
+
OPA656
CD
21pF
CPCB
0.3 pF
Oscilloscope
with 50
Inputs
-5 V
RF
100 k
CF + CPCB
0.52 pF
Figure 32. Wideband, High-Sensitivity, Transimpedance Amplifier
9.2.1 Design Requirements
Design a high-bandwidth, high-gain transimpedance amplifier with the design requirements shown in Table 1.
Table 1. Design Requirements
TARGET BANDWIDTH (MHz)
TRANSIMPEDANCE GAIN (KΩ)
PHOTODIODE CAPACITANCE (pF)
4
100
21
9.2.2 Detailed Design Procedure
Designs that require high bandwidth from a large area detector with relatively high transimpedance gain benefit
from the low input voltage noise of the OPA656. This input voltage noise is peaked up over frequency by the
diode source capacitance, and can, in many cases, become the limiting factor to input sensitivity. The key
elements to the design are the expected diode capacitance (CD) with the reverse bias voltage (VB) applied the
desired transimpedance gain, RF, and the GBP for the OPA656 (230 MHz). Figure 32 shows a transimpedance
circuit with the parameters as described in Table 1. With these three variables set (and including the parasitic
input capacitance for the OPA656 and the PCB added to CD), the feedback capacitor value (CF) may be set to
control the frequency response. To achieve a maximally-flat second-order Butterworth frequency response, the
feedback pole should be set to:
1
=
2pRFCF
GBP
4pRFCD
(3)
The input capacitance of the amplifier is the sum of its common-mode and differential capacitance (0.7+2.8) pF.
The parasitic capacitance from the photo-diode package and the PCB is approximately 0.3 pF. This results in a
total input capacitance, CD {D should be a subscript} = 24.8 pF. From Equation 3, the feedback pole should be
set at 2.7 MHz. Setting the pole at 2.7 MHz requires a total feedback capacitance of 0.585 pF
The approximate –3-dB bandwidth of the transimpedance amplifier circuit is given by:
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f-3dB = GBP / (2pRFCD ) Hz
(4)
Equation 4 estimates a closed-loop bandwidth of 3.84 MHz. The total feedback capacitance for the circuit used in
the measurement is estimated to be 0.520 pF. The total feedback capacitance includes the physical 0.4-pF
feedback capacitor in parallel with 120-fF of parasitic capacitance due to the feedback resistor and PCB trace.
The parasitic capacitance from the PCB trace can be minimized by removing the ground and power planes in the
feedback path. A TINA SPICE simulation of the circuit in Figure 32 resulted in a closed-loop bandwidth of 4.2
MHz.
Figure 33 shows the measured output noise of the system. The low-frequency output noise of 45 nV/√Hz gets
input-referred to 0.45 pA/√Hz. The transimpedance gain resistor is the dominant noise source with the
operational amplifier itself contributing a negligible amount, reflecting one of the main benefits in using a JFET
input amplifier in a high-gain transimpedance application. If the total output noise of the TIA is bandlimited to a
frequency less than the feedback pole frequency, a very simple expression for the equivalent output noise
voltage can be derived as shown in Equation 5.
where
•
•
•
•
•
•
VOUTN = Equivalent output noise when band limited to F < 1 / (2 ΩRfCf)
IN = Input current noise for the operational amplifier inverting input
EN = Input voltage noise for the operational amplifier
CD = Diode capacitance including operational amplifier and PCB parasitic capacitance
F = Band-limiting frequency in Hz (usually a postfilter before further signal processing)
4 kT = 1.6 e – 20 J at T = 290°K
(5)
Figure 34 shows the measured pulse response to a 2-μA input current pulse. The output voltage measured on
the scope is 0.1 V because of the 50-Ω termination to the scope. The measured rise/fall time and overshoot
match very well with simulation. Based on the measured rise and fall time of 85 ns, the approximate bandwidth of
the circuit = 0.35/85 ns = 4.1 MHz,which also matches the theoretical value calculated using Equation 4.
9.2.3 Application Curves
20m
1000
-20m
Output Voltage (V)
Output Noise (nV/vHz)
0m
100
-40m
-60m
-80m
-100m
-120m
10
100
-140m
1k
10k
100k
1M
Frequency (Hz)
10M
100M
0
10
20
D002
30
Time (usec)
40
50
60
D003
Rise Time = 84.9 ns
Fall Time = 85.4 ns
Figure 33. Measured Total TIA Noise
20
Figure 34. Transient Pulse Response to 2-µA Input Current
Pulse
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10 Power Supply Recommendations
The OPA656 is intended for operation on ±5-V supplies. Single-supply operation is allowed with minimal change
from the stated specifications and performance from a single supply of 8 V to 12 V maximum. The limit to lower
supply voltage operation is the useable input voltage range for the JFET-input stage. Operating from a single
supply of 12 V can have numerous advantages. With the negative supply at ground, the DC errors due to the
–PSRR term can be minimized. Typically, AC performance improves slightly at 12-V operation with minimal
increase in supply current.
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11 Layout
11.1 Layout Guidelines
Achieving optimum performance with a high-frequency amplifier like the OPA656 requires careful attention to
board layout parasitics and external component types. Recommendations that will optimize performance include.
1. Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the
output and inverting input pins can cause instability—on the noninverting input, it can react with the source
impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the
signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground
and power planes should be unbroken elsewhere on the board.
2. Minimize the distance (< 0.25") from the power-supply pins to high-frequency 0.1-µF decoupling capacitors.
At the device pins, the ground and power plane layout should not be in close proximity to the signal I/O pins.
Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling
capacitors. The power-supply connections should always be decoupled with these capacitors. Larger (2.2-μF
to 6.8-μF) decoupling capacitors, effective at lower frequency, should also be used on the supply pins. These
may be placed somewhat farther from the device and may be shared among several devices in the same
area of the PC board.
3. Careful selection and placement of external components will preserve the high frequency
performance of the OPA656. Resistors should be a very low reactance type. Surface-mount resistors work
best and allow a tighter overall layout. Metal film and carbon composition axially leaded resistors can also
provide good high frequency performance. Again, keep their leads and PCB trace length as short as
possible. Never use wirewound type resistors in a high frequency application. Because the output pin and
inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series
output resistor, if any, as close as possible to the output pin. Other network components, such as
noninverting input termination resistors, should also be placed close to the package. Where double side
component mounting is allowed, place the feedback resistor directly under the package on the other side of
the board between the output and inverting input pins. Even with a low parasitic capacitance shunting the
external resistors, excessively high resistor values can create significant time constants that can degrade
performance. Good axial metal film or surface mount resistors have approximately 0.2 pF in shunt with the
resistor. For resistor values > 1.5 kΩ, this parasitic capacitance can add a pole and/or zero below 500 MHz
that can effect circuit operation. Keep resistor values as low as possible consistent with load driving
considerations. It has been suggested here that a good starting point for design would be to keep RF || RG <
250 Ω for voltage amplifier applications. Doing this will automatically keep the resistor noise terms low, and
minimize the effect of their parasitic capacitance. Transimpedance applications (see ) can use whatever
feedback resistor is required by the application as long as the feedback compensation capacitor is set
considering all parasitic capacitance terms on the inverting node.
4. Connections to other wideband devices on the board may be made with short direct traces or through
onboard transmission lines. For short connections, consider the trace and the input to the next device as a
lumped capacitive load. Relatively wide traces (50 mils to 100 mils) should be used, preferably with ground
and power planes opened up around them. Estimate the total capacitive load and set RS from the plot of
Recommended RS vs Capacitive Load. Low parasitic capacitive loads (< 5 pF) may not need an RS because
the OPA656 is nominally compensated to operate with a 2-pF parasitic load. Higher parasitic capacitive
loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin) If a
long trace is required, and the 6-dB signal loss intrinsic to a doubly-terminated transmission line is
acceptable, implement a matched impedance transmission line using microstrip or stripline techniques
(consult an ECL design handbook for microstrip and stripline layout techniques). A 50-Ω environment is
normally not necessary onboard, and in fact a higher impedance environment will improve distortion as
shown in the distortion versus load plots. With a characteristic board trace impedance defined based on
board material and trace dimensions, a matching series resistor into the trace from the output of the OPA656
is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the
terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the
destination device— this total effective impedance should be set to match the trace impedance. If the 6-dB
attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated
at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as
shown in the plot of Recommended RS vs Capacitive Load. This will not preserve signal integrity as well as a
doubly-terminated line. If the input impedance of the destination device is low, there will be some signal
attenuation due to the voltage divider formed by the series output into the terminating impedance.
22
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Layout Guidelines (continued)
5. Socketing a high speed part like the OPA656 is not recommended. The additional lead length and pinto-pin capacitance introduced by the socket can create an extremely troublesome parasitic network which
can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by
soldering the OPA656 onto the board.
11.1.1 Demonstration Fixtures
Two printed-circuit-boards (PCBs) are available to assist in the initial evaluation of circuit performance using the
OPA656 device in its two package options. Both of these are offered free of charge as unpopulated PCBs,
delivered with a user's guide. The summary information for these fixtures is shown in Table 2.
Table 2. Demonstration Fixtures by Package
PRODUCT
PACKAGE
ORDERING NUMBER
LITERATURE NUMBER
OPA656U
SO-8
DEM-OPA-SO-1A
SBOU009
OPA656N
SOT23-5
DEM-OPA-SOT-1A
SBOU010
The demonstration fixtures can be requested at the Texas Instruments website (www.ti.com) through the
OPA656 product folder.
11.2 Layout Example
Ground Plane
removed under VIN-
Feedback element
trace length minimized
Bypass Cap.
Bypass Cap.
Figure 35. Layout Recommendation
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11.3 Thermal Considerations
The OPA656 will not require heatsinking or airflow in most applications. Maximum allowed junction temperature
will set the maximum allowed internal power dissipation as described below. In no case should the maximum
junction temperature be allowed to exceed 150°C.
Operating junction temperature (TJ) is given by TA + PD × θJA. The total internal power dissipation (PD) is the sum
of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power.
Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL
will depend on the required output signal and load but would, for a grounded resistive load, be at a maximum
when the output is fixed at a voltage equal to 1/2 of either supply voltage (for equal bipolar supplies). Under this
condition PDL = VS 2/(4 × RL) where RL includes feedback network loading.
Note that it is the power in the output stage and not into the load that determines internal power dissipation.
As a worst-case example, compute the maximum TJ using an OPA656N (SOT23-5 package) in the circuit of
Figure 29 operating at the maximum specified ambient temperature of +85°C and driving a grounded 100-Ω load.
PD = 10 V × 16.1 mA + 52 /(4 × (100 Ω || 800 Ω)) = 231 mW
Maximum TJ = 85°C + (0.23 W × 150°C/W) = 120°C.
All actual applications will be operating at lower internal power and junction temperature.
12 Device and Documentation Support
12.1 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.2 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.3 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.4 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
24
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PACKAGE OPTION ADDENDUM
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24-Aug-2018
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
OPA656N/250
ACTIVE
SOT-23
DBV
5
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
B56
OPA656N/250G4
ACTIVE
SOT-23
DBV
5
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
B56
OPA656NB/250
ACTIVE
SOT-23
DBV
5
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
B56
OPA656U
ACTIVE
SOIC
D
8
75
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
OPA
656U
OPA656U/2K5
ACTIVE
SOIC
D
8
2500
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
OPA
656U
OPA656UB
ACTIVE
SOIC
D
8
75
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
OPA
656U
B
OPA656UB/2K5
ACTIVE
SOIC
D
8
2500
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
OPA
656U
B
OPA656UG4
ACTIVE
SOIC
D
8
75
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-3-260C-168 HR
-40 to 85
OPA
656U
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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