OPA683
OPA
683
OPA68
3
SBOS221E – NOVEMBER 2001 – REVISED JULY 2008
www.ti.com
Very Low-Power, Current Feedback
OPERATIONAL AMPLIFIER With Disable
FEATURES
APPLICATIONS
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REDUCED BANDWIDTH CHANGE VERSUS GAIN
150MHz BANDWIDTH G = +2
> 90MHz BANDWIDTH TO GAIN > +10
LOW DISTORTION: < –69dBc at 5MHz
HIGH OUTPUT CURRENT: 110mA
SINGLE +5V TO +12V SUPPLY OPERATION
DUAL ±2.5V TO ±6V SUPPLY OPERATION
LOW SUPPLY CURRENT: 0.94mA
LOW SHUTDOWN CURRENT: 100µA
DESCRIPTION
The OPA683 provides a new level of performance in very lowpower, wideband, current feedback amplifiers. This CFBplus amplifier is among the first to use an internally closed-loop input buffer
stage that enhances performance significantly over earlier lowpower CFB amplifiers. While retaining the benefits of very low
power operation, this new architecture provides many of the
advantages of a more ideal CFB amplifier. The closed-loop input
stage buffer gives a very low and linearized impedance path at the
inverting input to sense the feedback error current. This improved
inverting input impedance gives exceptional bandwidth retention to
much higher gains and improved harmonic distortion over earlier
solutions limited by inverting input linearity. Beyond simple high
gain applications, the OPA683 CFBplus amplifier can allow the gain
setting element to be set with considerable freedom from amplifier
bandwidth interaction. This allows frequency response peaking
elements to be added, multiple input inverting summing circuits to
LOW POWER BROADCAST VIDEO DRIVERS
EQUALIZING FILTERS
SAW FILTER HIGH GAIN POST AMPLIFIERS
SHORT LOOP ADSL CO DRIVERS
MULTICHANNEL SUMMING AMPLIFIERS
PROFESSIONAL CAMERAS
ADC INPUT DRIVERS
have greater bandwidth, and low-power line drivers to meet the
demanding requirements of studio cameras and broadcast video.
The output capability for the OPA683 also sets a new mark in
performance for very low-power current feedback amplifiers. Delivering a full ±4VPP swing on ±5V supplies, the OPA683 also has
the output current to support this swing into a 100Ω load. This
minimal output headroom requirement is complemented by a
similar 1.2V input stage headroom giving exceptional capability for
single +5V operation.
The OPA683’s low 0.94mA supply current is precisely trimmed at
25°C. This trim, along with low shift over temperature and supply
voltage, gives a very robust design over a wide range of operating
conditions. System power may be further reduced by using the
optional disable control pin. Leaving this disable pin open, or holding
it HIGH, gives normal operation. If pulled LOW, the OPA683 supply
current drops to less than 100µA while the I/O pins go to a high
impedance state.
V+
OPA683 BANDWIDTH vs GAIN
9
G=2
G=5
+
Normalized Gain (dB)
6
VO
Z(S) IERR
V–
IERR
RF
3
0
–3
–6
–9
G = 10
–12
G = 20
G = 50
–15
–18
RF = 1.2kΩ
–21
RG
Low-Power
1
Amplifier
G = 100
10
Frequency (MHz)
100
200
U.S. Patent No. 6,724,260
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright © 2001-2008, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
www.ti.com
ELECTROSTATIC
DISCHARGE SENSITIVITY
ABSOLUTE MAXIMUM RATINGS(1)
Power Supply ............................................................................... ±6.5VDC
Internal Power Dissipation ................................. See Thermal Information
Differential Input Voltage .................................................................. ±1.2V
Input Voltage Range ............................................................................ ±VS
Storage Temperature Range: ID, IDBV ......................... –65°C to +125°C
Lead Temperature (soldering, 10s) .............................................. +300°C
Junction Temperature (TJ ) ........................................................... +175°C
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes
could cause the device not to meet its published specifications.
NOTE: (1) Stresses above those listed under “Absolute Maximum Ratings”
may cause permanent damage to the device. Exposure to absolute maximum
conditions for extended periods may affect device reliability.
OPA683 RELATED PRODUCTS
SINGLES
DUALS
TRIPLES
QUADS
OPA684
OPA691
OPA685
OPA2683
OPA2691
—
OPA3684
OPA3691
—
OPA4684
—
—
FEATURES
Low-Power CFBplus
High Slew Rate CFB
> 500MHz CFB
PACKAGE/ORDERING INFORMATION(1)
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
SO-8
D
–40°C to +85°C
OPA683D
"
"
"
"
OPA683ID
OPA683IDR
Rails,100
Tape and Reel, 2500
SOT23-6
DBV
–40°C to +85°C
A83
OPA683IDBVT
Tape and Reel, 250
"
"
"
"
OPA683IDBVR
Tape and Reel, 3000
OPA683
"
OPA683
"
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this document, or see the TI website
at www.ti.com.
PIN CONFIGURATION
Top View
SO-8
NC
1
8
DIS
Inverting Input
2
7
+VS
Noninverting Input
3
6
Output
–VS
4
5
NC
Top View
SOT23-6
Output
1
6
+VS
–VS
2
5
DIS
Noninverting Input
3
4
Inverting Input
6
5
4
A83
NC = No Connection
1
2
3
Pin Orientation/Package Marking
OPA683
2
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SBOS221E
ELECTRICAL CHARACTERISTICS: VS = ±5V
Boldface limits are tested at +25°C.
RF = 1.2kΩ, RL = 1kΩ, and G = +2 (see Figure 1 for AC performance only), unless otherwise noted.
OPA683ID, IDBV
TYP
PARAMETER
AC PERFORMANCE (See Figure 1)
Small-Signal Bandwidth (VO = 0.5VPP)
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Noninverting Input Current Noise
Inverting Input Current Noise
Differential Gain
Differential Phase
DC PERFORMANCE(4)
Open-Loop Transimpedance Gain (ZOL)
Input Offset Voltage
Average Offset Voltage Drift
Noninverting Input Bias Current
Average Noninverting Input Bias Current Drift
Inverting Input Bias Current
Average Inverting Input Bias Current Drift
INPUT
Common-Mode Input Range(5) (CMIR)
Common-Mode Rejection Ratio (CMRR)
Noninverting Input Impedance
Inverting Input Resistance (RI)
OUTPUT
Voltage Output Swing
Current Output, Sourcing
Current Output, Sinking
Closed-Loop Output Impedance
DISABLE (Disabled LOW)
Power-Down Supply Current (+VS)
Disable Time
Enable Time
Off Isolation
Output Capacitance in Disable
Turn On Glitch
Turn Off Glitch
Enable Voltage
Disable Voltage
Control Pin Input Bias Current (DIS)
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage Range
Minimum Operating Voltage Range
Max Quiescent Current
Min Quiescent Current
Power-Supply Rejection Ratio (–PSRR)
TEMPERATURE RANGE
Specification: D, DBV
Thermal Resistance, θJA
D
SO-8
DBV SOT-23-6
+25°C(1)
0°C to
70°C(2)
–40°C to
+85°C(2)
124
121
117
15
6.5
14
7.7
14
8.0
450
345
450
338
430
336
–63
–65
–67
–74
4.4
5.1
11.6
0.06
0.03
–54
–55
–62
–67
5.0
5.8
11.9
–54
–55
–62
–66
5.5
6.4
12.3
±1.5
700
300
±3.5
±2.0
±4.0
±3.0
±10
±3.75
CONDITIONS
+25°C
G = +1, RF = 1.2kΩ
G = +2, RF = 1.2kΩ
G = +5, RF = 1.2kΩ
G = +10, RF = 1.2kΩ
G = +20, RF = 1.2kΩ
G = +2, VO = 0.5VPP, RF = 1.2kΩ
RF = 1.2kΩ, VO = 0.5VPP
G = +2, VO = 4VPP
G = –1, VO = 4V Step (see Figure 2)
G = +2,VO = 4V Step
G = +2, VO = 0.5V Step
G = +2, VO = 4VStep
G = +2, f = 5MHz, VO = 2VPP
RL = 100Ω
RL ≥ 1kΩ
RL = 100Ω
RL ≥ 1kΩ
f > 1MHz
f > 1MHz
f > 1MHz
G = +2, NTSC, VO = 1.4VP, RL = 150Ω
G = +2, NTSC, VO = 1.4VP, RL = 150Ω
200
150
121
94
72
37
1.8
63
540
400
4.6
7.8
VO = 0V, RL = 1kΩ
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
Open-Loop, DC
1kΩ Load
VO = 0
VO = 0
G = +2, f = 100kHz
VDIS = 0
VIN = +1, See Figure 1
VIN = +1, See Figure 1
G = +2, 5MHz
G = +2, RL = 150Ω, VIN = 0
G = +2, RL = 150Ω, VIN = 0
VDIS = 0V
UNITS
MIN/ TEST
MAX LEVEL(3)
MHz
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
V/µs
ns
ns
typ
min
typ
typ
typ
min
max
typ
min
min
typ
typ
C
B
C
B
C
B
B
C
B
B
C
C
–54
–55
–62
–66
5.8
6.7
12.4
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
pA/√Hz
%
deg
max
max
max
max
max
max
max
typ
typ
B
B
B
B
B
B
B
C
C
270
±4.1
±12
±4.6
±15
±11
±20
250
±4.3
±12
±4.8
±15
±11.5
±20
kΩ
mV
µV/°C
µA
nA/°C
µA
nA°/C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
±3.65
53
±3.65
52
±3.60
52
V
dB
kΩ || pF
Ω
min
min
typ
typ
A
A
C
C
±4.0
130
–100
±4.0
125
–95
±3.9
120
–90
V
mA
mA
Ω
min
min
min
typ
A
A
A
C
–150
–170
–180
3.5
1.7
120
3.6
1.6
130
3.7
1.5
135
µA
ms
ns
dB
pF
mV
mV
V
V
µA
typ
typ
typ
typ
typ
typ
typ
min
max
max
C
C
C
C
C
C
C
A
A
A
±6
±6
±6
1.03
0.85
55
1.04
0.80
54
1.05
0.77
54
V
V
V
mA
mA
dB
typ
max
min
max
min
typ
C
A
C
A
A
A
–40 to +85
°C
typ
C
125
150
°C/W
°C/W
typ
typ
C
C
60
50 2
4.5
±4.1
150
–110
0.007
–100
60
40
70
1.7
±70
±20
3.4
1.8
80
±5
VS = ±5V
VS = ±5V
Input Referred
MIN/MAX OVER TEMPERATURE
±1.4
0.94
0.94
62
Junction-to-Ambient
NOTES: (1) Junction temperature = ambient for 25°C tested specifications. (2) Junction temperature = ambient at low temperature limit: junction temperature = ambient
+2°C at high temperature limit for over temperature tested specifications. (3) Test levels: (A) 100% tested at 25°C. Over temperature limits by characterization and
simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out-of-node. VCM is the input
common-mode voltage. (5) Tested < 3dB below minimum specified CMR at ± CMIR limits.
OPA683
SBOS221E
3
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
RF = 1.4kΩ, RL = 1kΩ, and G = +2 (see Figure 3 for AC performance only), unless otherwise noted.
OPA683ID, IDBV
TYP
PARAMETER
AC PERFORMANCE (See Figure 3)
Small-Signal Bandwidth (VO = 0.2VPP)
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Noninverting Input Current Noise
Inverting Input Current Noise
Differential Gain
Differential Phase
DC PERFORMANCE(4)
Open-Loop Transimpedance Gain (ZOL)
Input Offset Voltage
Average Offset Voltage Drift
Noninverting Input Bias Current
Average Noninverting Input Bias Current Drift
Inverting Input Bias Current
Average Inverting Input Bias Current Drift
INPUT
Least Positive Input Voltage(5)
Most Positive Input Voltage(5)
Common-Mode Rejection Ratio (CMRR)
Noninverting Input Impedance
Inverting Input Resistance (RI )
OUTPUT
Most Positive Output Voltage
Least Positive Output Voltage
Current Output, Sourcing
Current Output, Sinking
Closed-Loop Output Impedance
DISABLE (Disabled LOW)
Power-Down Supply Current (+VS)
Off Isolation
Output Capacitance in Disable
Turn On Glitch
Turn Off Glitch
Enable Voltage
Disable Voltage
Control Pin Input Bias Current (DIS)
POWER SUPPLY
Specified Single-Supply Operating Voltage
Max Single-Supply Operating Voltage
Min Single-Supply Operating Voltage
Max Quiescent Current
Min Quiescent Current
Power-Supply Rejection Ratio (+PSRR)
TEMPERATURE RANGE
Specification: D, DBV
Thermal Resistance, θJA
D
SO-8
DBV SOT-23-6
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
70°C(2)
–40°C to
+85°C(2)
96
92
90
9
6
8
8
8
8
180
175
170
–60
–66
–59
–63
4.4
5.1
11.6
0.24
0.19
–54
–55
–58
–57
5.0
5.8
11.9
–53
–55
–58
–56
5.5
6.4
12.3
±1.0
700
300
±3.0
±2
±4
±3
±8
Open-Loop
1.1
3.9
58
50 2
4.8
RL = 1kΩ to VS/2
RL = 1kΩ to VS/2
VO = VS/2
VO = VS/2
G = +2, f = 100kHz
4.2
0.8
80
70
0.009
VDIS = 0
G = +2, 5MHz
100
70
1.7
±70
±20
3.4
1.8
80
CONDITIONS
+25°C
G = +1, RF = 1.4kΩ
G = +2, RF = 1.4kΩ
G = +5, RF = 1.4kΩ
G = +10, RF = 1.4kΩ
G = +20, RF = 1.4kΩ
G = +2, VO < 0.5VPP, RF = 1.2kΩ
RF = 1.4kΩ, VO < 0.5VPP
G = +2, VO = 2VPP
G = +2, VO = 2V Step
G = +2, VO = 0.5V Step
G = +2, VO = 2VStep
G = 2, f = 5MHz, VO = 2VPP
RL = 100Ω to VS/2
RL ≥ 1kΩ to VS/2
RL = 100Ω to VS/2
RL ≥ 1kΩ to VS/2
f > 1MHz
f > 1MHz
f > 1MHz
G = +2, NTSC, VO = 1.4VP, RL = 150Ω
G = +2, NTSC, VO = 1.4VP, RL = 150Ω
145
119
95
87
60
14
1
70
210
5.9
7.8
VO = VS/2, RL = 1kΩ to VS/2
VCM = VS/2
VCM = VS/2
VCM = VS/2
VCM = VS/2
VCM = VS/2
VCM = VS/2
VCM = VS/2
G = +2, RL = 150Ω, VIN = VS/2
G = +2, RL = 150Ω, VIN = VS/2
VDIS = 0V
MIN/ TEST
MAX LEVEL(3)
MHz
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
typ
min
typ
typ
typ
min
max
typ
min
typ
typ
B
C
C
C
B
B
C
B
C
C
–53
–55
–58
–56
5.8
6.7
12.4
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
pA/√Hz
%
deg
max
max
max
max
max
max
max
typ
typ
B
B
B
B
B
B
B
C
C
270
±3.6
±12
±4.6
±12
±8.7
±15
250
±3.8
±12
±4.8
±12
±8.9
±15
kΩ
mV
µV/°C
µA
nA/°C
µA
nA°/C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
1.25
3.75
51
1.29
3.73
50
1.34
3.67
50
V
V
dB
kΩ || pF
Ω
max
min
min
typ
typ
A
A
A
C
C
4.1
0.9
65
52
4.1
0.9
63
50
4.0
1.00
58
45
V
mA
mA
Ω
min
min
min
min
typ
A
A
A
A
C
µA
dB
pF
mV
mV
V
V
µA
typ
typ
typ
typ
typ
min
max
max
C
C
C
C
C
A
A
A
V
V
V
mA
mA
dB
typ
max
min
max
min
typ
C
A
C
A
A
C
–40 to +85
°C
typ
C
125
150
°C/W
°C/W
typ
typ
C
C
3.5
1.7
120
3.6
1.6
130
3.7
1.5
135
12
12
12
0.91
0.71
0.91
0.69
0.91
0.67
5
VS = +5V
VS = +5V
Input Referred
UNITS
2.8
0.82
0.82
65
Junction-to-Ambient
NOTES: (1) Junction temperature = ambient for 25°C tested specifications. (2) Junction temperature = ambient at low temperature limit: junction temperature = ambient
+2°C at high temperature limit for over temperature tested specifications. (3) Test levels: (A) 100% tested at 25°C. Over temperature limits by characterization and
simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out-of-node. VCM is the input
common-mode voltage. (5) Tested < 3dB below minimum specified CMR at ± CMIR limits.
OPA683
4
www.ti.com
SBOS221E
TYPICAL CHARACTERISTICS: VS = ±5V
TA = 25°C, RF = 1.2kΩ, RL = 1kΩ, and G = +2 (see Figure 1 for AC performance only), unless otherwise noted.
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
VO = 0.5VPP
RF = 1.2kΩ
3
INVERTING SMALL-SIGNAL FREQUENCY RESPONSE
3
G = 100
G=1
0
G=2
–3
–6
G = 10
G=5
–9
0
–6
G = –2
–9
See Figure 2
–12
1
10
100
200
1
10
NONINVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
VO = 0.5VPP
0
VO = 1VPP
VO = 1VPP
Gain (dB)
Gain (dB)
G = –1
RL = 1kΩ
VO = 0.5VPP
6
200
INVERTING LARGE-SIGNAL FREQUENCY RESPONSE
3
G = +2
RL = 1kΩ
100
Frequency (MHz)
Frequency (MHz)
9
G = –1
G = –5
–3
See Figure 1
–12
G = –10
G = –24
VO = 0.5VPP
RF = 1.2kΩ
G = 50
Normalized Gain (3dB/div)
Normalized Gain (3dB/div)
6
3
–3
VO = 5VPP
–6
VO = 2VPP
VO = 5VPP
0
–9
VO = 2VPP
See Figure 1
1
See Figure 2
–12
10
100
200
1
10
Frequency (MHz)
NONINVERTING PULSE RESPONSE
0.8
0.8
3.2
0.6
2.4
Large-Signal Right Scale
0.2
1.6
0.8
Small-Signal Left Scale
0
0
–0.2
–0.8
–0.4
–1.6
–0.6
Output Voltage (200mV/div)
G = –1
Output Voltage (800mV/div)
Output Voltage (200mV/div)
G = +2
–2.4
0.6
2.4
0.4
1.6
0.2
0.8
0
0
Small-Signal Left Scale
–0.2
–0.8
Large-Signal Right Scale
–0.4
–0.6
See Figure 1
–1.6
–2.4
See Figure 2
–0.8
–3.2
Time (10ns/div)
–0.8
–3.2
Time (10ns/div)
OPA683
SBOS221E
200
INVERTING PULSE RESPONSE
3.2
0.4
100
Frequency (MHz)
5
www.ti.com
Output Voltage (800mV/div)
–3
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = 25°C, RF = 1.2kΩ, RL = 1kΩ, and G = +2 (see Figure 1 for AC performance only), unless otherwise noted.
HARMONIC DISTORTION vs LOAD RESISTANCE
HARMONIC DISTORTION vs FREQUENCY
–50
–50
VO = 2VPP
f = 5MHz
G = +2
–60
VO = 2VPP
RL = 1kΩ
2nd-Harmonic
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–55
–65
–70
–75
–80
–85
–60
2nd-Harmonic
–70
3rd-Harmonic
–80
3rd-Harmonic
See Figure 1
–90
See Figure 1
–90
100
0.1
1k
1
Load Resistance (Ω)
HARMONIC DISTORTION vs OUTPUT VOLTAGE
f = 5MHz
RL = 1kΩ
–60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–50
2nd-Harmonic
–70
–80
3rd-Harmonic
See Figure 1
0.5
1
VO = 2VPP
RL = 1kΩ
5
2nd-Harmonic
–60
–70
–80
3rd-Harmonic
See Figure 1
–90
±2.5
±3
±3.5
–90
Output Voltage (VPP)
±4
±4.5
±5
Supply Voltage (±V)
±5.5
±6
HARMONIC DISTORTION vs INVERTING GAIN
HARMONIC DISTORTION vs NONINVERTING GAIN
–50
–50
–60
2nd-Harmonic
–65
–70
–75
3rd-Harmonic
–80
VO = 2VPP
f = 5MHz
RL = 1kΩ
–55
Harmonic Distortion (dBc)
VO = 2VPP
f = 5MHz
RL = 1kΩ
–55
Harmonic Distortion (dBc)
20
5MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE
–50
–60
2nd-Harmonic
–65
–70
–75
3rd-Harmonic
–80
–85
–85
–90
10
Frequency (MHz)
See Figure 2
See Figure 1
–90
1
10
1
20
10
20
Inverting Gain (–V/V)
Gain (V/V)
OPA683
6
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SBOS221E
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = 25°C, RF = 1.2kΩ, RL = 1kΩ, and G = +2 (see Figure 1 for AC performance only), unless otherwise noted.
2-TONE, 3RD-ORDER
INTERMODULATION DISTORTION
INPUT VOLTAGE AND CURRENT NOISE DENSITY
–45
Inverting Current Noise
11.6pA/√Hz
Noninverting Current Noise
5.2pA/√Hz
10
Voltage Noise
4.4nV/√Hz
3rd-Order Spurious Level (dBc)
Voltage Noise (nV/√Hz)
Current Noise (pA/√Hz)
100
20MHz
–55
10MHz
+5V
–65
PI
1kΩ
–5V
1.2kΩ
–75
1.2kΩ
1
1MHz
–85
102
103
104
105
106
107
0.1
0.4
1
VPP at 1kΩ Load (each tone)
Frequency (Hz)
DISABLE TIME
–40
VDIS
G = +2
VDIS = 0V
–50
VIN = 1VDC
See Figure 1
4
–60
Feedthru (dB)
VOUT and VDIS (V)
5
2
DISABLED FEEDTHRU
6
3
2
5MHz
PO
50Ω OPA683
VOUT
–70
–80
1
–90
See Figure 1
0
–100
0
10
20
30
40
50
60
70
80
90
100
0.1
1
Time (ms)
10
100
Frequency (MHz)
SMALL-SIGNAL BANDWIDTH vs CLOAD
RS vs CLOAD
160
9
0.5dB Peaking
10pF
140
6
Normalized Gain (dB)
120
RS (Ω)
100
80
60
40
3
+5V
RS
VI
0
VO
100pF
50Ω OPA683
CL
1kΩ
47pF
–5V
1.2kΩ
–3
20
1.2kΩ
22pF
0
–6
1
10
100
1
CLOAD (pF)
OPA683
SBOS221E
10
100
200
Frequency (MHz)
7
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TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = 25°C, RF = 1.2kΩ, RL = 1kΩ, and G = +2 (see Figure 1 for AC performance only), unless otherwise noted.
OPEN-LOOP TRANSIMPEDANCE GAIN AND PHASE
CMRR and PSRR vs FREQUENCY
CMRR
60
50
+PSRR
40
–PSRR
20
10
0
104
105
106
Frequency (Hz)
107
–30
80
–60
60
–90
∠ ZOL
40
–120
20
–150
0
–180
104
108
105
108
109
OUTPUT CURRENT AND VOLTAGE LIMITATIONS
COMPOSITE VIDEO DIFFERENTIAL GAIN/PHASE
Gain = +2
NTSC, Positive Video
4
1W Power
Limit
RL = 500Ω
5
0.2
3
0.15
2
dG
VO (V)
0.1
=
RL
50
Ω
1
0
–1
–2
0.05
–3
dP
–4
1
2
3
4
Number of 150Ω Video Loads
–150
0
IO (mA)
TYPICAL DC DRIFT OVER TEMPERATURE
SUPPLY AND OUTPUT CURRENT
vs TEMPERATURE
–50
50
100
150
1
Sourcing Output Current
3
2
Output Current (mA)
Input Bias Currents (µA)
and Offset Voltage (mV)
–100
200
4
1
1W Power
Limit
–5
0
Noninverting Input Bias Current
0
Input Offset Voltage
–1
–2
175
0.95
Supply Current
Right Scale
150
0.9
Sinking Output Current
125
0.85
–3
Inverting Input Bias Current
100
–4
–50
–25
0
25
50
75
100
125
0.8
–25
Ambient Temperature (°C)
0
25
50
75
Ambient Temperature (°C)
100
125
OPA683
8
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SBOS221E
Supply Current (mA)
Differential Gain (%)
Differential Phase (°)
106
107
Frequency (Hz)
=1
00Ω
103
100
L
102
0
20log (ZOL)
R
30
120
Open-Loop Phase (°)
Open-Loop Transimpedance Gain (dBΩ)
Common-Mode Rejection Ratio (dB)
Power-Supply Rejection Ratio (dB)
70
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = 25°C, RF = 1.2kΩ, RL = 1kΩ, and G = +2 (see Figure 1 for AC performance only), unless otherwise noted.
SETTLING TIME
DISABLED SUPPLY CURRENT vs TEMPERATURE
0.05
110
0.03
0.02
0.01
0
–0.01
–0.02
–0.03
100
90
80
70
–0.04
+VS Current
–0.05
60
0
10
20
30
Time (ns)
40
50
60
–50
0
25
50
75
Ambient Temperature (°C)
125
8.0
3.2
6.4
6.4
6.4
2.4
4.8
4.8
4.8
1.6
3.2
0.8
1.6
0
0
Output Voltage
Right Scale
–0.8
–1.6
See Figure 1
–1.6
–2.4
–3.2
–4.8
Input Voltage
Left Scale
–4.0
Input Voltage (1.6V/div)
8.0
Output Voltage (1.6V/div)
8.0
–3.2
3.2
3.2
Output Voltage
Right Scale
1.6
1.6
0
0
–1.6
–1.6
–3.2
–3.2
–4.8
–6.4
–6.4
–8.0
–8.0
–4.8
Input Voltage
Left Scale
See Figure 2
–6.4
–8.0
Time (100ns/div)
Time (100ns/div)
CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY
INPUT AND OUTPUT RANGE vs SUPPLY VOLTAGE
100
6
5
4
3
2
1
0
–1
–2
–3
–4
–5
–6
Input
Voltage
Range
Output Impedance (Ω)
Input and Output Voltage Range
100
INVERTING OVERDRIVE RECOVERY
NONINVERTING OVERDRIVE RECOVERY
4.0
Input Voltage (0.8V/div)
–25
Output
Voltage
Range
OPA683
10
ZO
1.2kΩ
1.2kΩ
1
0.01
0.001
±2
±3
±4
±5
±6
100
± Supply Voltage
10k
100k
1M
10M
100M
Frequency (Hz)
OPA683
SBOS221E
1k
9
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Input Voltage (1.6V/div)
% Error to Final Value
Disabled Supply Current (µA)
2V Step
See Figure 1
0.04
TYPICAL CHARACTERISTICS: VS = +5V
TA = 25°C, RF = 1.4kΩ, RL = 1kΩ, and G = +2 (see Figure 3 for AC performance only), unless otherwise noted.
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
3
Normalized Gain (dB)
0
INVERTING SMALL-SIGNAL FREQUENCY RESPONSE
3
G = 50
RF = 1.4kΩ
VO = 0.2VPP
RL = 1kΩ
G=1
Normalized Gain (3dB/div)
6
G=2
–3
–6
G=5
–9
G = 10
–12
–15
0
–3
RF = 1.4kΩ
VO = 0.2VPP
RL = 1kΩ
–6
G = 20
See Figure 3
–18
See Figure 4
–12
1
10
100
200
1
10
Frequency (MHz)
0.5VPP
6
G = –1
RL = 1kΩ
VO = 0.2VPP
0
VO = 0.5VPP
0.2VPP
Gain (dB)
Gain (dB)
200
INVERTING LARGE-SIGNAL FREQUENCY RESPONSE
3
G = +2
RL = 1kΩ
100
Frequency (MHz)
NONINVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
9
G = –1
G = –2
G = –5
G = –10
G = –28
–9
3
–3
VO = 1VPP
–6
1VPP
VO = 2VPP
0
–9
2VPP
–3
See Figure 4
–12
1
10
100
200
1
10
Frequency (MHz)
NONINVERTING PULSE RESPONSE
0.3
INVERTING PULSE RESPONSE
0.4
1.6
1.2
0.3
1.2
0.2
0.8
0.1
0.4
0.2
0.8
0.1
0.4
Small-Signal Left Scale
0
0
–0.1
–0.4
–0.2
–0.8
–0.3
Output Voltage (100mV/div)
Large-Signal Right Scale
200
1.6
Output Voltage (400mV/div)
Output Voltage (100mV/div)
0.4
100
Frequency (MHz)
–1.2
0
0
Small-Signal Left Scale
–0.1
–0.4
Large-Signal Right Scale
–0.2
–0.3
See Figure 3
–0.8
–1.2
See Figure 4
–0.4
–1.6
Time (10ns/div)
–0.4
–1.6
Time (10ns/div)
OPA683
10
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SBOS221E
Output Voltage (400mV/div)
See Figure 3
TYPICAL CHARACTERISTICS: VS = +5V (Cont.)
TA = 25°C, RF = 1.4kΩ, RL = 1kΩ, and G = +2 (see Figure 3 for AC performance only), unless otherwise noted.
HARMONIC DISTORTION vs FREQUENCY
HARMONIC DISTORTION vs LOAD RESISTANCE
–50
–50
3rd-Harmonic
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
VO = 2VPP
RL = 1kΩ
VO = 2VPP
f = 5MHz
–55
–60
–65
–70
2nd-Harmonic
–75
–80
3rd-Harmonic
–60
2nd-Harmonic
–70
–80
–85
See Figure 3
See Figure 3
–90
–90
100
0.1
1k
1
–45
G = +2
RL = 1kΩ
f = 5MHz
3rd-Order Spurious Level (dBc)
Harmonic Distortion (dBc)
–50
–60
2nd-Harmonic
–70
3rd-Harmonic
–80
See Figure 3
0.5
20MHz
–55
10MHz
–65
–75
See Figure 3
2
3
0.1
1
VPP at 1Ω Load (each tone)
SUPPLY AND OUTPUT CURRENT
vs TEMPERATURE
COMPOSITE VIDEO DIFFERENTIAL GAIN/PHASE
0.3
90
0.9
0.25
Sourcing Output Current
Left Scale
Supply Current
Right Scale
0.85
0.8
70
Left Scale
Sinking Output Current
0.75
60
0.7
50
–50
–25
0
25
50
75
Ambient Temperature (°C)
100
G = +2
NTSC, Positive Video
0.2
dP
0.15
0.1
dG
0.05
0
1
125
2
3
4
Number of 150Ω Video Loads
OPA683
SBOS221E
Differential Gain (%)
Differential Phase (°)
0.95
Supply Current (mA)
100
80
5MHz
–85
1
Output Voltage (VPP)
Output Current (mA)
20
2-TONE, 3RD-ORDER
INTERMODULATION DISTORTION
HARMONIC DISTORTION vs OUTPUT VOLTAGE
–90
10
Frequency (MHz)
Load Resistance (Ω)
11
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APPLICATIONS INFORMATION
VERY LOW POWER CURRENT-FEEDBACK
OPERATION
The OPA683 gives a new level of performance in very low
power current-feedback op amps. Using a new input stage
buffer architecture, the OPA683 CFBplus amplifier gives improved bandwidth to higher gains than previous < 1mA
supply current amplifiers. This closed-loop internal buffer
gives a very low and linearized impedance at the inverting
node—isolating the amplifier’s AC performance from gain
element variations. This allows both the bandwidth and
distortion to remain nearly constant over gain—moving closer
to the ideal current-feedback performance of Gain Bandwidth
independence. This low power amplifier also delivers exceptional output power—its ±4V swing on ±5V supplies with
> 100mA output drive gives excellent performance into
standard video loads or doubly-terminated 50Ω cables. Single
+5V supply operation is also supported with similar bandwidths, but reduced output power capability. For improved
harmonic distortion driving heavier loads, in a low power
CFBplus amplifier, consider the OPA684, while for even
higher output power, consider the OPA691.
Figure 2 shows the DC-coupled, gain of –1V/V, dual powersupply circuit used as the basis of the Inverting Typical
Characteristics. Inverting operation offers several performance benefits. Since there is no common-mode signal
across the input stage, the slew rate for inverting operation
is higher and the distortion performance is slightly improved.
An additional input resistor, RM, is included in Figure 2 to set
the input impedance equal to the 50Ω. The parallel combination of RM and RG set the input impedance. As the desired
gain increases for the inverting configuration, RG is adjusted
to achieved the desired gain and RM is also adjusted to hold
a 50Ω input match. A point will be reached where RG will
equal 50Ω, RM is then removed and the input match is set by
RG only. With RG fixed to achieve an input match to 50Ω, to
increase gain, RF is simply increased. This will, however,
quickly reduce the achievable bandwidth as the feedback
resistor increases from its recommended value of 1.2kΩ. If
the source does not require an input match to 50Ω, either
adjust RM to the get the desired load or remove it and let the
RG resistor alone provide the input load.
+5V
Figure 1 shows the DC-coupled, gain of +2, dual powersupply circuit used as the basis of the ±5V Electrical Characteristics and Typical Characteristics. For test purposes, the
input impedance is set to 50Ω with a resistor to ground while
the output load is a 1kΩ resistor. Voltage swings reported in
the specifications are taken directly at the input and output
pins while load powers (dBm) are interpreted as the voltage
swing at the output converted to dBm as if the load were
50Ω. For the circuit of Figure 1, the total effective load will be
1kΩ || 2.4kΩ = 706Ω. Gain changes are most easily accomplished by simply resetting the RG value—holding RF constant at its recommended value of 1.2kΩ. The disable control
line (DIS) is typically left open to ensure normal amplifier
operation. It may, however, be asserted LOW to reduce the
amplifier quiescent to 100µA typically.
0.1µF
+
6.8µF
OPA683
RS = 50Ω
VO
1kΩ
DIS
RF
1.2kΩ
RG
1.2kΩ
VI
RT
52.3Ω
0.1µF
+
6.8µF
–5V
FIGURE 2. DC-Coupled, G = –1V/V, Bipolar Supply, Specification and Test Circuit.
+5V
0.1µF
+
These circuits are showing ±5V operation. The same circuit
can be applied with bipolar supplies ranging from ±2.5V to
±6V. Internal supply independent biasing gives nearly the
same performance for the OPA683 over this wide range of
supplies. Generally, the optimum feedback resistor value (for
nominally flat frequency response at G = +2) will increase in
value as the total supply voltage across the OPA683 is
reduced.
6.8µF
VI
RG = 50Ω
50Ω
OPA683
VO
1kΩ
DIS
RF
1.2kΩ
RG
1.2kΩ
0.1µF
+
6.8µF
–5V
FIGURE 1. DC-Coupled, G = +2V/V, Bipolar Supply, Specification and Test Circuit.
See Figure 3 for the AC-coupled, single +5V supply, gain of
+2V/V circuit configuration used as a basis for the +5V only
Electrical Characteristics and Typical Characteristics. The
key requirement of broadband single-supply operation is to
maintain input and output signal swings within the usable
voltage ranges at both the input and the output. The circuit
of Figure 3 establishes an input midpoint bias using a simple
resistive divider from the +5V supply (two 12.5kΩ resistors)
to the noninverting input. The input signal is then AC-coupled
OPA683
12
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SBOS221E
+5V
0.1µF
50Ω Source
+
+5V
6.8µF
12.5kΩ
0.1µF
0.1µF
2.5V
VI
0.1µF
50Ω
12.5kΩ
OPA683
DIS
RF
1.4kΩ
+
6.8µF
12.5kΩ
2.5V
VO
0.1µF
0.1µF
1kΩ
50Ω Source
0.1µF
12.5kΩ
OPA683
DIS
RF
1.4kΩ
RG
1.4kΩ
VO
1kΩ
VI
RG
1.4kΩ
52.3Ω
0.1µF
FIGURE 3. AC-Coupled, G = +2V/V, Single-Supply, Specification and Test Circuit.
into this midpoint voltage bias. The input voltage can swing
to within 1.25V of either supply pin, giving a 2.5VPP input
signal range centered between the supply pins. The input
impedance of Figure 3 is set to give a 50Ω input match. If the
source does not require a 50Ω match, remove this and drive
directly into the blocking capacitor. The source will then see
the 6.25kΩ load of the biasing network. The gain resistor
(RG) is AC-coupled, giving the circuit a DC gain of +1—which
puts the noninverting input DC bias voltage (2.5V) on the
output as well. The feedback resistor value has been adjusted from the bipolar supply condition to re-optimize for a
flat frequency response in +5V only, gain of +2 operation. On
a single +5V supply, the output voltage can swing to within
1.0V of either supply pin while delivering more than 50mA
output current giving 3VPP output swing into an AC-coupled
100Ω load if required (8dBm maximum at the matched load).
The circuit of Figure 3 shows a blocking capacitor driving into
a 1kΩ load resistor. Alternatively, the blocking capacitor
could be removed if the load is tied to a supply midpoint or
to ground if the DC current required by the load is acceptable.
Figure 4 shows the AC-coupled, single +5V supply, gain of
–1V/V circuit configuration used as a basis for the +5V only
Typical Characteristics. In this case, the midpoint DC bias on
the noninverting input is also decoupled with an additional
0.1µF decoupling capacitor. This reduces the source impedance at higher frequencies for the noninverting input bias
current noise. This 2.5V bias on the noninverting input pin
appears on the inverting input pin and, since RG is DC
blocked by the input capacitor, will also appear at the output
pin. One advantage to inverting operation is that since there
is no signal swing across the input stage, higher slew rates
and operation to even lower supply voltages is possible. To
retain a 1VPP output capability, operation down to a 3V
supply is allowed. At a +3V supply, the input stage is
saturated, but for the inverting configuration of a currentfeedback amplifier, wideband operation is retained even
under this condition.
FIGURE 4. AC-Coupled, G = –1V/V, Single-Supply, Specification and Test Circuit.
The circuits of Figure 3 and 4 show single-supply operation
at +5V. These same circuits may be used up to single
supplies of +12V with minimal changes in the performance of
the OPA683.
LOW POWER, VIDEO LINE DRIVER
APPLICATIONS
For low power, video line driving, the OPA683 provides the
output current and linearity to support multiple load composite video signals. Figure 5 shows a typical ±5V supply video
line driver application. The improved 2nd-harmonic distortion
of the CFBplus architecture, along with the OPA683’s high
output current and voltage, gives exceptional differential gain
and phase performance in a very low power solution. As the
Typical Characteristics show, a single video load shows a
dG/dP of 0.06%/0.03°. Multiple loads may also be driven with
< 0.15%/0.1° dG/dP for up to 4 parallel video loads where the
amplifier is driving an equivalent load of 37.5Ω.
+5V
VIDEOIN
Supply Decoupling not shown.
75Ω
75Ω
Coax 75Ω Load
OPA683
1.2kΩ
1.2kΩ
–5V
FIGURE 5. Gain of +2 Video Cable Driver.
OPA683
SBOS221E
DIS
13
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VERY LOW POWER ACTIVE FILTER
HIGH GAIN HF AMPLIFIER
The OPA683 provides an exceptionally capable gain block
for implementing Sallen-Key type filters. Typically, the bandwidth interaction with gain setting for low power amplifiers,
constrain these filters to using unity-gain amplifiers. Since
the OPA683 CFBplus design holds very high bandwidth to
high gains, implementations that provide signal gain, as well
as the desired filter shape, are easily implemented. Figure 6
shows an example of a 5MHz 2nd-order low-pass filter where
the amplifier is providing a voltage gain of 4. This singlesupply implementation (applicable to single +12V operation
as well) consumes only 5.1mW quiescent power. The two
12.5kΩ resistors bias the input and output at the supply
midpoint while the three 0.1µF capacitors block off the DC
current paths to ground for this mid-scale operating point.
The filter resistors and capacitors have been adjusted
to provide a Butterworth (Q = 0.707) response with a
ωO = 2π • 5MHz. This gives a flat passband response with a
–3dB cutoff at 5MHz. Figure 7 shows the small-signal frequency response for the circuit of Figure 6.
Where high gains at moderate frequencies are required in an
HF receiver channel, the OPA683 can provide a very low
power solution with moderate input noise figure. Figure 8
shows a technique that can improve the noise figure with no
added power. An input transformer provides a noiseless
voltage gain at the cost of higher source impedance for the
amplifier’s noninverting input current noise. The circuit of
Figure 8, using a 1:4 turns ratio (1:16 impedance ratio)
transformer, reduces the input noise figure from about 20dB
for just the amplifier to 10.6dB in combination. The bandwidth
for this circuit will be principally set by the transformer since
the OPA683 will give > 80MHz for the gain of 20V/V shown
in Figure 8. The overall circuit gives a gain to a matched 50Ω
load of 32dB (40V/V) from the transformer input. This example circuit provides this gain using only 10mW of quiescent power with application from 500kHz to 30MHz.
+5V
PI
+5V
Supply
De-coupling
Not Shown
50Ω
1:4
50Ω
OPA683
800Ω
PO
50Ω
100pF
12.5kΩ
0.1µF
157Ω
1.2kΩ
446Ω
VI
0.1µF
150pF
OPA683
12.5kΩ
VO
10.6dB
Noise Figure
–5V
63Ω
1kΩ
PO
= 32dB
PI
0.01µF
1.4kΩ
FIGURE 8. Low Power, High Gain HF Amplifier.
467Ω
0.1µF
LOW POWER, ADC DRIVER
FIGURE 6. 5MHz, 2nd-Order Low Pass Filter.
LOW POWER 5MHz LP ACTIVE FILTER
15
12
Gain (dB)
9
6
3
0
–3
–6
–9
1k
10k
100k
1M
10M 20M
Frequency (Hz)
FIGURE 7. Low Power Active Filter Frequency Response.
Where a low power, single-supply interface to a single-ended
input +5V ADC is required, the circuit of Figure 9 can provide
a very flexible, high performance solution. Running in an ACcoupled inverting mode allows the noninverting input to be
used for the common-mode voltage from the ADS820 converter. This midpoint reference biases both the noninverting
converter input and the amplifier noninverting input. With an
AC-coupled gain path, this +2.5V DC bias has a gain of +1
to the output putting the output at the DC midpoint for the
converter. The output then drives through an isolating resistor (60Ω) to the inverting input of the converter which is
further decoupled by a 22pF external capacitance to add to
its 5pF input capacitance. This coupling network provides a
high cutoff low-pass while also giving a low source impedance at high frequencies for the converter. The gain for this
circuit is set by adjusting RG to the desired value. For a 2VPP
maximum output driving the light load of Figure 9, the
OPA683 will provide < –80dBc THD through 1MHz as shown
in the Typical Characteristics. One of the important advantages for this CFBplus amplifier is that this distortion does not
degrade significantly at higher gains.
OPA683
14
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SBOS221E
RI
1.4kΩ
+5V
DIS
RG
2.5VDC
VI
60Ω
OPA683
VO
IN
ADS820
10-Bit
20MSPS
22pF
VO = –
1.4kΩ
RG
VI
2VPP
Max
50Ω
IN
+2.5V
CM
0.1µF
FIGURE 9. Low Power, Single-Supply, ADC Driver.
DESIGN-IN TOOLS
DEMONSTRATION FIXTURES
VI
Two printed circuit boards (PCBs) are available to assist in
the initial evaluation of circuit performance using the OPA683
in its two package options. Both of these are offered free of
charge as unpopulated PCBs, delivered with a user's guide.
The summary information for these fixtures is shown in
Table I.
α
VO
RI
iERR
PRODUCT
OPA683ID
OPA683IDBQ
PACKAGE
ORDERING
NUMBER
LITERATURE
NUMBER
SO-8
SOT23-6
DEM-OPA-SO-1A
DEM-OPA-SOT-1A
SBOU009
SBOU010
Z(S) iERR
RF
RG
TABLE I. Demonstration Fixtures by Package.
The demonstration fixtures can be requested at the Texas
Instruments web site (www.ti.com) through the OPA683
product folder.
OPERATING SUGGESTIONS
SETTING RESISTOR VALUES TO OPTIMIZE
BANDWIDTH
Any current-feedback op amp like the OPA683 can hold high
bandwidth over signal gain settings with the proper adjustment of the external resistor values. A low-power part like the
OPA683 typically shows a larger change in bandwidth due to
the significant contribution of the inverting input impedance
to loop-gain changes as the signal gain is changed. Figure
10 shows a simplified analysis circuit for any current feedback amplifier.
The key elements of this current feedback op amp model are:
α ⇒ Buffer gain from the noninverting input to the inverting input
RI ⇒ Buffer output impedance
FIGURE 10. Current Feedback Transfer Function Analysis
Circuit.
The buffer gain is typically very close to 1.00 and is normally
neglected from signal gain considerations. It will, however
set the CMRR for a single op amp differential amplifier
configuration. For the buffer gain α < 1.0, the CMRR =
–20 • log(1 – α). The closed loop input stage buffer used in
the OPA683 gives a buffer gain more closely approaching
1.00 and this shows up in a slightly higher CMRR than any
previous current feedback op amp. The 60dB typical CMRR
shown in the Electrical Characteristics implies a buffer gain
of 0.9990.
RI, the buffer output impedance, is a critical portion of the
bandwidth control equation. The OPA683 reduces this element to approximately 4.5Ω using the loop-gain of the input
buffer stage. This significant reduction in buffer output impedance, on very low power, contributes significantly to
extending the bandwidth at higher gains.
iERR ⇒ Feedback error current signal
Z(s) ⇒ Frequency dependent open loop transimpedance gain
from iERR to VO
OPA683
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R
α 1 + F
R
VO
α NG
G
=
=
VI
RF 1 + RF + RI NG
RF + RI 1 +
Z (S )
RG
1+
Z (S )
R
NG = 1 + F
R G
This is written in a loop-gain analysis format where the errors
arising from a non-infinite open-loop gain are shown in the
denominator. If Z(s) was infinite over all frequencies, the
denominator of Equation 1 would reduce to 1 and the ideal
desired signal gain shown in the numerator would be achieved.
The fraction in the denominator of Equation 1 determines the
frequency response. Equation 2 shows this as the loop-gain
equation.
(2)
Z (S )
RF + RI NG
3900
325
VO = 0.5VPP
3400
= Loop Gain
275
–3dB Bandwidth
2900
225
2400
175
1900
125
RF
1400
75
900
25
2
The OPA683 is internally compensated to give a maximally
flat frequency response for RF = 1.2kΩ at NG = 2 on ±5V
supplies. That optimum value goes to 1.4kΩ on a single +5V
supply. Normally, with a current feedback amplifier, it is
possible to adjust the feedback resistor to hold this bandwidth up as the gain is increased. The CFBplus architecture
has reduced the contribution of the inverting input impedance
to provide exceptional bandwidth to higher gains without
adjusting the feedback resistor value. The Typical Characteristics show the small-signal bandwidth over gain with a fixed
feedback resistor.
5
10
20
50
100
Voltage Gain (V/V)
FIGURE 11. Bandwidth and RF Optimized vs Gain.
3
G=5
G=2
0
Normalized Gain (dB)
If 20 • log(RF + NG • RI) were drawn on top of the open-loop
transimpedance plot, the difference between the two would
be the loop gain at a given frequency. Eventually, Z(s) rolls
off to equal the denominator of Equation 2 at which point the
loop gain has reduced to 1 (and the curves have intersected).
This point of equality is where the amplifier’s closed-loop
frequency response given by Equation 1 will start to roll off,
and is exactly analogous to the frequency at which the noise
gain equals the open-loop voltage gain for a voltage feedback op amp. The difference here is that the total impedance
in the denominator of Equation 2 may be controlled somewhat separately from the desired signal gain (or NG).
Bandwidth (MHz)
(1)
At very high gains, 2nd-order effects in the buffer output
impedance cause the overall response to peak up. If desired,
it is possible to retain a flatter frequency response at higher
gains by adjusting the feedback resistor to higher values as
the gain is increased. Figure 11 shows the empirically determined feedback resistor and resulting –3dB bandwidth from
gains of +2 to +100 to hold a < 0.5dB peaked response.
Here, since a slight peaking was allowed, a lower nominal RF
is suggested at a gain of +2 giving > 250MHz bandwidth.
This exceeds that shown in the Electrical Characteristics due
to the slightly lower feedback resistor allowing a modest
peaking in the response. Figure 12 shows the measured
frequency response curves with the adjusted feedback resistor value. While the bandwidth for this low-power part does
reduce at higher gains, going over a 50:1 gain range gives
only a factor of 10 bandwidth reduction. The 25MHz bandwidth at a gain of 100V/V is equivalent to a 2.5GHz gain
bandwidth product voltage feedback amplifier capability. Even
better bandwidth retention to higher gains can be delivered
by the slightly higher quiescent power OPA684.
Feedback Resistor (Ω)
A current-feedback op amp senses an error current in the
inverting node (as opposed to a differential input error voltage for a voltage feedback op amp) and passes this on to the
output through an internal frequency dependent transimpedance gain. The Typical Characteristics show this open-loop
transimpedance response. This is analogous to the openloop voltage gain curve for a voltage feedback op amp.
Developing the transfer function for the circuit of Figure 10
gives Equation 1:
–3
G = 100
–6
G = 10
G = 50
–9
G = 20
–12
1
10
100
200
Frequency (MHz)
FIGURE 12. Small-Signal Frequency Response with Optimized RF.
OPA683
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SBOS221E
OUTPUT CURRENT AND VOLTAGE
The OPA683 provides output voltage and current capabilities
that can support the needs of driving doubly-terminated 50Ω
lines. Changing the 1kΩ load in Figure 1 to a 100Ω will give
a total load that is the parallel combination of the 100Ω load
and the 2.4kΩ total feedback network impedance. This 96Ω
load will require no more than 36mA output current to support
a ±3.5V output voltage swing. This is within the specified
minimum output current of +58mA/–45mA over the full temperature range.
The specifications described above, though familiar in the
industry, consider voltage and current limits separately. In
many applications, it is the voltage • current, or V-I product,
which is more relevant to circuit operation. Refer to the
“Output Voltage and Current Limitations” plot in the Typical
Characteristics. The X and Y axes of this graph show the
zero-voltage output current limit and the zero-current output
voltage limit, respectively. The four quadrants give a more
detailed view of the OPA683’s output drive capabilities.
Superimposing resistor load lines onto the plot shows the
available output voltage and current for specific loads.
The minimum specified output voltage and current over
temperature are set by worst-case simulations at the cold
temperature extreme. Only at cold startup will the output
current and voltage decrease to the numbers shown in the
Electrical Specifications. As the output transistors deliver
power, their junction temperatures will increase, decreasing
their VBE’s (increasing the available output voltage swing)
and increasing their current gains (increasing the available
output current). In steady state operation, the available
output voltage and current will always be greater than that
shown in the over-temperature specifications since the output stage junction temperatures will be higher than the
minimum specified operating ambient.
To maintain maximum output stage linearity, no output short
circuit protection is provided. This will not normally be a
problem since most applications include a series matching
resistor at the output that will limit the internal power dissipation if the output side of this resistor is shorted to ground.
However, shorting the output pin directly to the adjacent
positive power-supply pin (8-pin packages) will, in most
cases, destroy the amplifier. If additional short-circuit protection is required, consider a small series resistor in the powersupply leads. This will, under heavy output loads, reduce the
available output voltage swing. A 5Ω series resistor in each
power-supply lead will limit the internal power dissipation to
less than 1W for an output short circuit while decreasing the
available output voltage swing only 0.25V for up to 50mA
desired load currents. Always place the 0.1µF power-supply
decoupling capacitors after these supply current limiting
resistors directly on the supply pins.
DRIVING CAPACITIVE LOADS
One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADC—including additional
external capacitance which may be recommended to im-
prove ADC linearity. A high-speed, high open-loop gain
amplifier like the OPA683 can be very susceptible to decreased stability and closed-loop response peaking when a
capacitive load is placed directly on the output pin. When the
amplifier’s open-loop output resistance is considered, this
capacitive load introduces an additional pole in the signal
path that can decrease the phase margin. Several external
solutions to this problem have been suggested. When the
primary considerations are frequency response flatness, pulse
response fidelity and/or distortion, the simplest and most
effective solution is to isolate the capacitive load from the
feedback loop by inserting a series isolation resistor between
the amplifier output and the capacitive load. This does not
eliminate the pole from the loop response, but rather shifts it
and adds a zero at a higher frequency. The additional zero
acts to cancel the phase lag from the capacitive load pole,
thus increasing the phase margin and improving stability.
The Typical Characteristics show the recommended “RS vs
Capacitive Load” and the resulting frequency response at the
load. The 1kΩ resistor shown in parallel with the load
capacitor is a measurement path and may be omitted.
Parasitic capacitive loads greater than 3pF can begin to
degrade the performance of the OPA683. Long PC board
traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always
consider this effect carefully, and add the recommended
series resistor as close as possible to the OPA683 output pin
(see Board Layout Guidelines).
DISTORTION PERFORMANCE
The OPA683 provides low distortion in a very low power
amplifier. The CFBplus architecture also gives two significant
areas of distortion improvement. First, in operating regions
where the 2nd-harmonic distortion due to output stage
nonlinearities is very low (frequencies < 1MHz, low output
swings into light loads) the linearization at the inverting node
provided by the CFBplus design gives 2nd-harmonic distortions that extend into the –90dBc region. Previous current
feedback amplifiers have been limited to approximately
–85dBc due to the nonlinearities at the inverting input. The
second area of distortion improvement comes in a distortion
performance that is more gain independent than prior solutions. To the extent that the distortion at a particular output
power is output stage dependent, 2nd-harmonic particularly,
and to a lesser extend 3rd-harmonic distortion, is constant as
the gain is increased. This is due to the constant loop gain
versus signal gain provided by the CFBplus design. As shown
in the Typical Characteristics, while the 2nd-harmonic is
constant with gain, the 3rd-harmonic degrades at higher
gains.
Relative to alternative amplifiers with < 1mA supply current,
the OPA683 holds much lower distortion at higher frequencies (> 5MHz) and to higher gains. Generally, until the
fundamental signal reaches very high frequency or power
levels, the 2nd-harmonic will dominate the distortion with a
lower 3rd-harmonic component. Focusing then on the 2ndharmonic, increasing the load impedance improves distortion
slightly for the OPA683. Remember that the total load in-
OPA683
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cludes the feedback network—in the noninverting configuration (see Figure 1) this is the sum of RF + RG, while in the
inverting configuration it is just RF. Also, providing an additional supply decoupling capacitor (0.1µF) between the supply pins (for bipolar operation) improves the 2nd-order distortion slightly (3dB to 6dB).
In most op amps, increasing the output voltage swing increases harmonic distortion directly. A low-power part like the
OPA683 includes quiescent boost circuits to provide the fullpower bandwidth shown. These act to increase the bias in a
very linear fashion only when high slew rate or output power
are required. The Typical Characteristics show the 2nd-harmonic increasing slightly from 500mVPP to 5VPP outputs while
the 3rd-harmonics also increase with output power.
The OPA683 has an extremely low 3rd-order harmonic distortion—particularly for light loads and at lower frequencies. This
also gives low 2-tone, 3rd-order intermodulation distortion as
shown in the Typical Characteristics. Since the OPA683
includes internal power boost circuits to retain good full-power
performance at high frequencies and outputs, it does not show
a classical 2-tone, 3rd-order intermodulation intercept characteristic. Instead, it holds relatively low and constant 3rd-order
intermodulation spurious levels over power. The Typical Characteristics show this spurious level as a dBc below the carrier
at fixed center frequencies swept over single-tone voltage
swing at a 1kΩ load. Very light loads such as ADC inputs for
will see < –85dBc 3rd-order spurious to 1MHz for full-scale
inputs. For much lower 3rd-order intermodulation distortion
through 200MHz, consider the OPA685.
NOISE PERFORMANCE
Wideband current-feedback op amps generally have a higher
output noise than comparable voltage feedback op amps. The
OPA683 offers an excellent balance between voltage and
current noise terms to achieve low output noise in a low- power
amplifier. The inverting current noise (11.6pA/√Hz) is lower
than most other current feedback op amps while the input
voltage noise (4.4nV/√Hz) is lower than any unity-gain stable,
comparable slew rate, voltage feedback op amp. This low input
voltage noise was achieved at the price of higher noninverting
input current noise (5.1pA/√Hz). As long as the AC source
impedance looking out of the noninverting node is less than
300Ω, this current noise will not contribute significantly to the
total output noise. The op amp input voltage noise and the two
input current noise terms combine to give low output noise
under a wide variety of operating conditions. Figure 13 shows
the op amp noise analysis model with all the noise terms
included. In this model, all noise terms are taken to be noise
voltage or current density terms in either nV/√Hz or pA/√Hz.
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 3 shows the general form for the
output noise voltage using the terms shown in Figure 13.
(3)
2
2
EO = ENI2 + (IBNR S ) + 4kTRS GN2 + (IBIRF ) + 4kTRF GN
ENI
EO
OPA683
RS
IBN
ERS
RF
√ 4kTRS
4kT
RG
RG
IBI
√ 4kTRF
4kT = 1.6E –20J
at 290°K
FIGURE 13. Op Amp Noise Analysis Model.
Dividing this expression by the noise gain (NG = (1 + RF/RG))
will give the equivalent input referred spot noise voltage at the
noninverting input, as shown in Equation 4.
(4)
2
I R
4kTRF
2
EN = ENI2 + (IBNR S ) + 4kTRS + BI F +
GN
GN
Evaluating these two equations for the OPA683 circuit and
component values (see Figure 1) will give a total output spot
noise voltage of 17.6nV/√Hz and a total equivalent input spot
noise voltage of 8.8nV/√Hz. This total input referred spot
noise voltage is higher than the 4.4nV/√Hz specification for
the op amp voltage noise alone. This reflects the noise
added to the output by the inverting current noise times the
feedback resistor. As the gain is increased, this fixed output
noise power term contributes less to the total output noise
and the total input referred voltage noise given by Equation
3 will approach just the 4.4nV/√Hz of the op amp itself. For
example, going to a gain of +20 in the circuit of Figure 1,
adjusting only the gain resistor to 63.2Ω, will give a total input
referred noise of 4.6nV/√Hz. A more complete description of
op amp noise analysis can be found in the TI application note
AB-103 (SBOA066). Refer to Texas Instruments’ web site at
www.ti.com.
DC ACCURACY AND OFFSET CONTROL
A current-feedback op amp like the OPA683 provides exceptional bandwidth in high gains, giving fast pulse settling but
only moderate DC accuracy. The Electrical Characteristics
show an input offset voltage comparable to high slew rate
voltage-feedback amplifiers. However, the two input bias
currents are somewhat higher and are unmatched. Whereas
bias current cancellation techniques are very effective with
most voltage feedback op amps, they do not generally
reduce the output DC offset for wideband current-feedback
op amps. Since the two input bias currents are unrelated in
both magnitude and polarity, matching the source impedance looking out of each input to reduce their error contribution to the output is ineffective. Evaluating the configuration
of Figure 1, using worst case +25°C input offset voltage and
OPA683
18
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SBOS221E
the two input bias currents, gives a worst case output offset
range equal to:
±(NG • VOS(MAX)) ± (IBN • RS/2 • NG) ± (IBI • RF)
where
NG = noninverting signal gain
= ±(2 • 3.5mV) ± (4µA • 25Ω • 2) ± (1.2kΩ • 10µA)
= ±7mV ± 0.1mV ± 12mV
= ±19.1mV
While the last term, the inverting bias current error, is dominant
in this low-gain circuit, the input offset voltage will become the
dominant DC error term as the gain exceeds 4V/V. Where
improved DC precision is required in a high-speed amplifier,
consider the OPA642 single and OPA2822 dual voltagefeedback amplifiers.
DISABLE OPERATION
The OPA683 provides an optional disable feature that may
be used to reduce system power when channel operation is
not required. If the V DIS control pin is left unconnected, the
OPA683 will operate normally. To disable, the control pin
must be asserted LOW. Figure 14 shows a simplified internal
circuit for the disable control feature.
The OPA683 provides very high power gain on low quiescent
current levels. When disabled, internal high impedance nodes
discharge slowly which, with the exceptional power gain
provided, give a self powering characteristic that leads to a
slow turn off characteristic. Typical full turn off times to rated
100µA disabled supply current are 60ms. Turn on times are
very fast—less than 40ns.
THERMAL ANALYSIS
Operating junction temperature (TJ) is given by TA + PD • θJA.
The total internal power dissipation (PD) is the sum of
quiescent power (PDQ) and additional power dissipated in the
output stage (PDL) to deliver load power. Quiescent power is
simply the specified no-load supply current times the total
supply voltage across the part. PDL will depend on the
required output signal and load but would, for a grounded
resistive load, be at a maximum when the output is fixed at
a voltage equal to 1/2 either supply voltage (for equal bipolar
supplies). Under this condition PDL = VS2/(4 • RL) where RL
includes feedback network loading.
40kΩ
Q1
25kΩ
250kΩ
IS
Control
When disabled, the output and input nodes go to a high
impedance state. If the OPA683 is operating in a gain of +1
(with a 1.2kΩ feedback resistor still required for stability), this
will show a very high impedance (1.7pF || 1MΩ) at the output
and exceptional signal isolation. If operating at a gain greater
than +1, the total feedback network resistance (RF + RG) will
appear as the impedance looking back into the output, but
the circuit will still show very high forward and reverse
isolation. If configured as an inverting amplifier, the input and
output will be connected through the feedback network
resistance (RF + RG) giving relatively poor input to output
isolation.
The OPA683 will not require external heat-sinking for most
applications. Maximum desired junction temperature will set
the maximum allowed internal power dissipation as described below. In no case should the maximum junction
temperature be allowed to exceed 175°C.
+VS
VDIS
This shuts off the collector current out of Q1, turning the
amplifier off. The supply current in the disable mode are only
those required to operate the circuit of Figure 14.
–VS
Note that it is the power in the output stage and not into the
load that determines internal power dissipation.
FIGURE 14. Simplified Disable Control Circuit.
In normal operation, base current to Q1 is provided through
the 250kΩ resistor while the emitter current through the 40kΩ
resistor sets up a voltage drop that is inadequate to turn on
the two diodes in Q1’s emitter. As V DIS is pulled LOW,
additional current is pulled through the 40kΩ resistor eventually turning on these two diodes (≈ 33µA). At this point, any
further current pulled out of V DIS goes through those diodes
holding the emitter-base voltage of Q1 at approximately 0V.
As an absolute worst case example, compute the maximum
TJ using an OPA683IDBV (SOT23-6 package) in the circuit
of Figure 1 operating at the maximum specified ambient
temperature of +85°C and driving a grounded 100Ω load.
PD = 10V • 1.05mA + 52 /(4 • (100Ω || 2.4kΩ)) = 76mW
Maximum TJ = +85°C + (0.076W • 150°C/W) = 96°C.
This maximum operating junction temperature is well below
most system level targets. Most applications will be lower
than this since an absolute worst case output stage power
was assumed in this calculation.
OPA683
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BOARD LAYOUT GUIDELINES
is a good starting point for design. Note that a 1.2kΩ
feedback resistor, rather than a direct short, is required for
the unity-gain follower application. A current-feedback op
amp requires a feedback resistor even in the unity-gain
follower configuration to control stability.
Achieving optimum performance with a high frequency amplifier like the OPA683 requires careful attention to board
layout parasitics and external component types. Recommendations that will optimize performance include:
a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the
output and inverting input pins can cause instability: on
the noninverting input, it can react with the source impedance to cause unintentional band-limiting.. To reduce
unwanted capacitance, a window around the signal I/O
pins should be opened in all of the ground and power
planes around those pins. Otherwise, ground and power
planes should be unbroken elsewhere on the board.
b) Minimize the distance (< 0.25") from the power-supply
pins to high frequency 0.1µF decoupling capacitors.
At the device pins, the ground and power-plane layout
should not be in close proximity to the signal I/O pins.
Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors.
The power-supply connections should always be decoupled
with these capacitors. An optional supply decoupling
capacitor across the two power supplies (for bipolar operation) will improve 2nd-harmonic distortion performance.
Larger (2.2µF to 6.8µF) decoupling capacitors, effective at
lower frequency, should also be used on the main supply
pins. These may be placed somewhat farther from the
device and may be shared among several devices in the
same area of the PC board.
c) Careful selection and placement of external components will preserve the high frequency performance
of the OPA683. Resistors should be a very low reactance
type. Surface-mount resistors work best and allow a
tighter overall layout. Metal film and carbon composition
axially-leaded resistors can also provide good high frequency performance. Again, keep their leads and PC
board trace length as short as possible. Never use wirewound type resistors in a high-frequency application.
Since the output pin and inverting input pin are the most
sensitive to parasitic capacitance, always position the
feedback and series output resistor, if any, as close as
possible to the output pin. Other network components,
such as noninverting input termination resistors, should
also be placed close to the package. Where double side
component mounting is allowed, place the feedback resistor directly under the package on the other side of the
board between the output and inverting input pins. The
frequency response is primarily determined by the feedback resistor value as described previously. Increasing its
value will reduce the peaking at higher gains, while
decreasing it will give a more peaked frequency response
at lower gains. The 1.2kΩ feedback resistor used in the
Electrical Characteristics at a gain of +2 on ±5V supplies
d) Connections to other wideband devices on the board
may be made with short direct traces or through
onboard transmission lines. For short connections, consider the trace and the input to the next device as a
lumped capacitive load. Relatively wide traces (50mils to
100mils) should be used, preferably with ground and
power planes opened up around them. Estimate the total
capacitive load and set RS from the plot of recommended
RS versus capacitive load. Low parasitic capacitive loads
(< 5pF) may not need an RS since the OPA683 is
nominally compensated to operate with a 2pF parasitic
load. If a long trace is required, and the 6dB signal loss
intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission
line using microstrip or stripline techniques (consult an
ECL design handbook for microstrip and stripline layout
techniques). A 50Ω environment is normally not necessary on board, and in fact a higher impedance environment will improve distortion as shown in the distortion
versus load plots. With a characteristic board trace impedance defined based on board material and trace dimensions, a matching series resistor into the trace from the
output of the OPA683 is used as well as a terminating
shunt resistor at the input of the destination device.
Remember also that the terminating impedance will be
the parallel combination of the shunt resistor and the input
impedance of the destination device: this total effective
impedance should be set to match the trace impedance.
The high output voltage and current capability of the
OPA683 allows multiple destination devices to be handled
as separate transmission lines, each with their own series
and shunt terminations. If the 6dB attenuation of a doublyterminated transmission line is unacceptable, a long trace
can be series-terminated at the source end only. Treat the
trace as a capacitive load in this case and set the series
resistor value as shown in the plot of “RS vs Capacitive
Load”. This will not preserve signal integrity as well as a
doubly-terminated line. If the input impedance of the
destination device is LOW, there will be some signal
attenuation due to the voltage divider formed by the series
output into the terminating impedance.
e) Socketing a high-speed part like the OPA683 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it
almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the
OPA683 onto the board.
OPA683
20
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SBOS221E
INPUT AND ESD PROTECTION
The OPA683 is built using a very high speed complementary
bipolar process. The internal junction breakdown voltages
are relatively low for these very small geometry devices.
These breakdowns are reflected in the Absolute Maximum
Ratings table where an absolute maximum 13V across the
supply pins is reported. All device pins have limited ESD
protection using internal diodes to the power supplies as
shown in Figure 15.
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (e.g. in systems with ±15V supply parts
driving into the OPA683), current limiting series resistors
should be added into the two inputs. Keep these resistor
values as low as possible since high values degrade both
noise performance and frequency response.
+V CC
External
Pin
–V CC
FIGURE 15. Internal ESD Protection.
OPA683
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Internal
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21
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Revision History
DATE
REVISION
7/08
E
3/06
D
PAGE
SECTION
2
Abs Max Ratings
3, 4
Electrical Characteristics,
Power Supply
15
Design-In Tools
DESCRIPTION
Changed Storage Temperature Range from −40°C to +125C to
−65°C to +125C.
Added minimum supply voltage.
Board part number changed.
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
OPA683
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SBOS221E
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
OPA683ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
683
OPA683IDBVR
ACTIVE
SOT-23
DBV
6
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
A83
OPA683IDBVT
ACTIVE
SOT-23
DBV
6
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
A83
OPA683IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
683
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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