OPA832IDBVR

OPA832 SBOS266E − JUNE 2003 − REVISED AUGUST 2008 Low-Power, Single-Supply, Fixed-Gain Video Buffer Amplifier FEATURES DESCRIPTION D HIGH BANDWIDTH: 80MHz (G = +2) D LOW SUPPLY CURRENT: 3.9mA D FLEXIBLE SUPPLY RANGE: D D D D D The OPA832 is a low-power, high-speed, fixed-gain amplifier designed to operate on a single +3.3V or +5V supply. Operation on ±5V or +10V supplies is also supported. The input range extends below ground and to within 1V of the positive supply. Using complementary common-emitter outputs provides an output swing to within 30mV of ground and 130mV of the positive supply. The high output drive current and low differential gain and phase errors also make it ideal for single-supply consumer video products. +2.8V to +11V Single Supply ±1.4V to ±5.5V Dual Supply INPUT RANGE INCLUDES GROUND ON SINGLE SUPPLY 4.9VPP OUTPUT SWING ON +5V SUPPLY HIGH SLEW RATE: 350V/µsec LOW INPUT VOLTAGE NOISE: 9.3nV/√Hz Pb-FREE SOT23 PACKAGE Low distortion operation is ensured by the high gain bandwidth product (200MHz) and slew rate (850V/µs), making the OPA832 an ideal input buffer stage to 3V and 5V CMOS converters. Unlike other low-power, single-supply amplifiers, distortion performance improves as the signal swing is decreased. A low 9.3nV/√ Hz input voltage noise supports wide dynamic range operation. APPLICATIONS D D D D SINGLE-SUPPLY VIDEO LINE DRIVERS CCD IMAGING CHANNELS LOW-POWER ULTRASOUND PORTABLE CONSUMER ELECTRONICS The OPA832 is available in an industry-standard SO-8 package. The OPA832 is also available in an ultra-small SOT23-5 package. For gains other than +1, −1, or +2, consider using the OPA830. RELATED PRODUCTS DESCRIPTION SINGLES DUALS TRIPLES QUADS Medium Speed Medium Speed, Fixed Gain OPA830 OPA2830 — OPA4830 — OPA2832 OPA3832 — LARGE−SIGNAL BANDWIDTH (1VPP AT MATCHED LOAD) +3.3V 0 Video DAC 976Ω 80.6Ω 75Ω −3 VO OPA832 II 75Ω Load 400Ω VO 400Ω Gain (dB) VI −6 = 1V/V VI −9 Single-Supply, Low-Cost Video Line Driver 1 10 100 Frequency (MHz) Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. Copyright  2003−2008, Texas Instruments Incorporated
          
             
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"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ABSOLUTE MAXIMUM RATINGS(1) Power Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +12VDC Internal Power Dissipation . . . . . . . . . . . . . . See Thermal Analysis Differential Input Voltage(2) . . . . . . . . . . . . . . . . . . . . . . . . . . . ±1.2V ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. Input Voltage Range (Single Supply) . . . . . . . −0.5V to +VS + 0.3V Storage Temperature Range: D, DBV . . . . . . . . . −65°C to +125°C Lead Temperature (soldering, 10s) . . . . . . . . . . . . . . . . . . . . +300°C Junction Temperature (TJ) . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C ESD Rating: Human Body Model (HBM) . . . . . . . . . . . . . . . . . . . . . . . 2000V Charge Device Model (CDM) . . . . . . . . . . . . . . . . . . . . . 1500V Machine Model (MM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 200V (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not supported. (2) Noninverting input to internal inverting node. PACKAGE/ORDERING INFORMATION(1) PRODUCT PACKAGE-LEAD PACKAGE DESIGNATOR SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER TRANSPORT MEDIA, QUANTITY OPA832 SO-8 Surface-Mount D −40°C to +85°C OPA832 OPA832ID Rails, 100 Tape and Reel, 2500 ″ ″ ″ ″ ″ OPA832IDR OPA832 SOT23-5 DBV −40°C to +85°C A74 OPA832IDBVT Tape and Reel, 250 ″ ″ ″ ″ ″ OPA832IDBVR Tape and Reel, 3000 (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet. PIN CONFIGURATIONS Output 1 −VS 2 Noninverting Input 3 5 +VS 4 400Ω 400Ω NC 1 8 NC 400Ω Inverting Input 2 7 +VS Noninverting Input 3 6 Output −VS 4 5 NC 400Ω Inverting Input 3 1 A74 2 SO−8 NC = No Connection 4 5 SOT23−5 Pin Orientation/Package Marking 2 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 ELECTRICAL CHARACTERISTICS: VS = ±5V Boldface limits are tested at +25°C. At TA = 25°C, G = +2, and RL = 150Ω to GND, unless otherwise noted (see Figure 3). OPA832ID, IDBV TYP PARAMETER AC PERFORMANCE (see Figure 3) Small-Signal Bandwidth Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Input Current Noise NTSC Differential Gain NTSC Differential Phase DC PERFORMANCE(4) Gain Error CONDITIONS +25°C +25°C(1) 0°C to 70°C(2) −40°C to +85°C(2) G = +2, VO ≤ 0.5VPP G = −1, VO ≤ 0.5VPP VO ≤ 0.5VPP G = +2, 2V Step 0.5V Step 0.5V Step G = +2, 1V Step VO = 2VPP, 5MHz RL = 150Ω RL = 500Ω RL = 150Ω RL = 500Ω f > 1MHz f > 1MHz RL = 150Ω RL = 150Ω 80 99 4.2 350 4.6 4.9 45 55 57 54 56 54 56 230 230 220 −64 −66 −57 −73 9.2 2.2 0.10 0.16 −60 −63 −50 −64 −60 −63 −49 −61 G = +2 G = −1 ±0.3 ±0.2 ±1.5 ±1.5 400 400 455 345 ±1.4 — +5.5 ±7 Internal RF and RG Maximum Minimum Average Drift Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Bias Current Drift Input Offset Current Input Offset Current Drift INPUT Negative Input Voltage Range Positive Input Voltage Range Input Impedance Differential Mode Common-Mode OUTPUT Output Voltage Swing Current Output, Sinking Current Output, Sourcing Short-Circuit Current Closed-Loop Output Impedance POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (+PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance D SO-8 DBV SOT23-5 MIN/MAX OVER TEMPERATURE UNITS MIN/ MAX TEST LEVEL(3) MHz MHz dB V/µs ns ns ns min min typ min max max max B B C B B B B −60 −63 −48 −57 dBc dBc dBc dBc nV/√Hz pA/√Hz % ° max max max max max max typ typ B B B B B B C C ±1.6 ±1.6 ±1.7 ±1.7 % % min max A B 460 340 ±0.1 ±8 ±20 +12 ±12 ±2 ±10 462 338 ±0.1 ±8.5 ±20 +13 ±12 ±2.5 ±10 Ω Ω %/°C mV µV/°C µA nA/°C µA nA/°C max max max max max max max max max A A B A B A B A B −5.0 3.0 −4.9 2.9 V V max min B A kΩ  pF kΩ  pF typ typ C C V V mA mA mA Ω max max min min typ typ A A A A C C V V mA mA dB min max max min min B A A A A −40 to +85 °C typ C 125 150 °C/W °C/W typ typ C C +10 ±0.1 — ±1.5 −5.4 3.2 −5.2 3.1 10  2.1 400  1.2 Output Shorted to Either Supply G = +2, f ≤ 100kHz ±4.9 ±4.6 85 85 120 0.2 VS = ±5V VS = ±5V Input-Referred ±1.4 — 4.25 4.25 68 RL = 1kΩ to GND RL = 150Ω to GND ±4.8 ±4.5 65 65 ±4.75 ±4.45 60 60 ±4.75 ±4.4 55 55 ±5.5 4.7 4.0 63 ±5.5 5.3 3.6 62 ±5.5 5.9 3.3 61 (1) Junction temperature = ambient for +25°C specifications. (2) Junction temperature = ambient at low temperature limits; junction temperature = ambient +5°C at high temperature limit for over temperature specifications. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of node. 3 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 ELECTRICAL CHARACTERISTICS: VS = +5V Boldface limits are tested at +25°C. At TA = 25°C, G = +2, and RL = 150Ω to VCM = 2V, unless otherwise noted (see Figure 1). OPA832ID, IDBV TYP PARAMETER AC PERFORMANCE (see Figure 1) Small-Signal Bandwidth Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Input Current Noise NTSC Differential Gain NTSC Differential Phase DC PERFORMANCE(4) Gain Error Internal RF and RG, Maximum Minimum Average Drift Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Bias Current Drift Input Offset Current Input Offset Current Drift CONDITIONS +25°C +25°C(1) 0°C to 70°C(2) −40°C to +85°C(2) G = +2, VO ≤ 0.5VPP G = −1, VO ≤ 0.5VPP VO ≤ 0.5VPP G = +2, 2V Step 0.5V Step 0.5V Step G = +2, 1V Step VO = 2VPP, 5MHz RL = 150Ω RL = 500Ω RL = 150Ω RL = 500Ω f > 1MHz f > 1MHz RL = 150Ω RL = 150Ω 92 103 4.2 348 4.3 4.6 4.6 56 60 55 58 55 58 230 223 223 −59 −62 −56 −72 9.3 2.3 0.11 0.14 −56 −59 −50 −65 −56 −59 −49 −62 ±0.3 ±0.2 400 400 ±1.5 ±1.5 455 345 ±0.5 — 5.5 ±5 G = +2 G = −1 VCM = 2.0V VCM = 2.0V INPUT Least Positive Input Voltage Most Positive Input Voltage Input Impedance Differential-Mode Common-Mode OUTPUT Least Positive Output Voltage Most Positive Output Voltage Current Output, Sourcing Current Output, Sinking Short-Circuit Output Current Closed-Loop Output Impedance POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance D SO-8 DBV SOT23-5 (1) MIN/MAX OVER TEMPERATURE UNITS MIN/ MAX TEST LEVEL(3) MHz MHz dB V/µs ns ns ns min min typ min max max max B B C B B B B −55 −59 −47 −58 dBc dBc dBc dBc nV/√Hz pA/√Hz % ° max max max max max max typ typ B B B B B B C C ±1.6 ±1.6 460 340 0.1 ±6 ±20 +12 ±12 ±2 ±10 ±1.7 ±1.7 462 338 0.1 ±6.5 ±20 +13 ±12 ±2.5 ±10 % % Ω Ω %/°C mV µV/°C µA nA/°C µA nA/°C min max max max max max max max max max max A B A A B A B A B A B 0 3.1 +0.1 3.0 V V max min B B kΩ  pF kΩ  pF typ typ C C V V V V mA mA mA Ω max max min min min min typ typ A A A A A A C C V V mA mA dB typ max max min min C A A A A −40 to +85 °C typ C 125 150 °C/W °C/W typ typ C C +10 ±0.1 — ±1.5 −0.5 3.3 −0.2 3.2 10  2.1 400  1.2 RL = 1kΩ to 2.0V RL = 150Ω to 2.0V RL = 1kΩ to 2.0V RL = 150Ω to 2.0V Output Shorted to Either Supply G = +2, f ≤ 100kHz 0.03 0.18 4.94 4.86 80 80 100 0.2 VS = +5V VS = +5V Input-Referred +2.8 — 3.9 3.9 66 0.16 0.3 4.8 4.6 60 60 0.18 0.35 4.6 4.5 55 55 0.20 0.40 4.4 4.4 52 52 +11 4.1 3.7 61 +11 4.8 3.5 60 +11 5.5 3.2 59 Junction temperature = ambient for +25°C specifications. Junction temperature = ambient at low temperature limits; junction temperature = ambient +5°C at high temperature limit for over temperature. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of node. (2) 4 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 ELECTRICAL CHARACTERISTICS: VS = +3.3V Boldface limits are tested at +25°C. At TA = 25°C, G = +2, and RL = 150Ω to VCM = 0.75V, unless otherwise noted (see Figure 2). OPA832ID, IDBV TYP PARAMETER AC PERFORMANCE (see Figure 2) Small-Signal Bandwidth Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Input Current Noise DC PERFORMANCE(4) Gain Error Internal RF and RG Maximum Minimum Average Drift Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Bias Current Drift Input Offset Current Input Offset Current Drift CONDITIONS +25°C +25°C(1) 0°C to 70°C(2) G = +2, VO ≤ 0.5VPP G = −1, VO ≤ 0.5VPP VO ≤ 0.5VPP 1V Step 0.5V Step 0.5V Step 1V Step 5MHz RL = 150Ω RL = 500Ω RL = 150Ω RL = 500Ω f > 1MHz f > 1MHz 95 103 4.2 170 4 4.2 48 59 63 57 61 115 115 −71 −74 −66 −69 9.4 2.4 −64 −70 −60 −66 G = +2 G = −1 ±0.3 ±0.2 UNITS MIN/ MAX TEST LEVEL(3) MHz MHz dB V/µs ns ns ns min min typ min max max max B B C B B B B −62 −66 −55 −62 dBc dBc dBc dBc nV/√Hz pA/√Hz max max max max max max B B B B B B ±1.5 ±1.5 ±1.6 ±1.6 % % min max A B 400 400 455 345 ±1 ±7 VCM = 0.75V — 5.5 +10 VCM = 0.75V ±0.1 ±1.5 460 340 0.1 ±8 ±20 +12 ±12 ±2 ±10 Ω Ω %/°C mV µV/°C µA nA/°C µA nA/°C max max max max max max max max max A A B A B A B A B — INPUT Least Positive Input Voltage Most Positive Input Voltage Input Impedance, Differential-Mode Common-Mode OUTPUT Least Positive Output Voltage Most Positive Output Voltage Current Output, Sourcing Current Output, Sinking Short-Circuit Output Current Closed-Loop Output Impedance POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance D SO-8 DBV SOT23-5 MIN/MAX OVER TEMPERATURE −0.5 1.5 10  2.1 400  1.2 −0.3 1.4 −0.2 1.3 V V kΩ  pF kΩ  pF max min typ typ B B C C 0.16 0.3 2.8 2.8 25 25 0.18 0.35 2.6 2.6 20 20 Output Shorted to Either Supply See Figure 2, f < 100kHz 0.03 0.1 3 3 35 35 80 0.2 V V V V mA mA mA Ω max max min min min min typ typ B B B B A A C C VS = +3.3V VS = +3.3V Input-Referred +2.8 — 3.8 3.8 60 +11 4.0 3.4 +11 4.7 3.2 V V mA mA dB typ max max min typ C A A A C −40 to +85 °C typ C 125 150 °C/W °C/W typ typ C C RL = 1kΩ to 0.75V RL = 150Ω to 0.75V RL = 1kΩ to 0.75V RL = 150Ω to 0.75V (1) Junction temperature = ambient for +25°C specifications. (2) Junction temperature = ambient at low temperature limits; junction temperature = ambient +5°C at high temperature limit for over temperature. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of node. 5 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V At TA = 25°C, G = +2, and RL = 150Ω to GND, unless otherwise noted (see Figure 3). SMALL−SIGNAL FREQUENCY RESPONSE 3 −3 G = −1 −6 −9 G = +2 −6 VO = 4VPP −15 −15 100 VO = 0.5VPP −9 −12 10 VO = 1VPP −3 −12 1 RL = 150Ω G = +2V/V 0 Normalized Gain (dB) 0 Normalized Gain (dB) LARGE−SIGNAL FREQUENCY RESPONSE 3 VO = 0.2VPP RL = 150Ω 500 VO = 2VPP 1 10 Frequency (MHz) SMALL−SIGNAL PULSE RESPONSE G = +2V/V RL = 150Ω VO = 0.2VPP Output Voltage (500mV/div) Output Voltage (50mV/div) 1.5 100 50 0 −50 −100 −150 G = +2V/V RL = 150Ω VO = 2VPP 1.0 0.5 0 −0.5 −1.0 −1.5 Time (10ns/div) Time (10ns/div) FREQUENCY RESPONSE vs CAPACITIVE LOAD Normalized Gain to Capacitive Load (dB) REQUIRED RS vs CAPACITIVE LOAD 40 1dB Peaking Targeted 35 30 25 RS (Ω ) 400 LARGE−SIGNAL PULSE RESPONSE 150 20 15 10 5 0 10 100 Capacitive Load (pF) 6 100 Frequency (MHz) 1k 3 CL = 10pF 0 −3 CL = 1000pF −6 C L = 100pF −9 VI RS OPA832 −12 CL 1kΩ (1) NOTE: (1) 1kΩ is optional. −15 1 10 Frequency (MHz) 100 400 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V (continued) At TA = 25°C, G = +2, and RL = 150Ω to GND, unless otherwise noted (see Figure 3). HARMONIC DISTORTION vs LOAD RESISTANCE G = +2V/V VO = 2VPP f = 5MHz −50 −60 2nd−Harmonic −70 3rd−Harmonic −80 G = +2V/V RL = 500Ω f = 5MHz −60 −70 3rd−Harmonic −80 2nd−Harmonic −90 −100 −90 100 1 −40 2nd−Harmonic 3rd−Harmonic −90 −100 −110 0.1 1 10 −45 8 500Ω 400Ω −55 400Ω −60 −65 −70 20MHz −75 10MHz −80 Power Limit −26 4 RL = 50Ω RL = 100Ω −2 −3 Output 1W Internal Current Limit P ower Limit −80 −40 0 I O (mA) 40 80 120 160 Maximum Output Voltage (V) Output RL = 500Ω −120 −22 −18 −14 −10 −6 −2 2 6 OUTPUT SWING vs LOAD RESISTANCE Current Lim it 0 −1 5MHz −85 5 2 1 10 −90 20 1W In ternal 3 9 Single−Tone Load Power (2dBm/div) 4 VO (V) 7 PO 50Ω OPA832 OUTPUT VOLTAGE AND CURRENT LIMITATIONS −6 −160 6 PI −50 Frequency (MHz) −5 5 TWO−TONE, 3RD−ORDER INTERMODULATION SPURIOUS −80 −4 4 HARMONIC DISTORTION vs FREQUENCY −70 5 3 Output Swing (VPP) −60 6 2 Load Resistance (Ω) G = +2V/V RL = 500Ω VO = 2VPP −50 0 1k 3rd−Order Spurious Level (dBc) −40 Harmonic Distortion (dBc) HARMONIC DISTORTION vs OUTPUT VOLTAGE −50 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −40 G = +2V/V VS = ±5V 3 2 1 0 −1 −2 −3 −4 −5 10 100 1k RL (Ω ) 7 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V At TA = 25°C, G = +2, and RL = 150Ω to VCM = 2V, unless otherwise noted (see Figure 1). SMALL−SIGNAL FREQUENCY RESPONSE 3 R L = 150Ω G = +2V/V 0 −3 Normalized Gain (dB) 0 Normalized Gain (dB) LARGE−SIGNAL FREQUENCY RESPONSE 3 VO = 0.2VPP RL = 150Ω G = −1 −6 −9 G = +2 −12 VO = 2VPP −3 −6 −9 −12 −15 −15 1 10 100 400 1 10 Frequency (MHz) SMALL−SIGNAL PULSE RESPONSE LARGE−SIGNAL PULSE RESPONSE G = +2V/V RL = 150Ω VO = 0.2VPP Output Voltage (500mV/div) Output Voltage (50mV/div) 400 1.5 0.10 0.05 0 −0.05 −0.10 −0.15 G = +2V/V RL = 150Ω VO = 2VPP 1.0 0.5 0 −0.5 −1.0 −1.5 Time (10ns/div) Time (10ns/div) REQUIRED RS vs CAPACITIVE LOAD FREQUENCY RESPONSE vs CAPACITIVE LOAD 1dB Peaking Targeted 35 30 25 20 15 10 5 0 10 100 Capacitive Load (pF) 1k Normalized Gain to Capacitive Load (dB) 40 RS (Ω ) 100 Frequency (MHz) 0.15 8 VO = 1VPP VO = 0.5VPP 3 CL = 10pF 0 −3 CL = 1000pF −6 CL = 100pF −9 −12 VI RS −15 CL 1kΩ (1) NOTE: (1) 1kΩ is optional. −18 1 10 Frequency (MHz) 100 300 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V (continued) At TA = 25°C, G = +2, and RL = 150Ω to VCM = 2V, unless otherwise noted (see Figure 1). HARMONIC DISTORTION vs LOAD RESISTANCE G = +2, HARMONIC DISTORTION vs FREQUENCY −40 G = +2V/V VO = 2VPP f = 5MHz −50 −50 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −40 −60 2nd−Harmonic −70 3rd−Harmonic −80 G = +2V/V RL = 500Ω VO = 2VPP −60 2nd−Harmonic −70 −80 −90 3rd−Harmonic −100 −110 −90 100 1k 0.1 1 Load Resistance (Ω) −40 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −30 G = +2V/V RL = 500Ω f = 5MHz −50 −60 2nd−Harmonic −70 −80 3rd−Harmonic −90 G = −1V/V RL = 500Ω f = 5MHz −50 −60 −70 3rd−Harmonic −80 −90 2nd−Harmonic −100 −100 −110 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 0.1 1 Output Voltage Swing (VPP) −50 INPUT VOLTAGE AND CURRENT NOISE PI 50Ω PO OPA832 500Ω −55 −60 −65 −70 20MHz −75 −80 −85 −90 20 100 Input Voltage Noise (nV/√Hz) Input Current Noise (pA/√Hz) −45 10 Frequency (MHz) TWO−TONE, 3RD−ORDER INTERMODULATION SPURIOUS −40 3rd−Order Spurious Level (dBc) 20 G = −1, HARMONIC DISTORTION vs FREQUENCY HARMONIC DISTORTION vs OUTPUT VOLTAGE −40 10 Frequency (MHz) Voltage Noise (9.3nV/√Hz) 10 Current Noise (2.3nV/√Hz) 10MHz 1 5MHz 100 −24 −22 −20 −18 −16 −14 −12 −10 −8 −6 −4 −2 1k 10k 100k 1M 10M Frequency (Hz) Single−Tone Load Power (dBm) 9 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V (continued) At TA = 25°C, G = +2, and RL = 150Ω to VCM = 2V, unless otherwise noted (see Figure 1). COMPOSITE VIDEO dG/dP COMMON−MODE REJECTION RATIO AND POWER−SUPPLY REJECTION RATIO vs FREQUENCY 1.2 +5V 80 1.0 70 VI 60 0.8 50 dG/dP PSRR and CMRR (dB) Video Loads OPA832 CMRR +PSRR 40 dP 0.6 0.4 30 dG 20 0.2 10 0 0 1 100 1k 10k 100k 1M 10M 2 100M 3 4 Number of 150Ω Loads Frequency (Hz) CLOSED−LOOP OUTPUT IMPEDANCE vs FREQUENCY OUTPUT SWING vs LOAD RESISTANCE Maximum Output Voltage (V) 4.5 400Ω G = +2V/V VS = +5V Output Impedance (Ω) 5.0 100 4.0 3.5 3.0 2.5 2.0 1.5 +5V 400Ω 10 OPA832 ZO 200Ω 1 1.0 0.5 0.1 0 100 1k 1k 10k 100k RL (Ω) 4.5 0.8 4.0 0.6 Most Positive Output Voltage Most Positive Input Voltage 2.5 RL = 150Ω 2.0 1.5 1.0 Least Positive Output Voltage 0.5 0 −0.5 8 Bias Current (IB) 6 0.2 0 4 2 10 × Input Offset (IOS) 0 −0.2 −2 −0.4 −4 Input Offset Voltage (VOS) −0.6 −6 90 −8 −20 −10 0 20 40 60 80 Ambient Temperature (10_C/div) 100 120 130 50 Ambient Temperature (10_ C/div) 10 10 0.4 −1.0 −40 −1.0 0 100M −0.8 Least Positive Input Voltage −50 Input Offset Voltage (mV) Voltage Ranges (V) 1.0 3.0 10M TYPICAL DC DRIFT OVER TEMPERATURE VOLTAGE RANGES vs TEMPERATURE 5.0 3.5 1M Frequency (Hz) Input Bias and Offset Voltage (µA) 10 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V (continued) At TA = 25°C, G = +2, and RL = 150Ω to VCM = 2V, unless otherwise noted (see Figure 1). TYPICAL DC DRIFT OVER TEMPERATURE Input Offset Voltage (mV) 0.8 8 Bias Current (IB) 0.6 6 4 0.4 0.2 0 2 10 × Input Offset (IOS) 0 −2 −0.2 −0.4 −4 Input Offset Voltage (VOS) −0.6 −6 −8 −0.8 −10 −20 0 20 40 60 80 100 120 130 −1.0 −40 Input Bias and Offset Voltage (µA) 10 1.0 Ambient Temperature (10_C/div) 11 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +3.3V At TA = 25°C, G = +2, and RL = 150Ω to VCM = 0.75V, unless otherwise noted (see Figure 2). SMALL−SIGNAL FREQUENCY RESPONSE 3 LARGE−SIGNAL FREQUENCY RESPONSE 3 VO = 0.2VPP RL = 150Ω 0 0 Normalized Gain (dB) G = −1 Normalized Gain (dB) RL = 150Ω G = +2V/V −3 G = +2 −6 −9 −12 VO = 1VPP −3 VO = 0.5VPP −6 −9 −12 −15 VO = 2VPP −15 1 10 100 300 1 10 Frequency (MHz) SMALL−SIGNAL PULSE RESPONSE 2.1 G = +2V/V RL = 150Ω VO = 200mVPP 1.9 Output Voltage (V) Output Voltage (V) 1.60 1.55 1.50 1.45 1.40 G = +2V/V RL = 150Ω VO = 1VPP 1.7 1.5 1.3 1.1 1.35 0.9 Time (10ns/div) Time (10ns/div) REQUIRED RS vs CAPACITIVE LOAD FREQUENCY RESPONSE vs CAPACITIVE LOAD Normalized Gain to Capacitive Load (dB) 60 1dB Peaking Targeted 50 40 RS (Ω) 300 LARGE−SIGNAL PULSE RESPONSE 1.65 30 20 10 0 1 10 100 Capacitive Load (pF) 12 100 Frequency (MHz) 1k 3 CL = 10pF 0 −3 C L = 1000pF −6 CL = 100pF −9 −12 −15 1 10 Frequency (MHz) 100 300 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +3.3V (continued) At TA = 25°C, G = +2, and RL = 150Ω to VCM = 0.75V, unless otherwise noted (see Figure 2). HARMONIC DISTORTION vs OUTPUT VOLTAGE HARMONIC DISTORTION vs LOAD RESISTANCE −40 G = +2V/V VO = 1VPP f = 5MHz −55 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −50 −60 3rd−Harmonic −65 −70 2nd−Harmonic −75 100 1k −60 −70 2nd−Harmonic −80 −90 0.50 1.00 1.25 Output Voltage Swing (V) HARMONIC DISTORTION vs FREQUENCY TWO−TONE, 3RD−ORDER INTERMODULATION SPURIOUS −40 3rd−Order Spurious Level (dBc) G = +2V/V RL = 500Ω VO = 1VPP −60 −70 −80 2nd−Harmonic −90 −100 3rd−Harmonic −110 0.1 0.75 Load Resistance (Ω) 1 10 −45 1.50 PI −50 PO OPA832 50Ω 500Ω −55 −60 −65 −70 −75 20MHz −80 5MHz 10MHz −85 −90 20 −26 −24 Frequency (MHz) −22 −20 −18 −16 −14 −12 −10 −8 Single−Tone Load Power (dBm) OUTPUT SWING vs LOAD RESISTANCE 3.3 G = +2V/V VS = +3.3V 3.0 Maximum Output Voltage (V) Harmonic Distortion (dBc) −50 3rd−Harmonic −100 −80 −40 −50 G = +2V/V RL = 500Ω f = 5MHz 2.7 Most Positive Output Voltage 2.4 2.1 1.8 1.5 1.2 0.9 0.6 Least Positive Output Voltage 0.3 0 10 100 1k RL (Ω ) 13 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 APPLICATIONS INFORMATION WIDEBAND VOLTAGE-FEEDBACK OPERATION The OPA832 is a fixed-gain, high-speed, voltagefeedback op amp designed for single-supply operation (+3V to +10V). It features internal RF and RG resistors which make it easy to select a gain of +2, +1, and −1 without external resistors.The input stage supports input voltages below ground and to within 1.7V of the positive supply. The complementary common-emitter output stage provides an output swing to within 25mV of either supply pin. The OPA832 is compensated to provide stable operation with a wide range of resistive loads. Figure 1 shows the AC-coupled, gain of +2 configuration used for the +5V Specifications and Typical Characteristic Curves. The input impedance matching resistor (66.5Ω) used for testing is adjusted to give a 50Ω input match when the parallel combination of the biasing divider network is included. Voltage swings reported in the Electrical Characteristics are taken directly at the input and output pins. For the circuit of Figure 1, the total effective load on the output at high frequencies is 150Ω || 800Ω. The 332Ω and 499Ω resistors at the noninverting input provide the common-mode bias voltage. Their parallel combination equals the DC resistance at the inverting input (RF RG), reducing the DC output offset due to input bias current. VS = +5V 6.8µF + 499Ω 0.1µF VIN 66.5Ω 0.1µF VCM = 2V 332Ω VOUT OPA832 RL 150Ω RG 400Ω RF 400Ω VCM = 2V VCM = 2V Figure 1. AC-Coupled, G = +2, +5V Single-Supply Specification and Test Circuit Figure 2 shows the AC-coupled, gain of +2 configuration used for the +3.3V Specifications and Typical Characteristic Curves. The input impedance matching resistor (66.5Ω) used for testing is adjusted to give a 50Ω 14 input match when the parallel combination of the biasing divider network is included. Voltage swings reported in the Electrical Characteristics are taken directly at the input and output pins. For the circuit of Figure 2, the total effective load on the output at high frequencies is 150Ω || 800Ω. The 887Ω and 258Ω resistors at the noninverting input provide the common-mode bias voltage. Their parallel combination equals the DC resistance at the inverting input (RF RG), reducing the DC output offset due to input bias current. VS = +3.3V 6.8µF + 887Ω 0.1µF VIN 66.5Ω 0.1µF VCM = 0.75V 258Ω OPA832 VOUT RL 150Ω RG 400Ω RF 400Ω VCM = 0.75V VCM = 0.75V Figure 2. AC-Coupled, G = +2, +3.3V Single-Supply Specification and Test Circuit Figure 3 shows the DC-coupled, gain of +2, dual powersupply circuit configuration used as the basis of the ±5V Electrical Characteristics and Typical Characteristics. For test purposes, the input impedance is set to 50Ω with a resistor to ground and the output impedance is set to 150Ω with a series output resistor. Voltage swings reported in the specifications are taken directly at the input and output pins. For the circuit of Figure 3, the total effective load will be 150Ω || 800Ω. Two optional components are included in Figure 3. An additional resistor (175Ω) is included in series with the noninverting input. Combined with the 25Ω DC source resistance looking back towards the signal generator, this gives an input bias current cancelling resistance that matches the 200Ω source resistance seen at the inverting input (see the DC Accuracy and Offset Control section). In addition to the usual power-supply decoupling capacitors to ground, a 0.01µF capacitor is included between the two power-supply pins. In practical PC board layouts, this optional capacitor will typically improve the 2nd-harmonic distortion performance by 3dB to 6dB. 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 available through the TI web page (www.ti.com). The applications group is also available for design assistance. These models predict typical small signal AC, transient steps, DC performance, and noise under a wide variety of operating conditions. The models include the noise terms found in the electrical specifications of the data sheet. These models do not attempt to distinguish between the package types in their smallsignal AC performance. +5V 0.1µF 6.8µF + 0.01µF 50Ω Source 175Ω GAIN OF +2V/V VIDEO LINE DRIVER VIN 50Ω VOUT OPA832 150Ω RF 400Ω One of the most suitable applications for the OPA832 is a simple gain of +2 video line driver. Figure 4 shows how simple this circuit is to implement, shown as a ±5V implementation. Single +5V operation is similar with blocking caps and DC common-mode biasing provided. RG 400Ω + 6.8µF 0.1µF +5V Video In −5V Figure 3. DC-Coupled, G = +2, Bipolar Supply Specification and Test Circuit DESIGN-IN TOOLS DEMONSTRATION FIXTURES Two printed circuit boards (PCBs) are available to assist in the initial evaluation of circuit performance using the OPA832 in its two package options. Both of these are offered free of charge as unpopulated PCBs, delivered with a user’s guide. The summary information for these fixtures is shown in Table 1. Video Loads OPA832 −5V Optional 1.3kΩ Pull−Down Figure 4. Gain of +2V/V Video Line Driver One optional element is shown in Figure 4. A 1.3kΩ pull-down to the negative supply will improve the differential phase significantly and the differential gain slightly. Figure 5 shows measured dG/dP with and without that pull-down resistor from 1 to 4 video loads. 1.2 +5V Table 1. Demonstration Fixtures by Package LITERATURE PACKAGE NUMBER NUMBER OPA832ID SO-8 DEM-OPA-SO-1A SBOU009 OPA832IDBV SOT23-5 DEM-OPA-SOT-1A SBOU010 The demonstration fixtures can be requested at the Texas Instruments web site (www.ti.com) through the OPA832 product folder. 0.8 dG/dP ORDERING PRODUCT Video In 1.0 V ide o L oa ds OPA832 O ptio nal 1.3kΩ Pull−Down − 5V 0.6 dP dP 0.4 dG 0.2 dG 0 1 2 No Pull−Down With 1.3kΩPull−Down 3 4 Number of 150Ω Loads MACROMODEL AND APPLICATIONS SUPPORT Computer simulation of circuit performance using SPICE is often a quick way to analyze the performance of the OPA832 and its circuit designs. This is particularly true for video and RF amplifier circuits where parasitic capacitance and inductance can play a major role on circuit performance. A SPICE model for the OPA832 is Figure 5. dG/dP vs Video Loads 15 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 CML output impedance, and connected to the transformer center tap, biasing the OPA832s. This input bias voltage is then amplified to provide the correct common-mode voltage to the input of the ADC. Using only 25.1mW power (3.8mA × 2 amplifiers × 3.3V), this configuration (amplifier + ADC) provides greater than 59dB SNR and 70dB SFDR to 2MHz, with all the components running on a low +3.3V supply. SINGLE-SUPPLY ADC INTERFACE The circuit shown in Figure 6 uses the OPA832 as a differential driver followed by an RC filter. In this circuit, the single-ended to differential conversion is realized by a 1:1 transformer driving the noninverting inputs of the two OPA832s. The common-mode level (CML) of the ADS5203 is reduced to the appropriate input level of 0.885V by the network divider composed of R1 and the +3.3V RT 20Ω +3.3V RS 50Ω OPA832 VIN 1:1 RM 50Ω RG 400Ω 50Ω Source RF 400Ω IN 1/2 ADS5203 10−Bit 40MSPS C 15pF +3.3V RT 20Ω RS 50Ω OPA832 IN CML RG 400Ω 2.3kΩ Output Impedance RF 400Ω VCM = 0.885V RI 1.91kΩ Figure 6. Low-Power, Single-Supply ADC Driver 16 C1 0.1µF 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 This circuit removes the peaking by bootstrapping out any parasitic effects on RG. The input impedance is still set by RM as the apparent impedance looking into RG is very high. RM may be increased to show a higher input impedance, but larger values will start to impact DC output offset voltage. This circuit creates an additional input offset voltage as the difference in the two input bias current times the impedance to ground at VIN. Figure 8 shows a comparison of small-signal frequency response for the unity-gain buffer of Figure 2 (with VCM removed from RG) compared to the improved approach shown in Figure 7. +5V RO 75Ω RG 400Ω RF 400Ω VIN RM 50Ω Figure 7. Improved Unity-Gain Buffer UNITY-GAIN BUFFER This buffer can simply be realized by not connecting RG to ground. This type of realization shows a peaking in the frequency response. A similar circuit that holds a flat frequency response giving improved pulse fidelity is shown in Figure 7. 6 3 G = +1 Buffer RG Floating Gain (dB) 0 −3 −6 G = +1 Buffer Figure 5 −9 −12 1 10 GAIN SETTING Setting the gain for the OPA832 is very easy. For a gain of +2, ground the −IN pin and drive the +IN pin with the signal. For a gain of +1, either leave the −IN pin open and drive the +IN pin or drive both the +IN and −IN pins as shown in Figure 7. For a gain of −1, ground the +IN pin and drive the −IN pin with the input signal. An external resistor may be used in series with the −IN pin to reduce the gain. However, since the internal resistors (RF and RG) have a tolerance and temperature drift different than the external resistor, the absolute gain accuracy and gain drift over temperature will be relatively poor compared to the previously described standard gain connections using no external resistor. OUTPUT CURRENT AND VOLTAGES VOUT OPA832 OPERATING SUGGESTIONS 100 Frequency (MHz) Figure 8. Buffer Frequency Response Comparison 400 The OPA832 provides outstanding output voltage capability. For the +5V supply, under no-load conditions at +25°C, the output voltage typically swings closer than 60mV to either supply rail. The minimum specified output voltage and current specifications over temperature are set by worst-case simulations at the cold temperature extreme. Only at cold startup will the output current and voltage decrease to the numbers shown in the min/max tables. As the output transistors deliver power, their junction temperatures will increase, decreasing their VBEs (increasing the available output voltage swing) and increasing their current gains (increasing the available output current). In steady-state operation, the available output voltage and current will always be greater than that shown in the over-temperature specifications, since the output stage junction temperatures will be higher than the minimum specified operating ambient. To maintain maximum output stage linearity, no output short-circuit protection is provided. This will not normally be a problem, since most applications include a series matching resistor at the output that will limit the internal power dissipation if the output side of this resistor is shorted to ground. However, shorting the output pin directly to the adjacent positive power-supply pin (8-pin packages) will possibly destroy the amplifier. If additional short-circuit protection is required, consider a small series resistor in the power-supply leads. This will reduce the available output voltage swing under heavy output loads. DRIVING CAPACITIVE LOADS One of the most demanding and yet very common load conditions for an op amp is capacitive loading. Often, the capacitive load is the input of an ADC—including additional external capacitance which may be recommended to improve ADC linearity. A high-speed, high 17 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 The criterion for setting this RS resistor is a 1dB peaked frequency response at the load. Increasing the noise gain will also reduce the peaking (see Figure 7). DISTORTION PERFORMANCE The OPA832 provides good distortion performance into a 150Ω load. Relative to alternative solutions, it provides exceptional performance into lighter loads and/or operating on a single +3.3V supply. Generally, until the fundamental signal reaches very high frequency or power levels, the 2nd-harmonic will dominate the distortion with a negligible 3rd-harmonic component. Focusing then on the 2nd-harmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network; in the noninverting configuration (see Figure 3) this is sum of RF + RG, while in the inverting configuration, only RF needs to be included in parallel with the actual load. Figure 9 shows the 2nd- and 3rd-harmonic distortion versus supply voltage. In order to maintain the input signal within acceptable operating range, the input common-mode voltage is adjusted for each supply voltage. For example, the common-mode voltage is +2V for a single +5V supply, and the distortion is −66.5dBc for the 2nd-harmonic and −74.6dBc for the 3rd-harmonic. 5.0 −68 4.5 −69 4.0 −70 3.5 −71 3.0 2nd−Harmonic Left Scale −72 2.5 −73 −74 −76 G = +2V/V RL = 500Ω VO = 2VPP f = 5MHz 3rd−Harmonic Left Scale −75 2.0 1.5 1.0 0.5 5 6 7 8 9 10 11 Supply Voltage (V) Figure 9. 5MHz Harmonic Distortion vs Supply Voltage NOISE PERFORMANCE Unity-gain stable, rail-to-rail (RR) output, voltage-feedback op amps usually show a higher input noise voltage. The 9.2nV/√Hz input voltage noise for the OPA832 however, is much lower than comparable amplifiers. The input-referred voltage noise and the two input-referred current noise terms (2.8pA/√Hz) combine to give low output noise under a wide variety of operating conditions. Figure 10 shows the op amp noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. ENI EO OPA832 RS IBN ERS RF √ 4kTRS RG 4kT RG √ 4kTRF I BI 4kT = 1.6E − 20J at 290_K Figure 10. Noise Analysis Model The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 1 shows the general form for the output noise voltage using the terms shown in Figure 10: EO + 18 5.5 Common−Mode Voltage Right Scale −67 Common−Mode Voltage (V) The Typical Characteristic curves show the recommended RS versus capacitive load and the resulting frequency response at the load. Parasitic capacitive loads greater than 2pF can begin to degrade the performance of the OPA832. Long PC board traces, unmatched cables, and connections to multiple devices can easily exceed this value. Always consider this effect carefully, and add the recommended series resistor as close as possible to the output pin (see the Board Layout Guidelines section). −66 Harmonic Distortion (dBc) open-loop gain amplifier like the OPA832 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. Ǹǒ Ǔ E NI ) ǒI BNRSǓ ) 4kTRS NG 2 ) ǒI BIR FǓ ) 4kTRFNG 2 2 2 (1) 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 Dividing this expression by the noise gain (NG = (1 + RF/RG)) will give the equivalent input-referred spot noise voltage at the noninverting input, as shown in Equation 2: EN + Ǹ ENI ) ǒIBNR SǓ ) 4kTRS ) 2 2 ǒ Ǔ IBIRF NG 2 ) 4kTRF NG (2) Evaluating these two equations for the circuit and component values shown in Figure 1 will give a total output spot noise voltage of 19.3nV/√Hz and a total equivalent input spot noise voltage of 9.65nV/√Hz. This is including the noise added by the resistors. This total input-referred spot noise voltage is not much higher than the 9.2nV/√Hz specification for the op amp voltage noise alone. DC ACCURACY AND OFFSET CONTROL The balanced input stage of a wideband voltage-feedback op amp allows good output DC accuracy in a wide variety of applications. The power-supply current trim for the OPA832 gives even tighter control than comparable products. Although the high-speed input stage does require relatively high input bias current (typically 5µA out of each input terminal), the close matching between them may be used to reduce the output DC error caused by this current. This is done by matching the DC source resistances appearing at the two inputs. Evaluating the configuration of Figure 3 (which has matched DC input resistances), using worst-case +25°C input offset voltage and current specifications, gives a worstcase output offset voltage equal to: (NG = noninverting signal gain at DC) ±(NG × VOS(MAX)) ± (RF × IOS(MAX)) = ±(2 × 10mV) ± (400Ω × 1.5µA) = ±10.6mV A fine-scale output offset null, or DC operating point adjustment, is often required. Numerous techniques are available for introducing DC offset control into an op amp circuit. Most of these techniques are based on adding a DC current through the feedback resistor. In selecting an offset trim method, one key consideration is the impact on the desired signal path frequency response. If the signal path is intended to be noninverting, the offset control is best applied as an inverting summing signal to avoid interaction with the signal source. If the signal path is intended to be inverting, applying the offset control to the noninverting input may be considered. Bring the DC offsetting current into the inverting input node through resistor values that are much larger than the signal path resistors. This will insure that the adjustment circuit has minimal effect on the loop gain and hence the frequency response. THERMAL ANALYSIS Maximum desired junction temperature will set the maximum allowed internal power dissipation, as described below. In no case should the maximum junction temperature be allowed to exceed 150°C. Operating junction temperature (TJ) is given by TA + PD × θJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power. Quiescent power is simply the specified noload supply current times the total supply voltage across the part. PDL will depend on the required output signal and load; though, for resistive loads connected to mid-supply (VS/2), PDL is at a maximum when the output is fixed at a voltage equal to VS/4 or 3VS/4. Under this condition, PDL = VS2/(16 × RL), where RL includes feedback network loading. Note that it is the power in the output stage, and not into the load, that determines internal power dissipation. As a worst-case example, compute the maximum TJ using an OPA832 (SOT23-5 package) in the circuit of Figure 3 operating at the maximum specified ambient temperature of +85°C and driving a 150Ω load at midsupply. PD = 10V × 3.9mA + 52/(16 × (150Ω || 400Ω)) = 53.3mW Maximum TJ = +85°C + (0.053W × 150°C/W) = 93°C. Although this is still well below the specified maximum junction temperature, system reliability considerations may require lower ensured junction temperatures. The highest possible internal dissipation will occur if the load requires current to be forced into the output at high output voltages or sourced from the output at low output voltages. This puts a high current through a large internal voltage drop in the output transistors. BOARD LAYOUT GUIDELINES Achieving optimum performance with a high-frequency amplifier like the OPA832 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability: on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance ( < 0.25”) from the power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and power-plane layout 19 
"#$ www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. Each power-supply connection should always be decoupled with one of these capacitors. An optional supply decoupling capacitor (0.1µF) across the two power supplies (for bipolar operation) will improve 2ndharmonic distortion performance. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. the input impedance of the destination device; this total effective impedance should be set to match the trace impedance. If the 6dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the typical characteristic curve Recommended RS vs Capacitive Load. This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. c) Careful selection and placement of external components will preserve the high-frequency performance. Resistors should be a very low reactance type. Surfacemount resistors work best and allow a tighter overall layout. Metal film or carbon composition axially-leaded resistors can also provide good high-frequency performance. Again, keep their leads and PC board traces as short as possible. Never use wire-wound type resistors in a high-frequency application. Since the output pin is the most sensitive to parasitic capacitance, always position the series output resistor, if any, as close as possible to the output pin. Other network components, such as noninverting input termination resistors, should also be placed close to the package. e) Socketing a high-speed part is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA832 onto the board. d) Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the typical characteristic curve Recommended RS vs Capacitive Load. Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA832 is nominally compensated to operate with a 2pF parasitic load. Higher parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin). If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary onboard, and in fact, a higher impedance environment will improve distortion as shown in the distortion versus load plots. With a characteristic board trace impedance defined (based on board material and trace dimensions), a matching series resistor into the trace from the output of the OPA832 is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and 20 INPUT AND ESD PROTECTION The OPA832 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power supplies, as shown in Figure 11. +VCC External Pin Internal Circuitry − VCC Figure 11. Internal ESD Protection These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (that is, in systems with ±15V supply parts driving into the OPA832), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible, since high values degrade both noise performance and frequency response. www.ti.com SBOS266E − JUNE 2003 − REVISED AUGUST 2008 Revision History DATE REV PAGE SECTION 8/08 E 2 Absolute Maximum Ratings 3/06 D 15 Design-In Tools DESCRIPTION Changed Storage Temperature minimum value from −40°C to −65°C. Board part number changed. NOTE: Page numbers for previous revisions may differ from page numbers in the current version. 21 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) OPA832ID ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA 832 OPA832IDBVR ACTIVE SOT-23 DBV 5 3000 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 A74 OPA832IDBVT ACTIVE SOT-23 DBV 5 250 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 A74 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of