OPA843
OPA
843
www.ti.com
SBOS268C – DECEMBER 2002 – DECEMBER 2008
Wideband, Low Distortion, Medium Gain,
Voltage-Feedback OPERATIONAL AMPLIFIER
FEATURES
DESCRIPTION
● HIGH BANDWIDTH: 260MHz (G = +5)
The OPA843 provides a level of speed and dynamic range
previously unattainable in a monolithic op amp. Using a high
Gain Bandwidth (GBW), two gain-stage design, the OPA843
gives a medium gain range device with exceptional dynamic
range. The “classic” differential input complements this high
dynamic range with DC precision beyond most high-speed
amplifier products. Very low input offset voltage and current,
high Common-Mode Rejection Ratio (CMRR) and PowerSupply Rejection Ratio (PSRR), and high open-loop gain
combine to give a high DC precision amplifier along with low
noise and high 3rd-order intercept.
● GAIN BANDWIDTH PRODUCT: 800MHz
● LOW INPUT VOLTAGE NOISE: 2.0nV/ √Hz
● VERY LOW DISTORTION: –96dBc (5MHz)
● HIGH OPEN-LOOP GAIN: 110dB
● FAST 12-BIT SETTLING: 10.5ns (0.01%)
● LOW INPUT OFFSET VOLTAGE: 300µV
● OUTPUT CURRENT: ±100mA
12- to 16-bit converter interfaces will benefit from this combination of features. High-speed transimpedance applications
can be implemented with exceptional DC precision as well.
Differential configurations using two OPA843s can deliver
very low distortion to high output voltages, as shown below.
APPLICATIONS
● ADC/DAC BUFFER AMPLIFIER
● LOW DISTORTION “IF” AMPLIFIER
● ACTIVE FILTERS
● LOW-NOISE RECEIVER
OPA843 RELATED PRODUCTS
● WIDEBAND TRANSIMPEDANCE
● TEST INSTRUMENTATION
● PROFESSIONAL AUDIO
● OPA643 UPGRADE
SINGLES
INPUT NOISE
VOLTAGE (nV/ √Hz )
GAIN-BANDWIDTH
PRODUCT (MHz)
OPA842
OPA846
OPA847
2.6
1.2
0.85
200
1750
3900
+5V
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
OPA843
–85
VI
50Ω
1:1
132Ω
402Ω
40.2Ω
402Ω
RL
400Ω
Harmonic Distortion (dBc)
–5V
40.2Ω
VO = 10VI
+5V
GD = 10
RL = 400Ω
F = 5MHz
–90
–95
2nd-Harmonic
–100
3rd-Harmonic
–105
OPA843
–110
1
–5V
10
Output Voltage Swing (Vp-p)
Very Low Distortion Differential Driver
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright © 2002-2008, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
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ELECTROSTATIC
DISCHARGE SENSITIVITY
ABSOLUTE MAXIMUM RATINGS(1)
Power Supply ............................................................................... ±6.5VDC
Internal Power Dissipation ...................................... See Thermal Analysis
Differential Input Voltage .................................................................. ±1.2V
Input Voltage Range ............................................................................ ±VS
Storage Voltage Range: D, DBV ................................... –65°C to +125°C
Lead Temperature (soldering, 10s) ............................................... +300°C
Junction Temperature (TJ) ............................................................ +150°C
ESD Rating (Human Body Model) .................................................. 2000V
(Charge Device Model) ............................................... 1500V
(Machine Model) ........................................................... 200V
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
NOTE: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.
PACKAGE/ORDERING INFORMATION(1)
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
SO-8
D
–40°C to +85°C
OPA843
OPA843ID
Rails, 100
"
"
"
"
OPA843IDR
Tape and Reel, 2500
SOT23-5
DBV
–40°C to +85°C
OARI
OPA843IDBVT
Tape and Reel, 250
"
"
"
"
OPA843IDBVR
Tape and Reel, 3000
OPA843
"
OPA843
"
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI website at www.ti.com.
PIN CONFIGURATIONS
8
NC
Inverting Input
2
7
+VS
Noninverting Input
3
6
Output
–VS
4
5
NC
Output
1
–VS
2
Noninverting Input
3
NC = No Connection
5
+VS
4
Inverting Input
1
OARI
3
1
SOT
2
NC
Top View
4
SO
5
Top View
Pin Orientation/Package Marking
2
OPA843
www.ti.com
SBOS268C
ELECTRICAL CHARACTERISTICS: VS = ±5V
Boldface limits are tested at +25°C.
At TA = +25°C, VS = ±5V, RF = 402Ω, RL = 100Ω, and G = +5, unless otherwise noted. See Figure 1 for AC performance.
OPA843ID, OPA843IDBV
TYP
PARAMETER
AC PERFORMANCE (see Figure 1)
Small-Signal Bandwidth (VO = 200mVPP)
Gain-Bandwidth Product
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +3
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
2-Tone, 3rd-Order Intercept
Input Voltage Noise
Input Current Noise
Rise-and-Fall Time
Slew Rate
Settling Time to 0.01%
0.1%
1.0%
Differential Gain
Differential Phase
DC PERFORMANCE(4)
Open-Loop Voltage Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Input Bias Current Drift
Input Offset Current
Input Offset Current Drift
INPUT
Common-Mode Input Range (CMIR)(5)
Common-Mode Rejection (CMRR)
Input Impedance
Differential-Mode
Common-Mode
OUTPUT
Output Voltage Swing
Current Output
Closed-Loop Output Impedance
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage
Minimum Operating Voltage
Max Quiescent Current
Min Quiescent Current
Power-Supply Rejection Ratio
(+PSRR, –PSRR)
THERMAL CHARACTERISTICS
Specified Operating Range: D, DBV
Thermal Resistance, θJA
D
SO-8
DBV SOT23-5
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C
+25°C(1)
0°C to
70°C
–40°C to
+85°C(2)
G = +3
G = +5
G = +10
G = +20
500
260
85
40
800
65
3.5
185
66
30
562
34
180
65
30
560
33
175
64
30
558
32
–74
–94
–100
–105
–72
–92
–98
–102
–70
–90
–95
–100
2.2
3.35
1.95
650
2.31
3.4
2.0
600
2.36
3.45
2.1
525
10
5.4
10.3
5.8
10.6
6.4
100
96
±1.4
±4
–36
25
±1.15
±2
±2.9
84
G = +5, RL = 100Ω, VO = 200mVPP
G = +5, f = 5MHz, VO = 2VPP
RL = 100Ω
RL = 500Ω
RL = 100Ω
RL = 500Ω
G = +5, f = 25MHz
f > 1MHz
f > 1MHz
0.2V Step
2V Step
2V Step
2V Step
2V Step
G = +4, NTSC, RL = 150Ω
G = +4, NTSC, RL = 150Ω
–76
–96
–102
–110
40
2.0
2.8
1.2
1000
10.5
7.5
3.2
0.001
0.012
VO = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
110
±0.30
±1.20
–20
–35
±0.25
±1.0
VCM = ±1V, Input Referred
±3.2
95
±3.0
VCM = 0V
VCM = 0V
12 || 1
3.2 || 1.2
RL > 1kΩ, Positive Output
RL > 1kΩ, Negative Output
RL = 100Ω, Positive Output
RL = 100Ω, Negative Output
VO = 0V
G = +5, f = 1kHz
3.2
–3.7
3.0
–3.5
±100
0.0001
85
UNITS
MIN/
MAX
TEST
LEVEL(3 )
MHz
MHz
MHz
MHz
MHz
MHz
dB
typ
min
min
min
min
min
typ
C
B
B
B
B
B
C
dBc
dBc
dBc
dBc
dBm
nV/√Hz
pA/√Hz
ns
V/µs
ns
ns
ns
%
deg
max
max
max
max
typ
max
max
max
min
typ
max
max
typ
typ
B
B
B
B
C
B
B
B
B
C
B
B
C
C
92
±1.5
±4
–37
25
±1.17
±2
dB
mV
µV/°C
µA
nA/°C
µA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
±2.8
82
V
dB
min
min
A
A
kΩ || pF
MΩ || pF
typ
typ
C
C
V
V
V
V
mA
Ω
min
min
min
min
min
typ
A
A
A
A
A
C
3.0
–3.5
2.8
–3.3
±90
2.9
–3.4
2.7
–3.2
±85
2.8
–3.3
2.6
–3.1
±80
±6
±4
±6
±4
22.5
18.3
V
V
V
mA
mA
typ
max
min
max
min
C
A
A
A
A
85
dB
min
A
–40 to +85
°C
typ
C
125
150
°C
°C
typ
typ
C
C
±5
VS = ±5V
VS = ±5V
20.2
20.2
20.8
19.6
±6
±4
22.2
19.1
|VS| = 4.5V to 5.5V, Input Referred
100
90
88
Junction-to-Ambient
NOTES: (1) Junction temperature = ambient temperature for 25°C min/max specifications. (2) Junction temperature = ambient at low temperature limit: junction
temperature = ambient +23°C at high temperature limit for over temperature min/max specifications. (3) Test Levels: (A) 100% tested at 25°C over-temperature
limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive outof-node. VCM is the input common-mode voltage. (5) Tested < 3dB below minimum specified CMRR at ±CMIR limits.
OPA843
SBOS268C
www.ti.com
3
TYPICAL CHARACTERISTICS: VS = ±5V
TA = +25°C, G = +5, RF = 402Ω, RG = 100Ω, and RL = 100Ω, unless otherwise noted.
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
6
VO = 0.2Vp-p
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
3
G = +3
3
Normalized Gain (dB)
Normalized Gain (dB)
G = +5
0
–3
G = +10
–6
G = +20
–9
–12
–3
RG = RS = 50Ω
VO = 0.2Vp-p
–6
–9
G = –16
–12
G = –32
–15
–15
See Figure 1
See Figure 2
–18
–18
106
17
14
107
109
108
106
107
Frequency (Hz)
NONINVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
21
RL = 100Ω
G = +5V/V
0.2Vp-p
1Vp-p
2Vp-p
18
200mVp-p
to 1Vp-p
RL = 100Ω
G = –8V/V
15
109
108
Frequency (Hz)
12
Gain (dB)
11
Gain (dB)
G = –4
G = –8
0
5Vp-p
8
2Vp-p
9
5Vp-p
6
3
5
0
2
–3
See Figure 1
108
107
109
108
Frequency (Hz)
Frequency (Hz)
NONINVERTING PULSE RESPONSE
INVERTING PULSE RESPONSE
Right Scale
100
G = –8
1.2
0.8
0.4
0
Small Signal ± 100mV
–100
Left Scale
–200
0
–0.4
–0.8
Output Voltage (100mV/div)
200
Large Signal ± 1V
Output Voltage (400mV/div)
G = +5
200
Large Signal ± 1V
Right Scale
100
1.2
0.8
0.4
0
Small Signal ± 100mV
–100
Left Scale
–200
0
–0.4
–0.8
–1.2
–1.2
See Figure 2
See Figure 1
Time (2ns/div)
4
109
Time (2ns/div)
OPA843
www.ti.com
SBOS268C
Output Voltage (400mV/div)
107
Output Voltage (100mV/div)
See Figure 2
–6
–1
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +5, RF = 402Ω, RG = 100Ω, and RL = 100Ω, unless otherwise noted.
5MHz HARMONIC DISTORTION vs LOAD RESISTANCE
1MHz HARMONIC DISTORTION vs LOAD RESISTANCE
–75
–75
VO = 2Vp-p
G = +5
VO = 5Vp-p
G = +5
–80
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–80
–85
2nd-Harmonic
–90
–95
–100
3rd-Harmonic
–105
–85
–90
2nd-Harmonic
–95
–100
3rd-Harmonic
–105
See Figure 1
See Figure 1
–110
–110
100
150
200
250
300
350
400
450
500
100
150
200
Resistance (Ω)
–70
2nd-Harmonic
–80
–90
3rd-Harmonic
450
500
–80
–85
2nd-Harmonic
–90
–95
3rd-Harmonic
–100
–105
See Figure 1
See Figure 1
–110
–110
1
10
0.1
100
1
10
Output Voltage Swing (Vp-p)
Frequency (MHz)
HARMONIC DISTORTION vs INVERTING GAIN
HARMONIC DISTORTION vs NONINVERTING GAIN
–75
–70
2nd-Harmonic
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
400
RL = 200Ω
F = 5MHz
G = +5
–75
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
VO = 2Vp-p
G = +5
RL = 200Ω
0.1
350
HARMONIC DISTORTION vs OUTPUT VOLTAGE
HARMONIC DISTORTION vs FREQUENCY
–100
300
Resistance (Ω)
–60
–70
250
–80
VO = 2Vp-p
RL = 200Ω
F = 5MHz
RF = 402Ω, RG Adjusted
–90
–100
3rd-Harmonic
10
15
–95
–105
3rd-Harmonic
–115
5
20
10
15
20
25
30
35
40
Gain (–V/V)
Gain (V/V)
OPA843
SBOS268C
VO = 2Vp-p
RL = 200Ω
F = 5MHz
RG = 50Ω, RF Adjusted
See Figure 2
See Figure 1
–110
5
2nd-Harmonic
–85
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5
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +5, RF = 402Ω, RG = 100Ω, and RL = 100Ω, unless otherwise noted.
2-TONE, 3RD-ORDER
INTERMODULATION INTERCEPT
INPUT VOLTAGE AND CURRENT NOISE DENSITY
10
55
PI
G = +5
Current Noise
Intercept Point (+dBm)
Voltage Noise nV/√Hz
Current Noise pA/√Hz
50
2.8pA/√Hz
Voltage Noise
2.0nV/√Hz
50Ω
PO
OPA843
50Ω
50Ω
402Ω
45
100Ω
40
35
30
1
25
102
103
104
105
106
107
10
15
20
25
30
NONINVERTING GAIN FLATNESS TUNE
VO = 200mVp-p
AV = +4
0.30
0.20
NG = 4
NG = 4.5
0
–0.10
NG = 5
–0.20
NG = 5.5
–0.30
55
60
65
70
1
G = –1
0
–1
–2
G = –2
–3
–4
External Compensation
See Figure 11
–5
G = –3
–6
1
10
100
1k
1
10
Frequency (MHz)
Normalized Gain to Capacitive Load (dB)
10
1
10
100
1k
FREQUENCY RESPONSE vs CAPACITIVE LOAD
G = +5
1
100
Frequency (MHz)
RECOMMENDED RS vs CAPACITIVE LOAD
100
RS (Ω)
50
VO = 200mVp-p
2
–0.40
17
RS adjusted to cap load.
14
C = 10pF
C = 100pF
11
C = 22pF
VI
8
50Ω
RS
VO
OPA843
CL
C = 47pF
1kΩ
402Ω
5
100Ω
1kΩ is optional.
2
106
1k
107
108
109
Frequency (Hz)
Capacitive Load (pF)
6
45
LOW GAIN INVERTING BANDWIDTH
External Compensation
See Figure 10
0.10
40
3
Normalized Gain (1dB/div)
Deviation from 12dB Gain (0.1dB/div)
0.40
35
Frequency (MHz)
Frequency (Hz)
OPA843
www.ti.com
SBOS268C
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +5, RF = 402Ω, RG = 100Ω, and RL = 100Ω, unless otherwise noted.
OPEN-LOOP GAIN AND PHASE
CMRR AND PSRR vs FREQUENCY
120
CMRR
80
–PSRR
60
40
–30
20log (AOL)
80
–60
∠AOL
60
20
40
–120
20
–150
0
–180
–20
102
103
104
105
106
107
103
104
107
108
CLOSED-LOOP OUTPUT IMPEDANCE
vs FREQUENCY
109
10
1
1W Internal
Power Limit
Output Impedance (Ω)
RL = 100
RL = 50
0
RL = 25
–1
1W Internal
Power Limit
–2
OPA843
ZO
0.1
402Ω
0.01
100Ω
0.001
0.0001
–3
0.00001
–0.1
–0.05
0
0.05
0.1
102
0.15
103
104
NONINVERTING OVERDRIVE RECOVERY
4
3
Output
Left Scale
106
107
108
RL = 100Ω
G=5
See Figure 1
INVERTING OVERDRIVE RECOVERY
1
5
0.8
4
0.6
3
0.4
1
0.2
0
0
–1
–0.2
–2
–0.4
–3
–0.6
–4
–5
Output Voltage (1V/div)
Input
Right Scale
Input Voltage (200mV/div)
5
105
Frequency (Hz)
IO (mA)
2
1
RL = 100Ω
G = –8
See Figure 2
Input
Right Scale
0.6
0.2
0
–1
0.8
0.4
1
0
Output
Left Scale
–0.2
–2
–0.4
–3
–0.6
–0.8
–4
–0.8
–1
–5
Time (40ns/div)
–1
Time (40ns/div)
OPA843
SBOS268C
106
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
2
2
105
Frequency (Hz)
3
–4
–0.15
102
Frequency (Hz)
4
1
–210
101
108
Input Voltage (200mV/div)
101
Output Voltage (1V/div)
–90
Open-Loop Phase (°)
100
100
0
VO (V)
0
+PSRR
Open-Loop Gain (dB)
Common-Mode Rejection Ratio (dB)
Power-Supply Rejection Ratio (dB)
120
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7
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +5, RF = 402Ω, RG = 100Ω, and RL = 100Ω, unless otherwise noted.
SETTLING TIME
VIDEO DIFFERENTIAL GAIN/DIFFERENTIAL PHASE
0.20
Differential Gain (%)
0.15
0.1
G = +4
0.10
0.05
0
–0.05
–0.10
–0.15
0.015
0.075
dP, Negative Video
0.01
0.05
dP, Positive Video
0.005
0.025
dG, Positive Video
–0.20
See Figure 1
–0.25
0
0
5
10
15
20
0
25
1
2
Time (ns)
TYPICAL DC DRIFT OVER TEMPERATURE
VIO
0
0
100 x IOS
–0.5
–12.5
IB
–1
Output Current (2mA/div)
12.5
25
50
75
100
22
106
20
Sourcing Output Current
98
19
Sinking Output Current
94
18
90
125
17
–50
–25
Ambient Temperature (°C)
0
25
50
75
100
125
Ambient Temperature (°C)
COMMON-MODE INPUT RANGE AND OUTPUT SWING
vs SUPPLY VOLTAGE
COMMON-MODE AND DIFFERENTIAL
INPUT IMPEDANCE
107
6
Impedance Magnitude (20log (Ω))
Positive Input
4
Voltage Range (V)
21
Supply Current
102
–25
0
5
110
Input Bias and Offset Current (µA)
Input Offset Voltage (mV)
0.5
–25
4
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
25
–50
3
Video Loads (150Ω each)
1
2
Positive Output
0
–2
Negative Output
–4
Negative Input
Common-Mode
106
105
Differential
104
103
102
–6
3
4
5
103
6
104
105
106
107
108
Frequency (Hz)
Supply Voltage (±V)
8
dG, Negative Video
OPA843
www.ti.com
SBOS268C
Supply Current (1mA/div)
Percent of Final Value (%)
0.02
RL = 100Ω
VO = 2V step
G = +5
Differential Phase (°)
0.25
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, GD = 10, RF = 1kΩ, RG = 100Ω, and RL = 100Ω, unless otherwise noted.
DIFFERENTIAL PERFORMANCE
TEST CIRCUIT
DIFFERENTIAL SMALL-SIGNAL
FREQUENCY RESPONSE
3
VO = 400mVp-p
+5V
GD = 5
0
OPA843
RG
100Ω
VI
RG
100Ω
Normalized Gain (dB)
RF
GD =
100Ω
–5V
RF
RL
RF
VO
+5V
GD = 10
–3
GD = 16
–6
GD = 32
–9
–12
–15
OPA843
–18
1
10
DIFFERENTIAL LARGE-SIGNAL
FREQUENCY RESPONSE
VO = 400mVp-p to 5Vp-p
17
VO = 8Vp-p
14
11
VO = 4Vp-p
GD = 10
F = 5MHz
–75
Harmonic Distortion (dBc)
20
Gain (dB)
1k
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
–70
23
GD = 10V/V
100
Frequency (MHz)
–5V
–80
–85
2nd-Harmonic
–90
–95
–100
3rd-Harmonic
–105
–110
8
10
100
10
200
100
DIFFERENTIAL DISTORTION vs FREQUENCY
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
–60
GD = 10
RL = 400Ω
F = 5MHz
–85
Harmonic Distortion (dBc)
Gain (dB)
–80
VO = 4Vp-p
GD = 10
RL = 400Ω
–70
1k
Load Resistance (Ω)
Frequency (MHz)
–80
2nd-Harmonic
–90
–100
–90
–95
2nd-Harmonic
–100
3rd-Harmonic
–105
–110
3rd-Harmonic
–110
–115
1
10
20
1
Frequency (Hz)
OPA843
SBOS268C
10
Output Voltage Swing (Vp-p)
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9
APPLICATIONS INFORMATION
+5V
WIDEBAND NONINVERTING OPERATION
The OPA843’s combination of speed and dynamic range is
useful in a wide variety of application circuits, as long as
simple guidelines common to all high-speed amplifiers are
observed. For example, good power-supply decoupling, as
shown in Figure 1, is essential to achieve the lowest possible
harmonic distortion and smooth frequency response. Careful
PC board layout and component selection will maximize the
performance of the OPA843 in all applications, as discussed
in the following sections of this data sheet. Figure 1 shows
the gain of +5 configuration used as the basis for most of the
Typical Characteristics. Most of the curves were characterized using signal sources with 50Ω driving impedance and
with measurement equipment presenting 50Ω load impedance. In Figure 1, the 50Ω shunt resistor at the input terminal
matches the source impedance of the test generator, while
the 50Ω series resistor at the VO terminal provides a matching resistor for the measurement equipment load. Generally,
data sheet specifications refer to the voltage swing at the
output pin (VO in Figure 1) while those referring to load power
are at the 50Ω load. The total 100Ω load from the series and
shunt matching resistors, combined with the 502Ω total
feedback network load, presents the OPA843 with an effective output load of approximately 83Ω.
+5V
+VS
0.1µF
+
2.2µF
50Ω Source
VIN
50Ω
RS 50Ω Load
50Ω
VO
OPA843
RF
402Ω
RG
100Ω
0.1µF
+
2.2µF
–VS
–5V
FIGURE 1. Gain of +5, High-Frequency Application and
Characterization Circuit.
0.1µF
RT
50Ω Source
+
RS 50Ω Load
50Ω
VO
OPA843
2.2µF
RF
402Ω
RG
50Ω
VI
RM
(optional)
0.1µF
+
2.2µF
–5V
FIGURE 2. Inverting G = –8 Specification and Test Circuit.
both the input termination resistor and the gain setting
resistor for the circuit. Although the signal gain for the circuit
of Figure 2 is equal to –8V/V (versus the +5V/V for Figure 1),
their noise gains are equal when the 50Ω source resistor is
included. This has the interesting effect of nearly doubling
the equivalent Gain Bandwidth Product (GBP) for the amplifier. This can be seen in comparing the G = +5 and G = –8
small-signal frequency response curves. Both show approximately 260MHz bandwidth, but the inverting configuration of
Figure 2 is giving 4dB higher signal gain. If the signal source
is actually the low impedance output of another amplifier, RG
is increased to the minimum value allowed at the output of
that amplifier and RF is adjusted to get the desired gain. It is
critical for stable operation of the OPA843 that this driving
amplifier show a very low output impedance through frequencies exceeding the expected closed-loop bandwidth for the
OPA843.
An optional input termination resistor is also shown in Figure 2.
This RM resistor may be used to adjust the input impedance to
lower values when RG needs to be adjusted higher. This might
be desirable at lower gains where increasing RF will reduce the
output loading improving harmonic distortion performance. For
instance, at a gain of –4 an RG set to 50Ω will require a 200Ω
feedback resistor. In this case, adjusting RF to 400Ω, setting RG
to 100Ω, and then adding a 100Ω RM resistor will deliver a gain
of –4 with a 50Ω input match.
BUFFERING HIGH-PERFORMANCE ADCs
WIDEBAND, INVERTING GAIN OPERATION
There can be significant benefits to operating the OPA843 as
an inverting amplifier. This is particularly true when a matched
input impedance is required. Figure 2 shows the inverting
gain circuit used as a starting point for the typical characteristics showing inverting mode performance.
Driving this circuit from a 50Ω source, and constraining the
gain resistor, RG, to equal 50Ω will give both a signal
bandwidth and noise advantage. RG in this case is acting as
10
A single-channel interface using the OPA843 can provide a low
noise/distortion interface to emerging 14-bit Analog-to-Digital
Converters (ADCs) through approximately 5MHz for medium
gain applications. Since the dominant distortion mechanism is
2nd-harmonic distortion, differential circuits using the OPA843
can extend this frequency range and/or power level to much
higher levels. The example on the front page of this data sheet,
for instance, shows better than 93dB SFDR at 5MHz for up to
8VPP signals. This is still being limited by the 2nd-harmonic with
OPA843
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SBOS268C
the 3rd-harmonic much lower. 2-tone 3rd-order intermodulation
terms will be much lower than most other solutions using the
circuit shown on the front page. The differential typical characteristic curves also show that a 4VPP output will have
> 80dBc SFDR through 20MHz using this differential approach.
1dB through 50MHz. For narrowband IF’s in the 44MHz
region, this configuration of the OPA843 will show a 3rd-order
intercept of 33dBm while dissipating only 200mW (23dBm)
power from ±5V supplies.
PHOTODIODE TRANSIMPEDANCE AMPLIFIER
WIDE DYNAMIC RANGE “IF” AMPLIFIER
The OPA843 offers an attractive alternative to standard fixedgain IF amplifier stages. Narrowband systems will benefit from
the exceptionally high 2-tone 3rd-order intermodulation intercept, as shown in the Typical Characteristics. Op amps with
high open-loop gain, like the OPA843, provide an intercept
that decreases with frequency along with the loop gain. The
OPA843’s 3rd-order intercept shows a decreasing intercept
with frequency. The OPA843’s intercept is > 30dBm up to
50MHz but improves to > 50dBm as the operating frequency
is reduced below 10MHz. Broadband systems will also benefit
from the very low even-order harmonics and intermodulation
components produced by the OPA843. Compared to standard
fixed-gain IF amplifiers, the OPA843 operating at IF’s below
50MHz provides much higher intercepts for its quiescent
power dissipation (200mW), superior gain accuracy, higher
reverse isolation, and lower I/O return loss. The noise figure
for the OPA843 will be higher than alternative fixed-gain
stages. If the application comes late in the amplifier chain with
significant gain in prior stages, this higher noise figure may be
acceptable. Figure 3 shows an example of a noninverting
configuration for the OPA843 used as an IF amplifier.
High Gain Bandwidth Product (GBP) and low input voltage
and current noise make the OPA843 an ideal wideband
transimpedance amplifier for low to moderate gains. Note
that unity-gain stability is not required for transimpedance
applications. Figure 4 shows an example photodiode amplifier circuit. The key parameters of this design are the estimated diode capacitance (CD) at the applied DC reverse bias
voltage (–VB), the desired transimpedance gain (RF), and the
GBP for the OPA843 (800MHz). With these three variables
set (and adding the OPA843’s parasitic input capacitance to
the value of CD to get CS), the feedback capacitor value (CF)
is selected to provide stability for the transimpedance frequency response.
+5V
Power-supply decoupling
not shown.
0.01µF
10kΩ
–5V
λ
+5V
50Ω Source
ID
Power-supply
decoupling not shown.
VO = IDRF
OPA843
RF
10kΩ
CF
0.75pF
CD
20pF
0.01µF
PI
VO
52.3Ω
1kΩ
OPA843
+5V
RS 50Ω Load
50Ω
P0
–VB
FIGURE 3. High Dynamic Range IF Amplifier.
To achieve a maximally flat 2nd-order Butterworth frequency
response, the feedback pole should be set to:
RF
1kΩ
1
=
2πRF CF
RG
144Ω
GBP
4 πRF CS
CS = CD + CI
(1)
0.01µF
1
PI
RF
Gain = P = 20log 2 1 + R dB = 12dB with values shown
O
G
FIGURE 3. High Dynamic Range IF Amplifier.
The input signal and the gain resistor are AC-coupled through
the 0.01µF blocking capacitors. This holds the DC input and
output operating point at ground independent of source impedance and gain setting. The RG value in Figure 3 (144Ω),
sets the gain to the matched load at 12dB. Using standard 1%
tolerance resistors for RF and RG will hold the gain to a ±0.2dB
tolerance. This example will give a –3dB bandwidth of approximately 100MHz while maintaining gain flatness within
Adding the OPA843’s common-mode and differential mode
input capacitances CI = (1.0 + 1.2)pF to the 20pF diode
source capacitance of Figure 4, and targeting a 10kΩ transimpedance gain using the 800MHz GBP for the OPA843,
the required feedback pole frequency is 16.9MHz. This will
require a total feedback capacitance of 0.94pF. Typical
surface-mount resistors have a parasitic capacitance of
0.2pF, leaving the required 0.75pF value shown in Figure 4
to get the required feedback pole.
This will set the –3dB bandwidth according to:
F −3dB ≅
(2)
The example of Figure 4 will give approximately 24MHz
–3dB bandwidth using the 0.75pF feedback compensation.
OPA843
SBOS268C
GBP
Hz
2πRF CS
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11
WIDEBAND INVERTING SUMMING AMPLIFIER
100pF
One common application for a wideband op amp like the
OPA843 is to sum a number of signal sources together.
Figure 5 shows the inverting summing configuration that is
most often used. This circuit offers the benefit that each input
sees an input impedance set only by its individual input
resistor, since the summing junction (inverting op amp node)
is a virtual ground. Each input is non-interactive with every
other. However, the bandwidth from any input to the summed
output is set by the op amp noise gain (NG), which is equal
to the noninverting voltage gain. Therefore, each inverting
channel may have a low gain to the output (like the –1 shown
in Figure 5); this noise gain will set the frequency response
and the loop stability. The noninverting gain for Figure 5 is
equal to +5, which will give a 260MHz bandwidth at a gain of
–1 for each of the input signals.
0Ω
Source
VI
61Ω
150Ω
VO
220pF
OPA843
402Ω
100Ω
FIGURE 6. 10MHz Butterworth Low-Pass Filter.
10MHz Low-Pass Filter
15
12
+5V
9
Power-supply decoupling not shown.
0.1µF
81.8Ω
OPA843
VO = –(V1 + V2 + V3 + V4)
Gain (dB)
6
0
–3
–6
–5V
402Ω
3
–9
RF
402Ω
–12
V1
–15
100k
402Ω
1M
V2
402Ω
100M
FIGURE 7. Frequency Response for Figure 6.
V3
402Ω
V4
FIGURE 5. Wideband Inverting Summing Amplifier.
2nd-Order Filter Topology
High-speed amplifiers like the OPA843 are good choices for
2nd-order filter building blocks as part of ADC driver channels. These can provide noise bandlimiting to improve the
SNR for the amplifier/converter combination. The circuit of
Figure 6 shows an example of a 10MHz Butterworth lowpass filter where the amplifier provides a low frequency gain
of 5 and a 2nd-order cutoff at 10MHz. The resistor values
have been adjusted slightly to account for the amplifier
bandwidth. Figure 7 shows the small-signal frequency response for this filter.
EQUALIZING FILTER APPLICATION
In sensor receiver applications, where the pickup is a sensor
or cable giving a bandlimited frequency response, an equalizing filter can sometimes be used to extend the useable
frequency range for the sensor. This is done mathematically
by taking the inverse of the rolloff transfer function and
implementing that as the amplifier frequency response. See
Figure 8 for one example of a wideband equalizer where two
stages of the OPA843 are used. This example is set to
12
10M
Frequency (MHz)
transition from a unity gain receiver at lower frequencies
(through the R5 path) to a gain of 20dB (10V/V) through the
R1 path at higher frequencies. The component values have
been selected to set the peak gain at approximately 30MHz.
A unique feature for this circuit is an independent tune on the
width of the peaking (Q of the response) by adjusting RG.
See Figure 9 for the effect of adjusting RG over the range of
20Ω to 100Ω.
DESIGN-IN TOOLS
DEMONSTRATION FIXTURES
Two printed circuit boards (PCBs) are available to assist in
the initial evaluation of circuit performance using the OPA843
in its two package options. Both of these are offered free of
charge as unpopulated PCBs, delivered with a user's guide.
The summary information for these fixtures is shown in the
table below.
PRODUCT
PACKAGE
ORDERING
NUMBER
LITERATURE
NUMBER
OPA843U
OPA843N
SO-8
SOT23-5
DEM-OPA-SO-1A
DEM-OPA-SOT-1A
SBOU009
SBOU010
The demonstration fixtures can be requested at the Texas
Instruments web site (www.ti.com) through the OPA843
product folder.
OPA843
www.ti.com
SBOS268C
+5V
+5V
VCC
VCC
OPA843
OPA843
Power-supply
decoupling not shown.
VEE
VEE
–5V
R4
600Ω
C2
41.125pF
–5V
RF
1.2kΩ
R1
120Ω
VOUT
RLOAD
1kΩ
R2
1.2kΩ
VIN
C1
5.2pF
RG
R5
1.2kΩ
FIGURE 8. Adjustable Equalizer.
value should be between 200Ω and 1kΩ. Below 200Ω, the
feedback network will present additional output loading that
can degrade the harmonic distortion performance of the
OPA843. Above 1kΩ, the typical parasitic capacitance (approximately 0.2pF) across the feedback resistor may cause
unintentional band limiting in the amplifier response.
40
(dB)
20
0
–20
–40
100kHz
1MHz
10MHz
100MHz
1GHz
Frequency
FIGURE 9. Equalizer Plot, Multiple Settings.
MACROMODELS AND APPLICATIONS SUPPORT
Computer simulation of circuit performance using SPICE is
often a quick way to analyze the performance of the OPA843
and its circuit designs. This is particularly true for video and
RF amplifier circuits where parasitic capacitance and inductance can play a major role on circuit performance. A SPICE
model for the OPA843 is available through the TI web page
(http://www.ti.com). The applications department is also available for design assistance. These models predict typical
small-signal AC, transient steps, DC performance, and noise
under a wide variety of operating conditions. The models
include the noise terms found in the electrical specifications
of this data sheet. These models do not attempt to distinguish between the package types in their small-signal AC
performance.
OPERATING SUGGESTIONS
A good rule of thumb is to target the parallel combination of
RF and RG (see Figure 1) to be less than about 200Ω. The
combined impedance RF || RG interacts with the inverting
input capacitance, placing an additional pole in the feedback
network, and thus a zero in the forward response. Assuming
a 2pF total parasitic on the inverting node, holding RF || RG
< 200Ω will keep this pole above 400MHz. By itself, this
constraint implies that the feedback resistor RF can increase
to several kΩ at high gains. This is acceptable as long as the
pole formed by RF and any parasitic capacitance appearing
in parallel is kept out of the frequency range of interest.
In the inverting configuration, an additional design consideration must be noted. RG becomes the input resistor and,
therefore, the load impedance to the driving source. If impedance matching is desired, RG may be set equal to the
required termination value. However, at low inverting gains
the resultant feedback resistor value can present a significant load to the amplifier output. For example, an inverting
gain of –4 with a 50Ω input matching resistor (= RG) would
require a 200Ω feedback resistor, which would contribute to
output loading in parallel with the external load. In such a
case, it would be preferable to increase both the RF and RG
values, and then achieve the input matching impedance with
a third resistor to ground; see Figure 2. The total input
impedance becomes the parallel combination of RG and the
additional shunt resistor.
BANDWIDTH vs GAIN
OPTIMIZING RESISTOR VALUES
Since the OPA843 is a voltage-feedback op amp, a wide
range of resistor values may be used for the feedback and
gain setting resistors. The primary limits on these values are
set by dynamic range (noise and distortion) and parasitic
capacitance considerations. Usually, the feedback resistor
Voltage-feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the GBP shown in the Electrical
Characteristics. Ideally, dividing GBP by the noninverting
signal gain (also called the Noise Gain, or NG) will predict the
closed-loop bandwidth. In practice, this only holds true when
OPA843
SBOS268C
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13
the phase margin approaches 90°, as it does in high-gain
configurations. At low signal gains, most amplifiers will exhibit a more complex response with lower phase margin. The
OPA843 is optimized to give a maximally flat 2nd-order
Butterworth response in a gain of 5. In this configuration, the
OPA843 has approximately 60° of phase margin and will
show a typical –3dB bandwidth of 260MHz. When the phase
margin is 60°, the closed-loop bandwidth is approximately √2
greater than the value predicted by dividing GBP by the noise
gain. Increasing the gain will cause the phase margin to
approach 90° and the bandwidth to more closely approach
the predicted value of (GBP/NG). At a gain of +20, the
40MHz bandwidth shown in the Electrical Characteristics
agrees with that predicted using the simple formula and the
typical GBP of 800MHz.
LOW GAIN OPERATION
Decreasing the operating gain for the OPA843 from the
nominal design point of +5 will decrease the phase margin.
This will increase the Q for the closed-loop poles, peak up
the frequency response, and extend the bandwidth. A peaked
frequency response will show overshoot and ringing in the
pulse response as well as a higher integrated output noise.
Operating at a noise gain less than +3 runs the risk of
sustained oscillation (loop instability). However, operation at
low gains would be desirable to take advantage of the much
higher slew rate and lower input noise voltage available in
the OPA843, as compared to the performance offered by
unity-gain stable op amps. Numerous external compensation
techniques have been suggested for operating a high-gain
op amp at low gains. Most of these give zero/pole pairs in the
closed-loop response that cause long term settling tails in the
pulse response and/or phase nonlinearity in the frequency
response. Figure 10 shows an external compensation method
for a noninverting configuration that does not suffer from
these drawbacks.
tune the flatness by adjusting RI. The Typical Characteristics
show a signal gain of +4 with the noise gain adjusted for
flatness using different values for R1.
Where low gain is desired, and inverting operation is acceptable, a new external compensation technique may be used to
retain the full slew rate and noise benefits of the OPA843 while
maintaining the increased loop gain and the associated improvement in distortion offered by the decompensated architecture. This technique shapes the noise gain for good stability
while giving an easily controlled 2nd-order low-pass frequency
response. Figure 11 shows this circuit. Considering only the
noise gain for the circuit of Figure 11, the low-frequency noise
gain (NG1) will be set by the resistor ratios while the highfrequency noise gain (NG2) will be set by the capacitor ratios.
The capacitor values set both the transition frequencies and
the high-frequency noise gain. If this noise gain, determined by
NG2 = 1 + CS/CF, is set to a value greater than the recommended minimum stable gain for the op amp and the noise
gain pole (set by 1/RFCF) is placed correctly, a very well
controlled 2nd-order low-pass frequency response will result.
+5V
Power-supply
decoupling not shown.
280Ω
0.1µF
OPA843
VO
–5V
RF
806Ω
402Ω
V1
RS = 0Ω
CS
12.6pF
CF
1.9pF
FIGURE 10. Noninverting Low Gain Circuit.
To choose the values for both CS and CF, two parameters
and only three equations need to be solved. The first parameter is the target high-frequency noise gain, NG2, which
should be greater than the minimum stable gain for the
OPA843. Here, a target NG2 of 7.5 will be used. The second
parameter is the desired low-frequency signal gain, which
also sets the low-frequency noise gain, NG1. To simplify this
discussion, we will target a maximally flat 2nd-order low-pass
Butterworth frequency response (Q = 0.707). The signal gain
of –2 shown in Figure 11 will set the low-frequency noise gain
to NG1 = 1 + RF/RG (= 3 in this example). Then, using only
these two gains and the GBP for the OPA843 (800MHz), the
key frequency in the compensation is determined by:
+5V
50Ω Source
VI
RT
50Ω
R1
133Ω
VO
RS
50Ω
OPA843
50Ω Load
+5V
RF
402Ω
RG
402Ω
FIGURE 10. Noninverting Low Gain Circuit.
The R1 resistor across the two inputs will increase the noise
gain (i.e., decrease the loop gain) without changing the
signal gain. This approach will retain the full slew rate to the
output but will give up some of the low-noise benefit of the
OPA843. Assuming a low source impedance, set R1 so that
1 + RF/(RG || RI) is ≥ +3. This approach may also be used to
14
Z0 =
GBP
NG1
NG1
1−
− 1− 2
2
NG
NG
2
2
NG 1
(11)
Physically, this Z0 (13.6MHz for the values shown in Figure 11)
is set by 1/(2π • RF (CF + CS)) and is the frequency at which
the rising portion of the noise gain would intersect unity gain
if projected back to 0dB gain. The actual zero in the noise gain
OPA843
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SBOS268C
occurs at NG1 • Z0 and the pole in the noise gain occurs at
NG2 • Z0. Since GBP is expressed in Hz, multiply Z0 by 2π and
use this to get CF by solving:
CF =
1
2πRF Z0 NG2
(12)
Finally, since CS and CF set the high-frequency noise gain,
determine CS by:
CS = (NG2 – 1)CF
(13)
The resulting closed-loop bandwidth will be approximately
equal to:
f −3dB ≅ Z 0 GBP
(14)
For the values shown in Figure 10, the f–3dB will be approximately 105MHz. This is less than that predicted by simply
dividing the GBP product by NG1. The compensation network
controls the bandwidth to a lower value while providing full
slew rate and exceptional distortion performance due to increased loop gain at frequencies below NG1 • Z0. The capacitor values shown in Figure 10 are calculated for NG1 = 3 and
NG2 = 7.5 with no adjustment for parasitics.
OUTPUT DRIVE CAPABILITY
The OPA843 has been optimized to drive the demanding load
of a doubly-terminated transmission line. When a 50Ω line is
driven, a series 50Ω into the cable and a terminating 50Ω load
at the end of the cable are used. Under these conditions, the
impedance of the cable appears resistive over a wide frequency range and the total effective load on the OPA843 is
100Ω in parallel with the resistance of the feedback network.
The Electrical Characteristics show a 6.1VPP swing into a
100Ω load—which is then reduced to a 3VPP swing at the
termination resistor. The ±85mA output drive over temperature provides adequate current drive margin for this load.
A common IF amplifier specification, which describes available output power is the –1dB compression point. This is
usually defined at a matched 50Ω load to be the sinusoidal
power where the gain has compressed by –1dB vs the gain
seen at very low power levels. This compression level is
frequency dependent for an op amp, due to both bandwidth
and slew rate limitations. For frequencies well within the
bandwidth and slew rate limit of the OPA843, the –1dB
compression at a matched 50Ω load will be > 13dBm based
on the minimum available 3Vp-p swing at the load. One
common use for the –1dB compression is to predict
intermodulation intercept. This is normally 10dB greater than
the –1dB compression power for a standard RF amplifier. This
simple rule of thumb does NOT apply to the OPA843. The high
open-loop gain and Class AB output stage of the OPA843
produce a much higher intercept than the –1dB compression
would predict, as shown in the Typical Characteristics.
DRIVING CAPACITIVE LOADS
One of the most demanding, and yet very common, load
conditions for an op amp is capacitive loading. A high-speed,
high open-loop gain amplifier like the OPA843 can be very
susceptible to decreased stability and closed-loop response
peaking when a capacitive load is placed directly on the
output pin. In simple terms, the capacitive load reacts with
the open-loop output resistance of the amplifier to introduce
an additional pole into the loop and thereby decrease the
phase margin. This issue has become a popular topic of
application notes and articles, and several external solutions
to this problem have been suggested. When the primary
considerations are frequency-response flatness, pulse response fidelity, and/or distortion, the simplest and most
effective solution is to isolate the capacitive load from the
feedback loop by inserting a series isolation resistor between
the amplifier output and the capacitive load. This does not
eliminate the pole from the loop response, but rather shifts it
and adds a zero at a higher frequency. The additional zero
acts to cancel the phase lag from the capacitive load pole,
thus increasing the phase margin and improving stability.
The Typical Characteristics show the recommended RS vs
Capacitive Load and the resulting frequency response at the
load. The criterion for setting the recommended resistor is
maximum bandwidth and flat frequency response at the load.
Since there is now a passive low-pass filter between the
output pin and the load capacitance, the response at the
output pin itself is typically somewhat peaked, and becomes
flat after the roll off action of the RC network. This is not a
concern in most applications, but can cause clipping if the
desired signal swing at the load is very close to the amplifier’s
swing limit.
Parasitic capacitive loads greater than 2pF can begin to
degrade the performance of the OPA843. Long PC board
traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always
consider this effect carefully and add the recommended
series resistor as close as possible to the OPA843 output pin
(see Board Layout section).
DISTORTION PERFORMANCE
The OPA843 is capable of delivering an exceptionally low
distortion signal at high frequencies and medium gains. The
distortion plots in the Typical Characteristics show the typical
distortion under a wide variety of conditions. Most of these
plots are limited to 100dB dynamic range. The OPA843’s
distortion does not rise above –100dBc until either the signal
level exceeds 0.5Vp-p and/or the fundamental frequency
exceeds 500kHz.
Distortion in the audio band is < –120dBc.
Generally, until the fundamental signal reaches very high
frequencies or powers, the 2nd-harmonic will dominate the
distortion with a negligible 3rd-harmonic component. Focusing then on the 2nd-harmonic, increasing the load impedance improves distortion directly. Remember that the total
load includes the feedback network—in the noninverting
configuration this is the sum of RF + RG, whereas in the
inverting configuration this is just RF (see Figure 1). Increasing output voltage swing increases harmonic distortion directly. A 6dB increase in output swing will generally increase
OPA843
SBOS268C
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15
the 2nd-harmonic 12dB and the 3rd-harmonic 18dB. Increasing the signal gain will also increase the 2nd-harmonic
distortion. Again, a 6dB increase in gain will increase the
2nd- and 3rd-harmonic by 6dB even with a constant output
power and frequency. Finally, the distortion increases as the
fundamental frequency increases due to the roll off in the
loop gain with frequency. Conversely, the distortion will
improve going to lower frequencies down to the dominant
open-loop pole at approximately 3kHz. Starting from the
–100dBc 2nd-harmonic for 2VPP into 200Ω, G = +5 distortion
at 500kHz (from the Typical Characteristics), the 2nd-harmonic distortion at 20kHz should be approximately:
shows the general form for this output noise voltage using the
terms presented in Figure 12.
EO =
(E
2
NI
ENI
√4kTRS
4kT
RG
16
RF
RG
IBI
√4kTRF
4kT = 1.6E – 20J
at 290°K
FIGURE 12. Op Amp Noise Analysis Model.
Dividing this expression by the noise gain (NG = 1 + RF/RG)
will give the equivalent input referred spot noise voltage at
the noninverting input, as shown in Equation 16.
2
4kTRF
I R
2
EN = ENI
+ (IBN RS )2 + 4kTRS + BI F +
NG
NG
(16)
Evaluating these two equations for the OPA843 circuit presented in Figure 1 will give a total output spot noise voltage
of 12.4nV/√Hz and an equivalent input spot noise voltage of
2.48nV/√Hz.
DC OFFSET CONTROL
The OPA843 can provide excellent DC signal accuracy due to
its high open-loop gain, high common-mode rejection, high
power supply rejection, and low input offset voltage and bias
current offset errors. To take full advantage of this low input
offset voltage, careful attention to input bias current cancellation is also required. The high-speed input stage for the
OPA843 has a relatively high input bias current (20µA typical
into the pins) but with a very close match between the two
input currents—typically 0.17µA input offset current. Figures
13 and 14 show typical distribution of input offset voltage and
current for the OPA843.
1000
NOISE PERFORMANCE
800
Mean = 0.38mV
Standard Deviation = 0.31mV
Total Count = 5572
Count
700
600
500
400
300
200
100
0
< –1.20
< –1.08
< –0.96
< –0.84
< –0.72
< –0.60
< –0.48
< –0.36
< –0.24
< –0.12
< –0.00
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