OPA
OPA847
847
www.ti.com
SBOS251E – JULY 2002 – REVISED DECEMBER 2008
Wideband, Ultra-Low Noise, Voltage-Feedback
OPERATIONAL AMPLIFIER with Shutdown
FEATURES
DESCRIPTION
●
●
●
●
●
●
●
●
The OPA847 combines very high gain bandwidth and large
signal performance with an ultra-low input noise voltage
(0.85nV/√Hz) while using only 18mA supply current. Where
power saving is critical, the OPA847 also includes an optional power shutdown pin that, when pulled low, disables the
amplifier and decreases the supply current to < 1% of the
powered-up value. This optional feature may be left disconnected to ensure normal amplifier operation when no powerdown is required.
HIGH GAIN BANDWIDTH: 3.9GHz
LOW INPUT VOLTAGE NOISE: 0.85nV/ √Hz
VERY LOW DISTORTION: –105dBc (5MHz)
HIGH SLEW RATE: 950V/µs
HIGH DC ACCURACY: VIO < ±100µV
LOW SUPPLY CURRENT: 18.1mA
LOW SHUTDOWN POWER: 2mW
STABLE FOR GAINS ≥ 12
APPLICATIONS
● HIGH DYNAMIC RANGE ADC PREAMPS
● LOW NOISE, WIDEBAND, TRANSIMPEDANCE
AMPLIFIERS
● WIDEBAND, HIGH GAIN AMPLIFIERS
● LOW NOISE DIFFERENTIAL RECEIVERS
● ULTRASOUND CHANNEL AMPLIFIERS
● IMPROVED UPGRADE FOR THE OPA687,
CLC425, AND LMH6624
The combination of very low input voltage and current noise,
along with a 3.9GHz gain bandwidth product, make the
OPA847 an ideal amplifier for wideband transimpedance
applications. As a voltage gain stage, the OPA847 is optimized for a flat frequency response at a gain of +20V/V and
is stable down to gains as low as +12V/V. New external
compensation techniques allow the OPA847 to be used at
any inverting gain with excellent frequency response control.
Using this technique in a differential Analog-to-Digital Converter (ADC) interface application, shown below, can deliver
one of the highest dynamic-range interfaces available.
OPA847 RELATED PRODUCTS
SINGLES
+5V
0.001µF
100Ω
+5V
20Ω
OPA847
OPA842
OPA843
OPA846
1.7pF
< 5.1dB
Noise
Figure
2kΩ
VCM ADS5500
14-Bit
125MSPS
0.1µF
850Ω
2kΩ
+5V
100Ω
1.7pF
0.001µF
20Ω
INN
100pF
OPA847
–5V
200
800
1750
2VPP, at converter input.
850Ω
39pF
2.6
2.0
1.2
–70
100pF
24.6dB Gain
–75
Harmonic Distortion (dBc)
1:2
39pF
GAIN BANDWIDTH
PRODUCT (MHz)
DIFFERENTIAL OPA847 DRIVER DISTORTION
INP
–5V
50Ω Source
INPUT NOISE
VOLTAGE (nV/ √Hz )
–80
–85
–90
2nd-Harmonic
–95
3rd-Harmonic
–100
–105
Ultra-High Dynamic Range
Differential ADC Driver
–110
10
20
30
40
50
Frequency (MHz)
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
Copyright © 2002-2008, Texas Instruments Incorporated
www.ti.com
ABSOLUTE MAXIMUM RATINGS(1)
ELECTROSTATIC
DISCHARGE SENSITIVITY
Power Supply ............................................................................... ±6.5VDC
Internal Power Dissipation ........................ See Thermal Analysis Section
Differential Input Voltage .................................................................. ±1.2V
Input Voltage Range ............................................................................ ±VS
Storage Temperature Range: D, DBV ........................... –65°C to +125°C
Lead Temperature (soldering, 10s) .............................................. +300°C
Junction Temperature (TJ ) ........................................................... +150°C
ESD Rating (Human Body Model) .................................................. 1500V
(Charge Device Model) ............................................... 1500V
(Machine Model) ........................................................... 100V
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
ESD damage can range from subtle performance degradation
to complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric
changes could cause the device not to meet its published
specifications.
NOTE: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.
PACKAGE/ORDERING INFORMATION(1)
PRODUCT
OPA847
PACKAGE-LEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
SO-8
D
–40°C to +85°C
OPA847
"
"
"
"
SOT23-6
DBV
–40°C to +85°C
OATI
"
"
"
"
OPA847ID
OPA847IDR
OPA847IDBVT
OPA847IDBVR
Rails, 100
Tape and Reel, 2500
Tape and Reel, 250
Tape and Reel, 3000
"
OPA847
"
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this document, or see the TI web site
at www.ti.com.
PIN CONFIGURATIONS
DIS
Inverting Input
2
7
+VS
Noninverting Input
3
6
Output
–VS
4
5
NC
1
6
+VS
–VS
2
5
DIS
Noninverting Input
3
4
Inverting Input
4
8
Output
OATI
1
NC = No Connection
3
1
SOT
2
NC
Top View
5
SO
6
Top View
Pin Orientation/Package Marking
2
OPA847
www.ti.com
SBOS251E
ELECTRICAL CHARACTERISTICS: VS = ±5V
Boldface limits are tested at +25°C.
RL = 100Ω, RF = 750Ω, RG = 39.2Ω, and G = +20 (see Figure 1 for AC performance only), unless otherwise noted.
OPA847ID, IDBV
TYP
PARAMETER
AC PERFORMANCE (see Figure 1)
Closed-Loop Bandwidth
Gain Bandwidth Product (GBP)
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +12
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
2-Tone, 3rd-Order Intercept
Input Voltage Noise Density
Input Current Noise Density
Pulse Response
Rise-and-Fall Time
Slew Rate
Settling Time to 0.01%
0.1%
1%
DC PERFORMANCE(4)
Open-Loop Voltage Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Input Bias Current Drift (magnitude)
Input Offset Current
Input Offset Current Drift
INPUT
Common-Mode Input Range (CMIR)(5)
Common-Mode Rejection Ratio (CMRR)
Input Impedance
Differential
Common-Mode
OUTPUT
Output Voltage Swing
Current Output, Sourcing
Current Output, Sinking
Closed-Loop Output Impedance
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio
+PSRR, –PSRR
POWER-DOWN (disabled low)
Power-Down Quiescent Current (+VS)
On Voltage (enabled high or floated)
Off Voltage (disabled asserted low)
Power-Down Pin Input Bias Current
Power-Down Time
Power-Up Time
Off Isolation
THERMAL
Specification ID, IDBV
Thermal Resistance, θJA
D
SO-8
DBV SOT23
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C
+25°C(1)
0°C to
70°C(2)
–40°C to
+85°C(2)
UNITS
G = +12, RG = 39.2Ω, VO = 200mVPP
G = +20, RG = 39.2Ω, VO = 200mVPP
G = +50, RG = 39.2Ω, VO = 200mVPP
G ≥ +50
G = +20, RL = 100Ω
600
350
78
3900
60
4.5
230
63
3100
40
7
210
60
3000
35
10
195
57
2800
30
12
MHz
MHz
MHz
MHz
MHz
dB
typ
min
min
min
min
max
C
B
B
B
B
B
G = +20, f = 5MHz, VO = 2VPP
RL = 100Ω
RL = 500Ω
RL = 100Ω
RL = 500Ω
G = +20, f = 20MHz
f > 1MHz
f > 1MHz
–74
–105
–103
–110
39
0.85
2.5
–70
–90
–96
–105
37
0.92
3.5
–69
–89
–91
–100
36
0.98
3.6
–68
–88
–88
–90
35
1.0
3.7
dBc
dBc
dBc
dBc
dBm
nV/√Hz
pA/√Hz
max
max
max
max
min
max
max
B
B
B
B
B
B
B
0.2V Step
2V Step
2V Step
2V Step
2V Step
1.2
950
20
10
6
1.75
700
2.0
625
2.2
535
12
8
14
10
18
12
ns
V/µs
ns
ns
ns
max
min
typ
max
max
B
B
C
B
B
VO = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
98
±0.1
±0.25
–19
–15
±0.1
±0.1
90
89
±0.58
±1.5
–41
–40
±0.7
±2
88
±0.60
±1.5
–42
–70
±0.85
±3.5
dB
mV
µV/°C
µA
nA/°C
µA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
VCM = ±0.5V, Input-Referred
±3.3
110
±3.0
93
±2.9
90
V
dB
min
min
A
A
VCM = 0V
VCM = 0V
2.7 || 2.0
2.3 || 1.7
kΩ || pF
MΩ || pF
typ
typ
C
C
≥ 400Ω Load
100Ω Load
VO = 0V
VO = 0V
G = +20, f = < 100kHz
±3.5
±3.4
100
–75
0.003
V
V
mA
mA
Ω
min
min
min
min
typ
A
A
A
A
C
VS = ±5V
VS = ±5V
±5
±6
18.1
18.1
|VS| = 4.5V to 5.5V, Input-Referred
±0.5
±0.25
–39
–15
±0.6
±0.1
±3.1
95
±3.3
±3.2
MIN/ TEST
MAX LEVEL(3)
±3.1
±3.0
56
–56
±3.0
±2.9
52
–52
18.4
17.8
±6
18.7
17.5
±6
18.9
17.1
V
V
mA
mA
typ
max
max
min
C
A
A
A
100
95
93
90
dB
min
A
–200
3.5
1.8
150
200
60
70
–270
3.75
1.7
190
–320
3.85
1.6
200
–370
3.95
1.5
210
µA
V
V
µA
ns
ns
dB
max
min
max
max
typ
typ
typ
A
A
A
A
C
C
C
–40 to +85
°C
typ
C
125
150
°C/W
°C/W
typ
typ
C
C
60
–60
±6
(Pin 8 on SO-8; Pin 5 on SOT23-6)
(VDIS = 0)
5MHz, Input to Output
Junction-to-Ambient
NOTES: (1) Junction temperature = ambient for +25°C specifications. (2) Junction temperature = ambient at low temperature limit: junction temperature = ambient +23°C
at high temperature limit for over temperature specifications. (3) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation.
(B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of node. VCM is the input common-mode
voltage. (5) Tested < 3dB below minimum specified CMRR at ±CMIR limits.
OPA847
SBOS251E
www.ti.com
3
TYPICAL CHARACTERISTICS: VS = ±5V
TA = 25°C, G = +20V/V, RG = 39.2Ω, and RL = 100Ω, unless otherwise noted.
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
6
0
VO = 0.2VPP
RL = 100Ω
RG = RS = 50Ω
RF Adjusted
G = +12
3
Normalized Gain (dB)
–3
G = +20
–6
G = +30
–9
G = +50
0
–3
–6
–9
G = –40
–12
–12
–15
1
10
100
1000
1
10
100
Frequency (MHz)
NONINVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
35
RG = 39.2Ω
RL = 100Ω
G = +20V/V
VO = 200mVPP
26
See Figure 2
32
VO = 0.2VPP
VO = 1VPP
29
Gain (dB)
23
20
17
VO = 2VPP
26
23
VO = 5VPP
VO = 1VPP
20
14
VO = 2VPP
11
17
VO = 5VPP
See Figure 1
10
100
10
1000
NONINVERTING PULSE RESPONSE
Large Signal ± 1V
Right Scale
0.15
0.10
Small Signal ± 100mV
0.05
Left Scale
0
0.25
1.00
0.20
0.75
0.50
0.25
0
0.15
0.10
0.05
0
–0.05
–0.25
–0.10
–0.50
–0.15
–0.75
–0.20
–1.00
–0.20
–1.25
–0.25
See Figure 1
–0.25
1000
INVERTING PULSE RESPONSE
1.25
Output Voltage (50mV/div)
G = +20V/V
Output Voltage (250mV/div)
0.25
100
Frequency (MHz)
Frequency (MHz)
Output Voltage (50mV/div)
RL = 100Ω
RG = RS = 50Ω
G = –40V/V
14
8
Large Signal ± 1V
Right Scale
Small Signal ± 100mV
Left Scale
–0.05
–0.15
1.25
1.00
0.75
0.50
0.25
0
–0.25
–0.10
Time (5ns/div)
4
1000
Frequency (MHz)
29
Gain (dB)
G = –50
See Figure 2
See Figure 1
–15
0.20
G = –30
G = –20
–0.50
G = –40V/V
RG = RS = 50Ω
RL = 100Ω
–0.75
See Figure 2
–1.00
–1.25
Time (5ns/div)
OPA847
www.ti.com
SBOS251E
Output Voltage (250mV/div)
Normalized Gain (dB)
6
VO = 0.2VPP
RG = 39.2Ω
RL = 100Ω
RF Adjusted
3
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = 25°C, G = +20V/V, RG = 39.2Ω, and RL = 100Ω, unless otherwise noted.
1MHz HARMONIC DISTORTION vs LOAD RESISTANCE
5MHz HARMONIC DISTORTION vs LOAD RESISTANCE
–70
Harmonic Distortion (dBc)
–75
Harmonic Distortion (dBc)
–75
G = +20V/V
VO = 2VPP
–80
–85
–90
2nd-Harmonic
–95
–100
3rd-Harmonic
–105
G = +20V/V
VO = 5VPP
–80
2nd-Harmonic
–85
–90
–95
3rd-Harmonic
–100
–110
See Figure 1
See Figure 1
–115
–105
100
150
200
250
300
350
400
450
500
100
150
200
–75
G = +20V/V
VO = 2VPP
RL = 200Ω
–80
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
300
350
400
450
500
HARMONIC DISTORTION vs OUTPUT VOLTAGE
HARMONIC DISTORTION vs FREQUENCY
–65
–75
250
Load Resistance (Ω)
Load Resistance (Ω)
2nd-Harmonic
–85
–95
3rd-Harmonic
–105
G = +20V/V
F = 5MHz
RL = 200Ω
–85
2nd-Harmonic
–90
–95
–100
3rd-Harmonic
–105
–110
See Figure 1
See Figure 1
–115
–115
0.1
1
10
0.1
100
1
HARMONIC DISTORTION vs INVERTING GAIN
HARMONIC DISTORTION vs NONINVERTING GAIN
–70
–75
–75
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–80
2nd-Harmonic
–85
–90
–95
10
Output Voltage Swing (VPP)
Frequency (MHz)
VO = 2VPP
RL = 200Ω
F = 5MHz
RF = 750Ω
RG Adjusted
–100
3rd-Harmonic
–105
–80
–85
–90
–95
–100
2nd-Harmonic
VO = 2VPP
RL = 200Ω
F = 5MHz
RG = 50Ω
RF Adjusted
3rd-Harmonic
–105
See Figure 2
See Figure 1
–110
–110
15
20
25
30
35
40
45
50
55
20
50
OPA847
SBOS251E
25
30
35
40
45
50
Gain –V/V
Gain (V/V)
www.ti.com
5
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = 25°C, G = +20V/V, RG = 39.2Ω, and RL = 100Ω, unless otherwise noted.
INPUT VOLTAGE AND CURRENT NOISE
2-TONE, 3RD-ORDER INTERMODULATION INTERCEPT
50
G = +20V/V
20dB to matched load.
45
2.7pA/√Hz
Current Noise
Intercept Point (+dBm)
Voltage Noise (nV/√Hz)
Current Voise (pA/√Hz)
10
1
0.85nV/√Hz
Voltage Noise
40
35
50Ω
PI
30
PO
50Ω OPA847
50Ω
750Ω
25
39.2Ω
20
0
101
102
103
104
105
106
107
5
10
15
20
Frequency (Hz)
VO = 200mVPP
AV = +12V/V
NG = Noise Gain
0.4
0.3
NG = 12
40
45
50
0
NG = 14
Normalized Gain (1dB)
Deviation from 21.58dB Gain (0.1dB)
35
1
NG = 16
0.2
0.1
0
–0.1
NG = 18
–0.2
NG = 20
External Compensation
See Figure 8
–0.4
G = –8
–1
–2
–3
VO = 0.2VPP
RF = 750Ω
–4
G = –1
–5
–6
G = –2
–7
–0.3
G = –4
External Compensation
See Figure 6
–8
–9
–0.5
1
10
100
1000
1
10
Normalized Gain to Capacitive Load (dB)
G = +20V/V
10
1
10
100
1000
FREQUENCY RESPONSE vs CAPACITIVE LOAD
RECOMMENDED RS vs CAPACITIVE LOAD
100
1
100
Frequency (MHz)
Frequency (MHz)
RS (Ω)
30
LOW GAIN INVERTING BANDWIDTH
NONINVERTING GAIN FLATNESS TUNE
0.5
1000
Capacitive Load (pF)
6
25
Frequency (MHz)
29
RS adjusted for capacitive load.
C = 10pF
26
C = 22pF
23
C = 47pF
C = 100pF
RS
VI
20
VO
50Ω OPA847
CL
1kΩ
750Ω
17
(1kΩ is optional.)
39.2Ω
14
1
10
100
1000
Frequency (MHz)
OPA847
www.ti.com
SBOS251E
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = 25°C, G = +20V/V, RG = 39.2Ω, and RL = 100Ω, unless otherwise noted.
COMMON-MODE REJECTION RATIO AND
POWER-SUPPLY REJECTION RATIO vs FREQUENCY
CMRR
0
+PSRR
110
–30
100
100
Open-Loop Gain (dB)
CMRR and PSRR (dB)
OPEN-LOOP GAIN AND PHASE
120
90
80
–PSRR
70
60
50
40
30
20log (AOL)
80
60
–90
40
–120
20
–150
0
–180
–210
–20
20
102
103
104
105
106
107
108
102
103
104
105
106
107
108
109
Frequency (Hz)
Frequency (Hz)
CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
4
10
VDIS
3
Output Impedance (Ω)
RL = 100Ω
2
1
VO (V)
–60
∠AOL
Open-Loop Phase (°)
120
RL = 50Ω
0
RL = 25Ω
–1
–2
G = +20V/V
ZO
OPA847
1
750Ω
0.1
39.2Ω
0.01
–3
0.001
–100
–50
0
50
100
103
150
104
10
0.4
8
0.3
6
0.2
Output
Left Scale
2
0.1
0
0
Output Voltage (V)
4
G = +20V/V
RL = 100Ω
0.5
Input Voltage (mV)
Output Voltage (V)
6
4
Input
Right Scale
0.20
0.15
0.10
0.05
0
–2
–0.1
–0.2
–6
–0.3
–6
–8
–0.4
–8
–0.5
–10
–4
–0.05
–0.10
Output
Left Scale
–0.15
–0.20
See Figure 2
–0.25
Time (40ns/div)
Time (40ns/div)
OPA847
SBOS251E
0.25
G = –40V/V
RG = 50Ω
RL = 100Ω
0
–4
–10
108
2
–2
See Figure 1
107
INVERTING OVERDRIVE RECOVERY
NONINVERTING OVERDRIVE RECOVERY
10
Input
Right Scale
106
Frequency (Hz)
IO (mA)
8
105
www.ti.com
7
Input Voltage (mV)
–4
–150
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = 25°C, G = +20V/V, RG = 39.2Ω, and RL = 100Ω, unless otherwise noted.
PHOTODIODE TRANSIMPEDANCE
FREQUENCY RESPONSE
SETTLING TIME
0.25
0.15
Transimpedance Gain (dBΩ)
0.10
0.05
0
–0.05
–0.10
–0.15
–0.20
RF = 20kΩ
CF Adjusted
86
83
80
0.01µF
77
20kΩ
IO
CF
CDIODE
[CD]
74
5
10
15
20
25
30
35
1
40
10
TYPICAL DC DRIFT OVER TEMPERATURE
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
0.2
100
12.5
VIO
0
0
–0.1
–12.5
Ib
–0.2
Supply Current
0
25
50
75
100
18
Sourcing Output Current
80
16
Sinking Output Current
70
14
60
12
50
–25.0
–25
20
90
Output Current (mA)
100 x IOS
0.1
Input Bias and Offset Current (µA)
25.0
–50
100
Frequency (MHz)
Time (ns)
Input Offset Voltage (mV)
VO
20kΩ OPA847
71
0
125
10
–50
–25
0
25
50
75
100
Ambient Temperature (°C)
Ambient Temperature (°C)
COMMON-MODE INPUT RANGE AND OUTPUT SWING
vs SUPPLY VOLTAGE
COMMON-MODE AND DIFFERENTIAL
INPUT IMPEDANCE
RL = 100Ω
3
Common-Mode
(2.3MΩ, DC)
106
Input Impedance (Ω)
Positive Output
2
1
Positive Input
0
125
107
5
Voltage Range (V)
CD = 20pF
CD = 50pF
CD = 100pF
See Figure 1
–0.25
4
CD = 10pF
[20log 20kΩ]
–1
Negative Input
–2
105
Differential
(2.7kΩ, DC)
104
103
–3
Negative Output
–4
102
102
6.00
5.75
5.50
5.25
5.00
4.75
4.50
4.25
4.00
3.75
3.50
3.25
3.00
2.75
2.50
–5
103
104
105
106
107
108
Frequency (Hz)
Supply Voltage (±V)
8
OPA847
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SBOS251E
Supply Current (mA)
Percent of Final Value (%)
89
G = +20V/V
RL = 100Ω
VO = 2V Step
0.20
TYPICAL CHARACTERISTICS: VS = ±5V
TA = 25°C, GD = 40V/V, RG = 50Ω, and RL = 400Ω, unless otherwise noted.
DIFFERENTIAL SMALL-SIGNAL
FREQUENCY RESPONSE
DIFFERENTIAL PERFORMANCE TEST CIRCUIT
3
+5V
0
Normalized Gain (dB)
DIS
OPA847
GD =
–5V
RG
50Ω
RF
RG
50Ω
VI
VO
R
= F
VI
RG
RL
RF
VO
GD = +20V/V
–3
GD = +30V/V
–6
–12
RG = 50Ω
VO = 400mVPP
–15
+5V
GD = +40V/V
GD = +50V/V
–9
RF Adjusted
–18
10
OPA847
100
1000
Frequency (MHz)
–5V
DIS
DIFFERENTIAL LARGE-SIGNAL
FREQUENCY RESPONSE
GD = 40V/V
GD = 40V/V
VO = 4VPP
F = 5MHz
Harmonic Distortion (dBc)
–60
32
Gain (dB)
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
–55
35
VO = 400mVPP
VO = 5VPP
29
VO = 8VPP
26
–65
–70
–75
2nd-Harmonic
–80
–85
–90
–95
3rd-Harmonic
–100
–105
–110
23
1
10
100
1000
50
100
150
Frequency (MHz)
350
400
450
500
GD = 40V/V
RL = 400Ω
F = 5MHz
–80
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
300
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
–75
GD = 40V/V
RL = 400Ω
VO = 4VPP
–75
250
Resistance (Ω)
DIFFERENTIAL DISTORTION vs FREQUENCY
–65
200
2nd-Harmonic
–85
–95
3rd-Harmonic
–105
–85
2nd-Harmonic
–90
–95
–100
3rd-Harmonic
–105
–115
–110
1
10
100
1
Frequency (MHz)
OPA847
SBOS251E
10
Differential Output Voltage Swing (VPP)
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9
APPLICATIONS INFORMATION
WIDEBAND, NONINVERTING OPERATION
The OPA847 provides a unique combination of a very low
input voltage noise along with a very low distortion output
stage to give one of the highest dynamic range op amps
available. Its very high gain bandwidth product (GBP) can be
used to either deliver high signal bandwidths at high gains, or
to deliver very low distortion signals at moderate frequencies
and lower gains. To achieve the full performance of the
OPA847, careful attention to PC board layout and component selection is required, as discussed in the following
sections of this data sheet.
Figure 1 shows the noninverting gain of a +20V/V circuit used
as the basis for most of the Typical Characteristics. Most of
the curves are characterized using signal sources with a 50Ω
driving impedance and with measurement equipment presenting a 50Ω load impedance. In Figure 1, the 50Ω shunt
resistor at the VI terminal matches the source impedance of
the test generator, while the 50Ω series resistor at the VO
terminal provides a matching resistor for the measurement
equipment load. Generally, data sheet voltage swing specifications are at the output pin (VO in Figure 1) while output
power specifications are at the matched 50Ω load. The total
100Ω load at the output combined with the 790Ω total
feedback network load presents the OPA847 with an effective output load of 89Ω for the circuit of Figure 1.
Voltage-feedback op amps, unlike current-feedback designs,
can use a wide range of resistor values to set their gain. The
circuit of Figure 1, and the specifications at other gains, use an
RG set to 39.2Ω and RF adjusted to get the desired gain. Using
this guideline ensures that the noise added at the output due
to the Johnson noise of the resistors does not significantly
increase the total over that due to the 0.85nV/√Hz input
voltage noise for the op amp itself. This RG is suggested as a
good starting point for design. Other values are certainly
acceptable, if required by the design.
WIDEBAND, INVERTING GAIN OPERATION
There can be significant benefits to operating the OPA847 as
an inverting amplifier. This is particularly true when a matched
input impedance is required. Figure 2 shows the inverting
gain of a –40V/V circuit used as a starting point for the
Typical Characteristics showing inverting mode performance.
Driving this circuit from a 50Ω source, and constraining the gain
resistor (RG) to equal 50Ω, gives both a signal bandwidth and
a noise advantage. RG, in this case, acts as both the input
termination resistor and the gain setting resistor for the circuit.
Although the signal gain for the circuit of Figure 2 is double that
for Figure 1, their noise gains are nearly equal when the 50Ω
source resistor is included. This has the interesting effect of
approximately doubling the equivalent GBP for the amplifier.
This can be seen by observing that the gain of –40 bandwidth
of 240MHz shown in the Typical Characteristics implies a gain
bandwidth product of 9.6GHz, giving a far higher bandwidth at
a gain of –40 than at a gain of +40. While the signal gain from
RG to the output is –40, the noise gain for bandwidth setting
purposes is 1 + RF/(2 • RG). In the case of a –40V/V gain, using
an RG = RS = 50Ω gives a noise gain = 1 + 2kΩ/100Ω = 21. This
inverting gain of –40V/V therefore has a frequency response
that more closely matches the gain of a +20 frequency response.
If the signal source is actually the low impedance output of
another amplifier, RG should be increased to be greater than
the minimum value allowed at the output for that amplifier
and RF adjusted to get the desired gain. It is critical for stable
operation of the OPA847 that this driving amplifier show a
very low output impedance through frequencies exceeding
the expected closed-loop bandwidth for the OPA847.
+5V
+VS
0.1µF
+5V
+VS
6.8µF
+
0.1µF
50Ω Source
VDIS
VI
50Ω
50Ω Load
VO
OPA847
+
6.8µF
VDIS
50Ω
0.01µF
RF
750Ω
50Ω Source
RG
50Ω
VO
95.3Ω
50Ω Load
50Ω
OPA847
RF
2kΩ
VI
RG
39.2Ω
+
6.8µF
0.1µF
0.1µF
6.8µF
–VS
–5V
–VS
–5V
FIGURE 1. Noninverting G = +20 Specification and Test Circuit.
10
+
FIGURE 2. Noninverting G = –40 Specification and Test Circuit.
OPA847
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SBOS251E
WIDEBAND, HIGH SENSITIVITY,
TRANSIMPEDANCE DESIGN
Equation 2 gives the approximate –3dB bandwidth that
results if CF is set using Equation 1.
The high GBP and low input voltage and current noise for the
OPA847 make it an ideal wideband transimpedance amplifier for low to moderate transimpedance gains. Very high
transimpedance gains (> 100kΩ) will benefit from the low
input noise current of a JFET input op amp such as the
OPA657. Unity-gain stability in the op amp is not required for
application as a transimpedance amplifier. Figure 3 shows
one possible transimpedance design example that would be
particularly suitable for the 155Mbit data rate of an OC-3
receiver. Designs that require high bandwidth from a large
area detector with relatively low transimpedance gain will
benefit from the low input voltage noise for the OPA847. The
amplifier’s input voltage noise is peaked up over frequency
by the diode source capacitance, and can (in many cases)
become the limiting factor to input sensitivity. The key elements to the design are the expected diode capacitance (CD)
with the reverse bias voltage (–VB) applied, the desired
transimpedance gain (RF), and the GBP for the OPA847
(3900MHz). With these three variables set (including the
parasitic input capacitance for the OPA847 added to CD), the
feedback capacitor value (CF) can be set to control the
frequency response.
+5V
Power-supply
decoupling not shown.
100pF
0.1µF
12kΩ
OPA847
VDIS
–5V
λ
RF
12kΩ
CF
0.18pF
1pF
Photodiode
–VB
FIGURE 3. Wideband, High Sensitivity, OC-3 Transimpedance
Amplifier.
To achieve a maximally flat 2nd-order Butterworth frequency
response, set the feedback pole as shown in Equation 1.
1
=
2πRF CF
GBP
4πRF CD
(1)
Adding the common-mode and differential mode input capacitance (1.2 + 2.5)pF to the 1pF diode source capacitance
of Figure 3, and targeting a 12kΩ transimpedance gain using
the 3900MHz GBP for the OPA847 requires a feedback pole
set to 74MHz to get a nominal Butterworth frequency response design. This requires a total feedback capacitance of
0.18pF. That total is shown in Figure 3, but recall that typical
surface-mount resistors have a parasitic capacitance of 0.2pF,
leaving no external capacitor required for this design.
f −3dB =
(2)
The example of Figure 3 gives approximately 104MHz flat
bandwidth using the 0.18pF feedback compensation capacitor. This bandwidth easily supports an OC-3 receiver with
exceptional sensitivity.
If the total output noise is bandlimited to a frequency less
than the feedback pole frequency, a very simple expression
for the equivalent input noise current is shown as Equation 3.
(3)
4kT
iEQ = iN2 +
RF
2
+
(EN 2πCD F)2
3
where:
iEQ = Equivalent input noise current if the output noise is
bandlimited to f < 1/2πRFCF
iN = Input current noise for the op amp inverting input
eN = Input voltage noise for the op amp
CD = Total Inverting Node Capacitance
f = Bandlimiting frequency in Hz (usually a post filter prior
to further signal processing)
Evaluating this expression up to the feedback pole frequency
at 74MHz for the circuit of Figure 3 gives an equivalent input
noise current of 3.0pA/√Hz. This is slightly higher than the
2.5pA/√Hz input current noise for the op amp. This total
equivalent input current noise is slightly increased by the last
term in the equivalent input noise expression. It is essential
in this case to use a low-voltage noise op amp. For example,
if a slightly higher input noise voltage, but otherwise identical,
op amp were used instead of the OPA847 in this application
(say 2.0nV/√Hz), the total input referred current noise would
increase to 3.7pA/√Hz. Low input voltage noise is required
for the best sensitivity in these wideband transimpedance
applications. This is often unspecified for dedicated transimpedance amplifiers with a total output noise for a specified
source capacitance given instead. It is the relatively high
input voltage noise for those components that cause higher
than expected output noise if the source capacitance is
higher than specified.
The output DC error for the circuit of Figure 3 is minimized by
including a 12kΩ to ground on the noninverting input. This
reduces the contribution of input bias current errors (for total
output offset voltage) to the offset current times the feedback
resistor. To minimize the output noise contribution of this
resistor, 0.01µF and 100pF capacitors are included in parallel. Worst-case output DC error for the circuit of Figure 3 at
25°C is:
VOS = ±0.5mV (input offset voltage) ± 0.6µA (input offset
current) • 12kΩ = ±7.2mV
Worst-case output offset DC drift (over the 0°C to 70°C span) is:
dVOS/dT = ±1.5µV/°C (input offset drift) ± 2nA/°C (input
offset current drift) • 12kΩ = ±21.5µV/°C.
OPA847
SBOS251E
GBP
(Hz)
2πR F C D
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11
Even with bias current cancellation, the output DC errors are
dominated in this example by the offset current term. Improved output DC precision and drift are possible, particularly
at higher transimpedance gains, using the JFET input
OPA657. The JFET input removes the input bias current
from the error equation (eliminating the need for the resistor
to ground on the noninverting input), leaving only the input
offset voltage and drift as an output DC error term.
Included in the Typical Characteristics are transimpedance
frequency response curves for a fixed 20kΩ gain over various detector diode capacitance settings. These curves are
repeated in Figure 4, along with the test circuit. As the
photodiode capacitance changes, the feedback capacitor
must change to maintain a stable and flat frequency response. Using Equation 1, CF is adjusted to give the
Butterworth frequency responses shown in Figure 4.
Considering only the noise gain (which is the same as the
noninverting signal gain) for the circuit of Figure 5, the lowfrequency noise gain (NG1) is set by the resistor ratio, while
the high-frequency noise gain (NG2) is set by the capacitor
ratio. The capacitor values set both the transition frequencies
and the high-frequency noise gain. If the high-frequency
noise gain, determined by NG2 = 1 + CS/CF, is set to a value
greater than the recommended minimum stable gain for the
op amp, and the noise gain pole (set by 1/RFCF) is placed
correctly, a very well controlled 2nd-order low-pass fre-
+5V
VDIS
OPA847
RG
200Ω
PHOTODIODE TRANSIMPEDANCE
FREQUENCY RESPONSE
Transimpedance Gain (dBΩ)
89
CD = 10pF
80
0.01µF
FIGURE 5. Broadband, Low-Inverting Gain External
Compensation.
20kΩ
IO
CD
74
quency response results.
CF
71
1
10
100
Frequency (MHz)
FIGURE 4. Transimpedance Bandwidth vs CD.
LOW-GAIN COMPENSATION FOR IMPROVED SFDR
Where a low gain is desired, and inverting operation is
acceptable, a new external compensation technique can be
used to retain the full slew rate and noise benefits of the
OPA847, while giving increased loop gain and the associated distortion improvements offered by a non-unity-gain
stable op amp. This technique shapes the loop gain for good
stability, while giving an easily controlled 2nd-order low-pass
frequency response. This technique is used for the circuit on
the front page of this data sheet in a differential configuration
to achieve extremely low distortion through high frequencies
(< –90dBc through 30MHz). The amplifier portion of this
circuit is set up for a differential gain of 8.5V/V from a
differential input signal to the output. Using the input transformer shown improves the noise figure and translates from
a single-ended to a differential signal. If the source is differential already, it can be fed directly into the gain setting
resistors. To set the compensation capacitors (CS and CF),
consider the half circuit of Figure 5, where the 50Ω source is
reflected through the 1:2 transformer, then cut in half, and
grounded to give a total impedance to the AC ground for the
circuit on the front page equal to 200Ω.
12
CF
1.7pF
–5V
VO
20kΩ OPA847
77
CS
39pF
CD = 20pF
CD = 50pF
CD = 100pF
83
RF
850Ω
VI
RF = 20kΩ
CF Adjusted [20 log(20kΩ)]
86
VO
To choose the values for both CS and CF, two parameters and
only three equations need to be solved. The first parameter is
the target high-frequency noise gain (NG 2), which should be
greater than the minimum stable gain for the OPA847. Here, a
target of NG2 = 24 is used. The second parameter is the desired
low-frequency signal gain, which also sets the low-frequency
noise gain (NG1). To simplify this discussion, we will target a
maximally flat, 2nd-order, low-pass Butterworth frequency response (Q = 0.707). The signal gain shown in Figure 5 sets the
low-frequency noise gain to NG1 = 1 + RF/RG (= 5.25 in this
example). Then, using only these two gains and the GBP for the
OPA847 (3900MHz), the key frequency in the compensation is
set by Equation 4.
ZO =
GBP
NG1
NG1
1−
− 1− 2
2
NG2
NG2
NG 1
(4)
Physically, this ZO (4.4MHz for the values shown above) is
set by 1/(2πRF(CF + CS)) and is the frequency at which the
rising portion of the noise gain would intersect the unity gain
if projected back to a 0dB gain. The actual zero in the noise
gain occurs at NG1 • ZO and the pole in the noise gain occurs
at NG2 • ZO. That pole is physically set by 1/(RFCF). Since
GBP is expressed in Hz, multiply ZO by 2π and use to get CF
by solving Equation 5.
CF =
1
(= 1.76pF)
2πRF ZO NG2
(5)
OPA847
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SBOS251E
Finally, since CS and CF set the high-frequency noise gain,
determine CS using Equation 6 (solving for CS by using NG2 = 24):
CS = (NG2 − 1)CF
LOW GAIN INVERTING BANDWIDTH
1
(6)
0
Normalized Gain (1dB)
which gives CS = 40.6pF.
Both of these calculated values have been reduced slightly
in Figure 5 to account for parasitics. The resulting closedloop bandwidth is approximately equal to Equation 7.
f –3dB ≅ ZO • GBP
(7)
For the values shown in Figure 5, f–3dB is approximately
131MHz. This is less than that predicted by simply dividing
the GBP product by NG1. The compensation network controls
the bandwidth to a lower value, while providing the full slew
rate at the output and an exceptional distortion performance
due to increased loop gain at frequencies below NG1 • ZO.
–1
–2
G = –8
VO = 0.2VPP
–3
–4
G = –1
–5
–6
G = –2
–7
–8
–9
1
10
100
+5V
VDIS
OPA847
The Typical Characteristics show the exceptional bandwidth
control possible using this technique at low inverting gains.
Figure 6 repeats the measured results with the test circuit shown.
LOW-NOISE FIGURE,
HIGH DYNAMIC RANGE AMPLIFIER
The low input noise voltage of the OPA847 and its very high
2-tone, 3rd-order intermodulation intercept can be used to
good advantage as a fixed-gain amplifier. While input noise
1000
Frequency (MHz)
Using this low-gain inverting compensation, along with the
differential structure for the circuit shown on the front page of
this data sheet, gives a significant reduction in harmonic
distortion. The measured distortion at 2VPP output does not
rise above –95dB until frequencies > 20MHz are applied.
The compensation capacitors, CS and CF, are set by targeting
a high-frequency noise gain of 21 and using equations 4 through
6. This approach allows relatively low inverting gain applications
to use the full slew rate and low input noise of the OPA847.
G = –4
RG
–5V
VO
RF
750Ω
VI
0Ω Source
CS
CF
FIGURE 6. Low-Gain Inverting Performance.
figures in the 10dB range (for a matched 50Ω input) are
easily achieved with just the OPA847, Figure 7 illustrates a
technique to reduce the noise figure even further, while
providing a broadband, high-gain HF amplifier stage using
two stages of the OPA847.
6.19kΩ
+5V
Input match
set by this
feedback path
+5V
–5V
50Ω Source
PI
750Ω
1:2
1.5kΩ
> 55dBm
intercept
to 30MHz
OPA847
200Ω
4.3dB
Noise
Figure
PO
OPA847
10pF
1.6pF
–5V
46pF
420Ω
30.1Ω
Overall Gain
PO
PI
= 35.6dB
FIGURE 7. Very High Dynamic Range HF Amplifier.
OPA847
SBOS251E
www.ti.com
13
This circuit uses two stages of forward gain with an overall
feedback loop to set the input impedance match. The input
transformer provides both a noiseless voltage gain and a
signal inversion to retain an overall noninverting signal path
from PI to PO. The second amplifier stage is inverting to
provide the correct feedback polarity through the 6.19kΩ
resistor. To achieve a 50Ω input match at the primary of the
1:2 transformer, the secondary must see a 200Ω load impedance. At higher frequencies, the match is provided by the
200Ω resistor in series with 10pF. The low-noise figure
(4.3dB) for this circuit is achieved by using the transformer,
the low-voltage noise OPA847, and the input match set by
the feedback at lower frequencies intended for this HF
design. The 1st-stage amplifier provides a gain of +15V/V.
The very high SFDR is provided by operating the output
stage at a low signal gain of –2 and using the inverting
compensation technique to shape the noise gain to hold it
stable. This 2nd-stage compensation is set to intentionally
bandlimit the overall response to approximately 100MHz. For
output loads > 400Ω, this circuit can give a 2-tone SFDR that
exceeds 90dB through 30MHz. In narrowband applications,
the 3rd-order intercept exceeds 55dBm. Besides offering a
very high dynamic range, this circuit improves on standard
HF amplifiers by offering a precisely controlled gain and a
very flexible output interface capability.
+5V
VI
Figure 8 shows a resistor-based compensation technique
that allows the flatness at low noninverting signal gains to be
controlled separately from the signal gain. This approach
retains the full slew rate to the output but gives up some of
the low-noise benefit of the OPA847. Including the effect of
the total source impedance (25Ω in Figure 8), tuning resistor
R1 can be set using Equation 8.
R1 =
RF + RS A V
NG − A V
where:
AV = desired signal gain (+12V/V in Figure 8)
NG = target noise gain (adjusted in Figure 9)
RS = total source impedance
14
(8)
50Ω
50Ω
R1
50Ω
OPA847
VO
–5V
RF
750Ω
RG
66.5Ω
FIGURE 8. Low Noninverting Gain Flatness Trim.
The effect of this noninverting gain flatness tune is shown in
Figure 9. At an NG of 12, R1 is removed and only RF and RG
are present in Figure 8. The peaking is typically 4.5dB, as
shown in the small-signal frequency response curves versus
gain curves at this setting. As R1 is decreased, the operating
noise gain (NG) increases, reducing the peaking and bandwidth until the nominal design point of +20 noise gain gives
a non-peaked response.
NONINVERTING GAIN FLATNESS TUNE
NONINVERTING GAIN FLATNESS COMPENSATION
0.5
Deviation from 21.58dB Gain (0.1dB)
Decreasing the operating gain from the nominal design point of
+20 decreases the phase margin. This increases Q for the
closed-loop poles, peaks up the frequency response, and
extends the bandwidth. A peaked frequency response shows
overshoot and ringing in the pulse response, as well as higher
integrated output noise. When operating the OPA847 at a
noninverting gain < +12V/V, increased peaking and possible
sustained oscillations may result. However, operation at low
gains may be desirable to take advantage of the higher slew
rate and exceptional DC precision of the OPA847. Numerous
external compensation techniques are suggested for operating
a high-gain op amp at low gains. Most of these give zero/pole
pairs in the closed-loop response that cause long term settling
tails in the pulse response and/or phase nonlinearity in the
frequency response.
VDIS
VO = 200mVPP
AV = +12V/V
NG = Noise Gain
0.4
0.3
NG = 12
NG = 14
NG = 16
0.2
0.1
0
–0.1
NG = 18
–0.2
NG = 20
–0.3
–0.4
–0.5
1
10
100
1000
Frequency (MHz)
FIGURE 9. Frequency Response Flatness with External
Tuning Resistor.
DIFFERENTIAL OPERATION
Operating two OPA847 amplifiers in a differential inverting
configuration can further suppress even-order harmonic terms.
The Typical Characteristics show measured performance for
this condition. These measurements were done at the relatively
high gain of 40V/V. Even lower distortion is possible operating
at lower gains using the external inverting compensation techniques, as discussed previously. For the distortion data presented in Figure 10, the output swing is increased to 4V PP into
400Ω to allow direct comparison to the single-channel data at
2VPP into 200Ω. Comparing the 2nd- and 3rd-harmonics at
20MHz in Figure 10 to the gain of +20, 2V PP, 200Ω data, shows
the 2nd-harmonic is reduced to –76dBc (from –67dBc) and the
3rd-harmonic is reduced from –80dBc to –85dBc. Using the two
OPA847
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SBOS251E
Harmonic Distortion (dBc)
–65
supply of +5V and up to a single supply of +12V. If shutdown
is desired for single-supply operation, it is important to realize
that the shutdown pin is referenced from the positive supply
pin. Open collector (drain) interfaces are suggested for
single-supply operation above +5V.
GD = 40V/V
RL = 400Ω
VO = 4VPP
–75
2nd-Harmonic
–85
DESIGN-IN TOOLS
–95
DEMONSTRATION FIXTURES
3rd-Harmonic
–105
–115
1
10
100
Frequency (MHz)
FIGURE 10. Differential Distortion vs Frequency.
amplifiers in this configuration has significantly reduced the
2nd-harmonic, even after doubling the output voltage swing (to
4VPP) and the gain (to 40V/V).
The OPA847 can be operated from a single power supply if
system constraints require it. Operation from a single +5V to
+12V supply is possible with minimal change in AC performance. The Typical Characteristics show the input and
output voltage ranges for a bipolar supply range from ±2.5V
to ±6.0V. The Common-Mode Input Range and Output Swing
vs Supply Voltage curve shows that the required headroom
on both the input and output pins remains at approximately
1.5V over this entire range. On a single +5V supply, for
instance, this means the noninverting input should remain
centered at +2.5V ± 1V, as should the output pin. Figure 11
shows an example application biasing the noninverting input
at mid-supply and running an AC-coupled input to the inverting gain path. Since the gain resistor is blocked off for DC,
the bias point on the noninverting input appears at the output,
centering up the output as well as on the power supply. The
OPA847 can support this mode of operation down to a single
+5V
2RF
Power-supply decoupling
not shown.
V
R
VO = CC – VI F
OPA847
2
RG
RF
VI
FIGURE 11. Single-Supply Inverting Amplifier.
LITERATURE
NUMBER
SO-8
SOT23-6
DEM-OPA-SO-1B
DEM-OPA-SOT-1B
SBOU026
SBOU027
The demonstration fixtures can be requested at the Texas
Instruments web site (www.ti.com) through the OPA847
product folder.
MACROMODELS AND APPLICATIONS SUPPORT
Computer simulation of circuit performance using SPICE is
often a quick way to analyze the performance of the OPA847
in its intended application. This is particularly true for video
and RF amplifier circuits where parasitic capacitance and
inductance can play a major role in circuit performance. A
SPICE model for the OPA847 is available through the TI web
site (www.ti.com). These models do a good job of predicting
small-signal AC and transient performance under a wide
variety of operating conditions. They do not do as well in
predicting the harmonic distortion characteristics. These
models do not attempt to distinguish between the package
types in their small-signal AC performance.
The OPA847 provides a very low input noise voltage while
requiring a low 18.1mA of quiescent current. To take full
advantage of this low input noise, careful attention to the other
possible noise contributors is required. See Figure 12 for the
op amp noise analysis model with all the noise terms included.
In this model, all the noise terms are taken to be noise voltage
or current density terms in either nV/√Hz or pA/√Hz.
The total output spot noise voltage is computed as the
square root of the squared contributing terms to the output
noise power. This computation adds all the contributing noise
powers at the output by superposition, then takes the square
OPA847
SBOS251E
ORDERING
NUMBER
SETTING RESISTOR VALUES TO MINIMIZE NOISE
+12V
Range
VDIS
RG
OPA847ID
OPA847IDBV
PACKAGE
OPERATING SUGGESTIONS
+VCC
0.01µF
PRODUCT
TABLE I. Demonstration Fixtures by Package.
SINGLE-SUPPLY OPERATION
2RF
Two printed circuit boards (PCBs) are available to assist in
the initial evaluation of circuit performance using the OPA847
in its two package options. Both of these are offered free
of charge as unpopulated PCBs, delivered with a user’s
guide. The summary information for these fixtures is shown
in Table I.
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15
ENI
EO
OPA847
RS
IBN
ERS
RF
√4kTRS
IBI
RG
4kT
RG
√4kTRF
4kT = 1.6E – 20J
at 290°K
FIGURE 12. Op Amp Noise Analysis Model.
root to get back to a spot noise voltage. Equation 9 shows the
general form for this output noise voltage using the terms
illustrated in Figure 11.
(9)
EO =
(E
2
NI
)
+ (IBN RS )2 + 4kTRS NG2 + (IBI RF )2 + 4kTRF NG
Dividing this expression by the noise gain (NG = 1 + RF/RG)
gives the equivalent input-referred spot noise voltage at the
noninverting input, as shown in Equation 10.
(10)
I R 2 4kTRF
2
EN = ENI
+ (IBN RS )2 + 4kTRS + BI F +
NG
NG
Putting high resistor values into Equation 10 can quickly
dominate the total equivalent input-referred noise. A 45Ω
source impedance on the noninverting input adds a Johnson
voltage noise term equal to the amplifier’s voltage noise by
itself. As a simplifying constraint, set RG = RS in Equation 10
and assume an RS/2 source impedance at the noninverting
input, where RS is the signal source impedance and another
matching RS to ground is at the noninverting input. This
results in Equation 11, where NG > 12 is assumed to further
simplify the expression.
2
EN = ENI
+
5
(IB RS )2 + 4kT 3R2S
4
(11)
Evaluating this expression for RS = 50Ω gives a total equivalent input noise of 1.4nV/√Hz. Note that at these higher
gains, the simplified input referred spot noise expression of
Equation 11 does not include the gain. This is a good
approximation for NG > 12, as is typically required by stability
considerations.
FREQUENCY RESPONSE CONTROL
Voltage-feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the Electrical Characteristics. Ideally, dividing GBP by the noninverting signal gain (also called the
Noise Gain, or NG) predicts the closed-loop bandwidth. In
16
practice, this only holds true when the phase margin approaches 90°, as it does in high-gain configurations. At low
gains (increased feedback factors), most high-speed amplifiers exhibit a more complex response with lower phase
margin. The OPA847 is compensated to give a maximally flat
2nd-order Butterworth closed-loop response at a noninverting
gain of +20 (see Figure 1). This results in a typical gain of
+20 bandwidth of 350MHz, far exceeding that predicted by
dividing the 3900MHz GBP by 20. Increasing the gain causes
the phase margin to approach 90° and the bandwidth to more
closely approach the predicted value of (GBP/NG). At a gain
of +50, the OPA847 very nearly matches the 78MHz bandwidth predicted using the simple formula and the typical GBP
of 3900MHz.
Inverting operation offers some interesting opportunities to
increase the available GBP. When the source impedance is
matched by the gain resistor (see Figure 2), the signal gain
is (1 + RF/RG), while the noise gain for bandwidth purposes
is (1 + RF/2RG). This cuts the noise gain almost in half,
increasing the minimum operating gain for inverting operation under these condition to –22 and the equivalent gain
bandwidth product to > 7.8GHz.
DRIVING CAPACITIVE LOADS
One of the most demanding, and yet very common, load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADC, including additional
external capacitance that may be recommended to improve
ADC linearity. A high-speed, high open-loop gain amplifier
like the OPA847 can be very susceptible to decreased
stability and may give closed-loop response peaking when a
capacitive load is placed directly on the output pin. When the
amplifier’s open-loop output resistance is considered, this
capacitive load introduces an additional pole in the signal
path that can decrease the phase margin. Several external
solutions to this problem are suggested. When the primary
considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most
effective solution is to isolate the capacitive load from the
feedback loop by inserting a series isolation resistor between
the amplifier output and the capacitive load. This does not
eliminate the pole from the loop response, but rather shifts it
and adds a zero at a higher frequency. The additional zero
acts to cancel the phase lag from the capacitive load pole,
thus increasing the phase margin and improving stability.
The Typical Characteristics help the designer pick a recommended RS versus capacitive load. The resulting frequency
response curves show a flat response for several selected
capacitive loads and recommended RS combinations. Parasitic capacitive loads greater than 2pF can begin to degrade
the performance of the OPA847. Long PCB traces, unmatched cables, and connections to multiple devices can
easily cause this value to be exceeded. Always consider this
effect carefully and add the recommended series resistor as
close as possible to the OPA847 output pin (see the Board
Layout section).
OPA847
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SBOS251E
The OPA847 has an extremely low 3rd-order harmonic
distortion. This also gives a high 2-tone 3rd-order
intermodulation intercept, as shown in the Typical Characteristics. This intercept curve is defined at the 50Ω load when
driven through a 50Ω matching resistor to allow direct comparisons to RF devices. This matching network attenuates
the voltage swing from the output pin to the load by 6dB. If
the OPA847 drives directly into the input of a high-impedance device, such as an ADC, this 6dB attenuation is not
taken. Under these conditions, the intercept as reported in
the Typical Characteristics increases by a minimum of 6dBm.
The intercept is used to predict the intermodulation spurious
power levels for two closely spaced frequencies. If the two
test frequencies, f1 and f2, are specified in terms of average
and delta frequency, fO = (f1 + f2)/2 and ∆f = f2 – f1 /2, the
two 3rd-order, close-in spurious tones appear at fO ± 3 • ∆f.
The difference between the two equal test-tone power levels
and these intermodulation spurious power levels is given by
∆dBc = 2(IM3 – PO), where IM3 is the intercept taken from
the Typical Characteristics and PO is the power level in dBm
at the 50Ω load for one of the two closely spaced test
frequencies. For instance, at 30MHz, the OPA847 at a gain
of +20 has an intercept of 34dBm at a matched 50Ω load.
The OPA847 can provide excellent DC signal accuracy due
to its high open-loop gain, high common-mode rejection, high
power-supply rejection, and low input offset voltage and bias
current offset errors. To take full advantage of its low ±0.5mV
input offset voltage, careful attention to the input bias current
cancellation is also required. The low-noise input stage for
the OPA847 has a relatively high input bias current (19µA
typical into the pins), but with a very close match between the
two input currents—typically ±100nA input offset current.
Figures 13 and 14 show typical distributions of input offset
voltage and current for the OPA847.
1200
Mean = 48µV
Standard Deviation = 110µV
Total Count = 4040
1000
800
600
400
200
0
< –600
< –540
< –480
< –420
< –360
< –300
< –240
< –180
< –120
< –60
0
< 60
< 120
< 180
< 240
< 300
< 360
< 420
< 480
< 540
< 600
> 600
Generally, until the fundamental signal reaches very high
frequencies or powers, the 2nd-harmonic dominates the distortion with a negligible 3rd-harmonic component. Focusing
then on the 2nd-harmonic, increasing the load impedance
improves distortion directly. Remember that the total load
includes the feedback network—in the noninverting configuration this is the sum of RF + RG, while in the inverting
configuration this is only RF (see Figure 2). Increasing the
output voltage swing increases harmonic distortion directly. A
6dB increase in output swing generally increases the 2ndharmonic 12dB and the 3rd-harmonic 18dB. Increasing the
signal gain also increases the 2nd-harmonic distortion. Finally,
the distortion increases as the fundamental frequency increases due to the rolloff in the loop gain with frequency.
Conversely, the distortion improves going to lower frequencies
down to the dominant open-loop pole at approximately 80kHz.
DC ACCURACY AND OFFSET CONTROL
µV
FIGURE 13. Input Offset Voltage Distribution in µV.
900
800
700
Mean = 50nA
Standard Deviation = 120nA
Total Count = 4040
600
500
400
300
200
100
0
< –600
< –540
< –480
< –420
< –360
< –300
< –240
< –180
< –120
< –60
0
< 60
< 120
< 180
< 240
< 300
< 360
< 420
< 480
< 540
< 600
> 600
The OPA847 is capable of delivering an exceptionally low
distortion signal at high frequencies over a wide range of
gains. The distortion plots in the Typical Characteristics show
the typical distortion under a wide variety of conditions. Most
of these plots are limited to a 110dB dynamic range. The
OPA847’s distortion driving a 200Ω load does not rise above
–90dBc until either the signal level exceeds 2.0VPP and/or
the fundamental frequency exceeds 5MHz. Distortion in the
audio band is < –130dBc.
Count
DISTORTION PERFORMANCE
If the full envelope of the two frequencies needs to be 2VPP,
this requires each tone to be 4dBm. The 3rd-order
intermodulation spurious tones will then be 2(34 – 4) = 60dBc
below the test-tone power level (–56dBm). If this same 2VPP
2-tone envelope is delivered directly into the input of an ADC
without the matching loss or the loading of the 50Ω network,
the intercept would increase to at least 40dBm.
With the same signal and gain conditions, but now driving
directly into a light load, the spurious tones will then be at
least 2(40 – 4) = 72dBc below the 4dBm test-tone power
levels centered on 30MHz. Tests have shown that they are
in fact much lower due to the lighter loading presented by
most ADCs.
Count
The criterion for setting the RS resistor is a maximum bandwidth, flat frequency response at the load. For the OPA847
operating in a gain of +20, the frequency response at the
output pin is very flat to begin with, allowing relatively small
values of RS to be used for low capacitive loads. As the signal
gain is increased, the unloaded phase margin also increases.
Driving capacitive loads at higher gains requires lower RS
values than those shown for a gain of +20.
nA
FIGURE 14. Input Offset Current Distribution in nA.
OPA847
SBOS251E
www.ti.com
17
The total output offset voltage can be considerably reduced
by matching the source impedances looking out of the two
inputs. For example, one way to add bias current cancellation to the circuit of Figure 1 is to insert a 12.1Ω series
resistor into the noninverting input from the 50Ω terminating
resistor. When the 50Ω source resistor is DC-coupled, this
increases the source impedance for the noninverting input
bias current to 37.1Ω. Since this is now equal to the impedance looking out of the inverting input (RF || RG) for Figure 1,
the circuit cancels the gains for the bias currents to the
output, leaving only the offset current times the feedback
resistor as a residual DC error term at the output. Using the
750Ω feedback resistor, this output error is now less than
±0.85µA • 750Ω = ±640µV over the full temperature range for
the circuit of Figure 1, with a 12.1Ω resistor added as
described. The output DC offset is then dominated by the
input offset voltage multiplied by the signal gain. For the
circuit of Figure 1, this is a worst-case output DC offset of
±0.6mV • 20 = ±12mV over the full temperature range.
A fine-scale output offset null, or DC operating point adjustment, is sometimes required. Numerous techniques are
available for introducing a DC offset control into an op amp
circuit. Most of these techniques eventually reduce to setting
up a DC current through the feedback resistor. One key
consideration to selecting a technique is to ensure that it has
a minimal impact on the desired signal path frequency
response. If the signal path is intended to be noninverting,
the offset control is best applied as an inverting summing
signal to avoid interaction with the signal source. If the signal
path is intended to be inverting, applying the offset control to
the noninverting input can be considered. For a DC-coupled
inverting input signal, this DC offset signal sets up a DC
current back into the source that must be considered. An
offset adjustment placed on the inverting op amp input can
also change the noise gain and frequency response flatness.
Figure 15 shows one example of an offset adjustment for a
DC-coupled signal path that has minimum impact on the
signal frequency response.
In this case, the input is brought into an inverting gain resistor
with the DC adjustment as an additional current summed into
the inverting node. The resistor values setting this offset
adjustment are much larger than the signal path resistors.
This ensures that this adjustment has minimal impact on the
loop gain and, hence, the frequency response.
POWER SHUTDOWN OPERATION
The OPA847 provides an optional power shutdown feature
that can be used to reduce system power. If the V DIS control
pin is left unconnected, the OPA847 operates normally. This
shutdown is intended only as a power saving feature. Forward path isolation is very good for small signals. Large
signal isolation is not ensured. Using this feature to multiplex
two or more outputs together is not recommended. Large
signals applied to the shutdown output stages can turn on
parasitic devices, degrading signal linearity for the desired
channel.
Turn-on time is very quick from the shutdown condition,
typically < 60ns. Turn-off time is strongly dependent on the
external circuit configuration, but is typically 200ns for the
circuit of Figure 1. Using the OPA847 with higher external
resistor values, such has high-gain transimpedance circuits,
slows the shutdown time since the time constants for the
internal nodes to discharge are longer.
To shutdown, the control pin must be asserted low. This logic
control is referenced to the positive supply, as shown in the
simplified circuit of Figure 16.
+VS
8kΩ
Q1
+5V
VCC
Power-supply decoupling
not shown.
48Ω
0.1µF
OPA847
17kΩ
VDIS
VEE
–5V
+5V
RG
50Ω
VI
±200mV Output Adjustment
100Ω
0.1µF
5kΩ
VO
VI
=–
RF
RG
= –20V/V
–5V
FIGURE 15. DC-Coupled, Inverting Gain of –20 with Output
Offset Adjustment.
18
IS
Control
–VS
FIGURE 16. Simplified Shutdown Control Circuit.
RF
1kΩ
5kΩ
20kΩ
120kΩ
VO
In normal operation, base current to Q1 is provided through
the 120kΩ resistor, while the emitter current through the 8kΩ
resistor sets up a voltage drop that is inadequate to turn on
the two diodes in Q1’s emitter. As V DIS is pulled low,
additional current is pulled through the 8kΩ resistor, eventually turning on these two diodes (≈ 180µA). At this point,
any further current pulled out of V DIS goes through those
diodes holding the emitter-base voltage of Q1 at approximately 0V. This shuts off the collector current out of Q1,
turning the amplifier off. The supply current in the shutdown
mode is only that required to operate the circuit of Figure 16.
OPA847
www.ti.com
SBOS251E
The shutdown feature for the OPA847 is a positive-supply
referenced, current-controlled interface. Open-collector (or drain)
interfaces are most effective, as long as the controlling logic
can sustain the resulting voltage (in open mode) that appears
at the V DIS pin. The V DIS pin voltage is one diode below the
positive supply voltage applied to the OPA847 if the logic
voltage is open. For voltage output logic interfaces, the on/off
voltage levels described in the Electrical Characteristics apply
only for a +5V supply. An open-drain interface is recommended
for a shutdown operation using a higher positive supply and/or
logic families with inadequate high-level voltage swings.
THERMAL ANALYSIS
The OPA847 does not require heatsinking or airflow in most
applications. Maximum desired junction temperature sets the
maximum allowed internal power dissipation, as described
here. In no case should the maximum junction temperature
be allowed to exceed 150°C.
Operating junction temperature (TJ) is given by TA + PD • θJA.
The total internal power dissipation (PD) is the sum of
quiescent power (PDQ) and additional power dissipated in the
output stage (PDL) to deliver load power. Quiescent power is
simply the specified no-load supply current times the total
supply voltage across the part. PDL depends on the required
output signal and load but would, for a grounded resistive
load, be at a maximum when the output is fixed at a voltage
equal to half either supply voltage (for equal bipolar supplies). Under this worst-case condition, PDL = VS2/(4 • RL),
where RL includes feedback network loading. This is the
absolute highest power that can be dissipated for a given RL.
All actual applications dissipate less power in the output
stage.
Note that it is the power in the output stage and not into the
load that determines internal power dissipation.
As a worst-case example, compute the maximum TJ using an
OPA847IDBV (SOT23-6 package) in the circuit of Figure 1
operating at the maximum specified ambient temperature of
+85°C and driving a grounded 100Ω load. Maximum internal
power is:
PD = 10V • 18.9mA + 52/(4(100Ω || 789Ω)) = 259mW
Maximum TJ = +85°C + (0.26W • 150°C/W) = 124°C
All actual applications will operate at a lower junction temperature than the 124°C computed above. Compute your
actual output stage power to get an accurate estimate of
maximum junction temperature, or use the results shown
here as an absolute maximum.
BOARD LAYOUT
Achieving optimum performance with a high-frequency amplifier like the OPA847 requires careful attention to board
layout parasitics and external component types. Recommendations that will optimize performance include:
a) Minimize parasitic capacitance to any AC ground for all
of the signal I/O pins. Parasitic capacitance on the output and
inverting input pins can cause instability: on the noninverting
input, it can react with the source impedance to cause
unintentional bandlimiting. To reduce unwanted capacitance,
create a window around the signal I/O pins in all of the
ground and power planes around these pins. Otherwise,
ground and power planes should be unbroken elsewhere on
the board.
b) Minimize the distance (< 0.25") from the power-supply
pins to high-frequency 0.1µF decoupling capacitors. At the
device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power-supply
connections should always be decoupled with these capacitors. Larger (2.2µF to 6.8µF) decoupling capacitors, effective
at lower frequencies, should also be used on the main supply
pins. These can be placed somewhat further from the device
and can be shared among several devices in the same area
of the PC board.
c) Careful selection and placement of external components preserves the high-frequency performance of the
OPA847. Use resistors that have low reactance at high
frequencies. Surface-mount resistors work best and allow a
tighter overall layout. Metal film and carbon composition
axially leaded resistors can also provide good high-frequency performance. Again, keep their leads and PCB trace
length as short as possible. Never use wirewound-type
resistors in a high-frequency application. Since the output pin
and inverting input pin are the most sensitive to parasitic
capacitance, always position the feedback and series output
resistor, if any, as close as possible to the output pin. Other
network components, such as noninverting input termination
resistors, should also be placed close to the package. Where
double-side component mounting is allowed, place the feedback resistor directly under the package on the other side of
the board between the output and inverting input pins. Even
with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant
time constants that can degrade performance. Good axial
metal film or surface-mount resistors have approximately
0.2pF in shunt with the resistor. For resistor values > 2.0kΩ,
this parasitic capacitance can add a pole and/or zero below
400MHz that can effect circuit operation. Keep resistor values as low as possible, consistent with load driving considerations. It has been suggested here that a good starting
point for design would be to set RG to 39.2Ω. Doing this
automatically keeps the resistor noise terms low, and minimizes the effect of their parasitic capacitance. Transimpedance applications can use much higher resistor values. The
compensation techniques described in this data sheet allow
excellent frequency response control, even with very high
feedback resistor values.
d) Connections to other wideband devices on the board
can be made with short, direct traces or through onboard
transmission lines. For short connections, consider the trace
and the input to the next device as a lumped capacitive load.
Relatively wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up around
them. Estimate the total capacitive load and set RS from the
plot of Recommended RS vs Capacitive Load. Low parasitic
OPA847
SBOS251E
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19
capacitive loads (< 4pF) may not need an RS, since the
OPA847 is nominally compensated to operate with a 2pF
parasitic load. Higher parasitic capacitive loads without an RS
are allowed as the signal gain increases from +20V/V (increasing the unloaded phase margin). If a long trace is
required, and the 6dB signal loss intrinsic to a doublyterminated transmission line is acceptable, implement a
matched impedance transmission line using microstrip or
stripline techniques (consult an ECL design handbook for
microstrip and stripline layout techniques). A 50Ω environment is normally not necessary onboard and, in fact, a higher
impedance environment improves distortion, as shown in the
distortion versus load plots. With a characteristic board trace
impedance defined based on board material and trace dimensions, a matching series resistor into the trace from the
output of the OPA847 is used, as well as a terminating shunt
resistor at the input of the destination device. Remember
also that the terminating impedance is the parallel combination of the shunt resistor and the input impedance of the
destination device; this total effective impedance should be
set to match the trace impedance. If the 6dB attenuation of
a doubly-terminated transmission line is unacceptable, a
long trace can be series-terminated at the source-end only.
Treat the trace as a capacitive load in this case and set the
series resistor value as shown in the plot of Recommended
RS vs Capacitive Load. This does not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some
signal attenuation due to the voltage divider formed by the
series output into the terminating impedance.
e) Socketing a high-speed part like the OPA847 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an extremely troublesome parasitic network that can make it
20
almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the OPA847
onto the board.
INPUT AND ESD PROTECTION
The OPA847 is built using a very high-speed complementary
bipolar process. The internal junction breakdown voltages are
relatively low for these very small geometry devices. These
breakdowns are reflected in the Absolute Maximum Ratings
table. All device pins are protected with internal ESD protection diodes to the power supplies, as shown in Figure 17.
+VCC
External
Pin
Internal
Circuitry
–VCC
FIGURE 17. Internal ESD Protection.
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (for example, in systems with ±15V
supply parts driving into the OPA847), current limiting series
resistors should be added into the two inputs. Keep these
resistor values as low as possible, since high values degrade
both noise performance and frequency response.
OPA847
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SBOS251E
Revision History
DATE
REVISION
PAGE
12/08
E
2
4/06
D
15
SECTION
DESCRIPTION
Absolute Maximum Ratings Changed minimum Storage Temperature Range from −40°C to −65°C.
Design-In Tools
Board part number changed.
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
OPA847
SBOS251E
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21
PACKAGE OPTION ADDENDUM
www.ti.com
6-Feb-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
OPA847ID
ACTIVE
SOIC
D
8
75
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
847
OPA847IDBVR
ACTIVE
SOT-23
DBV
6
3000
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OATI
OPA847IDBVT
ACTIVE
SOT-23
DBV
6
250
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OATI
OPA847IDBVTG4
ACTIVE
SOT-23
DBV
6
250
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OATI
OPA847IDG4
ACTIVE
SOIC
D
8
75
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
847
OPA847IDR
ACTIVE
SOIC
D
8
2500
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
847
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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