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OPA861IDBVR

OPA861IDBVR

  • 厂商:

    BURR-BROWN(德州仪器)

  • 封装:

    SOT23-6

  • 描述:

    IC TRANSCOND 1 CIRCUIT SOT23-6

  • 数据手册
  • 价格&库存
OPA861IDBVR 数据手册
OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 Wide Bandwidth Operational Transconductance Amplifier (OTA) Check for Samples: OPA861 FEATURES 1 • • • • • Wide Bandwidth (80MHz, Open-Loop, G = +5) High Slew Rate (900V/µs) High Transconductance (95mA/V) External IQ-Control Low Quiescent Current (5.4mA) APPLICATIONS • • • • • • • Video/Broadcast Equipment Communications Equipment High-Speed Data Acquisition Wideband LED Drivers Control Loop Amplifiers Wideband Active Filters Line Drivers The OTA or voltage-controlled current source can be viewed as an ideal transistor. Like a transistor, it has three terminals—a high impedance input (base), a low-impedance input/output (emitter), and the current output (collector). The OPA861, however, is selfbiased and bipolar. The output collector current is zero for a zero base-emitter voltage. AC inputs centered about zero produce an output current, which is bipolar and centered about zero. The transconductance of the OPA861 can be adjusted with an external resistor, allowing bandwidth, quiescent current, and gain trade-offs to be optimized. Used as a basic building block, the OPA861 simplifies the design of AGC amplifiers, LED driver circuits for fiber optic transmission, integrators for fast pulses, fast control loop amplifiers and control amplifiers for capacitive sensors, and active filters. The OPA861 is available in SO-8 and SOT23-6 surface-mount packages. DESCRIPTION The OPA861 is a versatile monolithic component designed for wide-bandwidth systems, including high performance video, RF and IF circuitry. The OPA861 is a wideband, bipolar operational transconductance amplifier (OTA). 0 −10 R C1 R V IN V OUT C2 Gain (dB) −20 −30 10MHz Low−Pass Filter −40 20kHz Low−Pass Filter −50 −60 −70 −80 1k 10k 100k 1M 10M 100M 1G Frequency (Hz) Low−Pass Negative Impedance Converter (NIC) Filter Frequency Response of 20kHz and 10MHz Low−Pass NIC Filters 1 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2005–2013, Texas Instruments Incorporated OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION (1) (1) SPECIFIED TEMPERATURE RANGE PACKAGE MARKING PRODUCT PACKAGE PACKAGE DESIGNATOR OPA861 SO-8 D –45°C to +85°C OPA861 OPA861 SOT23-6 DBV –45°C to +85°C N5R ORDERING NUMBER TRANSPORT MEDIA, QUANTITY OPA861ID Rails, 75 OPA861IDR Tape and Reel, 2500 OPA861IDBVT Tape and Reel, 250 OPA861IDBVR Tape and Reel, 3000 For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. ABSOLUTE MAXIMUM RATINGS (1) Power Supply ±6.5VDC Internal Power Dissipation See Thermal Information Differential Input Voltage ±1.2V Input Common-Mode Voltage Range ±VS Storage Temperature Range: D –65°C to +125°C Lead Temperature (soldering, 10s) +260°C Junction Temperature (TJ) +150°C ESD Rating: (1) (2) Human Body Model (HBM) (2) 1500V Charge Device Model (CDM) 1000V Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operations of the device at these and any other conditions beyond those specified is not supported. Pin 2 for the SO-8 package > 500V HBM. Pin 4 for the SOT23-6 package > 500V HBM. Figure 1. PIN CONFIGURATION Top View I Q Adjust 1 8 C E 2 7 V+ = +5V B 3 6 NC V− = −5V 4 5 NC I Q Adjust 1 6 +VS −VS 2 5 C B 3 4 E SOT23−6 SO−8 2 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 ELECTRICAL CHARACTERISTICS: VS = ±5V RL = 500Ω and RADJ = 250Ω, unless otherwise noted. OPA861ID, IDBV TYP PARAMETER MIN/MAX OVER TEMPERATURE CONDITIONS +25°C +25°C (2) 0°C to 70°C (3) –40°C to +85°C (3) UNITS MIN/ MAX TEST LEVEL (1) G = +5, VO = 200mVPP, RL = 500Ω 80 77 75 74 MHz min B G = +5, VO = 1VPP 80 MHz typ C G = +5, VO = 5VPP 80 MHz typ C G = +5, VO = 5V Step 900 V/µs min B VO = 1V Step 4.4 ns typ C OTA—Open-Loop (see Figure 33) AC PERFORMANCE Bandwidth Slew Rate Rise Time and Fall Time Harmonic Distortion 860 850 840 G = +5, VO = 2VPP, 5MHz 2nd-Harmonic RL = 500Ω –68 –55 –54 –53 dB max B 3rd-Harmonic RL = 500Ω –57 –52 –51 –49 dB max B Base Input Voltage Noise f > 100kHz 2.4 3.0 3.3 3.4 nV/√Hz max B Base Input Current Noise f > 100kHz 1.7 2.4 2.45 2.5 pA/√Hz max B Emitter Input Current Noise f > 100kHz 5.2 15.3 16.6 17.5 pA/√Hz max B Minimum OTA Transconductance (gm) VO = ±10mV, RC = 50Ω, RE = 0Ω 95 80 77 75 mA/V min A Maximum OTA Transconductance (gm) VO = ±10mV, RC = 50Ω, RE = 0Ω 95 150 155 160 mA/V max A VB = 0V, RC = 0Ω, RE = 100Ω ±3 ±12 ±15 ±20 mV max A ±67 ±120 μV/°C max B ±6 ±6.6 μA max A ±20 ±25 nA/°C max B ±125 ±140 μA max A ±500 ±600 nA/°C max B ±30 ±38 μA max A ±250 ±300 nA/°C max B ±3.6 ±3.6 OTA DC PERFORMANCE (4) (see Figure 33) B-Input Offset Voltage Average B-Input Offset Voltage Drift B-Input Bias Current Average B-Input Bias Current Drift E-Input Bias Current Average E-Input Bias Current Drift C-Output Bias Current Average C-Output Bias Current Drift VB = 0V, RC = 0Ω, RE = 100Ω VB = 0V, RC = 0Ω, RE = 100Ω ±1 ±5 VB = 0V, RC = 0Ω, RE = 100Ω VB = 0V, VC = 0V ±30 ±100 VB = 0V, VC = 0V VB = 0V, VC = 0V ±5 ±18 VB = 0V, VC = 0V OTA INPUT (see Figure 33) B-Input Voltage Range ±4.2 B-Input Impedance ±3.7 455 || 2.1 V min B kΩ || pF typ C Min E-Input Resistance 10.5 12.5 13.0 13.3 Ω max B Max E-Input Resistance 10.5 6.7 6.5 6.3 Ω min B IE = ±1mA ±4.2 ±3.7 ±3.6 ±3.6 V min A VE = 0 ±15 ±10 ±9 ±9 mA min A IC = ±1mA ±4.7 ±4.0 ±3.9 ±3.9 V min A VC = 0 ±15 ±10 ±9 ±9 mA min A kΩ || pF typ C OTA OUTPUT E-Output Voltage Compliance E-Output Current, Sinking/Sourcing C-Output Voltage Compliance C-Output Current, Sinking/Sourcing C-Output Impedance (1) (2) (3) (4) 54 || 2 Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. Junction temperature = ambient for +25°C specifications. Junction temperature = ambient at low temperature limit; junction temperature = ambient + 7°C at high temperature limit for over temperature specifications. Current is considered positive out of node. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 3 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com ELECTRICAL CHARACTERISTICS: VS = ±5V (continued) RL = 500Ω and RADJ = 250Ω, unless otherwise noted. OPA861ID, IDBV TYP MIN/MAX OVER TEMPERATURE +25°C (2) 0°C to 70°C (3) –40°C to +85°C (3) Maximum Operating Voltage ±6.3 ±6.3 ±6.3 Minimum Operating Voltage ±2.0 ±2.0 PARAMETER CONDITIONS +25°C MIN/ MAX TEST LEVEL (1) V typ C V max A ±2.0 V min B UNITS POWER SUPPLY Specified Operating Voltage ±5 Maximum Quiescent Current RADJ = 250Ω 5.4 5.9 7.0 7.4 mA max A Minimum Quiescent Current RADJ = 250Ω 5.4 4.9 4.3 3.4 mA min A ΔIC/ΔVS ±20 ±50 ±60 ±65 µA/V max A –40 to +85 °C typ C OTA Power-Supply Rejection Ratio (+PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance θ JA D SO-8 Junction-to-Ambient +125 °C/W typ C DBV SOT23-6 Junction-to-Ambient +150 °C/W typ C 4 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 ELECTRICAL CHARACTERISTICS: VS = +5V RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted. OPA861ID, IDBV TYP MIN/MAX OVER TEMPERATURE CONDITIONS +25°C +25°C (2) 0°C to 70°C (3) –40°C to +85°C (3) UNITS MIN/ MAX TEST LEVEL (1) Bandwidth G = +5, VO = 200mVPP, RL = 500Ω 73 72 72 70 MHz min B G = +5, VO = 1VPP 73 MHz typ C Slew Rate G = +5, VO = 2.5V Step 410 395 390 390 V/µs min B VO = 1V Step 4.4 ns typ C PARAMETER OTA—Open-Loop (see Figure 33) AC PERFORMANCE Rise Time and Fall Time Harmonic Distortion G = +5, VO = 2VPP, 5MHz 2nd-Harmonic RL = 500Ω –67 –55 –54 –54 dB max B 3rd-Harmonic RL = 500Ω –57 –50 –49 –48 dB max B Base Input Voltage Noise f > 100kHz 2.4 3.0 3.3 3.4 nV/√Hz max B Base Input Current Noise f > 100kHz 1.7 2.4 2.45 2.5 pA/√Hz max B Emitter Input Current Noise f > 100kHz 5.2 15.3 16.6 17.5 pA/√Hz max B Minimum OTA Transconductance (gm) VO = ±10mV, RC = 50Ω, RE = 0Ω 85 70 67 65 mA/V min A Maximum OTA Transconductance (gm) VO = ±10mV, RC = 50Ω, RE = 0Ω 85 140 145 150 mA/V max A VB = 0V, RC = 0Ω, RE = 100Ω ±3 ±12 ±15 ±20 mV max A ±67 ±120 μV/°C max B ±6 ±6.6 μA max A ±20 ±25 nA/°C max B ±125 ±140 μA max A ±500 ±600 nA/°C max B μA typ C B OTA DC PERFORMANCE (4) (see Figure 33) B-Input Offset Voltage Average B-Input Offset Voltage Drift B-Input Bias Current Average B-Input Bias Current Drift E-Input Bias Current Average E-Input Bias Current Drift C-Output Bias Current VB = 0V, RC = 0Ω, RE = 100Ω VB = 0V, RC = 0Ω, RE = 100Ω ±1 ±5 VB = 0V, RC = 0Ω, RE = 100Ω VB = 0V, VC = 0V ±30 ±100 VB = 0V, VC = 0V VB = 0V, VC = 0V ±15 OTA INPUT (see Figure 33) Most Positive B-Input Voltage 4.2 3.7 3.6 3.6 V min Least Positive B-Input Voltage 0.8 1.3 1.4 1.4 V max B kΩ || pF typ C B-Input Impedance 455 || 2.1 Min E-Input Resistance 11.8 14.4 14.9 15.4 Ω max B Max E-Input Resistance 11.8 7.1 6.9 6.7 Ω min B OTA OUTPUT Maximum E-Output Voltage Compliance IE = ±1mA 4.2 3.7 3.6 3.6 V min A Minimum E-Output Voltage Compliance IE = ±1mA 0.8 1.3 1.4 1.4 V max A VE = 0 ±8 ±7 ±6.5 ±6.5 mA min A Maximum C-Output Voltage Compliance IC = ±1mA 4.7 4.0 3.9 3.9 V min A Minimum C-Output Voltage Compliance IC = ±1mA 0.3 1.0 1.1 1.1 V max A VC = 0 ±8 ±7 ±6.5 ±6.5 mA min A kΩ || pF typ C E-Output Current, Sinking/Sourcing C-Output Current, Sinking/Sourcing C-Output Impedance (1) (2) (3) (4) 54 || 2 Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. Junction temperature = ambient for +25°C specifications. Junction temperature = ambient at low temperature limit; junction temperature = ambient + 3°C at high temperature limit for over temperature specifications. Current is considered positive out of node. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 5 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com ELECTRICAL CHARACTERISTICS: VS = +5V (continued) RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted. OPA861ID, IDBV TYP MIN/MAX OVER TEMPERATURE +25°C (2) 0°C to 70°C (3) –40°C to +85°C (3) Maximum Operating Voltage 12.6 12.6 12.6 Minimum Operating Voltage 4 4 PARAMETER CONDITIONS +25°C MIN/ MAX TEST LEVEL (1) V typ C V max A 4 V min B UNITS POWER SUPPLY Specified Operating Voltage 5 Maximum Quiescent Current RADJ = 250Ω 4.7 5.2 6.0 6.4 mA max A Minimum Quiescent Current RADJ = 250Ω 4.7 4.2 3.4 3.0 mA min A ΔIC/ΔVS ±20 ±50 ±60 ±65 µA/V max A –40 to +85 °C typ C OTA Power-Supply Rejection Ratio (+PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance θ JA D SO-8 Junction-to-Ambient +125 °C/W typ C DBV SOT23-6 Junction-to-Ambient +150 °C/W typ C 6 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 TYPICAL CHARACTERISTICS: VS = ±5V At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted. OTA TRANSCONDUCTANCE vs FREQUENCY OTA TRANSCONDUCTANCE vs QUIESCENT CURRENT 150 1000 IO UT VIN = 100mVPP RL = 50Ω VIN = 10mVPP 50Ω Transconductance (mA/V) Transconductance (mA/V) VI N 50Ω I Q = 5.4mA (102mA/V) IQ = 6.5mA (117mA/V) 100 IQ = 1.9mA (51mA/V) 120 2 gm = -0.8265.IQ + 24.197.IQ - 1.466 90 IOUT 60 VIN 50W 30 50W IQ = 3.4mA (79mA/V) 10 0 1M 10M 100M 1G 0 1 2 Figure 2. 140 6 IQ = 6.5mA 100 IQ = 3.4mA 80 60 IQ = 1.9mA 40 Small signal around input voltage. −40 −30 −20 −10 0 2 IQ = 3.4mA 0 IQ = 1.9mA −2 VIN −4 50Ω 50Ω 20 30 −70 −60 −50 −40 −30 −20 −10 40 10 20 30 40 50 60 70 Figure 5. OTA LARGE-SIGNAL PULSE RESPONSE 3 0.6 0.2 0 G = +5V/V RL = 500Ω VIN = 0.25VPP f IN = 20MHz See Figure 48 Output Voltage (V) 2 0.4 −0.8 0 OTA Input Voltage (mV) OTA SMALL-SIGNAL PULSE RESPONSE Output Voltage (V) IOUT −8 10 0.8 −0.6 8 IQ = 5.4mA Figure 4. −0.4 7 4 Input Voltage (mV) −0.2 6 IQ = 6.5mA −6 20 0 5 OTA TRANSFER CHARACTERISTICS 8 OTA Output Current (mA) Transconductance (mA/V) OTA TRANSCONDUCTANCE vs INPUT VOLTAGE IQ = 5.4mA 4 Figure 3. 160 120 3 Quiescent Current (mA) Frequency (Hz) 1 0 −1 −2 G = +5V/V RL = 500Ω VIN = 1VPP fIN = 20MHz See Figure 48 −3 Time (10ns/div) Time (10ns/div) Figure 6. Figure 7. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 7 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com TYPICAL CHARACTERISTICS: VS = ±5V (continued) At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted. B-INPUT RESISTANCE vs QUIESCENT CURRENT C-OUTPUT RESISTANCE vs QUIESCENT CURRENT 120 OTA C- Output Resistance (kW ) OTA B- Input Resistance (kW ) 500 490 480 470 460 450 440 430 110 100 90 80 70 60 50 40 0 1 2 3 4 5 6 7 0 8 1 2 Figure 8. E-OUTPUT RESISTANCE vs QUIESCENT CURRENT 6 7 8 INPUT VOLTAGE AND CURRENT NOISE DENSITY Input Voltage Noise Density (nV/√Hz) Input Current Noise Density (pA/√Hz) OTA E- Output Resistance (W ) 5 100 50 40 30 20 10 0 E−Input Current Noise (5.2pA/√Hz) 10 B−Input Voltage Noise (2.4nV/√Hz) B−Input Current Noise (1.65pA/√Hz) 1 0 1 2 3 4 5 6 7 100 8 Quiescent Current (mA) 1k 10k 100k 1M 10M Frequency (Hz) Figure 10. Figure 11. QUIESCENT CURRENT vs RADJ 1MHz OTA VOLTAGE AND CURRENT NOISE DENSITY vs QUIESCENT CURRENT ADJUST RESISTOR 16 Input Voltage Noise Density (nV/√Hz) Input Current Noise Density (pA/√Hz) 8 7 Quiescent Current (mA) 4 Figure 9. 60 6 5 4 3 2 1 E−Input Current Noise (pA/√Hz) 14 12 10 8 B−Input Voltage Noise (nV/√Hz) 6 B−Input Current Noise (pA/√Hz) 4 2 0 0 0.1 1 10 100 1k 10k 100k 0 200 400 600 800 1000 1200 1400 1600 1800 2000 Quiescent Current Adjust Resistor (Ω ) Quiescent Current Adjust Resistor (Ω) Figure 12. 8 3 Quiescent Current (mA) Quiescent Current (mA) Figure 13. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 TYPICAL CHARACTERISTICS: VS = ±5V (continued) At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted. QUIESCENT CURRENT vs TEMPERATURE 3 9 4 2 8 2 1 B−Input Offset Voltage 0 0 −2 −1 −4 B−Input Bias Current −6 −40 −20 0 20 40 60 80 100 Quiescent Current (mA) 6 Input Bias Current (µA) Offset Voltage (mV) B-INPUT OFFSET VOLTAGE AND BIAS CURRENT vs TEMPERATURE 7 6 5 −2 4 −3 3 −40 120 −20 0 Ambient Temperature (_ C) 20 Figure 14. Five Representative Units IQ 300 20 100 120 IADJ 4 3 2 = -5E-18 x RADJ + 1E-12 x RADJ - 7E-08 x RADJ + 0.0046 x RADJ + 37.8 250 IQ/IADJ Ratio OTA C−Output Bias Current (µA) 80 IQ/IADS Ratio vs RADJ 350 30 10 0 −10 200 150 −20 100 −30 50 −40 −40 60 Figure 15. C-OUTPUT BIAS CURRENT vs TEMPERATURE 40 40 Ambient Temperature (_ C) −20 0 20 40 60 80 100 IQ = Quiescent Current. IADJ = Current flowing out of IQ adjust pin. 0 0.01 120 0.1 1 10 100 1k 10k 100k Quiescent Current Adjust Resistor (W ) Ambient Temperature (_ C) Figure 16. Figure 17. QUIESCENT CURRENT vs ADJUST PIN BIAS CURRENT IQ Adjust Pin Bias Current (mA) 250 200 150 100 50 0 0.01 0.1 1 10 100 1k 10k 100k Quiescent Current Adjust Resistor (W) Figure 18. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 9 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com TYPICAL CHARACTERISTICS: VS = +5V At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted. OTA TRANSCONDUCTANCE vs FREQUENCY 100 OTA TRANSCONDUCTANCE vs IQ 150 OTA Transconductance (mA/V) IQ = 5.8mA (93mA/V) Transconductance (mA/V) IQ = 4.7mA (80mA/V) IQ = 3.1mA (60mA/V) IQ = 1.65mA (37mA/V) I OU T VIN 50Ω 50Ω IOUT VIN 120 50Ω 50Ω 90 60 30 RL = 50Ω VIN = 10mVPP VIN = 100mVPP 10 0 1 10 100 1k 0 1 Frequency (Hz) OTA TRANSCONDUCTANCE vs INPUT VOLTAGE 5 6 7 OTA TRANSFER CHARACTERISTICS 6 IQ = 5.8mA IQ = 5.8mA IQ = 4.7mA 100 80 OTA Output Current (mA) Transconductance (mA/V) 4 Figure 20. 120 IQ = 3.1mA 60 IQ = 1.65mA 40 20 Small−signal around input voltage. 0 −30 −20 −10 0 10 20 4 IQ = 3.1mA 2 IQ = 4.7mA I OUT −2 V IN −4 −6 −50 −40 −30 −20 −10 30 1.5 0.10 1.0 −0.15 −0.20 Output Voltage (V) 0.15 −0.10 20 30 40 50 OTA LARGE-SIGNAL PULSE RESPONSE 2.0 G = +5V/V R L = 500Ω VIN = 0.07VPP f IN = 20MHz 10 Figure 22. OTA SMALL-SIGNAL PULSE RESPONSE −0.05 0 OTA Input Voltage (mV) 0.20 0 50 Ω 50 Ω Figure 21. 0.05 IQ = 1.65mA 0 Input Voltage (mV) Output Voltage (V) 3 Quiescent Current (mA) Figure 19. 0.5 0 −0.5 −1.0 −1.5 −2.0 Time (10ns/div) G = +5V/V R L = 500Ω VIN = 0.7VPP fIN = 20MHz Time (10ns/div) Figure 23. 10 2 Figure 24. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 TYPICAL CHARACTERISTICS: VS = +5V (continued) At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted. C-OUTPUT RESISTANCE vs QUIESCENT CURRENT 120 490 110 OTA C−Output Resistance (kΩ ) OTA B−Input Resistance (kΩ ) B-INPUT RESISTANCE vs QUIESCENT CURRENT 500 480 470 460 450 440 430 100 90 80 70 60 50 40 420 0 1 2 3 4 5 6 7 0 1 2 3 Figure 25. 5 6 7 Figure 26. E-OUTPUT RESISTANCE vs QUIESCENT CURRENT QUIESCENT CURRENT vs RADJ 60 7 50 6 Quiescent Current (mA) OTA E−Output Resistance (Ω ) 4 Quiescent Current (mA) Quiescent Current (mA) 40 30 20 10 5 4 3 2 1 0 0 0 1 2 3 4 5 6 7 0.1 Quiescent Current (mA) 1 10 100 1k 10k 100k Quiescent Current Adjust Resistor (Ω) Figure 27. Figure 28. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 11 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com APPLICATION INFORMATION The OPA861 is a versatile monolithic transconductance amplifier designed for widebandwidth systems, including high-performance video, RF, and IF circuitry. The operation of the OPA861 is discussed in the OTA (Operational Transconductance Amplifier) section of this data sheet. Over the years and depending on the writer, the OTA section of an op amp has been referred to as a Diamond Transistor, Voltage-Controlled Current source, Transconductor, Macro Transistor, or positive second-generation current conveyor (CCII+). Corresponding symbols for these terms are shown in Figure 29. C 3 VIN1 B 1 IOUT 2 VIN2 E Diamond Transistor Transconductor (used here) Voltage−Controlled Current Source C VIN1 Z CCII+ VIN2 B IOUT E Current Conveyor II+ TRANSCONDUCTANCE (OTA) SECTION—AN OVERVIEW The symbol for the OTA section is similar to a transistor (see Figure 29). Applications circuits for the OTA look and operate much like transistor circuits—the transistor is also a voltage-controlled current source. Not only does this characteristic simplify the understanding of application circuits, it aids the circuit optimization process as well. Many of the same intuitive techniques used with transistor designs apply to OTA circuits. The three terminals of the OTA are labeled B, E, and C. This labeling calls attention to its similarity to a transistor, yet draws distinction for clarity. While the OTA is similar to a transistor, one essential difference is the sense of the C-output current: it flows out the C terminal for positive B-to-E input voltage and in the C terminal for negative B-to-E input voltage. The OTA offers many advantages over a discrete transistor. The OTA is self-biased, simplifying the design process and reducing component count. In addition, the OTA is far more linear than a transistor. Transconductance of the OTA is constant over a wide range of collector currents—this feature implies a fundamental improvement of linearity. BASIC CONNECTIONS Macro Transistor Figure 29. Symbols and Terms Regardless of its depiction, the OTA section has a high-input impedance (B-input), a low-input/output impedance (E-input), and a high-impedance current source output (C-output). Figure 30 shows basic connections required for operation. These connections are not shown in subsequent circuit diagrams. Power-supply bypass capacitors should be located as close as possible to the device pins. Solid tantalum capacitors are generally best. RQ = 250W, roughly sets IQ = 5.4mA. RC 1 8 RE RS (25W to 200W) RADJ 250W 3 6 4 5 (1) 0.1mF 2.2mF Solid Tantalum -VS (1) 7 + VIN -5V 2 +5V +VS 0.1mF + 2.2mF Solid Tantalum NOTE: (1) VS = ±6.5V absolute maximum. Figure 30. Basic Connections 12 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 QUIESCENT CURRENT CONTROL PIN The quiescent current of the transconductance portion of the OPA861 is set with a resistor, RADJ, connected from pin 1 to –VS. The maximum quiescent current is 6mA. RADJ should be set between 50Ω and 1kΩ for optimal performance of the OTA section. This range corresponds to the 5mA quiescent current for RADJ = 50Ω, and 1mA for RADJ = 1kΩ. If the IQ adjust pin is connected to the negative supply, the quiescent current will be set by the 250Ω internal resistor. Reducing or increasing the quiescent current for the OTA section controls the bandwidth and AC behavior as well as the transconductance. With RADJ = 250Ω, this sets approximately 5.4mA total quiescent current at 25°C. It may be appropriate in some applications to trim this resistor to achieve the desired quiescent current or AC performance. Applications circuits generally do not show the resistor RQ, but it is required for proper operation. With a fixed RADJ resistor, quiescent current increases with temperature (see Figure 12 in the Typical Characteristics section). This variation of current with temperature holds the transconductance, gm, of the OTA relatively constant with temperature (another advantage over a transistor). It is also possible to vary the quiescent current with a control signal. The control loop in Figure 31 shows 1/2 of a REF200 current source used to develop 100mV on R1. The loop forces 125mV to appear on R2. Total quiescent current of the OPA861 is approximately 37 × I1, where I1 is the current made to flow out of pin 1. OPA861 R1 1.25kΩ BASIC APPLICATIONS CIRCUITS Most applications circuits for the OTA section consist of a few basic types, which are best understood by analogy to a transistor. Used in voltage-mode, the OTA section can operate in three basic operating states—common emitter, common base, and common collector. In the current-mode, the OTA can be useful for analog computation such as current amplifier, current differentiator, current integrator, and current summer. Common-E Amplifier or Forward Amplifier Figure 32 compares the common-emitter configuration for a BJT with the common-E amplifier for the OTA section. There are several advantages in using the OTA section in place of a BJT in this configuration. Notably, the OTA does not require any biasing, and the transconductance gain remains constant over temperature. The output offset voltage is close to 0, compared with several volts for the common-emitter amplifier. The gain is set in a similar manner as for the BJT equivalent with Equation 1: R G+ 1 L gm ) R E V+ 1/2 REF200 100µA With this control loop, quiescent current will be nearly constant with temperature. Since this method differs from the temperature-dependent behavior of the internal current source, other temperature-dependent behavior may differ from that shown in the Typical Characteristics. The circuit of Figure 31 will control the IQ of the OPA861 somewhat more accurately than with a fixed external resistor, RQ. Otherwise, there is no fundamental advantage to using this more complex biasing circuitry. It does, however, demonstrate the possibility of signal-controlled quiescent current. This capability may suggest other possibilities such as AGC, dynamic control of AC behavior, or VCO. IQ Adjust (1) 1 I1 R2 425Ω TLV2262 Figure 31. Optional Control Loop for Setting Quiescent Current Just as transistor circuits often use emitter degeneration, OTA circuits may also use degeneration. This option can be used to reduce the effects that offset voltage and offset current might otherwise have on the DC operating point of the OTA. The E-degeneration resistor may be bypassed with a large capacitor to maintain high AC gain. Other circumstances may suggest a smaller value capacitor used to extend or optimize high-frequency performance. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 13 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com The forward amplifier shown in Figure 33 and Figure 34 corresponds to one of the basic circuits used to characterize the OPA861. Extended characterization of this topology appears in the Typical Characteristics section of this datasheet. V+ RL RS VO VO VI Inverting Gain VOS = Several Volts R1 160Ω RE RS VI 8 C 3 B OPA861 E 2 V− VI 8 C 3 B VO OPA861 Figure 33. Forward Amplifier Configuration and Test Circuit RL E 2 RE G = 5V/V IQ = 5.4mA RE 78Ω (a) Transistor Common−Emitter Amplifier Transconductance varies over temperature. 100Ω RC 500Ω RL1 Noninverting Gain VOS = 0V VO Network Analyzer 8 3 (b) OTA Common−E Amplifier Transconductance remains constant over temperature. R1 100W Figure 32. Common-Emitter vs Common-E Amplifier (2) A positive voltage at the B-input, pin 3, causes a positive current to flow out of the C-input, pin 8. This gives a noninverting gain where the circuit of Figure 32a is inverting. Figure 32b shows an amplifier connection of the OPA861, the equivalent of a common-emitter transistor amplifier. Input and output can be ground-referenced without any biasing. The amplifier is non-inverting because of the sense of the output current. 14 RL2 rE 2 VI The transconductance of the OTA with degeneration can be calculated by Equation 2: g m_deg + 1 1 gm ) R E RIN 50W OTA RL = RL1 + RL2 || RIN RE G = RL RE + rE At IQ = 5.4mA G= RL RE + 10.5W rE = rE = 1 gm 1 95mA/V = 10.5W at IQ = 5.4mA Figure 34. Forward Amplifier Design Equations Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 Common-C Amplifier Figure 35b shows the OPA861 connected as an Efollower—a voltage buffer. It is interesting to notice that the larger the RE resistor, the closer to unity gain the buffer will be. If the OPA861 is to be used as a buffer, use RE ≥ 500Ω for best results. For the OPA861 used as a buffer, the gain is given by Equation 3: 1 G+ [1 1 ) g 1R m (3) E This low impedance can be converted to a high impedance by inserting the buffer amplifier in series. Current-Mode Analog Computations As mentioned earlier, the OPA861 can be used advantageously for analog computation. Among the application possibilities are functionality as a current amplifier, current differentiator, current integrator, current summer, and weighted current summer. Table 1 lists these different uses with the associated transfer functions. These functions can easily be combined to form active filters. Some examples using these currentmode functions are shown later in this document. V+ G=1 VOS = 0.7V VI V+ VO RL RE VO Noninverting Gain VOS = Several Volts V− (a) Transistor Common−Collector Amplifier (Emitter Follower) G+ 100Ω VI OPA861 1)g R1 +1 1 VIN RE m ǒ R O + g1 ø R E m 8 C 3 B 1 RE Ǔ V(a) Transistor Common-Base Amplifier G=1 VOS = 0V E 2 RE RL G= RE + VO (b) OTA Common−C Amplifier (Buffer) 100W 8 C 3 B =- RL RE VO OPA861 E 2 Figure 35. Common-Collector vs Common-C Amplifier 1 gm Inverting Gain VOS = 0V RL RE A low value resistor in series with the B-input is recommended. This resistor helps isolate trace parasitic from the inputs, reduces any tendency to oscillate, and controls frequency response peaking. Typical resistor values are from 25Ω to 200Ω. VIN (b) OTA Common-B Amplifier Figure 36. Common-Base Transistor vs Common-B OTA Common-B Amplifier Figure 36 shows the Common-B amplifier. This configuration produces an inverting gain and a low impedance input. Equation 4 shows the gain for this configuration. RL R G+ [* L 1 RE RE ) g m (4) Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 15 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com Table 1. Current-Mode Analog Computation Using the OTA Section FUNCTIONAL ELEMENT TRANSFER FUNCTION IMPLEMENTATION WITH THE OTA SECTION IOUT IIN Current Amplifier I OUT + R1 R2 R1 I IN R2 IOUT Current Integrator IIN 1 I OUT + C C ŕ I dt R R IN IOUT Current Summer n I OUT + 1 S I j j+1 I1 I2 In I OUT Weighted Current Summer n I OUT + 1 S I j j+1 Rj R R R1 I1 R Rn In OPA861 APPLICATIONS Control-Loop Amplifier DC-Restore Circuit A new type of control loop amplifier for fast and precise control circuits can be designed with the OPA861. The circuit of Figure 37 illustrates a series connection of two voltage control current sources that have an integral (and at higher frequencies, a proportional) behavior versus frequency. The control loop amplifiers show an integrator behavior from DC to the frequency represented by the RC time constant of the network from the C-output to GND. Above this frequency, they operate as an amp with constant gain. The series connection increases the overall gain to about 110dB and thus minimizes the control loop deviation. The differential configuration at the inputs enables one to apply the measured output signal and the reference voltage to two identical high-impedance inputs. The output buffer decouples the C-output of the second OTA in order to insure the AC performance and to drive subsequent output stages. The OPA861 can be used advantageously with an operational amplifier, here the OPA656, as a DCrestore circuit. Figure 38 illustrates this design. Depending on the collector current of the transconductance amplifier (OTA) of the OPA861, a switching function is realized with the diodes D1 and D2. 16 When the C-output is sourcing current, the capacitor C1 is being charged. When the C-output is sinking current, D1 is turned off and D2 is turned on, letting the voltage across C1 be discharged through R2. The condition to charge C1 is set by the voltage difference between VREF and VOUT. For the OTA Coutput to source current, VREF has to be greater than VOUT. The rate of charge of C1 is set by both R1 and C1. The discharge rate is given by R2 and C1. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 6 5 8 BUF602 VOUT 3 8 180Ω 2 10pF VREF 10pF 3 2 10Ω 180Ω 33Ω 10Ω 33Ω 6 VIN Figure 37. Control-Loop Amplifier Using Three OPA861s C1 100pF 20Ω JFET−Input, Wideband VIN D 1, D2 = 1N4148 RQ = 1kΩ OPA656 R2 100kΩ D1 VOUT 20Ω D2 CCII 8 C The OTA amplifier works as a current conveyor (CCII) in this circuit, with a current gain of 1. R1 and C1 set the DC restoration time constant. D1 adds a propagation delay to the DC restoration. R2 and C1 set the decay time constant. E 2 R1 40.2Ω B 3 R2 100Ω VREF Figure 38. DC Restorer Circuit Negative Impedance Converter Filter: Low-Pass Filter The OPA861 can be used as a negative impedance converter to realize the low-pass filer shown in Figure 39. The transfer function is shown in Equation 5: VOUT 1 = VIN 1 + sR(C1 + C2) + s2C1C2R2 (5) with: w0 + R C1 R VIN 1 ǸC1C 2 R C 1C 2 VOUT Q= C2 C1 + C2 Figure 39. Low-Pass Negative Impedance Converter Filter Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 17 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com Differential Line Driver/Receiver The input impedance is shown in Equation 6: Z IN + 1 ) R 1 ) sRC 2sC 1 ) 2sRC (6) Figure 40 shows the frequency responses for lowpass, Butterworth filters set at 20kHz and 10MHz. For the 20kHz filter, set R to 1kΩ and C 1 + 1 C 2 + 5.6mF 2 . For the 10MHz filter, the parasitic capacitance at the output pin needs to be taken into consideration. In the example of Figure 40, the parasitic is 3pF, which gives us the settings of R = 1.13kΩ, C1 = 10pF, and C2 = 17pF. The wide bandwidth and high slew rate of the OPA861 current-mode amplifier make it an ideal line driver. The circuit in Figure 42 makes use of two OPA861s to realize a single-ended to differential conversion. The high-impedance current source output of the OPA861 allows it to drive lowimpedance or capacitive loads without series resistances and avoids any attenuation that would have otherwise occured in the resistive network. The OPA861 used as a differential receiver exhibits excellent common-mode rejection ratio, as can be seen in Figure 41. Common−Mode Rejection Ratio (dBc) 0 −10 Gain (dB) −20 −30 −40 −50 −60 −70 −80 1k 10k 100k 1M 10M 100M 1G 0 −10 −20 −30 −40 −50 −60 −70 −80 −90 −100 0.001 0.01 Frequency (Hz) 0.1 1 10 100 Frequency (MHz) Figure 40. Small-Signal Frequency Response for a Low-Pass Negative Impedance Converter Filter Figure 41. Differential Driver Common-Mode Rejection Ratio for 2VPP Input Signals To 50Ω Load 50Ω VIN 50Ω 10Ω 50Ω 100Ω 50Ω 10Ω 50Ω Figure 42. Twisted-Pair Differential Driver and Receiver with the OPA861 18 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 ACTIVE FILTERS USING THE OPA861 IN CURRENT CONVEYOR STRUCTURE One further example of the versatility of the Diamond Transistor and Buffer is the construction of highfrequency (> 10MHz) active filters. Here, the Current Conveyor structure, shown in Figure 43, is used with the Diamond Transistor as a Current Conveyor. VOUT +1 C IOUT E CCII− B C VIN C R R R R of the operational amplifier becomes a negative second type of Current Conveyor (CCII), as shown in Figure 43. Both arrangements have identical transfer functions and the same level of sensitivity to deviations. The most recent implementation of active filters in a Current-Conveyor structure produced a second-order Bi-Quad filter. The value of the resistance in the emitter of the Diamond Transistor controls the filter characteristic. For more information, refer to application note SBOS047, New Ultra HighSpeed Circuit Techniques with Analog ICs. IIN C/2 C/2 Reciprocal Networks T(s) = VOUT VIN = IOUT IIN = + 4KQ2/R2C2 s2 + 2/RC[2Q(1 − K) + 1]s + 4KQ2/R2C2 VIN VOUT N I OUT N I IN NA I IN − Figure 43. Current Conveyor VOUT = I OUT VIN IIN Interreciprocal Networks The method of converting RC circuit loops with operational amplifiers in Current Conveyor structures is based upon the adjoint network concept. A network is reversible or reciprocal when the transfer function does not change even when the input and output have been exchanged. Most networks, of course, are nonreciprocal. The networks of Figure 44, perform interreciprocally when the input and output are exchanged, while the original network, N, is exchanged for a new network NA. In this case, the transfer function remains the same, and NA is the adjoing network. It is easy to construct an adjoint network for any given circuit, and these networks are the base for circuits in Current-Conveyor structure. Individual elements can be interchanged according to the list in Figure 45. Voltage sources at the input become short circuits, and the current flowing there becomes the output variable. In contrast, the voltage output becomes the input, which is excitated by a current source. The following equation describes the interreciprocal features of the circuit: VOUT/VIN = IOUT/IIN. Resistances and capacitances remain unchanged. In the final step, the operational amplifier with infinite input impedance and 0Ω output impedance is transformed into a current amplifier with 0Ω input impedance and infinite output impedance. A Diamond Transistor with the base at ground comes quite close to an ideal current amplifier. The wellknown Sallen-Key low-pass filter with positive feedback, is an example of conversion into CurrentConveyor structure, see Figure 46. The positive gain + VIN VOUT N I OUT − Figure 44. Networks Element 1 − VOUT + R 1 Passive Elements Controlled Sources C 1 1 1 Adjoint VIN 1 Signal Sources + V − 2 1 2 1 2 1 2 1 IOUT 2 IIN R C 2 2 3 3 µV 2 µI I 4 4 Figure 45. Individual Elements in the Current Conveyor R3 R2 VIN BUF602 C1 R1 RB1 C2 R1M RB2 R1S R2S VOUT R2M RB3 R3S Figure 46. Universal Active Filter Transfer Function Filter Characteristics The transfer function of the universal active filter of Figure 46 is shown in Equation 7. Five filter types can be made with this structure: • For a low-pass filter, set R2 = R3 = ∞, • For a high-pass filter, set R1 = R2 = ∞, • For a bandpass filter, set R1 = R3 = ∞, • For a band rejection filter, set R2 = ∞; R1 = R3, • For an all-pass filter, set R1 = R1S; R2 = R2S; and R3 = R3S. R R 1M R 1R s 2C1C 2R 1M R2M ) sC 1 R1M ) R1 V OUT 2 1 3 F(p) + + R R VIN s2C C R 2M ) sC 1M ) 1 1 2 3S 2S R 1S (7) Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 19 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com A few designs for a low-pass filter are shown in Figure 47 and Table 2. Gain (dB) Table 2. Component Values for Filters Shown In Figure 47 fO R RO CO 1MHz 150 100 2nF 20MHz 150 100 112.5pF 50MHz 150 100 25pF 3 0 -3 -6 -9 -12 -15 -18 -21 -24 -27 -30 -33 -36 -39 -42 -45 -48 50MHz Filter High-CMRR, Moderate Precision, Differential I/O ADC Driver The circuit shown in Figure 48 depicts an ADC driver implemented with two OPA861s. Since the gain is set here by the ratio of the internal 600Ω resistors and RE, its accuracy will only be as good as the input resistor of the ADS5272. The small-signal frequency response for this circuit has 150MHz at –3dB bandwidth for a gain of approximately 5.6dB, as shown in Figure 49. The advantage of this circuit lies in its high CMRR to 100kHz; see Figure 50. This circuit also has more than 10 bits of linearity. VIN1 20MHz Filter 100k 1M RE 600Ω 100M 1G VIN2 The advantages of building active filters using a Current Conveyor structure are: • The increase in output resistance of operational amplifiers at high frequencies makes it difficult to construct feedback filter structures (decrease in stop-band attenuation). • All filter coefficients are represented by resistances, making it possible to adjust the filter frequency response without affecting the filter coefficients. The capacitors which determine the frequency are located between the ground and the current source outputs and are thus grounded on one side. Therefore, all parasitic capacitances can be viewed as part of these capacitors, making them easier to comprehend. The features which determine the frequency characteristics are currents, which charge the integration capacitors. This situation is similar to the transfer characteristic of the Diamond Transistor. 600Ω Figure 48. High CMRR, Moderate Precision, Differential I/O ADC Driver 6 5.6dB 3 Gain (dB) Figure 47. Butterworth Low-Pass Filter with the Universal Active Filter 20 VCM OPA861 10M Frequency (Hz) • 600Ω For All Filters: R2 = R3 = ¥ R1 = R1S = R2S = 1/2 R3S = R R1M = R2M = R0 C1 = C2 = C0 10k • ADS5272 OPA861 1MHz Filter 0 −3 −6 −9 1M 10M 100M 1G Frequency (Hz) Figure 49. ADC Driver, Small-Signal Frequency Response Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 NOISE PERFORMANCE Common−Mode Rejection Ratio(dB) 75 Input−Referred 70 65 60 55 50 45 The OTA noise model consists of three elements: a voltage noise on the B-input; a current noise on the B-input; and a current noise on the E-input. Figure 51 shows the OTA noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. 40 35 en 30 VO 25 RL 20 1k 10k 100k 1M 10M 100M 1G RS Frequency (Hz) √4kTRS Figure 50. CMRR of the ADC Driver ibn RG ibi √4kTRS DESIGN-IN TOOLS Figure 51. OTA Noise Analysis Model DEMONSTRATION BOARDS A printed circuit board (PCB) is available to assist in the initial evaluation of circuit performance using the OPA861. This module is available free, as an unpopulated PCB delivered with descriptive documentation. The summary information for the board is shown below: The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 8 shows the general form for the output noise voltage using the terms shown in Figure 51. eO = PRODUCT PACKAGE BOARD PART NUMBER OPA861ID SO-8 DEM-OTA-SO-1A The board can be requested Instruments web site (www.ti.com). on LITERATURE REQUEST NUMBER SBOU035 the Texas MACROMODELS AND APPLICATIONS SUPPORT Computer simulation of circuit performance using SPICE is often useful when analyzing the performance of analog circuits and systems. This principle is particularly true for Video and RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. A SPICE model for the OPA861 is available through the Texas Instruments web page (www.ti.com). These models do a good job of predicting small-signal AC and transient performance under a wide variety of operating conditions. They do not do as well in predicting the harmonic distortion. These models do not attempt to distinguish between the package types in their small-signal AC performance. [eN2 + (RSiBN)2 + 4kTRS + (RGiBI)2 + 4kTRG] 2 RL RG + 1 gm (8) THERMAL ANALYSIS Maximum desired junction temperature will set the maximum allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed 150°C. Operating junction temperature (TJ) is given by TA + PD × θ JA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver output current. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for the OPA861 be at a maximum when the maximum IO is being driven into a voltage source that puts the maximum voltage across the output stage. Maximum IO is 15mA times a 9V maximum across the output. Note that it is the power in the output stage and not into the load that determines internal power dissipation. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 21 OPA861 SBOS338G – AUGUST 2005 – REVISED MAY 2013 www.ti.com As a worst-case example, compute the maximum TJ using an OPA861IDBV in the circuit of Figure 32b operating at the maximum specified ambient temperature of +85°C and driving a –1V voltage reference. PD = 10V × 5.4mA + (15mA × 9V) = 185mW Maximum TJ = +85°C + (0.19W × 150°C/W) = 114°C. Although this is still well below the specified maximum junction temperature, system reliability considerations may require lower tested junction temperatures. The highest possible internal dissipation will occur if the load requires current to be forced into the output for positive output voltages or sourced from the output for negative output voltages. This puts a high current through a large internal voltage drop in the output transistors. BOARD LAYOUT GUIDELINES Achieving optimum performance with a highfrequency amplifier like the OPA861 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the inverting input pin can cause instability: on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance (< 0.25") from the powersupply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and powerplane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power-supply connections should always be decoupled with these capacitors. An optional supply decoupling capacitor (0.1µF) across the two power supplies (for bipolar operation) will improve 2nd-harmonic distortion performance. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. 22 c) Careful selection and placement of external components will preserve the high-frequency performance of the OPA861. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal film or carbon composition, axially-leaded resistors can also provide good high-frequency performance. Again, keep their leads and PC board traces as short as possible. Never use wirewound type resistors in a high-frequency application. d) Connections to other wideband devices on the board may be made with short, direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. e) Socketing a high-speed part like the OPA861 is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network that makes it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA861 onto the board. INPUT AND ESD PROTECTION The OPA861 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power supplies as shown in Figure 52. +VCC External Pin Internal Circuitry −VCC Figure 52. Internal ESD Protection These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (for example, in systems with ±15V supply parts driving into the OPA861), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible since high values degrade both noise performance and frequency response. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 OPA861 www.ti.com SBOS338G – AUGUST 2005 – REVISED MAY 2013 REVISION HISTORY NOTE: Page numbers for previous revisions may differ from page numbers in the current version. Changes from Revision F (May 2011) to Revision G • Page Changed transfer function equations in Negative Impedance Converter Filter: Low-Pass Filter section .......................... 17 Changes from Revision E (August 2008) to Revision F Page • Updated Figure 30 .............................................................................................................................................................. 12 • Updated Equation 8 ............................................................................................................................................................ 21 Changes from Revision D (August 2006) to Revision E • Page Changed storage temperature range rating in Absolute Maximum Ratings table from –40°C to +125°C to –65°C to +125°C .................................................................................................................................................................................. 2 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: OPA861 23 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) OPA861ID ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA 861 OPA861IDBVT ACTIVE SOT-23 DBV 6 250 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 NSR (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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