SN10503PWP

DBV-5 D-8 DGN-8 DGK-8 D-14 SN10501 SN10502 SN10503 PWP-14 www.ti.com .................................................................................................................................................. SLOS408B – MARCH 2003 – REVISED JANUARY 2009 HIGH-SPEED RAIL-TO-RAIL OUTPUT VIDEO AMPLIFIERS FEATURES 1 • High Speed – 100 MHz Bandwidth (–3 dB, G = 2) – 900 V/s Slew Rate • Excellent Video Performance – 50 MHz Bandwidth (0.1 dB, G = 2) – 0.007% Differential Gain – 0.007 Differential Phase • Rail-to-Rail Output Swing – VO = –4.5 / 4.5 (RL= 150 Ω) • High Output Drive, IO = 100 mA (typ) • Ultralow Distortion – HD2 = –78 dBc (f = 5 MHz, RL = 150 Ω) – HD3 = –85 dBc (f = 5 MHz, RL = 150 Ω) • Wide Range of Power Supplies – VS = 3 V to 15 V VIDEO DRIVE CIRCUIT 2 • • • • • Video Line Driver Imaging DVD / CD ROM Active Filtering General Purpose Signal Chain Conditioning + 10 µF Video In 3 4 75 Ω 0.1 µF 5 SN10501 75 Ω 1 + VO − 2 75 Ω + VS− 1.43 kΩ 10 µF 0.1 µF 1.43 kΩ 6.3 VO = 0.1 VPP −0.1 dB at 49 MHz 6.2 6.1 Signal Gain − dB APPLICATIONS VS+ 6.0 5.9 5.8 VO = 2 VPP −0.1 dB at 51 MHz 5.7 5.6 5.5 5.4 Gain = 2 RL = 150 Ω to GND VS = ±5 V RF = 1.43 kΩ 5.3 100 k DESCRIPTION 1M 10 M 100 M 1G f − Frequency − Hz The SN1050x family is a set of rail-to-rail output single, dual, and triple low-voltage, high-output swing, low-distortion high-speed amplifiers ideal for driving data converters, video switching, or low distortion applications. This family of voltage-feedback amplifiers can operate from a single 15-V power supply down to a single 3-V power supply while consuming only 14 mA of quiescent current per channel. In addition, the family offers excellent ac performance with 100-MHz bandwidth, 900-V/µs slew rate and harmonic distortion (THD) at –78 dBc at 5 MHz. DEVICE DESCRIPTION SN10501 Single SN10502 Dual SN10503 Triple 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. owerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2003–2009, Texas Instruments Incorporated SN10501 SN10502 SN10503 SLOS408B – MARCH 2003 – REVISED JANUARY 2009 .................................................................................................................................................. www.ti.com This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ABSOLUTE MAXIMUM RATINGS operating free-air temperature range unless otherwise (1) UNIT Supply voltage, VS 16.5 V Input voltage, VI ±VS Output current, IO 150 mA Differential input voltage, VID 4V Continuous power dissipation See Dissipation Rating Table Maximum junction temperature, TJ 150°C Maximum junction temperature, continuous operation, longterm reliability, TJ (2) 125°C Storage temperature range, Tstg –65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds (1) (2) 300°C The absolute maximum ratings under any condition is limited by the constraints of the silicon process. Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not implied. The maximum junction temperature for continuous operation is limited by package constraints. Operation above this temperature may result in reduced reliability and/or lifetime of the device. PACKAGE DISSIPATION RATINGS (1) (2) (3) 2 POWER RATING (2) PACKAGE θJC(°C/W) (1) θJA(°C/W) TA ≤ 25°C TA = 85°C DBV (5) 55 255.4 391 mW 156 mW D (8) 38.3 97.5 1.02 W 410 mW D (14) 26.9 66.6 1.5 W 600 mW DGK (8) 54.2 260 385 mW 154 mW DGN (8) (3) 4.7 58.4 1.71 W 685 mW PWP (14) (3) 2.07 37.5 2.67 W 1.07 W This data was taken using the JEDEC standard High-K test PCB. Power rating is determined with a junction temperature of 125°C. This is the point where distortion starts to substantially increase. Thermal management of the final PCB should strive to keep the junction temperature at or below 125°C for best performance and long term reliability. The SN10501, SN10502, and SN10503 may incorporate a PowerPAD™ on the underside of the chip. This acts as a heatsink and must be connected to a thermally dissipating plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction temperature which could permanently damage the device. See TI Technical Brief SLMA002 for more information about utilizing the PowerPAD™ thermally enhanced package. Submit Documentation Feedback Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 SN10501 SN10502 SN10503 www.ti.com .................................................................................................................................................. SLOS408B – MARCH 2003 – REVISED JANUARY 2009 RECOMMENDED OPERATING CONDITIONS MIN Supply voltage,(VS+ and VS-) MAX UNIT Dual supply 1.35 8 Single supply 2.7 16 VS- + 1.1 VS+ - 1.1 Input common-mode voltage range V V PACKAGE ORDERING INFORMATION PACKAGED DEVICES PACKAGE TYPE TRANSPORT MEDIA, QUANTITY — SOT-23-5 Tape and Reel, 250 — — SOT-23-5 Tape and Reel, 3000 SN10502DGK — MSOP-8 Rails, 75 SN10502DGKR — MSOP-8 Tape and Reel, 2500 SN10501DGN SN10502DGN — MSOP-8-PP Rails, 75 SN10501DGNR SN10502DGNR — MSOP-8-PP Tape and Reel, 2500 SN10501D SN10502D SN10503D SOIC Rails, 75 SN10501DR SN10502DR SN10503DR SOIC Tape and Reel, 2500 — — SN10503PWP TSSOP-14-PP Rails, 75 — — SN10503PWPR TSSOP-14-PP Tape and Reel, 2000 SINGLE DUAL TRIPLE SN10501DBVT — SN10501DBVR SN10501DGK SN10501DGKR PIN ASSIGNMENTS PACKAGE DEVICES SN10501 DBV PACKAGE (TOP VIEW) VOUT VS− IN+ 1 5 SN10501 D, DGK, DGN PACKAGE (TOP VIEW) VS+ NC IN− IN+ VS− 2 3 4 IN − 1 8 2 7 3 6 4 5 NC VS+ VOUT NC SN10502 D, DGK, DGN PACKAGE (TOP VIEW) 1OUT 1IN− 1IN+ VS− 1 8 2 7 3 6 4 5 VS+ 2OUT 2IN− 2IN+ NC − No internal connection SN10503 D, PWP PACKAGE (TOP VIEW) NC NC NC VS+ 1IN+ 1IN− 1OUT 1 14 2 13 3 12 4 11 5 10 6 9 7 8 2OUT 2IN− 2IN+ VS− 3IN+ 3IN− 3OUT NC − No internal connection Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 Submit Documentation Feedback 3 SN10501 SN10502 SN10503 SLOS408B – MARCH 2003 – REVISED JANUARY 2009 .................................................................................................................................................. www.ti.com ELECTRICAL CHARACTERISTICS VS = 5 V, RL = 150 Ω, and G = 2 unless otherwise noted TYP PARAMETER TEST CONDITIONS 25°C OVER TEMPERATURE 0°C to 70°C 25°C –40°C to 85°C UNITS MIN/MAX AC PERFORMANCE G = 1, VO = 100 mVPP 170 MHz Typ G = 2, VO = 100 mVPP, Rf = 1 kΩ 100 MHz Typ G = 10, VO = 100 mVPP, Rf = 1 kΩ 12 MHz Typ 0.1 dB flat bandwidth G = 2, VO = 100 mVPP, Rf = 1.43 kΩ 50 MHz Typ Gain bandwidth product G > 10, f = 1 MHz, Rf = 1 kΩ 120 MHz Typ Full-power bandwidth( (1)) G = 2, VO = ±2.5 VPP 57 MHz Typ Slew rate G = 2, VO = ±2.5 VPP 900 V/µs Min 25 ns Typ 52 ns Typ –78 dBc Typ –85 dBc Typ 0.007 % Typ 0.007 ° Typ 13 nV/√Hz Typ 0.8 pA/√Hz Typ f = 5 MHz Ch-to-Ch –90 dB Typ VO = ±2 V 100 80 75 75 dB Min 12 25 30 30 mV Max Small signal bandwidth Settling time to 0.1% Settling time to 0.01% G = -2, VO = ±2 VPP Harmonic distortion Second harmonic distortion Third harmonic distortion Differential gain (NTSC, PAL) Differential phase (NTSC, PAL) Input voltage noise Input current noise Crosstalk (dual and triple only) G = 2, VO = 2 VPP, f = 5 MHz, RL = 150 Ω G = 2, R = 150 Ω f = 1 MHz DC PERFORMANCE Open-loop voltage gain (AOL) Input offset voltage Input bias current 0.9 3 5 5 µA Max 100 500 700 700 nA Max –4 / 4 –3.9 / 3.9 V Min 94 70 65 65 dB Min 33 MΩ Typ 1 / 0.5 pF Max RL = 150 Ω –4.5 / 4.5 V Typ RL = 499 Ω –4.7 / 4.7 –4.5 / 4.5 –4.4 / 4.4 -4.4 / 4.4 V Min Min VCM = 0 V Input offset current INPUT CHARACTERISTICS Common-mode input range Common-mode rejection ratio VCM = 2 V Input resistance Input capacitance Common-mode / differential OUTPUT CHARACTERISTICS Output voltage swing Output current (sourcing) Output current (sinking) Output impedance RL = 10 Ω f = 1 MHz 100 92 88 88 mA -100 -92 -88 -88 mA Min Ω Typ 0.09 POWER SUPPLY Specified operating voltage Maximum quiescent current Per channel Power supply rejection (±PSRR) (1) 4 ±5 ±8 ±8 ±8 V Max 14 18 20 22 mA Max 75 62 60 60 dB Min Full-power bandwidth = SR / 2πVpp Submit Documentation Feedback Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 SN10501 SN10502 SN10503 www.ti.com .................................................................................................................................................. SLOS408B – MARCH 2003 – REVISED JANUARY 2009 ELECTRICAL CHARACTERISTICS VS = 5 V, RL = 150 Ω, and G = 2 unless otherwise noted TYP PARAMETER TEST CONDITIONS 25°C OVER TEMPERATURE 25°C 0°C to 70C -40°C to 85C UNITS MIN/MAX AC PERFORMANCE Small signal bandwidth G = 1, VO = 100 mVPP 170 MHz Typ G = 2, VO = 100 mVPP, Rf = 1.5 kΩ 100 MHz Typ G = 10, VO = 100 mVPP, Rf = 1.5 kΩ 12 MHz Typ 0.1 dB flat bandwidth G = 2, VO = 100 mVPP, Rf = 1.24 kΩ 50 MHz Typ Gain bandwidth product G > 10, f = 1 MHz, Rf = 1.5 kΩ 120 MHz Typ 60 MHz Typ 750 V/µs Min 27 ns Typ 48 ns Typ –82 dBc Typ –88 dBc Typ 0.014 % Typ 0.011 ° Typ 13 nV/√Hz Typ 0.8 pA/√Hz Typ f = 5 MHz Ch-to-Ch –90 dB Typ VO = 1.5 V to 3.5 V 100 80 75 75 dB Min 12 25 30 30 mV Max VCM = 2.5 V 0.9 3 5 5 µA Max 100 500 700 700 nA Max 1/4 1.1 / 3.9 V Min 96 70 65 65 dB Min 33 MΩ Typ 1 / 0.5 pF Max RL = 150 Ω 0.5 / 4.5 V Typ RL = 499 Ω 0.2 / 4.8 Full-power bandwidth( (1)) Slew rate Settling time to 0.1% Settling time to 0.01% G = 2, VO = 4 V step G = -2, VO = 2 V Harmonic distortion Second harmonic distortion Third harmonic distortion Differential gain (NTSC, PAL) Differential phase (NTSC, PAL) Input voltage noise Input current noise Crosstalk (dual and triple only) G = 2, VO = 2 VPP, f = 5 MHz, RL = 150 Ω G = 2, R = 150 Ω f = 1 MHz DC PERFORMANCE Open-loop voltage gain (AOL) Input offset voltage Input bias current Input offset current INPUT CHARACTERISTICS Common-mode input range Common-mode rejection ratio VCM = 1.5 V to 3.5 V Input resistance Input capacitance Common-mode / differential OUTPUT CHARACTERISTICS Output voltage swing Output current (sourcing) Output current (sinking) Output impedance RL = 10 Ω f = 1 MHz 0.4 / 4.6 V Min 95 0.3 / 4.7 0.4 / 4.6 85 80 80 mA Min –95 -85 –80 –80 mA Min Ω Typ 0.09 POWER SUPPLY Specified operating voltage Maximum quiescent current Per channel Power supply rejection (±PSRR) (1) 5 16 16 16 V Max 12 15 17 19 mA Max 70 62 60 60 dB Min Full-power bandwidth = SR / 2πVpp Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 Submit Documentation Feedback 5 SN10501 SN10502 SN10503 SLOS408B – MARCH 2003 – REVISED JANUARY 2009 .................................................................................................................................................. www.ti.com TYPICAL CHARACTERISTICS TABLE OF GRAPHS FIGURE Frequency response 1– 8 Small signal frequency response 9, 10 Large signal frequency response 11 Slew rate vs Output voltage step 12, 13 Harmonic distortion vs Frequency 14, 15 Voltage and current noise vs Frequency Differential gain vs Number of loads 17, 18 Differential phase vs Number of loads 19, 20 Quiescent current vs Supply voltage 21 Output voltage vs Load resistance 22 Open-loop gain and phase vs Frequency 23 Rejection ratio vs Frequency 24 Rejection ratio vs Case temperature Common-mode rejection ratio vs Input common-mode range 26, 27 Output impedance vs Frequency 28, 29 Crosstalk vs Frequency Input bias and offset current vs Case temperature FREQUENCY RESPONSE 8 VO = 0.1 VPP −0.1 dB at 49 MHz 6 3 2 −2 100 k 1M 5.9 5.8 VO = 2 VPP −0.1 dB at 51 MHz 5.7 5.5 5.4 100 M 1G 1M FREQUENCY RESPONSE VO = 2 VPP −0.1 dB at 14 MHz Signal Gain − dB 5 VO = 0.1 VPP −0.1 dB at 14 MHz 5.7 5.3 1M 10 M 1G 100 k 1M −2 100 M 1G f − Frequency − Hz Figure 4. Submit Documentation Feedback FREQUENCY RESPONSE 6.2 1M 10 M 6.0 5.9 5.8 5.7 5.5 5.4 5.3 100 M 1G f − Frequency − Hz VO = 2 VPP −0.1 dB at 58 MHz 6.1 5.6 Gain = 2 RL = 150 Ω to VS/2 VS = 5 V RF = 1.24 kΩ 100 k 1G 6.3 2 −1 100 M Figure 3. 3 0 10 M f − Frequency − Hz VO = 0.1 VPP −3 dB at 99 MHz 4 1 Gain = 2 RL = 150 Ω to GND VS = ±5 V RF = 301 Ω 100 k 100 M VO = 2 VPP −3 dB at 99 MHz 7 6.0 5.4 −1 FREQUENCY RESPONSE 6 5.5 10 M 8 6.1 5.6 Gain = 2 RL = 150 Ω to GND VS = ±5 V RF = 301 Ω Figure 2. 6.3 5.8 2 f − Frequency − Hz Figure 1. 5.9 VO = 0.1 VPP −3 dB at 99 MHz 3 −2 100 k f − Frequency − Hz 6.2 4 0 5.3 10 M 5 1 Gain = 2 RL = 150 Ω to GND VS = ±5 V RF = 1.43 kΩ Signal Gain − dB −1 6.0 5.6 Gain = 2 RL = 150 Ω to GND VS = ±5 V RF = 1.43 kΩ 0 Signal Gain − dB VO = 0.1 VPP −3 dB at 99 MHz 4 VO = 2 VPP −3 dB at 99 MHz 7 6.1 5 1 Signal Gain − dB FREQUENCY RESPONSE 8 6.2 Signal Gain − dB Signal Gain − dB 30 31, 32 FREQUENCY RESPONSE 6 6 25 6.3 VO = 2 VPP −3 dB at 99 MHz 7 16 VO = 0.1 VPP −0.1 dB at 48 MHz Gain = 2 RL = 150 Ω to VS/2 VS = 5 V RF = 1.24 kΩ 100 k 1M 10 M 100 M 1G f − Frequency − Hz Figure 5. Figure 6. Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 SN10501 SN10502 SN10503 www.ti.com .................................................................................................................................................. SLOS408B – MARCH 2003 – REVISED JANUARY 2009 FREQUENCY RESPONSE FREQUENCY RESPONSE VO = 2 VPP −3 dB at 89 MHz 7 6.2 6 5 VO = 0.1 VPP −3 dB at 84 MHz 4 3 2 1 Gain = 2 RL = 150 Ω to VS/2 VS = 5 V RF = 301Ω 0 −1 −2 100 k 1M 5.9 5.7 5.6 5.4 5.3 100 M VO = 2 VPP −0.1 dB at 16 MHz 5.8 1G f − Frequency − Hz 1M FREQUENCY RESPONSE 100 M 1 −2 1G RL = 499 Ω RF = 1.5 kΩ VO = 100 mVPP VS = 5 V 4 3 2 2 0 Gain = 1 −1 −2 100 k 1 1200 1000 VS = ±5 V Gain = 2 RL = 150 Ω RF = 1 kΩ VO = 2 VPP VS = ±5 V Fall 600 400 200 0 1M 10 M 100 M 1G 0 1 2 3 4 5 6 7 Figure 10. Figure 11. Figure 12. SLEW RATE vs OUTPUT VOLTAGE STEP HARMONIC DISTORTION vs FREQUENCY HARMONIC DISTORTION vs FREQUENCY −10 Rise 500 400 300 200 100 0 0.5 1 1.5 2 2.5 3 3.5 4 VO − Output Voltage Step − V Figure 13. −10 Gain = 2 RL = 150 Ω VO = 2 VPP VS = ±5 V −20 −30 Harmonic Distortion − dBc Harmonic Distortion − dBc Fall 8 0 0 0 800 VO − Output Voltage Step − V Gain = 2 RL = 150 Ω RF = 1 kΩ VS = 5 V 600 Rise f − Frequency − Hz 800 700 Gain = 2 RL = 150 Ω RF = 1 kΩ VS = ±5 V VS = 5 V 100 k 1G 1G FREQUENCY RESPONSE 0 1M 10 M 100 M f − Frequency − Hz 100 M SLEW RATE vs OUTPUT VOLTAGE STEP 4 3 10 M Figure 9. 5 1 1M Figure 8. SR − Slew Rate − V/ µ s Signal Gain − dB 5 100 k f − Frequency − Hz 7 Gain = 2 Gain = 1 f − Frequency − Hz 6 Signal Gain − dB 2 0 6 SR − Slew Rate − V/ µ s 3 8 7 RL = 150 Ω RF = 1 kΩ VO = 100 mVPP VS = ±5 V 4 −1 10 M Figure 7. 8 5 Gain = 2 RL = 150 Ω to VS/2 VS = 5 V RF = 301 Ω 100 k Gain = 2 6 6.0 5.5 10 M 7 VO = 0.1 VPP −0.1 dB at 16 MHz 6.1 Signal Gain − dB Signal Gain − dB FREQUENCY RESPONSE 8 6.3 Signal Gain − dB 8 −40 −50 HD2 −60 −70 HD3 −80 −20 −30 −40 −50 −60 −70 −80 −90 −90 −100 −100 0.1 0.1 1 10 f − Frequency − MHz 100 Figure 14. Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 Gain = 2 RL = 150 Ω VO = 2 VPP VS = 5 V HD2 HD3 1 10 f − Frequency − MHz 100 Figure 15. Submit Documentation Feedback 7 SN10501 SN10502 SN10503 SLOS408B – MARCH 2003 – REVISED JANUARY 2009 .................................................................................................................................................. www.ti.com VOLTAGE AND CURRENT NOISE vs FREQUENCY DIFFERENTIAL GAIN vs NUMBER OF LOADS 0.20 Hz 1 In Differential Gain − % 10 0.16 I n − Current Noise − pA/ Hz Vn − Voltage Noise − nV/ Vn 0.4 Gain = 2 Rf = 1.5 kΩ 40 IRE − NTSC Worst Case ±100 IRE Ramp 0.18 0.14 0.12 0.10 VS = 5 V 0.08 0.06 VS = ±5 V 0.1 10 M 1 10 k 100 k 1M VS = 5 V 0.2 0.15 VS = ±5 V 0.1 0 0 0.20 3 4 0 5 0 4 5 DIFFERENTIAL PHASE vs NUMBER OF LOADS QUIESCENT CURRENT vs SUPPLY VOLTAGE 0.06 0.3 20 0.25 VS = 5 V 0.2 0.15 VS = ±5 V 0.1 VS = ±5 V 0.04 Gain = 2 Rf = 1.5 kΩ 40 IRE − PAL Worst Case ±100 IRE Ramp 0.35 VS = 5 V 0.08 22 0.4 0.10 0 1 2 3 4 16 12 10 TA = −40°C 8 6 4 2 0 0 Number of Loads − 150 Ω TA = 25°C 14 0 5 TA = 85°C 18 0.05 0.02 1 2 3 4 5 1.5 2 2.5 3 3.5 4 Figure 19. Figure 20. Figure 21. OUTPUT VOLTAGE vs LOAD RESISTANCE OPEN-LOOP GAIN AND PHASE vs FREQUENCY REJECTION RADIO vs FREQUENCY 100 VS = ±5 V, 5 V, and 3.3 V 90 2 1 TA = −40 to 85°C −1 −2 −3 −4 100 1k RL − Load Resistance − Ω 10 k Figure 22. Submit Documentation Feedback 90 180 80 160 70 140 60 120 50 100 40 80 30 60 20 40 10 20 0 −10 −5 100 200 100 5 VS = ±5 V, 5 V, and 3.3 V 80 Phase − ° Open-Loop Gain − dB 3 220 Rejection Ratios − dB 110 4 4.5 VS − Supply Voltage − ±V Number of Loads − 150 Ω 5 10 3 DIFFERENTIAL GAIN vs NUMBER OF LOADS 0.12 0 2 Number of Loads − 150 Ω Figure 18. 0.14 0 1 Figure 17. Differential Phase − ° 0.16 2 Figure 16. Gain = 2 Rf = 1.5 kΩ 40 IRE − PAL Worst Case ±100 IRE Ramp 0.18 1 Number of Loads − 150 Ω f − Frequency − Hz Differential Gain − % 0.25 0.05 Quiescent Current − mA/Ch 1k VO − Output Voltage − V 0.3 0.04 0.02 8 Gain = 2 Rf = 1.5 kΩ 40 IRE − NTSC Worst Case ±100 IRE Ramp 0.35 Differential Phase − ° 10 100 DIFFERENTIAL PHASE vs NUMBER OF LOADS 70 CMMR 60 50 PSRR 40 30 20 10 0 −20 1 k 10 k 100 k 1 M 10 M 100 M 1 G f − Frequency − Hz 0 0.1 Figure 23. 1 10 f − Frequency − MHz 100 Figure 24. Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 SN10501 SN10502 SN10503 www.ti.com .................................................................................................................................................. SLOS408B – MARCH 2003 – REVISED JANUARY 2009 Rejection Ratio − dB 80 PSRR 70 60 50 40 −40−30−20−10 0 10 20 30 40 50 60 70 80 90 90 80 70 60 50 40 30 20 0 −6 TC − Case Temperature − °C −2 0 2 4 80 70 60 50 40 30 20 0 6 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 VICR − Input Common-Mode Voltage Range − V Figure 26. Figure 27. OUTPUT IMPEDANCE vs FREQUENCY OUTPUT IMPEDANCE vs FREQUENCY CROSSTALK vs FREQUENCY 120 100 Gain = 2 RL = 150 Ω to GND VO = 2 VPP VS = ±5 V RF = 301 Ω 0.1 Gain = 2 RL = 150 Ω to VS/2 VO = 2 VPP VS = 5 V 10 Crosstalk all Channels 100 RF = 301 Ω 1 80 60 40 0.1 20 RF = 1.43 kΩ 0.01 100 k 1M 10 M RF = 1.24 kΩ 100 M 0.01 100 k 1G f − Frequency − Hz 1M 10 M 100 M 0 100 k 1G 1M Figure 29. 0.84 0.9 IIB+ 0.78 −5 IIB− 0.76 −10 −15 0.74 0.72 −20 −25 0.7 −40−30−20−10 0 10 20 30 40 50 60 70 80 90 5 VS = ±5 V 0.88 I IB − Input Bias Current − µ A 0 I OS − Input Offset Current − nA IOS 1G INPUT BIAS AND OFFSET CURRENT vs CASE TEMPERATURE 10 5 100 M Figure 30. VS = 5 V 0.82 10 M f − Frequency − Hz INPUT BIAS AND OFFSET CURRENT vs CASE TEMPERATURE 0.8 VS = ±5 V, 5 V, and 3.3 V Gain = 1 RL = 150 Ω VIN= −1 dB TA = 25°C f − Frequency − Hz Figure 28. I IB − Input Bias Current − µ A VS = 5 V TA = 25°C 10 Figure 25. ZO − Output Impedance − Ω ZO − Output Impedance − Ω 1 −4 90 VICR − Input Common-Mode Voltage Range − V 100 10 VS = ±5 V TA = 25°C 10 100 0 IOS 0.86 −5 IIB+ 0.84 0.82 −10 −15 0.8 −20 IIB− 0.78 −25 I OS − Input Offset Current − nA CMMR 90 100 CMRR − Common-Mode Rejection Ratio − dB VS = ±5 V, 5 V, and 3.3 V COMMON-MODE REJECTION RATIO vs INPUT COMMON-MODE RANGE Crosstalk − dB 100 COMMON-MODE REJECTION RATIO vs INPUT COMMON-MODE RANGE CMRR − Common-Mode Rejection Ratio − dB REJECTION RATIO vs CASE TEMPERATURE −30 0.76 −40−30−20−10 0 10 20 30 40 50 60 70 80 90 Case Temperature − °C Case Temperature − °C Figure 31. Figure 32. Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 Submit Documentation Feedback 9 SN10501 SN10502 SN10503 SLOS408B – MARCH 2003 – REVISED JANUARY 2009 .................................................................................................................................................. www.ti.com APPLICATION INFORMATION HIGH-SPEED OPERATIONAL AMPLIFIERS The SN1050x operational amplifiers are a family of single, dual, and triple rail-to-rail output voltage feedback amplifiers. The SN1050x family combines both a high slew rate and a rail-to-rail output stage. WIDEBAND, INVERTING OPERATION Applications Section Contents • • • • • • • • • • • decrease the loading effect of the feedback network on the output of the amplifier, but this enhancement comes at the expense of additional noise and potentially lower bandwidth. Feedback-resistor values between 1 kΩ and 2 kΩ are recommended for most situations. Wideband, Noninverting Operation Wideband, Inverting Gain Operation Video Drive Circuits Single Supply Operation Power Supply Decoupling Techniques Recommendations Active Filtering With the SN1050x Driving Capacitive Loads Board Layout Thermal Analysis Additional Reference Material Mechanical Package Drawings and Since the SN1050x family are general-purpose, wideband voltage-feedback amplifiers, several familiar operational-amplifier applications circuits are available to the designer. Figure 34 shows a typical inverting configuration where the input and output impedances and noise gain from Figure 33 are retained in an inverting circuit configuration. Inverting operation is one of the more common requirements and offers several performance benefits. The inverting configuration shows improved slew rates and distortion due to the pseudo-static voltage maintained on the inverting input. 5V +VS + WIDEBAND, NONINVERTING OPERATION The SN1050x is a family of unity gain stable rail-to-rail output voltage feedback operational amplifiers designed to operate from a single 3-V to 15-V power supply. Figure 33 is the noninverting gain configuration of 2V/V used to demonstrate the typical performance curves. 5V 100 pF 499 Ω 50 Ω Source VI Rg Rf 1.3 kΩ RM 52.3 Ω 1.3 kΩ 0.1 µF 49.9 Ω −5 V 0.1 µF 6.8 µF 499 Ω 1.3 kΩ 0.1 µF 6.8 µF 100 pF −5 V + −VS Figure 33. Wideband, Noninverting Gain Configuration Voltage-feedback amplifiers, unlike current-feedback designs, can use a wide range of resistors values to set their gain with minimal impact on their stability and frequency response. Larger-valued resistors 10 −VS VO _ Rg + Figure 34. Wideband, Inverting Gain Configuration Rf 1.3 kΩ 6.8 µF 100 pF + VI VO _ + 50 Ω Source 6.8 µF + RT 649 Ω CT 0.1 µF +VS 100 pF 0.1 µF Submit Documentation Feedback In the inverting configuration, some key design considerations must be noted. One is that the gain resistor (Rg) becomes part of the signal channel input impedance. If input impedance matching is desired (beneficial when the signal is coupled through a cable, twisted pair, long PC-board trace, or other transmission-line conductors), Rg may be set equal to the required termination value and Rf adjusted to give the desired gain. However, care must be taken when dealing with low inverting gains, because the resulting feedback-resistor value can present a significant load to the amplifier output. For an inverting gain of 2, setting Rg to 49.9 Ω for input matching eliminates the need for RM but requires a 100-Ω feedback resistor. This has the advantage that the noise gain becomes equal to 2 for a 50-Ω source impedance—the same Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 SN10501 SN10502 SN10503 www.ti.com .................................................................................................................................................. SLOS408B – MARCH 2003 – REVISED JANUARY 2009 as the noninverting circuit in Figure 33. However, the amplifier output now sees the 100-Ω feedback resistor in parallel with the external load. To eliminate this excessive loading, increase both Rg and Rf, values, as shown in Figure 34, and then provide the input-matching impedance with a third resistor (RM) to ground. The total input impedance becomes the parallel combination of Rg and RM. The last major consideration to discuss in inverting amplifier design is setting the bias-current cancellation resistor on the noninverting input. If the resistance is set equal to the total dc resistance looking out of the inverting terminal, the output dc error, due to the input bias currents, is reduced to the input-offset current multiplied by Rf in Figure 34. The dc source impedance looking out of the inverting terminal is 1.3 kΩ || (1.3 kΩ + 25.6 Ω) = 649 Ω. To reduce the additional high-frequency noise introduced by the resistor at the noninverting input, and power-supply feedback, RT is bypassed with a capacitor to ground. Video Drive Circuits Most video-distribution systems are designed with 75-Ω series resistors to drive a matched 75-Ω cable. In order to deliver a net gain of 1 to the 75-Ω matched load, the amplifier is typically set up for a voltage gain of 2, compensating for the 6-dB attenuation of the voltage divider formed by the series and shunt 75-Ω resistors at either end of the cable. The circuit shown in Figure 36 meets this requirement. The SN1050x gain flatness and differential gain/phase performance provide exceptional results in video distribution applications. VS+ + 10 µF Video In 0.1 µF 5 3 4 75 Ω + 75 Ω 1 − VO 2 75 Ω SINGLE SUPPLY OPERATION The SN1050x family is designed to operate from a single 3-V to 15-V power supply. When operating from a single power supply, care must be taken to ensure that the input signal and amplifier are biased appropriately to allow for the maximum output voltage swing. The circuits shown in Figure 35 demonstrate methods to configure an amplifier for single-supply operation. +VS 50 Ω Source + VI 49.9 Ω RT VO _ 499 Ω +VS Rf 2 Rg 1.3 kΩ 1.43 kΩ 1.43 kΩ Figure 36. Cable Drive Application Differential gain and phase measure the change in overall small-signal gain and phase for the color subcarrier frequency (3.58 MHz in NTSC systems) vs changes in the large-signal output level (which represents luminance information in a composite video signal). The SN1050x, with the typical 150-Ω load of a single matched video cable, shows less than 0.007% / 0.007° differential gain/phase errors over the standard luminance range for a positive video (negative sync) signal. 1.3 kΩ VS+ 0.1 µF 52.3 Ω Video In 1.3 kΩ _ +VS +VS 2 2 75 Ω 5 3 4 1.3 kΩ RT VO 10 µF Rf VS Rg 75 Ω + 2 VI 10 µF 0.1 µF +VS 50 Ω Source + VS− + 75 Ω 1 − 75 Ω VO 2 75 Ω VO + 499 Ω 1.43 kΩ 1.43 kΩ 0.1 µF 75 Ω VO VS− Figure 35. DC-Coupled Single Supply Operation + 10 µF 75 Ω Figure 37. Video Distribution Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 Submit Documentation Feedback 11 SN10501 SN10502 SN10503 SLOS408B – MARCH 2003 – REVISED JANUARY 2009 .................................................................................................................................................. www.ti.com Similar performance is observed for negative video signals. In practice, similar performance is achieved even with three video loads as shown in Figure 37 due to the linear high-frequency output impedance of the SN1050x. This circuit is suitable for driving video cables, provided that the length does not exceed a few feet. If longer cables are driven, the gain of the SN1050x can be increased to compensate for cable loss. Configuring the SN1050x for single-supply video applications is easily done, but attention must be given to input and output bias voltages to ensure proper system operation. Unlike some video amplifiers, the SN1050x input common-mode voltage range does not include the negative power supply, but rather it is about 1-V from each power supply. For split supply configurations, this is very beneficial. For single-supply systems, there are some design constraints that must be observed. Figure 38 shows a single-supply video configuration illustrating the dc bias voltages acceptable for the SN1050x. The lower end of the input common-mode range is specified as 1 V. The upper end is limited to 4 V with the 5-V supply shown, but the output range and gain of 2 limit the highest acceptable input voltage to 4.5 V / 2 = 2.25 V. The 4.5-V output is what is typically expected with a 150-Ω load. It is easily seen that the input and output voltage ranges are limiting factors in the total system. Both specifications must be taken into account when designing a system. 1.24 kΩ 1.24 kΩ Input Range = 1 V to 2.25 V − 75 Ω + 75 Ω VO Range = 1 V to 2.25 V 75 Ω RT Figure 39. AC-Coupled Output Single-Supply Video Amplifier In some systems, the physical size and/or cost of a 470-µF capacitor can be prohibitive. One way to circumvent this issue is to use two smaller capacitors in a feedback configuration as shown in Figure 40. This is commonly known as SAG correction. This circuit increases the gain of the amplifier up to 3 V/V at low frequencies to counteract the increased impedance of the capacitor placed at the amplifier output. One issue that must be resolved is that the gain at low frequencies is typically limited by the power-supply voltage and the output swing of the amplifier. Therefore, it is possible to saturate the amplifier at these low frequencies if full analysis is not done on this system which includes both input and output requirements. 1.24 kΩ RT 22 µF 1.24 kΩ Input Range = 1 V to 2.25 V 1.24 kΩ 75 Ω + RT + VO Range = 0 V to 1.25V 470 µF 5V Output Range = 2 V to 4.5 V Output Range = 2 V to 4.5 V 5V 1.24 kΩ 5V Input Range = 1 V to 2.25 V 1.24 kΩ VO Range = 0 V to 1.25V 22 µF Output Range = 2 V to 4.5 V 75 Ω 75 Ω Figure 40. AC-Coupled SAG Corrected Output Single-Supply Video Amplifier Figure 38. DC-Coupled Single-Supply Video Amplifier In most systems, this may be acceptable because most receivers are ac-coupled and set the black level to the desired system value, typically 0 V (0-IRE). But, to ensure full compatibility with any system, it is often desirable to place an ac coupling capacitor on the output as shown in Figure 39. This removes the dc-bias voltage appearing at the amplifier output. To minimize field tilt, the size of this capacitor is typically 470 µF, although values as small as 220 µF have been used with acceptable results. 12 Submit Documentation Feedback Many times the output of the video encoder or DAC does not have the capability to output the 1-V to 2.25-V range, but rather a 0-V to 1.25-V range. In this instance, the signal must be ac-coupled to the amplifier input as shown in Figure 41. Note that it does not matter what the voltage output of the DAC is, but rather the voltage swing should be kept less than 1.25 VPP. Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 SN10501 SN10502 SN10503 www.ti.com .................................................................................................................................................. SLOS408B – MARCH 2003 – REVISED JANUARY 2009 1.24 kΩ 5V 5V DAC Output = 0 V to 1.25V 68 µF 1.24 kΩ 4.64 kΩ Output Range = 2 V to 4.5 V 75 Ω + VO Range = 0 V to 1.25V DAC Output = 0 V to 2V 5V 470 µF 1.24 kΩ 5V 5V DAC Output = 0 V to 2V 3.01 kΩ Output Range = 0.5 V to 4.5V 75 Ω + VO Range = 0 V to 2V 470 µF Input = 2.5 V Figure 44. Inverting AC-Coupled Wide Output Swing Single-Supply Video Amplifier APPLICATION CIRCUITS Active Filtering With the SN1050x High-frequency active filtering with the SN1050x is achievable due to the amplifier's high slew rate, wide bandwidth, and voltage feedback architecture. Several options are available for high-pass, low-pass, bandpass, and bandstop filters of varying orders. A simple two-pole, low-pass filter is presented in Figure 45 as an example, with two poles at 25 MHz. 4.7 pF VO Range = 0 V to 2V 50 Ω Source 1.3 kΩ 470 µF 47 µF 75 Ω 3.01 kΩ 75 Ω 75 Ω 10 kΩ To further increase dynamic range at the output, the output dc bias should be centered around 2.5 V for the 5-V system shown. However, a wide output range requires a wide input range, and should be centered around 2.5 V. The best ways to accomplish this are to ac-couple the gain resistor or bias it at 2.5 V with a reference supply as shown in Figure 42 and Figure 43. Output Range = 0.5 V to 4.5V − + Input Range = 1 V to 2.25V 1.24 kΩ 5V 10 µF 75 Ω Figure 41. AC-Coupled Input and Output Single-Supply Video Amplifier 68 µF 2.49 kΩ 10 kΩ 47 µF 2.26 kΩ 1.24 kΩ VI 1.3 kΩ 52.3 Ω Input Range = 1.5 V to 3.5V 5V _ 49.9 Ω VO Figure 42. AC-Coupled Wide Output Swing Single-Supply Video Amplifier + 33 pF −5 V 1.24 kΩ 1.24 kΩ Figure 45. A Two-Pole Active Filter With Two Poles Between 90 MHz and 100 MHz 2.5 V 5V 5V DAC Output = 0 V to 2V 3.01 kΩ + Output Range = 0.5 V to 4.5V 75 Ω 470 µF 47 µF 3.01 kΩ VO Range = 0 V to 2V 75 Ω Input Range = 1.5 V to 3.5V Figure 43. AC-Coupled Wide Output Swing Single-Supply Video Amplifier Using Voltage Reference Another beneficial configuration is to use the amplifier in an inverting configuration as shown in Figure 44. Driving Capacitive Loads A demanding, yet very common application for an op amp is capacitive loading. Often, this load is the input of an A/D converter, including additional external capacitance, sometimes recommended to improve A/D linearity. A high-speed, high open-loop gain amplifier like the SN1050x can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the amplifier's open-loop output resistance is considered, the capacitance introduces an additional pole in the signal path that can decrease the phase margin. When the primary considerations are frequency-response flatness, pulse response fidelity, or distortion, the simplest and most effective solution is to isolate the capacitive load Copyright © 2003–2009, Texas Instruments Incorporated Product Folder Link(s): SN10501 SN10502 SN10503 Submit Documentation Feedback 13 SN10501 SN10502 SN10503 SLOS408B – MARCH 2003 – REVISED JANUARY 2009 .................................................................................................................................................. www.ti.com from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. This does not eliminate the pole from the loop response, but rather shifts it and adds a zero at a higher frequency. The additional zero cancels the phase lag from the capacitive-load pole, thus increasing the phase margin and improving stability. Power Supply Decoupling Techniques and Recommendations Power-supply decoupling is a critical aspect of any high-performance amplifier design process. Careful decoupling provides higher-quality ac performance, most notably improved distortion performance. The following guidelines ensure the highest level of performance. 1. Place decoupling capacitors as close to the power-supply inputs as possible, with the goal of minimizing the inductance of the path from ground to the power supply 2. Placement priority; locate the smallest-value capacitors nearest to the device. 3. Solid power and ground planes are recommended to reduce the inductance along power-supply return-current paths, with the exception of the areas underneath the input and output pins. 4. Recommended values for power supply decoupling include a bulk decoupling capacitor (6.8 to 22 µF), a mid-range decoupling capacitor (0.1 µF) and a high frequency decoupling capacitor (1000 pF) for each supply. A 100 pF capacitor can be used across the supplies as well for extremely high-frequency return currents, but often is not required. BOARD LAYOUT Achieving optimum performance with a high-frequency amplifier like the SN1050x requires careful attention to board layout parasitics and external component types. Recommendations to optimize performance include: 1. Minimize parasitic capacitance to any ac ground for all signal I/O pins. Parasitic capacitance on the output and inverting-input pins can cause instability: on the noninverting input, it can react with the source impedance to cause unintentional band limiting. To reduce unwanted capacitance, open a window in all ground and power planes around the signal I/O pins. Keep ground and power planes unbroken elsewhere on the board. 2. Minimize the distance (< 0.25”) from the power-supply pins to high frequency 0.1-µF decoupling capacitors. At the device pins, the 14 Submit Documentation Feedback ground- and power-plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power supply connections should always be decoupled with these capacitors. Larger (2.2-µF to 6.8-µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. 3. Careful selection and placement of external components preserves the high frequency performance of the SN1050x. Choose low-reactance resistors. Surface-mount resistors work best, and allow a tighter overall layout. Metal-film and carbon-composition axial-lead resistors can also provide good high-frequency performance. Again, keep component leads and PC-board trace length as short as possible. Never use wirewound resistors in a high frequency application. Since the output pin and inverting-input pin are the most sensitive to parasitic capacitance, always position the feedback and series-output resistor, if any, as close as possible to the output pin. Other network components, such as noninverting-input termination resistors, should also be placed close to the package. Where double-sided component mounting is allowed, place the feedback resistor directly under the package on the other side of the board between the output and inverting input pins. Even with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant time constants that can degrade performance. Good axial-lead metal-film or surface-mount resistors have approximately 0.2 pF in shunt with the resistor. For resistor values > 2.0 kΩ, this parasitic capacitance can add a pole and/or a zero below 400 MHz that can affect circuit operation. Keep resistor values as low as possible, consistent with load-driving considerations. A good starting point for design is to set the Rf to 1.3 kΩ for low-gain, noninverting applications. This automatically keeps the resistor noise terms low, and minimizes the effect of their parasitic capacitance. 4. Connections to other wideband devices on the board may be made with short, direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Use relatively wide traces (50 mils to 100 mils), preferably with ground and power planes opened up around them. Low parasitic capacitive loads (