THS3092DDAR

THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 HIGH-VOLTAGE, LOW-DISTORTION, CURRENT-FEEDBACK OPERATIONAL AMPLIFIERS FEATURES • • • • • • • DESCRIPTION Low Distortion – 66 dBc HD2 at 10 MHz, RL = 100 Ω – 76 dBc HD3 at 10 MHz, RL = 100 Ω Low Noise – 13 pA/√Hz Noninverting Current Noise – 13 pA/√Hz Inverting Current Noise – 2 nV/√Hz Voltage Noise High Slew Rate: 5700 V/µs (G = 5, VO = 20 VPP) Wide Bandwidth: 160 MHz (G = 5, RL = 100 Ω) High Output Current Drive: ±250 mA Wide Supply Range: ±5 V to ±15 V Power-Down Feature: (THS3096 Only) APPLICATIONS • • • • High-Voltage Arbitrary Waveform Power FET Driver Pin Driver VDSL Line Driver Total Harmonic Distortion − dBc −30 −40 G = 5, RF = 715 Ω, RL = 100 Ω, VS = ±15 V −50 The THS3096 features a power-down pin (PD) that puts the amplifier in low power standby mode, and lowers the quiescent current from 9.5 mA to 500 µA. The wide supply range combined with total harmonic distortion as low as -66 dBc at 10 MHz, in addition, to the high slew rate of 5700 V/µs makes the THS3092/6 ideally suited for high-voltage arbitrary waveform driver applications. Moreover, having the ability to handle large voltage swings driving into high-resistance and high-capacitance loads while maintaining good settling time performance makes the THS3092/6 ideal for Pin driver and PowerFET driver applications. The THS3092 is offered in an 8-pin SOIC (D), and the 8-pin SOIC (DDA) packages with PowerPAD™. The THS3096 is offered in the 8-pin SOIC (D) and the 14-pin TSSOP (PWP) packages with PowerPAD. TYPICAL ARBITARY WAVEFORM GENERATOR OUTPUT DRIVE CIRCUIT TOTAL HARMONIC DISTORTION vs FREQUENCY −20 The THS3092 and THS3096 are dual high-voltage, low-distortion, high speed, current-feedback amplifiers designed to operate over a wide supply range of ±5 V to ±15 V for applications requiring large, linear output signals such as Pin, Power FET, and VDSL line drivers. VO = 20 VPP VOUT IOUT1 DAC5686 IOUT2 VO = 10 VPP − + − + THS3092 THS4271 −60 −70 VO = 5 VPP − + −80 VO = 2 VPP −90 100 k 1M 10 M THS3092 100 M f − Frequency − Hz Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. UNLESS OTHERWISE NOTED this document contains PRODUCTION DATA information current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2003–2006, Texas Instruments Incorporated THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. TOP VIEW D, DDA TOP VIEW D, PWP THS3096 THS3092 1VOUT 1VIN − 1VIN + VS− 1 8 2 7 3 6 4 5 1VOUT 1VIN− 1VIN+ VS− NC REF NC VS+ 2VOUT 2VIN− 2VIN+ NC = No Internal Connection 1 14 2 13 3 12 4 11 5 10 6 9 7 8 VS+ 2VOUT 2VIN− 2VIN+ NC PD NC NC = No Internal Connection See Note A. Note A: The devices with the power down option defaults to the ON state if no signal is applied to the PD pin. Additionallly, the REF pin functional range is from VS− to (VS+ − 4 V). ORDERING INFORMATION PART NUMBER THS3092D PACKAGE TYPE SOIC-8 THS3092DR THS3092DDA SOIC-8-PP (1) THS3092DDAR TRANSPORT MEDIA, QUANTITY Rails, 75 Tape and Reel, 2500 Rails, 75 Tape and Reel, 2500 Power-down THS3096D SOIC-8 THS3096DR THS3096PWP TSSOP-14-PP (1) THS3096PWPR (1) Rails, 75 Tape and Reel, 2500 Rails, 90 Tape and Reel, 2000 The PowerPAD is electrically isolated from all other pins. DISSIPATION RATING TABLE (1) (2) (3) 2 POWER RATING (2) PACKAGE ΘJC (°C/W) ΘJA (°C/W) (1) TA≤ 25°C TA = 85°C D-8 38.3 97.5 1.02 W 410 mW DDA-8 (3) 9.2 45.8 2.18 W 873 mW PWP-14 (3) 2.07 37.5 2.67 W 1.07 W This data was taken using the JEDEC standard High-K test PCB. Power rating is determined with a junction temperature of 125°C. This is the point where distortion starts to substantially increase. Thermal management of the final PCB should strive to keep the junction temperature at or below 125°C for best performance and long term reliability. The THS3092 and THS3096 may incorporate a PowerPAD™ on the underside of the chip. This acts as a heatsink and must be connected to a thermally dissipating plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction temperature which could permanently damage the device. See TI Technical Brief SLMA002 for more information about utilizing the PowerPAD™ thermally enhanced package. Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 RECOMMENDED OPERATING CONDITIONS Supply voltage MIN MAX Dual supply ±5 ±15 Single supply 10 30 -40 85 Operating free-air temperature, TA UNIT V °C ABSOLUTE MAXIMUM RATINGS over operating free-air temperature (unless otherwise noted) (1) UNIT Supply voltage, VS- to VS+ 33 V Input voltage, VI ± VS Differential input voltage, VID ±4V Output current, IO 350 mA Continuous power dissipation See Dissipation Ratings Table Maximum junction temperature, TJ 150°C Maximum junction temperature, continuous operation, long term reliability, TJ (2) 125°C Storage temperature, Tstg -65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 300°C ESD ratings: (1) (2) HBM 2000 CDM 1500 MM 150 The absolute maximum ratings under any condition is limited by the constraints of the silicon process. Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not implied. The maximum junction temperature for continuous operation is limited by package constraints. Operation above this temperature may result in reduced reliability and/or lifetime of the device. Submit Documentation Feedback 3 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 ELECTRICAL CHARACTERISTICS VS = ±15 V, RF = 909 Ω, RL = 100 Ω, and G = 2 (unless otherwise noted) TYP PARAMETER TEST CONDITIONS 25°C OVER TEMPERATURE 25°C 0°C to 70°C -40°C to 85°C UNIT MIN/TYP/ MAX MHz TYP V/µs TYP ns TYP ns TYP dBc TYP AC PERFORMANCE Small-signal bandwidth, -3 dB G = 1, RF = 1.1 kΩ, VO = 200 mVPP 135 G = 2, RF = 909 Ω, VO = 200 mVPP 145 G = 5, RF = 715 Ω, VO = 200 mVPP 160 G = 10, RF = 604 Ω, VO = 200 mVPP 145 0.1 dB bandwidth flatness G = 2, RF = 909 Ω, VO = 200 mVPP 50 Large-signal bandwidth G = 5, RF = 715 Ω , VO = 5 VPP 150 G = 2, VO = 10-V step, RF = 909 Ω 4000 G = 5, VO = 20-V step, RF = 715 Ω 5700 Slew rate (25% to 75% level) Rise and fall time G = 2, VO = 5-VPP, RF = 909 Ω 5 Settling time to 0.1% G = -2, VO = 2 VPP step 42 Settling time to 0.01% G = -2, VO = 2 VPP step 72 Harmonic distortion 2nd Harmonic distortion RL = 100Ω 66 RL = 1 kΩ 66 RL = 100Ω 76 RL = 1 kΩ 78 3rd Harmonic distortion G = 2, RF = 909 Ω , VO = 2 VPP, f = 10 MHz Input voltage noise f > 10 kHz 2 nV / √Hz TYP Noninverting input current noise f > 10 kHz 13 pA / √Hz TYP Inverting input current noise f > 10 kHz 13 pA / √Hz TYP Differential gain Differential phase Crosstalk G = 2, RL = 150 Ω, RF = 909 Ω G = 2, RL = 100 Ω, f = 10 MHz NTSC 0.013% PAL 0.011% NTSC 0.020° PAL 0.026° Ch 1 to 2 60 Ch 2 to 1 56 TYP dB DC PERFORMANCE Transimpedance VO = ±7.5 V, Gain = 1 Input offset voltage 850 350 300 300 kΩ MIN 0.9 3 4 4 mV MAX ±10 ±10 µV/°C TYP 4 15 20 20 µA MAX ±20 ±20 µA/°C TYP 3.5 15 20 20 µA MAX ±20 ±20 µA/°C TYP 1.7 10 15 15 µA MAX ±20 ±20 µA/°C TYP V MIN Average offset voltage drift Noninverting input bias current Average bias current drift Inverting input bias current VCM = 0 V Average bias current drift Input offset current Average offset current drift INPUT CHARACTERISTICS Common-mode input range Common-mode rejection ratio ±13.3 ±13 ±13 78 68 65 65 dB MIN Noninverting input resistance 1.3 MΩ TYP Noninverting input capacitance 0.1 pF TYP Inverting input resistance 30 Ω TYP Inverting input capacitance 1.4 pF TYP 4 VCM = ±10 V ±13.6 Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 ELECTRICAL CHARACTERISTICS (continued) VS = ±15 V, RF = 909 Ω, RL = 100 Ω, and G = 2 (unless otherwise noted) TYP PARAMETER TEST CONDITIONS OVER TEMPERATURE 25°C 25°C 0°C to 70°C -40°C to 85°C UNIT MIN/TYP/ MAX RL = 1 kΩ ±13.2 ±12.8 ±12.5 ±12.5 RL = 100 Ω ±12.5 ±12.1 ±11.8 ±11.8 V MIN Output current (sourcing) RL = 40 Ω 280 225 200 Output current (sinking) RL = 40 Ω 250 200 175 200 mA MIN 175 mA Output impedance f = 1 MHz, Closed loop 0.06 MIN Ω TYP OUTPUT CHARACTERISTICS Output voltage swing POWER SUPPLY Specified operating voltage ±15 ±16 ±16 ±16 V MAX Maximum quiescent current 9.5 10.5 11 11 mA MAX 9.5 8.5 8 8 mA MIN Minimum quiescent current Per channel Power supply rejection (+PSRR) VS+ = 15.5 V to 14.5 V, VS- = 15 V 75 70 65 65 dB MIN Power supply rejection (-PSRR) VS+ = 15 V, VS- = -15.5 V to -14.5 V 73 68 65 65 dB MIN POWER-DOWN CHARACTERISTICS (THS3096 ONLY) VS+– 4 REF voltage range (1) Power-down voltage level (1) Power-down quiescent current VPD quiescent current V Enable PD ≥ REF + 2 Disable PD ≤ REF + 0.8 PD = 0V 500 700 800 800 VPD = 0 V, REF = 0 V, 11 15 20 20 VPD = 3.3 V, REF = 0 V 11 15 20 20 Turnon time delay 90% of final value 60 Turnoff time delay 10% of final value 150 (1) MAX VS– MIN MIN MAX µA MAX µA MAX µs TYP For detailed information on th behavior of the powerdown circuit, see the Powerdown and Powerdown Reference sections in the Application Information of this data sheet. Submit Documentation Feedback 5 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 ELECTRICAL CHARACTERISTICS VS = ±5 V, RF = 909 Ω, RL = 100 Ω, and G = 2 (unless otherwise noted) TYP PARAMETER TEST CONDITIONS 25°C OVER TEMPERATURE 25°C 0°C to 70°C -40°C to 85°C UNIT MIN/TYP/ MAX MHz TYP V/µs TYP ns TYP ns TYP dBc TYP AC PERFORMANCE Small-signal bandwidth, -3 dB G = 1, RF = 1.1 kΩ, VO = 200 mVPP 125 G = 2, RF = 909 Ω, VO = 200 mVPP 140 G = 5, RF = 715 Ω, VO = 200 mVPP 145 G = 10, RF = 604 Ω, VO = 200 mVPP 135 0.1 dB bandwidth flatness G = 2, RF = 909 Ω, VO = 200 mVPP 42 Large-signal bandwidth G = 2, RF = 909 Ω , VO = 5 VPP 125 G = 2, VO= 5-V step, RF = 909 Ω 1050 G = 5, VO= 5-V step, RF = 715 Ω 1350 Slew rate (25% to 75% level) Rise and fall time G = 2, VO = 5-V step, RF = 909 Ω 5 Settling time to 0.1% G = -2, VO = 2 VPP step 35 Settling time to 0.01% G = -2, VO = 2 VPP step 73 Harmonic distortion 2nd Harmonic distortion RL = 100Ω 64 RL = 1 kΩ 67 RL = 100Ω 75 RL = 1 kΩ 75 3rd Harmonic distortion G = 2, RF = 909 Ω , VO = 2 VPP, f = 10 MHz Input voltage noise f > 10 kHz 2 nV / √Hz TYP Noninverting input current noise f > 10 kHz 13 pA / √Hz TYP Inverting input current noise f > 10 kHz 13 pA / √Hz TYP Differential gain Differential phase Crosstalk G = 2, RL = 150 Ω, RF = 909 Ω G = 2, RL = 100 Ω, f = 10 kHz NTSC 0.027% PAL 0.025% NTSC 0.04° PAL 0.05° Ch 1 to 2 60 Ch 2 to 1 56 TYP dB DC PERFORMANCE Transimpedance VO = ±2.5 V, Gain = 1 Input offset voltage 700 250 200 200 kΩ MIN 0.3 2 3 3 mV MAX ±10 ±10 µV/°C TYP 2 15 20 20 µA MAX ±20 ±20 µA/°C TYP 5 15 20 20 µA MAX ±20 ±20 µA/°C TYP 1 10 15 15 µA MAX ±20 ±20 µA/°C TYP V MIN Average offset voltage drift Noninverting input bias current Average bias current drift Inverting input bias current VCM = 0 V Average bias current drift Input offset current Average offset current drift INPUT CHARACTERISTICS Common-mode input range Common-mode rejection ratio ±3.3 ±3 ±3 66 60 57 57 dB MIN Noninverting input resistance 1.1 MΩ TYP Noninverting input capacitance 1.2 pF TYP Inverting input resistance 32 Ω TYP Inverting input capacitance 1.5 pF TYP 6 VCM = ±2.0 V, VO = 0 V ±3.6 Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 ELECTRICAL CHARACTERISTICS (continued) VS = ±5 V, RF = 909 Ω, RL = 100 Ω, and G = 2 (unless otherwise noted) TYP PARAMETER TEST CONDITIONS OVER TEMPERATURE 25°C 25°C 0°C to 70°C -40°C to 85°C UNIT MIN/TYP/ MAX RL = 1 kΩ ±3.4 ±3.1 ±2.8 ±2.8 RL = 100 Ω ±3.1 ±2.7 ±2.5 ±2.5 V MIN Output current (sourcing) RL = 10 Ω 200 160 140 Output current (sinking) RL = 10 Ω 180 150 125 140 mA MIN 125 mA Output impedance f = 1 MHz, Closed loop 0.09 MIN Ω TYP OUTPUT CHARACTERISTICS Output voltage swing POWER SUPPLY Specified operating voltage ±5 ±4.5 ±4.5 ±4.5 V MAX Maximum quiescent current 8.2 9 9.5 9.5 mA MAX 8.2 7 6.5 6.5 mA MIN Minimum quiescent current Per channel Power supply rejection (+PSRR) VS+ = 5.5 V to 4.5 V, VS- = -5 V 73 68 63 63 dB MIN Power supply rejection (-PSRR) VS+ = 5 V, VS- = -4.5 V to 5.5 V 71 65 60 60 dB MIN POWER-DOWN CHARACTERISTICS (THS3096 ONLY) VS+ – 4 REF voltage range (1) Power-down voltage level (1) Power-down quiescent current VPD quiescent current PD ≥ REF + 2 Disable PD ≤ REF + 0.8 PD = 0V 300 500 600 600 VPD = 0 V, REF = 0 V, 11 15 20 20 VPD = 3.3 V, REF = 0 V 11 15 20 20 90% of final value 60 Turnoff time delay 10% of final value 150 MIN V Enable Turnon time delay (1) MAX VS– MIN MAX µA MAX µA MAX µs TYP For detailed information on th behavior of the powerdown circuit, see the Powerdown and Powerdown Reference sections in the Application Information of this data sheet. TYPICAL CHARACTERISTICS TABLE OF GRAPHS FIGURE ±15-V graphs Noninverting frequency response 1, 2 Inverting frequency response 3 0.1 dB flatness 4 Noninverting frequency response 5 Inverting frequency response 6 Frequency response capacitive load 7 Recommended RISO vs Capacitive load 2nd Harmonic distortion vs Frequency 3rd Harmonic distortion vs Frequency Slew rate vs Output voltage step Noise vs Frequency Settling time 8 9, 11 10, 12 13, 14, 15 16 17, 18 Quiescent current vs Supply voltage 19 Output voltage vs Load resistance 20 Input bias and offset current vs Case temperature 21 Input offset voltage vs Case temperature 22 Submit Documentation Feedback 7 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 TYPICAL CHARACTERISTICS (continued) FIGURE Transimpedance vs Frequency 23 Rejection ratio vs Frequency 24 Noninverting small signal transient response 25 Inverting large signal transient response 26, 27 Overdrive recovery time 28 Differential gain vs Number of loads 29 Differential phase vs Number of loads 30 Closed loop output impedance vs Frequency 31 Crosstalk vs Frequency 32 Power-down quiescent current vs Supply voltage 33 Turnon and turnoff time delay 8 34 Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 TABLE OF GRAPHS FIGURE ±5-V graphs Noninverting frequency response 35 Inverting frequency response 36 0.1 dB flatness 37 Noninverting frequency response 38 Inverting frequency response 39 Settling time 40 2nd Harmonic distortion vs Frequency 3rd Harmonic distortion vs Frequency 41 Slew rate vs Output voltage step 42 43, 44, 45 Noninverting small signal transient response 46 Output voltage load resistance 47 Input bias and offset current vs Case temperature 48 Overdrive recovery time 49 Rejection ratio vs Frequency 50 Crosstalk vs Frequency 51 TYPICAL CHARACTERISTICS (±15 V) NONINVERTING FREQUENCY RESPONSE RF = 499 Ω RF = 909 Ω 7 Noninverting Gain − dB Noninverting Gain − dB 8 6 5 RF = 1.2 k Ω 4 3 2 1 Gain = 2, RL = 100 Ω, VO = 200 mVPP, VS = ±15 V 0 1M 10 M 100 M f − Frequency − Hz Figure 1. 1G 24 22 20 18 16 14 12 10 INVERTING FREQUENCY RESPONSE RL = 100 Ω, VO = 200 mVPP, VS = ±15 V G = 10, RF = 604 Ω Inverting Gain − dB 9 NONINVERTING FREQUENCY RESPONSE G = 5, RF = 715 Ω G =2, RF = 909 Ω 8 6 4 2 0 −2 −4 G =1, RF = 1.1 kΩ 1M 10 M 100 M f − Frequency − Hz Figure 2. Submit Documentation Feedback 1G 24 22 20 18 16 14 12 10 8 6 4 2 0 −2 −4 RL = 100 Ω, VO = 200 mVPP, VS = ±15 V G = −10, RF = 604 Ω G = −5, RF = 715 Ω G = −2, RF = 806 Ω G = −1, RF = 909 Ω 1M 10 M 100 M 1G f − Frequency − Hz Figure 3. 9 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 TYPICAL CHARACTERISTICS (±15 V) (continued) 0.1 dB FLATNESS NONINVERTING FREQUENCY RESPONSE 16 Gain = 2, RF = 909 Ω, RL = 200 Ω, VO = 200 mVPP, VS = ±15 V 6.1 6 5.9 5.8 12 VO = 10VPP VO = 5VPP VO = 1VPP Gain = 5, RF = 715 Ω, RL = 100 Ω, VS = ±15 V 4 100 M 1G 1M Figure 6. FREQUENCY RESPONSE CAPACITIVE LOAD RECOMMENDED RISO vs CAPTIVATE LOAD 2ND HARMONIC DISTORTION vs FREQUENCY R(ISO) = 30.9 Ω CL = 22 pF 8 R(ISO) = 20 Ω CL = 47 pF R(ISO) = 15 Ω CL = 100 pF Gain = 5 RL = 100 Ω VS =±15 V 2nd Harmonic Destortion − dBc Recommended R ISO − Ω 10 −40 Gain = 5, VS = ±15 V 40 12 0 VO = 2 VPP Figure 5. 45 2 VO = 5 VPP Figure 4. 100 M R(ISO) = 39.2 Ω CL = 10 pF 4 Gain = −5, RF = 715 Ω, RL = 100 Ω, VS = ±15 V f − Frequency − Hz 14 35 30 25 20 15 715 Ω 178 Ω 10 RISO − CL + 5 100 M 1G VS = ±15 V, VO = 2 VPP 1G G=1 RF = 1.1 kΩ, RL = 100 Ω −50 −60 −70 G=1 RF = 1.1 kΩ, RL = 1 kΩ G=2 RF = 909 Ω, RL = 1 kΩ −80 G=2 RF = 909 Ω, RL = 100 Ω −90 0 10 M 10 100 k 100 1M 10 M 100 M f − Frequency − Hz CL − Capacitive Load − pF f − Frequency − Hz Figure 7. Figure 8. Figure 9. 3RD HARMONIC DISTORTION vs FREQUENCY 2ND HARMONIC DISTORTION vs FREQUENCY 3RD HARMONIC DISTORTION vs FREQUENCY −40 −30 −50 −60 −70 −30 VO = 20 VPP G=1 RF = 1.1 kΩ, RL = 1 kΩ −40 G=1 RF = 1.1 kΩ, RL = 100 Ω G=2 RF = 909 Ω, RL = 100 Ω −50 −60 −70 VO = 10 VPP G=5 RF = 715 Ω, RL = 100 Ω, Vs = ±15 V VO = 5 VPP −80 −80 −90 100 k G=2 RF = 909 Ω, RL = 1 kΩ 1M 10 M f − Frequency − Hz Figure 10. 100 M VO = 20 VPP −40 VO = 10 VPP −50 −60 −70 G=5 RF = 715 Ω, RL = 100 Ω, Vs = ±15 V −80 VO = 5 VPP −90 −90 −100 100 k Harmonic Distortion −dBc VS = ±15 V, VO = 2 VPP Harmonic Distortion −dBc 3rd Harmonic Distortion − dBc 2 10 M VO = 10 VPP 6 0 1M VO = 20 VPP 8 10 M 100 M f − Frequency − Hz 1M 10 M f − Frequency − Hz 16 6 10 4 0 100 k 10 12 8 6 VO = 1 VPP 14 VO = 20VPP 10 2 5.7 Signal Gain − dB VO = 2VPP 14 Noninverting Gain − dB Noninverting Gain − dB 6.2 16 Inverting Gain − dB 6.3 −2 INVERTING FREQUENCY RESPONSE VO = 2 VPP VO = 2 VPP −100 1M 10 M 100 M f − Frequency − Hz Figure 11. Submit Documentation Feedback 100 k 1M 10 M f − Frequency − Hz Figure 12. 100 M THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 TYPICAL CHARACTERISTICS (±15 V) (continued) SLEW RATE vs OUTPUT VOLTAGE STEP SLEW RATE vs OUTPUT VOLTAGE STEP 4000 700 6000 Gain = 2 RL = 100 Ω RF = 909 Ω VS = ±15 V 3500 600 SR − Slew Rate − V/ µ s Rise Fall 500 400 300 Gain = 1 RL = 100 Ω RF = 1.1 kΩ VS = ±15 V 200 100 3000 2500 2000 Rise 1500 1000 0.5 1 1.5 2 2.5 3 3.5 1 2 3 4 5 6 7 8 9 10 0 10 12 14 16 18 20 Figure 15. NOISE vs FREQUENCY SETTLING TIME SETTLING TIME Rising Edge 10 In+ Vn VO − Output Voltage − V VO − Output Voltage − V In− 0.5 0.25 Gain = −2 RL = 100 Ω RF =806 Ω VS = ±15 V 0 −0.25 −0.5 −0.75 Falling Edge −1 1 100 1k 10 k f − Frequency − Hz −1.25 100 k 0 1 2 3 4 5 6 7 8 9 10 4.5 4 3.5 3 2.5 2 1.5 1 0.5 0 −0.5 −1 −1.5 −2 −2.5 −3 −3.5 −4 −4.5 Rising Edge Gain = −2 RL = 100 Ω RF = 806 Ω VS = ±15 V Falling Edge 0 2 4 t − Time − ns 6 8 10 Figure 17. Figure 18. QUIESCENT CURRENT vs SUPPLY VOLTAGE OUTPUT VOLTAGE vs LOAD RESISTANCE INPUT BIAS AND OFFSET CURRENT vs CASE TEMPERATURE TA = 25 °C 9.5 9 TA = −40 °C 8.5 8 7.5 Per Channel 7 5 6 7 8 9 10 11 12 13 14 15 VS − Supply Voltage − ±V Figure 19. VO - Output Voltage - V TA = 85 °C 16 7 12 6.5 6 I IB - Input Bias Currents - µ A I OS - Input Offset Currents - µ A 11 8 4 VS = ±15 V TA = -40 to 85°C 0 -4 -8 -12 -16 10 100 RL - Load Resistance - Ω Figure 20. Submit Documentation Feedback 12 t − Time − ns Figure 16. 10.5 4 8 Figure 14. 100 3 6 Figure 13. 0.75 10 4 VO − Output Voltage −VPP 1.25 10 2 VO − Output Voltage −VPP 1 Hz Fall 0 0 4 1000 I n − Current Noise − pA/ Hz 2000 1000 VO − Output Voltage − VPP Vn − Voltage Noise − nV/ 3000 0 0 I Q− Quiescent Current − mA 4000 Fall 500 0 Rise Gain = 5 RL = 100 Ω RF = 715 Ω VS = ±15 V 5000 SR − Slew Rate − V/µ s 800 SR − Slew Rate − V/ µ s SLEW RATE vs OUTPUT VOLTAGE STEP 1000 VS = ±15 V IIB- 5.5 5 4.5 4 IIB+ 3.5 3 2.5 2 1.5 1 IOS 0.5 0 -40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90 TC - Case Temperature - °C Figure 21. 11 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 TYPICAL CHARACTERISTICS (±15 V) (continued) INPUT OFFSET VOLTAGE vs CASE TEMPERATURE TRANSIMPEDANCE vs FREQUENCY 70 100 2.5 2 VS = ±15 V 1.5 1 VS = ±5 V 0.5 90 VS = ±15 V and ±5 V 70 60 50 40 30 10 10 0 1M 10 M 100 M 1G 100 k 1M 10 M 100 M 1G f − Frequency − Hz Figure 24. NONINVERTING SMALL SIGNAL TRANSIENT RESPONSE INVERTING LARGE SIGNAL TRANSIENT RESPONSE INVERTING LARGE SIGNAL TRANSIENT RESPONSE 6 0.1 Input 0 −0.05 −0.1 Gain = 2, RL = 100 Ω, RF = 909 Ω, VS = ±15 V −0.15 −0.2 −0.25 0 10 20 30 3 2 1 Input 0 −1 −2 Gain = 2, RL = 100 Ω, RF = 715 Ω, VS = ±15 V −3 −4 −5 −6 −5 0 −0.3 −10 Output 4 40 50 60 70 VO − Output Voltage − V 0.15 0.05 12 10 5 Output VO − Output Voltage − V Gain = −5, RL = 100 Ω, RF = 715 Ω, VS = ±15 V 8 6 4 2 Input 0 −2 −4 −6 Output −8 −10 −12 5 10 15 20 25 30 35 40 45 50 55 60 −10 0 10 20 30 40 50 60 t − Time − ns t − Time − ns t − Time − ns Figure 25. Figure 26. Figure 27. OVERDRIVE RECOVERY TIME DIFFERENTIAL GAIN vs NUMBER OF LOADS DIFFERENTIAL PHASE vs NUMBER OF LOADS 20 0.10 4 G = 5, RL = 100 Ω, RF = 715 Ω, VS = ±15 V 3 1 0 0 −5 −1 −10 −2 −15 −3 Differential Gain − % 5 0.08 VI − Input Voltage − V 2 0.07 PAL 0.04 0.03 0.02 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 t − Time − µs Figure 28. 1 Gain = 2 RF = 909 Ω VS = ±15 V 40 IRE − NTSC and Pal Worst Case ±100 IRE Ramp 0.04 0.03 PAL 0.02 NTSC 0.01 NTSC 0.01 −20 ° 0.06 0.05 70 0.05 Gain = 2 RF = 909 Ω VS = ±15 V 40 IRE − NTSC and Pal Worst Case ±100 IRE Ramp 0.09 Differential Phase − VO − Output Voltage − V PSRR+ 20 Figure 23. 0.2 VO − Output Voltage − V 30 f − Frequency − Hz 0.3 12 CMRR 40 Figure 22. 0.25 10 50 20 TC - Case Temperature - °C 15 PSRR− 80 0 100 k 0 -40 -30 -20-10 0 10 20 30 40 50 60 70 80 90 VS = ±15 V 60 Rejection Ratio − dB Transimpedance Gain − dB Ohms VOS - Input Offset Voltage - mV 3 REJECTION RATIO vs FREQUENCY −4 0 0 1 2 3 4 5 6 7 8 0 0 1 2 3 4 5 6 Number of Loads − 150 Ω Number of Loads − 150 Ω Figure 29. Figure 30. Submit Documentation Feedback 7 8 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 TYPICAL CHARACTERISTICS (±15 V) (continued) CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY CROSSTALK vs FREQUENCY 10 −10 G= 5, CH1 to 2 1 909 Ω 600 VS = ±15 V, RL = 100 Ω −20 Crosstalk − dB Closed-Loop Output Impedance − Ω Gain = 2, RISO = 5.11 Ω, RF = 909 Ω, VS = ±15 V Powerdown Quiescent Current - µ A 0 100 POWER-DOWN QUIESCENT CURRENT vs SUPPLY VOLTAGE 909 Ω 0.1 −40 G= 5, CH2 to 1 −50 −60 G= 2, CH2 to 1 −70 −80 5.11 Ω VO − −30 + G= 2, CH1 to 2 −90 0.01 −100 1M 10 M 100 M 1G 500 TA = 85°C 400 TA = -40°C 300 TA = 25°C 200 100 0 100 k 1M 10 M 100 M f − Frequency − Hz f − Frequency − Hz Figure 31. 3 1G 4 5 6 7 8 9 10 11 12 13 14 15 VS - Supply Voltage - ±V Figure 32. Figure 33. TURNON AND TURNOFF TIME DELAY 6 5 4 Gain = 2, VI = 0.1 Vdc RL = 100 Ω VS = ±15 V and ±5 V 3 2 1 0 0.3 0.2 0.1 Power-on Pulse − V VO − Output Voltage Level − V Power-on Pulse Output Voltage 0 −0.1 0 1 2 3 4 5 6 7 t − Time − ms Figure 34. TYPICAL CHARACTERISTICS (±5 V) NONINVERTING FREQUENCY RESPONSE 8 6 4 2 0 −2 −4 G = 5, RF = 715 Ω RL = 100 Ω, VO = 200 mVPP, VS = ±5 V G = 2, RF = 909 Ω 10 M 6.3 G = −10, RF = 604 Ω 100 M f − Frequency − Hz Figure 35. 1G Gain = 2, RF = 909 Ω, RL = 100 Ω, VO = 200 mVPP, VS = ±5 V 6.2 G = −5, RF = 715 Ω RL = 100 Ω, VO = 200 mVPP, VS = ±5 V G = −2, RF = 806 Ω 2 0 −2 −4 G = 1, RF = 1.1 kΩ 1M 16 14 12 10 8 6 4 0.1 dB FLATNESS Noniverting Gain −dB 16 14 12 10 24 22 20 18 G = 10, RF = 604 Ω Inverting Gain − dB Noninverting Gain − dB 24 22 20 18 INVERTING FREQUENCY RESPONSE 6.1 6 5.9 5.8 G = −1, RF = 906 Ω 5.7 1M 10 M 100 M 1G 1 10 f − Frequency − Hz f − Frequency − MHz Figure 36. Figure 37. Submit Documentation Feedback 100 13 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 TYPICAL CHARACTERISTICS (±5 V) (continued) NONINVERTING FREQUENCY RESPONSE INVERTING FREQUENCY RESPONSE 16 16 1.25 VO = 2 VPP 12 10 VO = 1 VPP 8 VO = 5 VPP 6 G = 5, RF = 715 Ω, RL = 100 Ω, VS = ±5V 1M 10 8 VO = 2 VPP 4 G = 5, RF = 715 Ω, RL = 100 Ω, VS = ±5V 2 100 M 10 M 0 1G 0 1 2 3 5 6 7 8 9 SLEW RATE vs OUTPUT VOLTAGE STEP −40 G = 2, RF = 909 Ω, RL = 100 Ω −70 G =1, RF = 1.1 kΩ, RL = 1 kΩ −80 G = 2, RF = 909 Ω, RL = 1 kΩ 100 k 1M 10 M f − Frequency − Hz −50 G = 2, RF = 909Ω, RL = 1 kΩ G = 1, RF = 1.1 kΩ, RL = 1 kΩ −70 Gain =1 RL = 100 Ω RF = 1.1 kΩ VS = ±5 V 800 G = 1, RF = 1.1 kΩ, RL = 100 Ω −60 700 Rise 600 Fall 500 400 300 200 −80 VO = 2 VPP, VS = ±5 V 100 −90 100 k 100 M 10 900 G = 2, RF = 909Ω, RL = 100 Ω G =1, RF = 1.1 kΩ, RL = 100 Ω 1M 10 M 0 100 M 0 0.5 1 1.5 2 2.5 3 3.5 4 f − Frequency − Hz VO − Output Voltage −VPP Figure 41. Figure 42. Figure 43. SLEW RATE vs OUTPUT VOLTAGE STEP SLEW RATE vs OUTPUT VOLTAGE STEP NONINVERTING SMALL SIGNAL TRANSIENT RESPONSE 1400 Gain = 2 RL = 100 Ω RF = 909Ω VS = ±5 V Rise SR − Slew Rate − V/ µ s 1200 Fall 800 0.3 Gain = 5 RL = 100 Ω RF = 715 Ω VS = ±5 V 700 600 500 400 300 200 0.25 Output 0.2 Rise VO − Output Voltage − V 1200 1000 Fall 800 600 400 0.15 0.1 Input 0.05 0 −0.05 −0.1 Gain = 2 RL = 100 Ω RF = 909 Ω VS = ±5 V −0.15 −0.2 200 100 −0.25 0 0 4 t − Time − ns 3RD HARMONIC DISTORTION vs FREQUENCY 3rd Harmonic Distortion − dBc 2nd Harmonic Destortion − dBc −1.25 1G 2ND HARMONIC DISTORTION vs FREQUENCY −90 14 Falling Edge −1 Figure 40. −60 900 −0.5 Figure 39. −50 1000 Gain = −2 RL = 100 Ω RF = 806 Ω VS = ±5 V 0 Figure 38. VS = ±5 V, VO = 2 VPP 1100 0.5 0.25 −0.75 VO = 5 VPP 10 M 100 M f − Frequency − Hz 1M f − Frequency − Hz −40 Rising Edge 0.75 −0.25 6 SR − Slew Rate − V/ µ s 4 2 SR − Slew Rate − V/ µ s VO − Output Voltage − V 12 1 VO = 1 VPP 14 Inverting Gain − dB Noninverting Gain − dB 14 0 SETTLING TIME 0 VO − Output Voltage −VPP 1 2 3 4 VO − Output Voltage −VPP Figure 44. Figure 45. 1 2 3 4 5 0 Submit Documentation Feedback 5 −0.3 −10 0 10 20 30 40 t − Time − ns Figure 46. 50 60 70 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 TYPICAL CHARACTERISTICS (±5 V) (continued) INPUT BIAS AND OFFSET CURRENT vs CASE TEMPERATURE 2 1.5 1 0.5 VS = ±5 V TA = -40 to 85°C 0 -0.5 -1 -1.5 -2 -2.5 -3 5 VS = ±5 V IIB- 6 5 IOS 4 3 2 IIB+ 1 0 -40 -30 -20 -10 0 -3.5 100 1000 Figure 47. 0.8 0.6 0.4 1 0.2 0 0 −1 −0.2 −2 −0.4 −3 −0.6 −4 −0.8 −1 0 10 20 30 40 50 60 70 80 90 0.2 0.4 0.6 0.8 1 t − Time − µs Figure 48. Figure 49. REJECTION RATIO vs FREQUENCY CROSSTALK vs FREQUENCY 70 0 VS = ±5 V −10 60 VS = ±5 V, RL = 100 Ω −20 PSRR- G= 5, CH1 to 2 50 40 20 2 TC - Case Temperature - °C RL - Load Resistance - Ω 30 3 −5 Crosstalk − dB 10 1 Gain = 5, RL = 100 Ω, RF = 715 Ω, VS = ±5 V 4 VI − Input Voltage − V 7 OVERDRIVE RECOVERY TIME VO − Output Voltage − A 8 3 2.5 I IB - Input Bias Current - µ A I OS - Input Offset Current - µ A 3.5 Rejection Ratio - dB VO - Output Voltage - V OUTPUT VOLTAGE vs LOAD RESISTANCE CMRR PSRR+ −30 −40 G= 5, CH2 to 1 −50 −60 G= 2, CH2 to 1 −70 −80 10 0 100 k G= 2, CH1 to 2 −90 −100 1M 10 M f - Frequency - Hz 100 M 100 k Figure 50. Submit Documentation Feedback 1M 10 M 100 M f − Frequency − Hz 1G Figure 51. 15 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 APPLICATION INFORMATION WIDEBAND, NONINVERTING OPERATION The THS3092/6 are unity gain stable 135-MHz current-feedback operational amplifiers, designed to operate from a ±5-V to ±15-V power supply. Figure 52 shows the THS3092 in a noninverting gain of 2-V/V configuration typically used to generate the performance curves. Most of the curves were characterized using signal sources with 50-Ω source impedance, and with measurement equipment presenting a 50-Ω load impedance. 15 V + 0.1 µF + VI 49.9 Ω 6.8 µF 1 2 _ −VS −15 V Figure 52. Wideband, Noninverting Gain Configuration ±15 -- 1.1 k ±5 -- 1.1 k ±15 909 909 909 178 715 ±5 178 715 ±15 66.5 604 ±5 66.5 604 -1 ±15 and ±5 909 909 -2 ±15 and ±5 402 806 -5 ±15 and ±5 143 715 -10 ±15 and ±5 60.4 604 10 Current-feedback amplifiers are highly dependent on 16 RF (Ω) 909 6.8 µF + RG (Ω) ±5 5 0.1 µF SUPPLY VOLTAGE (V) ±15 50 Ω LOAD 909 Ω RG GAIN (V/V) 49.9 Ω RF 909 Ω Table 1. Recommended Resistor Values for Optimum Frequency Response THS3092 and THS3096 RF and RG values for minimal peaking with RL = 100 Ω +VS 50 Ω Source the feedback resistor RF for maximum performance and stability. Table 1 shows the optimal gain setting resistors RF and RG at different gains to give maximum bandwidth with minimal peaking in the frequency response. Higher bandwidths can be achieved, at the expense of added peaking in the frequency response, by using even lower values for RF. Conversely, increasing RF decreases the bandwidth, but stability is improved. Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 WIDEBAND, INVERTING OPERATION Figure 53 shows the THS3092 in a typical inverting gain configuration where the input and output impedances and signal gain from Figure 52 are retained in an inverting circuit configuration. +VS 50 Ω Source + VI 15 V +VS + 0.1 µF + VI RG RF 402 Ω RM 57.6 Ω 806 Ω 0.1 µF RG 909 Ω 6.8 µF +VS 2 + 50 Ω Source VI 57.6 Ω −15 V +VS 2 909 Ω RF VS RG _ 402 Ω RT + −VS Figure 53. Wideband, Inverting Gain Configuration 50 Ω LOAD RF 49.9 Ω 50 Ω LOAD 49.9 Ω _ +VS 2 6.8 µF _ 50 Ω Source 49.9 Ω RT 806 Ω 49.9 Ω 50 Ω LOAD +VS 2 Figure 54. DC-Coupled, Single-Supply Operation SINGLE SUPPLY OPERATION VDSL Driver Circuit The THS3092/6 have the capability to operate from a single supply voltage ranging from 10 V to 30 V. When operating from a single power supply, biasing the input and output at mid-supply allows for the maximum output voltage swing. The circuits shown in Figure 54 shows inverting and noninverting amplifiers configured for single supply operations. The THS3092 and THS3096 have the ability to drive over 200 mA of current with very high voltage swings. Using these amplifiers coupled with the very high slew rate, low distortion, and low noise required in VDSL applications, makes for a perfect match. In VDSL systems where the receive signal is critical, the use of a low transformer ratio is necessary. With this low ratio, the output swing required from the line driver amplifier must increase, especially when driving the VDSL’s full 14.5-dBm power onto the line. The line driver's low distortion and noise is critical for the VDSL as the receive bands are intertwined with the transmit frequency bands up to the 12-MHz VDSL limit. Figure 55 shows a traditional hybrid connection approach for achieving the 14.5-dBm line power utilizing a 1:1 transformer. Looking at the input to the amplifiers shows a low-pass filter consisting of two separate capacitors to ground. There is an argument that since the signal coming out of the DAC is fully-differential then a single capacitor (10 pF in this case) is perfectly acceptable. The problem with this idea is that many DACs have common-mode energy due to images around the sampling frequency which must be filtered before reaching the amplifier. An amplifier simply amplifies its input–including the DAC’s images at high frequencies–and pass it through to the transformer and ultimately to the line, possibly causing the system to fail EMC compliance. A single capacitor does not remove these Submit Documentation Feedback 17 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 20 V 10 V 0.01 F 4.99 k 1:1 330 pF 200  THS3092 24.9  *Hybrid Connection Not Shown For Simplicity 4.99 k Hybrid 0.022 F To RX 604
0.022 F 10 V 0.015 F THS3092 Figure 56. 100
− + 330 pF + 14.5 dBm Line Power 604
133
200
100  − DAC VIN− − 22 pF 1.21 k 14.5 dBm Line Power 6.8 F 49.9
+ 22 pF DAC VIN− 604  THS3092 330 pF 1.21 k 0.01 F 22 pF 0.01 F DAC VIN+ 1:1 604  191  26 V 4.99 k
− 22 pF 13 V 200
24.9  + 330 pF 0.022 F DAC VIN+ 6.8 F THS3092 200  0.022 F common-mode images, it only removes the differential signal images. However, two separate filter capacitors filter both the common-mode signals and the differential-mode signals. Be sure to place the ground connection point of the capacitors next to each other, and then tie a single ground point at the middle of this trace. Video Distribution 49.9
4.99 k
13 V Figure 55. Additionally, level shifting must be done to center the common-mode voltage appearing at the amplifier’s noninverting input to optimally the midpoint of the power supply. As a side benefit of the ac-coupling/level shifter, a simple high pass filter is formed. This is generally a good idea for VDSL systems where the transmit band is typically above 1 MHz, but can be as low as 25 kHz. One of the concerns about any DSL line driver is the power dissipation. One of the most common ways to reduce power is by using active termination, aka synthesized impedance. Refer to TI Application Note SLOA100 for more information on active termination. The drawback to active termination is the received signal is reduced by the same synthesis factor utilized in the system. Due to the very high attenuation of the line at up to 12 MHz, the receive signal can be severely diminished. Thus, the use of active termination should be kept to modest levels at best. Figure 56 shows an example of utilizing a simple active termination scheme with a synthesis factor of 2 to achieve the same line power, but with a reduced power supply voltage that ultimately saves power in the system. The wide bandwidth, high slew rate, and high output drive current of the THS3092/6 matches the demands for video distribution for delivering video signals down multiple cables. To ensure high signal quality with minimal degradation of performance, a 0.1-dB gain flatness should be at least 7x the passband frequency to minimize group delay variations from the amplifier. A high slew rate minimizes distortion of the video signal, and supports component video and RGB video signals that require fast transition times and fast settling times for high signal quality. 909 Ω 909 Ω 15 V − + VI 75 Ω 75-Ω Transmission Line −15 V 75 Ω n Lines VO(1) 75 Ω VO(n) 75 Ω 75 Ω Figure 57. Video Distribution Amplifier Application Driving Capacitive Loads Applications such as FET line drivers can be highly capacitive and cause stability problems for high-speed amplifiers. Figure 58 through Figure 63 show recommended methods for driving capacitive loads. The basic idea is to use a resistor or ferrite chip to isolate the phase shift at high frequency caused by the capacitive load from the amplifier’s feedback path. See Figure 58 for recommended resistor values versus capacitive load. 18 Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 45 Gain = 5, VS = ±15 V Recommended R ISO − Ω 40 35 30 25 20 15 715 Ω 178 Ω 10 − RISO CL + 5 0 10 100 CL − Capacitive Load − pF Figure 58. Recommended RISO vs Capacitive Load Figure 61 shows another method used to maintain the low frequency load independence of the amplifier while isolating the phase shift caused by the capacitance at high frequency. At low frequency, feedback is mainly from the load side of RISO. At high frequency, the feedback is mainly via the 27-pF capacitor. The resistor RIN in series with the negative input is used to stabilize the amplifier and should be equal to the recommended value of RF at unity gain. Replacing RIN with a ferrite of similar impedance at about 100 MHz as shown in Figure 62 gives similar results with reduced dc offset and low frequency noise. (See the ADDITIONAL REFERENCE MATERIAL section for Expanding the usability of current-feedback amplifiers.) RF 715 Ω 27 pF VS 178 Ω _ + 5.11 Ω VS 715 Ω RG 178 Ω 1 µF −VS RIN 100 Ω LOAD RISO 715 Ω VS _ 100 Ω LOAD 5.11 Ω + 49.9 Ω 1 µF −VS 49.9 Ω VS Figure 59. Figure 61. 715 Ω 178 Ω RF VS Ferrite Bead _ + −VS VS 1 µF 27 pF FIN 100 Ω LOAD RG 49.9 Ω 715 Ω 178 Ω FB VS _ + −VS VS Figure 60. Placing a small series resistor, RISO, between the amplifier’s output and the capacitive load, as shown in Figure 59, is an easy way of isolating the load capacitance. Using a ferrite chip in place of RISO, as shown in Figure 60, is another approach of isolating the output of the amplifier. The ferrite's impedance characteristic versus frequency is useful to maintain the low frequency load independence of the amplifier while isolating the phase shift caused by the capacitance at high frequency. Use a ferrite with similar impedance to RISO, 20 Ω - 50 Ω, at 100 MHz and low impedance at dc. 100 Ω LOAD 5.11 Ω 1 µF 49.9 Ω Figure 62. Figure 63 is shown using two amplifiers in parallel to double the output drive current to larger capacitive loads. This technique is used when more output current is needed to charge and discharge the load faster like when driving large FET transistors. Submit Documentation Feedback 19 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 rails and are given in the specification tables. Above the Enable Threshold Voltage, the device is on. Below the Disable Threshold Voltage, the device is off. Behavior in between these threshold voltages is not specified. 715 Ω VS 178 Ω 5.11 Ω _ 24.9 Ω + −VS 715 Ω VS VS 178 Ω _ 24.9 Ω 1 nF 5.11 Ω + −VS Figure 63. Figure 64 shows a push-pull FET driver circuit typical of ultrasound applications with isolation resistors to isolate the gate capacitance from the amplifier. VS VS Figure 65 shows the total system output impedance which includes the amplifier output impedance in parallel with the feedback plus gain resistors, which cumulate to 2420 Ω. Figure 52 shows this circuit configuration for reference. 2500 ZOPD − Powerdown Output Impedance − Ω 5.11 Ω + _ Note that this power-down functionality is just that; the amplifier consumes less power in power-down mode. The power-down mode is not intended to provide a high-impedance output. In other words, the power-down functionality is not intended to allow use as a 3-state bus driver. When in power-down mode, the impedance looking back into the output of the amplifier is dominated by the feedback and gain setting resistors, but the output impedance of the device itself varies depending on the voltage applied to the outputs. VS = ±15 V and ±5 V 2000 −VS 1500 604 Ω 1000 133 Ω 604 Ω VS _ 5.11 Ω 1.21 kΩ 500 1.21 kΩ − + 50 Ω VO 1M 10 M 0 100 k 100 M 1G f − Frequency − Hz + −VS Figure 65. Power-down Output Impedance vs Frequency −VS Figure 64. PowerFET Drive Circuit SAVING POWER WITH POWER-DOWN FUNCTIONALITY AND SETTING THRESHOLD LEVELS WITH THE REFERENCE PIN The THS3096 features a power-down pin (PD) which lowers the quiescent current from 9.5 mA down to 500 µA, ideal for reducing system power. The power-down pin of the amplifier defaults to the positive supply voltage in the absence of an applied voltage, putting the amplifier in the power-on mode of operation. To turn off the amplifier in an effort to conserve power, the power-down pin can be driven towards the negative rail. The threshold voltages for power-on and power-down are relative to the supply 20 As with most current feedback amplifiers, the internal architecture places some limitations on the system when in power-down mode. Most notably is the fact that the amplifier actually turns ON if there is a ±0.7 V or greater difference between the two input nodes (V+ and V–) of the amplifier. If this difference exceeds ±0.7 V, the output of the amplifier creates an output voltage equal to approximately [(V+ – V–) – 0.7 V] × Gain. This also implies that if a voltage is applied to the output while in power-down mode, the V- node voltage is equal to VO(applied) × RG/(RF + RG). For low gain configurations and a large applied voltage at the output, the amplifier may actually turn ON due to the aforementioned behavior. The time delays associated with turning the device on Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 and off are specified as the time it takes for the amplifier to reach either 10% or 90% of the final output voltage. The time delays are in the order of microseconds because the amplifier moves in and out of the linear mode of operation in these transitions. POWER-DOWN REFERENCE PIN OPERATION In addition to the power-down pin, the THS3096 features a reference pin (REF) which allows the user to control the enable or disable power-down voltage levels applied to the PD pin. In most split-supply applications, the reference pin is connected to ground. In either case, the user needs to be aware of voltage-level thresholds that apply to the power-down pin. The tables below show examples and illustrate the relationship between the reference voltage and the power-down thresholds. In the table, the threshold levels are derived by the following equations: PD ≤ REF + 0.8 V for disable PD ≥ REF + 2.0 V for enable where the usable range at the REF pin is VS– ≤ VREF≤ (VS+ – 4 V). The recommended mode of operation is to tie the REF pin to midrail, thus setting the enable/disable thresholds to Vmidrail + 2 V and Vmidrail + 0.8 V respectively. Table 2. Power-Down Threshold Voltage Levels SUPPLY VOLTAGE (V) REFERENCE PIN VOLTAGE (V) ENABLE LEVEL (V) DISABLE LEVEL (V) ±15, ±5 0.0 2.0 0.8 ±15 2.0 4.0 2.8 ±15 –2.0 0.0 –1.2 ±5 1.0 3.0 1.8 ±5 –1.0 1.0 –0.2 30 15 17 15.8 10 5.0 7.0 5.8 Note that if the REF pin is left unterminated, it will float to the positive rail and will fall outside of the recommended operating range given above VS– ≤ VREF ≤ (VS+ – 4 V). As a result, it will no longer serve as a reliable reference for the PD pin, and the enable/disable thresholds given above will no longer apply. If the PD pin is also left unterminated, it will also float to the positive rail and the device will be enabled. If balanced, split supplies are used (±VS) and the REF and PD pins are grounded, the device will be disabled. PRINTED-CIRCUIT BOARD LAYOUT TECHNIQUES FOR OPTIMAL PERFORMANCE Achieving optimum performance with high frequency amplifier, like the THS3092/6, requires careful attention to board layout parasitic and external component types. Recommendations that optimize performance include: • Minimize parasitic capacitance to any ac ground for all of the signal I/O pins. Parasitic capacitance on the output and input pins can cause instability. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. • Minimize the distance (< 0.25”) from the power supply pins to high frequency 0.1-µF and 100-pF decoupling capacitors. At the device pins, the ground and power plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power supply connections should always be decoupled with these capacitors. Larger (6.8 µF or more) tantalum decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. • Careful selection and placement of external components preserve the high frequency performance of the THS3092/6. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Again, keep their leads and PC board trace length as short as possible. Never use wirebound type resistors in a high frequency application. Since the output pin and inverting input pins are the most sensitive to parasitic capacitance, always position the feedback and series output resistors, if any, as close as possible to the inverting input pins and output pins. Other network components, such as input termination resistors, should be placed close to the gain-setting resistors. Even with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant time constants that can degrade performance. Good axial metal-film or surface-mount resistors have approximately 0.2 pF in shunt with the resistor. For resistor values > 2.0 kΩ, this parasitic capacitance can add a pole and/or a zero that can effect circuit operation. Keep resistor values as low as possible, consistent with load driving considerations. Submit Documentation Feedback 21 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 • • Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50 mils to 100 mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and determine if isolation resistors on the outputs are necessary. Low parasitic capacitive loads (< 4 pF) may not need an RS since the THS3092/6 are nominally compensated to operate with a 2-pF parasitic load. Higher parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin). If a long trace is required, and the 6-dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50-Ω environment is not necessary onboard, and in fact, a higher impedance environment improves distortion as shown in the distortion versus load plots. With a characteristic board trace impedance based on board material and trace dimensions, a matching series resistor into the trace from the output of the THS3092/6 is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance is the parallel combination of the shunt resistor and the input impedance of the destination device: this total effective impedance should be set to match the trace impedance. If the 6-dB attenuation of a doubly terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case. This does not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there is some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. Socketing a high speed part like the THS3092/6 are not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the THS3092/6 parts directly onto the board. leadframe upon which the die is mounted [see Figure 66(a) and Figure 66(b)]. This arrangement results in the lead frame being exposed as a thermal pad on the underside of the package [see Figure 66(c)]. Because this thermal pad has direct thermal contact with the die, excellent thermal performance can be achieved by providing a good thermal path away from the thermal pad. Note that devices such as the THS3092/6 have no electrical connection between the PowerPAD and the die. The PowerPAD package allows for both assembly and thermal management in one manufacturing operation. During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be soldered to a copper area underneath the package. Through the use of thermal paths within this copper area, heat can be conducted away from the package into either a ground plane or other heat dissipating device. The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of surface mount with the, heretofore, awkward mechanical methods of heatsinking. DIE DIE End View (b) 22 Bottom View (c) Figure 66. Views of Thermal Enhanced Package Although there are many ways to properly heatsink the PowerPAD package, the following steps illustrate the recommended approach. 0.300 0.100 0.035 Pin 1 0.026 0.010 0.030 0.060 0.140 0.050 0.176 0.060 0.035 0.010 vias PowerPAD™ DESIGN CONSIDERATIONS The THS3092/6 are available in a thermally-enhanced PowerPAD family of packages. These packages are constructed using a downset Thermal Pad Side View (a) 0.080 Top View Figure 67. DDA PowerPAD PCB Etch and Via Pattern Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 PowerPAD™ LAYOUT CONSIDERATIONS 1. PCB with a top side etch pattern as shown in Figure 67. There should be etch for the leads as well as etch for the thermal pad. 2. Place 13 holes in the area of the thermal pad. These holes should be 10 mils in diameter. Keep them small so that solder wicking through the holes is not a problem during reflow. 3. Additional vias may be placed anywhere along the thermal plane outside of the thermal pad area. This helps dissipate the heat generated by the THS3092/6 IC. These additional vias may be larger than the 10-mil diameter vias directly under the thermal pad. They can be larger because they are not in the thermal pad area to be soldered so that wicking is not a problem. 4. Connect all holes to the internal ground plane. Note that the PowerPAD is electrically isolated from the silicon and all leads. Connecting the PowerPAD to any potential voltage such as VS-, is acceptable as there is no electrical connection to the silicon. 5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection methodology. Web connections have a high thermal resistance connection that is useful for slowing the heat transfer during soldering operations. This makes the soldering of vias that have plane connections easier. In this application, however, low thermal resistance is desired for the most efficient heat transfer. Therefore, the holes under the THS3092/6 PowerPAD package should make their connection to the internal ground plane with a complete connection around the entire circumference of the plated-through hole. 6. The top-side solder mask should leave the terminals of the package and the thermal pad area with its 13 holes exposed. The bottom-side solder mask should cover the 13 holes of the thermal pad area. This prevents solder from being pulled away from the thermal pad area during the reflow process. 7. Apply solder paste to the exposed thermal pad area and all of the IC terminals. 8. With these preparatory steps in place, the IC is simply placed in position and run through the solder reflow operation as any standard surface-mount component. This results in a part that is properly installed. POWER DISSIPATION AND THERMAL CONSIDERATIONS The THS3092/6 incorporates automatic thermal shutoff protection. This protection circuitry shuts down the amplifier if the junction temperature exceeds approximately 160°C. When the junction temperature reduces to approximately 140°C, the amplifier turns on again. But, for maximum performance and reliability, the designer must take care to ensure that the design does not exeed a junction temperature of 125°C. Between 125°C and 150°C, damage does not occur, but the performance of the amplifier begins to degrade and long term reliability suffers. The thermal characteristics of the device are dictated by the package and the PC board. Maximum power dissipation for a given package can be calculated using the following formula. T
TA P Dmax  max
JA where: PDmax is the maximum power dissipation in the amplifier (W). Tmax is the absolute maximum junction temperature (°C). TA is the ambient temperature (°C). θJA = θJC + θCA θJC is the thermal coefficient from the silicon junctions to the case (°C/W). θCA is the thermal coefficient from the case to ambient air (°C/W). For systems where heat dissipation is more critical, the THS3092 is offered in an 8-pin SOIC (DDA) with PowerPAD package, and the THS3096 is offered in a 14-pin TSSOP (PWP) with PowerPAD package for even better thermal performance. The thermal coefficient for the PowerPAD packages are substantially improved over the traditional SOIC. Maximum power dissipation levels are depicted in the graph for the available packages. The data for the PowerPAD packages assume a board layout that follows the PowerPAD layout guidelines referenced above and detailed in the PowerPAD application note (literature number SLMA002). The following graph also illustrates the effect of not soldering the PowerPAD to a PCB. The thermal impedance increases substantially which may cause serious heat and performance issues. Be sure to always solder the PowerPAD to the PCB for optimum performance. Submit Documentation Feedback 23 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 DESIGN TOOLS PD − Maximum Power Dissipation − W 4 ΤJ = 125°C Evaluation Fixtures, Application Support 3.5 θJA = 45.8°C/W 3 θJA = 95°C/W 2 1.5 1 0.5 θJA = 158°C/W 0 −40 −20 0 20 40 60 80 100 TA − Free-Air Temperature − °C Results are With No Air Flow and PCB Size = 3”x 3” θJA = 45.8°C/W for 8-Pin SOIC w/PowerPAD (DDA) θJA = 58.4°C/W for 8-Pin MSOP w/PowerPAD (DGN) θJA = 95°C/W for 8-Pin SOIC High−K Test PCB (D) θJA = 158°C/W for 8-Pin MSOP w/PowerPAD w/o Solder Figure 68. Maximum Power Distribution vs Ambient Temperature When determining whether or not the device satisfies the maximum power dissipation requirement, it is important to not only consider quiescent power dissipation, but also dynamic power dissipation. Often times, this is difficult to quantify because the signal pattern is inconsistent, but an estimate of the RMS power dissipation can provide visibility into a possible problem. 24 Models, and Texas Instruments is committed to providing its customers with the highest quality of applications support. To support this goal an evaluation board has been developed for the THS3092/6 operational amplifier. The board is easy to use, allowing for straightforward evaluation of the device. The evaluation board can be ordered through the Texas Instruments web site, www.ti.com, or through your local Texas Instruments sales representative. θJA = 58.4°C/W 2.5 Spice Computer simulation of circuit performance using SPICE is often useful when analyzing the performance of analog circuits and systems. This is particularly true for video and RF-amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. A SPICE model for the THS3092/6 is available through either the Texas Instruments web site (www.ti.com) or as one model on a disk from the Texas Instruments Product Information Center (1–800–548–6132). The PIC is also available for design assistance and detailed product information at this number. These models do a good job of predicting small-signal ac and transient performance under a wide variety of operating conditions. They are not intended to model the distortion characteristics of the amplifier, nor do they attempt to distinguish between the package types in their small-signal ac performance. Detailed information about what is and is not modeled is contained in the model file itself. Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 THS3092 EVM GND J8 -5V J7 5V J9 FB1 FB2 C1 C2 5V -5 V C3 + + C4 TP1 TP2 R6 J2 R1 5V U1:A R2 J3 2 3 R3 8 1 J1 R5 Figure 70. THS3092 EVM Board Layout (Top Layer) R4 4 -5 V J4 R7 R8 R9 6 5 7 R10 U1:B J5 J6 R11 R12 Figure 69. THS3092 EVM Schematic Figure 71. THS3092 EVM Board Layout (Ground Plane) Submit Documentation Feedback 25 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 Figure 72. THS3092 EVM Board Layout (Power Plane) Figure 73. THS3092 EVM Board Layout (Bottom Layer) Table 3. THS3092 EVM Bill of Materials THS3092DGN EVM (1) 26 ITEM DESCRIPTION SMD SIZE REFERENCE DESIGNATOR PCB QTY MANUFACTURER'S PART NUMBER (1) 1 Bead, Ferrite, 3 A, 80 Ω 1206 FB1, FB2 2 (Steward) HI1206N800R-00 2 Cap. 22 µF, Tanatalum, 35 V, 10% D C1, C2 2 (AVX) TAJD685K035R 3 Cap. 0.1 µF, Ceramic, X7R, 16 V 0805 C3, C4 2 (AVX) 08055C104KAT2A 4 Resistor, 178 Ω, 1/8 W, 1% 0805 R1, R8 2 (KOA) RK73H2ALTD1780F 5 Resistor, 715 Ω, 1/8 W, 1% 0805 R6, R7 2 (KOA) RK73H2ALTD7150F 6 Open 1206 R4, R12 2 7 Resistor, 0 Ω, 1/4 W, 1% 1206 R2, R9 2 (KOA) RK73Z2BLTD 8 Resistor, 49.9 Ω, 1/4 W, 1% 1206 R1, R5, R10, R11 4 (KOA) RK73H2BLTD49R9F 9 Connector, edge, SMA PCB jack J1, J2, J3, J4, J5, J6 6 (Johnson) 142-0701-801 10 Jack, banana, 0.25" dia. hole J7, J8, J9 3 (SPC) 813 11 Test point, black TP1, TP2 2 (Keystone) 5001 12 IC, THS3092 U1 1 (TI) THS3092DDA 13 Board, printed-circuit 1 (TI) EDGE # 6446250 Rev. A The manufacturer's part numbers were used for test purposes only. Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 THS3096 EVM GND J8 5V J7 -5 V J9 FB1 5V FB2 C5 + C1 C3 -5 V C2 C4 + TP1 TP2 R4 J1 R3 5V U1:A R1 J2 2 3 R2 14 4 R5 1 15 J3 Figure 75. THS3096 EVM Board Layout (Top Layer) R6 -5 V J4 R9 R8 R7 12 11 J5 R12 U1:B 13 9 6 R11 J6 R10 R13 J10 JP1 R14 5V R15 Figure 74. THS3096 EVM Schematic Figure 76. THS3096 EVM Board Layout (Ground Plane) Submit Documentation Feedback 27 THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 Figure 77. THS3096 EVM Board Layout (Power Plane) Figure 78. THS3096 EVM Board Layout (Bottom Layer) Table 4. THS3096 EVM Bill of Materials THS3096PWP EVM DESCRIPTION SMD SIZE REFERENCE DESIGNATOR PCB QTY MANUFACTURER'S PART NUMBER 1 Bead, Ferrite, 3 A, 80 Ω 1206 FB1, FB2 2 (Steward) HI1206N800R-00 2 Cap. 22 µF, Tanatalum, 25 V, 10% D C1, C2 2 (AVX) TAJD226K025R 3 Cap. 0.1 µF, Ceramic, X7R, 50 V 0805 C3, C4 2 (AVX) 08055C104KAT2A 4 Cap. 0.1 µF, Ceramic, X7R, 50 V 1206 C5 1 (AVX) 12065C104KAT2A 5 Resistor, 100 Ω, 1/8W, 1% 0805 R13 1 (KOA) RK73H2ALTD1000F 6 Resistor, 178 Ω, 1/8 W, 1% 0805 R3, R8 2 (KOA) RK73H2ALTD1780F 7 Resistor, 715 Ω, 1/8 W, 1% 0805 R4, R9 2 (KOA) RK73H2ALTD7150F 8 Resistor, 20 kΩ, 1/8 W, 1% 0805 R14, R15 2 (KOA) RK73H2ALTD2002F 9 Open 1206 R6, R10 2 10 Resistor, 0 Ω, 1/4 W, 1% 1206 R1, R7 2 (KOA) RK73Z2BLTD 11 Resistor, 49.9 Ω, 1/4 W, 1% 1206 R2, R5, R11, R12 4 (KOA) RK73H2BLTD49R9F 12 Header, 0.1" ctrs, 0.025" sq. pins 2 pos. JP1 1 (Sullins) PZC36SAAN 13 Shunts JP1 1 (Sullins) SSC02SYAN 14 Connector, SMA PCB jack J1, J2, J3, J4, J5, J6 6 (Amphenol) 901-144-8RFX 15 Jack, banana, 0.25" dia. hole J7, J8, J9 3 (SPC) 813 16 Test point, red J10 1 (Keystone) 5000 17 Test point, black TP1, TP2 2 (Keystone) 5001 18 IC, THS3096 U1 1 (TI) THS3096PWP 19 Board, printed-circuit 1 (TI) EDGE # 6454586 Rev. A ITEM 28 Submit Documentation Feedback THS3092 THS3096 www.ti.com SLOS428B – DECEMBER 2003 – REVISED FEBRUARY 2006 ADDITIONAL REFERENCE MATERIAL • PowerPAD Made Easy, application brief (SLMA004) • PowerPAD Thermally Enhanced Package, technical brief (SLMA002) • Voltage Feedback vs Current Feedback Amplifiers, (SLVA051) • Current Feedback Analysis and Compensation (SLOA021) • Current Feedback Amplifiers: Review, Stability, and Application (SBOA081) • Effect of Parasitic Capacitance in Op Amp Circuits (SLOA013) • Expanding the Usability of Current-Feedback Amplifiers, by Randy Stephens, 3Q 2003 Analog Applications Journal www.ti.com/sc/analogapps). • Active Output Impedance for ADSL Line Drivers (SLOA100) Submit Documentation Feedback 29 PACKAGE OPTION ADDENDUM www.ti.com 13-Aug-2021 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) THS3092D ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 3092 THS3092DDA ACTIVE SO PowerPAD DDA 8 75 RoHS & Green SN Level-1-260C-UNLIM -40 to 85 3092 THS3092DDAG3 ACTIVE SO PowerPAD DDA 8 75 RoHS & Green SN Level-1-260C-UNLIM -40 to 85 3092 THS3092DDAR ACTIVE SO PowerPAD DDA 8 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 85 3092 THS3096D ACTIVE SOIC D 14 50 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 THS3096 THS3096PWP ACTIVE HTSSOP PWP 14 90 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 THS3096 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of