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THS3091, THS3095
SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
THS309x High-voltage, Low-distortion, Current-feedback Operational Amplifiers
1 Features
3 Description
•
The THS3091 and THS3095 are high-voltage, lowdistortion, high-speed, current-feedback amplifier
designed to operate over a wide supply range of ±5 V
to ±15 V for applications requiring large, linear output
signals such as Pin, Power FET, and VDSL line
drivers.
1
•
•
•
•
•
•
Low Distortion
– 77-dBc HD2 at 10 MHz, RL = 1 kΩ
– 69-dBc HD3 at 10 MHz, RL = 1 kΩ
Low Noise
– 14-pA/√Hz Noninverting Current Noise
– 17-pA/√Hz Inverting Current Noise
– 2-nV/√Hz Voltage Noise
High Slew Rate: 7300 V/μs (G = 5, VO = 20 VPP)
Wide Bandwidth: 210 MHz (G = 2, RL = 100 Ω)
High Output Current Drive: ±250 mA
Wide Supply Range: ±5 V to ±15 V
Power-Down Feature: THS3095 Only
2 Applications
•
•
•
•
High-Voltage Arbitrary Waveform Generators
Power FET Drivers
Pin Drivers
VDSL Line Drivers
The THS3095 features a power-down pin (PD) that
puts the amplifier in low power standby mode, and
lowers the quiescent current from 9.5 mA to 500 μA.
The wide supply range combined with total harmonic
distortion as low as –69 dBc at 10 MHz, in addition,
to the high slew rate of 7300 V/μs makes the
THS309x ideally suited for high-voltage arbitrary
waveform driver applications. Moreover, having the
ability to handle large voltage swings driving into
high-resistance and high-capacitance loads while
maintaining good settling time performance makes
the devices ideal for Pin driver and Power FET driver
applications.
The THS3091 and THS3095 are offered in an 8-pin
SOIC (D), and the 8-pin SOIC (DDA) packages with
PowerPAD™.
Device Information(1)
PART NUMBER
PACKAGE
THS309x
BODY SIZE (NOM)
SOIC(8)
4.90 mm × 3.91 mm
SO PowerPAD(8)
4.89 mm × 3.90 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
−
+
THS3091
VOUT
IOUT1
DAC5686
IOUT2
−
+
−
+
THS3091
THS4271
Total Harmonic Distortion vs Frequency
−20
Total Harmonic Distortion − dBc
Typical Arbitrary Waveform Generator Output
Drive Circuit
−30
−40
G = 5,
RF = 1 kW,
RL = 100 W,
VS = ±15 V
VO = 20 VPP
VO = 10 VPP
−50
−60
VO = 5 VPP
−70
−80
VO = 2 VPP
−
+
−90
100 k
THS3091
1M
10 M
100 M
f − Frequency − Hz
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. UNLESS OTHERWISE NOTED, this document contains PRODUCTION
DATA.
THS3091, THS3095
SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
6.9
6.10
7
1
1
1
2
3
4
Absolute Maximum Ratings ...................................... 4
ESD Ratings.............................................................. 4
Recommended Operating Conditions....................... 4
Thermal Information .................................................. 4
Electrical Characteristics THS3091 .......................... 5
Electrical Characteristics THS3095 .......................... 8
Dissipation Ratings Table ....................................... 11
Typical Characteristics ............................................ 12
Typical Characteristics (±15 V) ............................... 14
Typical Characteristics (±5 V) ............................... 21
Detailed Description ............................................ 25
7.1 Overview ................................................................. 25
7.2 Feature Description................................................. 25
7.3 Device Functional Modes........................................ 26
8
Application and Implementation ........................ 29
8.1 Application Information............................................ 29
8.2 Typical Application ................................................. 32
9 Power Supply Recommendations...................... 37
10 Layout................................................................... 37
10.1 Layout Guidelines ................................................. 37
10.2 Layout Example ................................................... 38
10.3 PowerPAD Design Considerations ....................... 41
11 Device and Documentation Support ................. 43
11.1
11.2
11.3
11.4
11.5
11.6
11.7
Device Support......................................................
Documentation Support ........................................
Related Links ........................................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
43
43
43
43
44
44
44
12 Mechanical, Packaging, and Orderable
Information ........................................................... 44
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision G (February, 2007) to Revision H
•
Page
Added Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and
Implementation section, Power Supply Recommendations section, Layout section, Device and Documentation
Support section, and Mechanical, Packaging, and Orderable Information section................................................................ 1
Changes from Revision F (February, 2007) to Revision G
Page
•
Changed common-mode rejection ratio specifications from 78 dB (typ) to 69 dB (typ); from 68 dB at +25°C to 62
dB; from 65 dB at (0°C to +70°C) and (–40°C to +85°C) to 59 dB. ....................................................................................... 6
•
Corrected load resistor value for output current specification (sourcing and sinking) from RL = 40 Ω to RL = 10 Ω ........... 10
•
Changed output current (sourcing) specifications from 200 mA (typ) to 180 mA (typ); from 160 mA at +25°C to 140
mA; from 140 mA at (0°C to +70°C) and (–40°C to +85°C) to 120 mA ............................................................................... 10
•
Corrected output current (sinking) specifications from 180 mA (typ) to –160 mA (typ); from 150 mA at +25°C to –140
mA; from 125 mA at (0°C to +70°C) and (–40°C to +85°C) to –120 mA ............................................................................. 10
2
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Copyright © 2003–2015, Texas Instruments Incorporated
Product Folder Links: THS3091 THS3095
THS3091, THS3095
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SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
5 Pin Configuration and Functions
D, DDA Package (THS3091)
8-Pin SOIC With PowerPAD
Top View
NC
VIN−
VIN+
VS−
1
8
NC
2
7
3
6
4
5
VS+
VOUT
NC
D, DDA Package (THS3095)
8-Pin SOIC With PowerPAD
Top View
REF
VIN−
VIN+
VS−
1
8
2
7
3
6
4
5
PD
VS+
VOUT
NC
Pin Functions (1)
PIN
NAME
SOIC
I/O
SOT-23
DESCRIPTION
THS3091
1
NC
5
—
—
No connection
8
Vin–
2
4
I
Inverting input
Vin+
3
3
I
Noninverting input
–Vs
4
2
POW
Vout
6
1
O
+Vs
7
5
POW
NC
5
—
—
PD
8
—
I
Amplifier power down, LOW - Amplifier disabled, HIGH (default) - Amplifier
enabled
REF
1
—
I
Voltage reference input to set PD threshold level
Vin–
2
4
I
Inverting input
Vin+
3
3
I
Noninverting input
Vout
6
1
O
Output of amplifier
–Vs
4
2
POW
Negative power supply
+Vs
7
5
POW
Positive power supply
Negative power supply
Output of amplifier
Positive power supply
THS3095
(1)
No connection
NC – No internal connection
Copyright © 2003–2015, Texas Instruments Incorporated
Product Folder Links: THS3091 THS3095
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3
THS3091, THS3095
SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
www.ti.com
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
VS- to VS+
Supply voltage
VI
Input voltage
VID
Differential input voltage
IO
Output current
MAX
UNIT
33
V
±VS
Continuous power dissipation
4
±V
350
mA
See ESD Ratings
TJ
Maximum junction temperature
150
°C
TJ (2)
Maximum junction temperature, continuous operation, long-term reliability
125
°C
Tstg
Storage temperature
150
°C
(1)
(2)
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
The maximum junction temperature for continuous operation is limited by package constraints. Operation above this temperature may
result in reduced reliability and/or lifetime of the device.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±1500
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
Supply voltage
TA
MIN
NOM
MAX
Dual supply
±5
±15
±16
Single supply
10
30
32
Operating free-air temperature
–40
85
UNIT
V
°C
6.4 Thermal Information
THS309x
THERMAL METRIC (1)
D (SOIC)
DDA (SO
PowerPAD) (2)
UNIT
8 PINS
8 PINS
RθJA
Junction-to-ambient thermal resistance
113.5
51.8
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
57.7
58.3
°C/W
RθJB
Junction-to-board thermal resistance
54.2
32.3
°C/W
ψJT
Junction-to-top characterization parameter
11.5
12.2
°C/W
ψJB
Junction-to-board characterization parameter
53.7
32.2
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
n/a
7.8
°C/W
(1)
(2)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
The THS3091 and THS3095 may incorporate a PowerPAD on the underside of the chip. This acts as a heatsink and must be connected
to a thermally dissipating plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction temperature
which could permanently damage the device. See TI Technical Brief SLMA002 for more information about utilizing the PowerPAD
thermally enhanced package.
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Copyright © 2003–2015, Texas Instruments Incorporated
Product Folder Links: THS3091 THS3095
THS3091, THS3095
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SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
6.5 Electrical Characteristics THS3091
VS = ±15 V, RF = 1.21 kΩ, RL = 100 Ω, and G = 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
AC PERFORMANCE
G = 1, RF = 1.78 kΩ, VO = 200 mVPP
TA = 25°C
235
G = 2, RF = 1.21 kΩ, VO = 200 mVPP
TA = 25°C
210
G = 5, RF = 1 kΩ, VO = 200 mVPP
TA = 25°C
190
G = 10, RF = 866 Ω, VO = 200 mVPP
TA = 25°C
180
0.1-dB Bandwidth flatness
G = 2, RF = 1.21 kΩ, VO = 200 mVPP
TA = 25°C
95
Large-signal bandwidth
G = 5, RF = 1 kΩ , VO = 4 VPP
TA = 25°C
135
G = 2, VO = 10-V step, RF = 1.21 kΩ
TA = 25°C
5000
G = 5, VO = 20-V step, RF = 1 kΩ
TA = 25°C
7300
Rise and fall time
G = 2, VO = 5-VPP, RF = 1.21 kΩ
TA = 25°C
5
Settling time to 0.1%
G = –2, VO = 2 VPP step
TA = 25°C
42
Settling time to 0.01%
G = –2, VO = 2 VPP step
TA = 25°C
72
Small-signal bandwidth, –3 dB
Slew rate (25% to 75% level)
MHz
V/μs
ns
ns
HARMONIC DISTORTION
2nd Harmonic distortion
G = 2, RF = 1.21 kΩ,
VO = 2 VPP, f = 10 MHz
3rd Harmonic distortion
RL = 100Ω
TA = 25°C
66
RL = 1 kΩ
TA = 25°C
77
RL = 100 Ω
TA = 25°C
74
RL = 1 kΩ
dBc
TA = 25°C
69
Input voltage noise
f > 10 kHz
TA = 25°C
2
nV / √Hz
Noninverting input current
noise
f > 10 kHz
TA = 25°C
14
pA / √Hz
Inverting input current noise
f > 10 kHz
pA / √Hz
Differential gain
G = 2, RL = 150 Ω, RF = 1.21 kΩ
Differential phase
TA = 25°C
17
NTSC
TA = 25°C
0.013%
PAL
TA = 25°C
0.011%
NTSC
TA = 25°C
0.020°
PAL
TA = 25°C
0.026°
DC PERFORMANCE
TA = 25°C
Transimpedance
VO = ±7.5 V, Gain = 1
350
TA = 0°C to 70°C
300
TA = –40°C to 85°C
300
TA = 25°C
Input offset voltage
VCM = 0 V
850
TA = 25°C
kΩ
0.9
TA= 25°C
3
TA = 0°C to 70°C
4
TA = –40°C to 85°C
Average offset voltage drift
VCM = 0 V
Noninverting input bias current
VCM = 0 V
4
TA = 0°C to 70°C
±10
TA = –40°C to 85°C
±10
TA = 25°C
Inverting input bias current
VCM = 0 V
VCM = 0 V
4
15
TA = 0°C to 70°C
20
VCM = 0 V
±20
TA = –40°C to 85°C
±20
TA = 25°C
3.5
nA/°C
TA= 25°C
15
TA = 0°C to 70°C
20
±20
TA = –40°C to 85°C
±20
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μA
20
TA = 0°C to 70°C
Copyright © 2003–2015, Texas Instruments Incorporated
μA
20
TA = 0°C to 70°C
–40°C to 85°C
Average bias current drift
μV/°C
TA= 25°C
TA = –40°C to 85°C
Average bias current drift
mV
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nA/°C
5
THS3091, THS3095
SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
www.ti.com
Electrical Characteristics THS3091 (continued)
VS = ±15 V, RF = 1.21 kΩ, RL = 100 Ω, and G = 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TA = 25°C
Input offset current
VCM = 0 V
TYP
TA= 25°C
10
TA = 0°C to 70°C
15
TA = –40°C to 85°C
Average offset current drift
VCM = 0 V
MAX
UNIT
1.7
μA
15
TA = 0°C to 70°C
±20
TA = –40°C to 85°C
±20
nA/°C
INPUT CHARACTERISTICS
TA = 25°C
±13.6
TA= 25°C
Common-mode input range
±13.3
TA = 0°C to 70°C
±13
TA = –40°C to 85°C
±13
TA = 25°C
Common-mode rejection ratio
VCM = ±10 V
V
69
TA= 25°C
62
TA = 0°C to 70°C
59
TA = –40°C to 85°C
59
dB
Noninverting input resistance
TA = 25°C
1.3
MΩ
Noninverting input capacitance
TA = 25°C
0.1
pF
Inverting input resistance
TA = 25°C
30
Ω
Inverting input capacitance
TA = 25°C
1.4
pF
6
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SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
Electrical Characteristics THS3091 (continued)
VS = ±15 V, RF = 1.21 kΩ, RL = 100 Ω, and G = 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OUTPUT CHARACTERISTICS
TA = 25°C
RL = 1 kΩ
Output voltage swing
±13.2
TA = 25°C
±12.8
TA = 0°C to 70°C
±12.5
TA = –40°C to 85°C
±12.5
TA = 25°C
RL = 100 Ω
TA= 25°C
±12.1
TA = 0°C to 70°C
±11.8
TA = –40°C to 85°C
±11.8
TA= 25°C
RL = 40 Ω
Output current (sourcing)
280
TA = 25°C
225
TA = 0°C to 70°C
200
TA = –40°C to 85°C
200
TA = 25°C
Output current (sinking)
RL = 40 Ω
Output impedance
f = 1 MHz, Closed loop
V
±12.5
mA
250
TA= 25°C
200
TA = 0°C to 70°C
175
TA = –40°C to 85°C
175
mA
TA = 25°C
0.06
TA = 25°C
±15
Ω
POWER SUPPLY
Specified operating voltage
TA= 25°C
±16
TA = 0°C to 70°C
±16
TA = –40°C to 85°C
±16
TA = 25°C
9.5
TA= 25°C
Maximum quiescent current
10.5
TA = 0°C to 70°C
11
TA = –40°C to 85°C
TA= 25°C
9.5
8.5
TA = 0°C to 70°C
8
TA = –40°C to 85°C
8
TA = 25°C
Power supply rejection
(+PSRR)
VS+ = 15.5 V to 14.5 V, VS– = 15 V
VS+ = 15 V, VS– = –15.5 V to –14.5 V
mA
75
TA= 25°C
70
TA = 0°C to 70°C
65
TA = –40°C to 85°C
65
TA = 25°C
Power supply rejection
(–PSRR)
mA
11
TA = 25°C
Minimum quiescent current
V
dB
73
TA= 25°C
68
TA = 0°C to 70°C
65
TA = –40°C to 85°C
65
dB
POWER-DOWN CHARACTERISTICS (THS3091 ONLY)
TA = 25°C
VS+ –4
TA = 25°C
VS–
Enable
TA = 25°C
PD ≥
REF +2
Disable
TA = 25°C
PD ≤
REF +.8
REF voltage range (1)
Power-down voltage level
(1)
(1)
V
V
For detailed information on the behavior of the power-down circuit, see the power-down functionality and power-down reference sections
in the Application Information section of this data sheet.
Copyright © 2003–2015, Texas Instruments Incorporated
Product Folder Links: THS3091 THS3095
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www.ti.com
Electrical Characteristics THS3091 (continued)
VS = ±15 V, RF = 1.21 kΩ, RL = 100 Ω, and G = 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TA = 25°C
Power-down quiescent current
PD = 0V
TYP
TA= 25°C
700
TA = 0°C to 70°C
800
TA = –40°C to 85°C
VPD quiescent current
μA
11
TA= 25°C
15
TA = 0°C to 70°C
20
TA = –40°C to 85°C
20
TA = 25°C
VPD = 3.3 V, REF = 0 V
UNIT
800
TA = 25°C
VPD = 0 V, REF = 0 V,
MAX
500
11
TA= 25°C
15
TA = 0°C to 70°C
20
TA = –40°C to 85°C
μA
20
Turnon time delay
90% of final value
TA = 25°C
60
Turnoff time delay
10% of final value
TA = 25°C
150
μs
6.6 Electrical Characteristics THS3095
VS = ±5 V, RF = 1.15 kΩ, RL = 100 Ω, and G = 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
AC PERFORMANCE
G = 1, RF = 1.78 kΩ, VO = 200 mVPP
TA= 25°C
190
G = 2, RF = 1.15 kΩ, VO = 200 mVPP
TA= 25°C
180
G = 5, RF = 1 kΩ, VO = 200 mVPP
TA= 25°C
160
G = 10, RF = 866 Ω, VO = 200 mVPP
TA= 25°C
150
0.1-dB Bandwidth flatness
G = 2, RF = 1.15 kΩ, VO = 200 mVPP
TA= 25°C
65
Large-signal bandwidth
G = 2, RF = 1.15 kΩ , VO = 4 VPP
TA= 25°C
160
G = 2, VO= 5-V step, RF = 1.21 kΩ
TA= 25°C
1400
G = 5, VO= 5-V step, RF = 1 kΩ
TA= 25°C
1900
Rise and fall time
G = 2, VO = 5-V step, RF = 1.21 kΩ
TA= 25°C
5
Settling time to 0.1%
G = –2, VO = 2 VPP step
TA= 25°C
35
Settling time to 0.01%
G = –2, VO = 2 VPP step
TA= 25°C
73
Small-signal bandwidth, –3 dB
Slew rate (25% to 75% level)
MHz
V/μs
ns
ns
HARMONIC DISTORTION
2nd Harmonic distortion
G = 2, RF = 1.15 kΩ,
VO = 2 VPP, f = 10 MHz
3rd Harmonic distortion
RL = 100 Ω
TA = 25°C
77
RL = 1 kΩ
TA = 25°C
73
RL = 100 Ω
TA = 25°C
70
RL = 1 kΩ
TA = 25°C
68
dBc
Input voltage noise
f > 10 kHz
TA = 25°C
2
nV / √Hz
Noninverting input current noise
f > 10 kHz
TA = 25°C
14
pA / √Hz
Inverting input current noise
f > 10 kHz
TA = 25°C
17
pA / √Hz
NTSC
TA = 25°C
0.027%
PAL
TA = 25°C
0.025%
NTSC
TA = 25°C
0.04°
PAL
TA = 25°C
0.05°
Differential gain
G = 2, RL = 150 Ω,
RF = 1.15 kΩ
Differential phase
DC PERFORMANCE
TA = 25°C
Transimpedance
8
VO = ±2.5 V, Gain = 1
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700
TA= 25°C
250
TA= 0°C to 70°C
200
TA= –40°C to 85°C
200
kΩ
Copyright © 2003–2015, Texas Instruments Incorporated
Product Folder Links: THS3091 THS3095
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SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
Electrical Characteristics THS3095 (continued)
VS = ±5 V, RF = 1.15 kΩ, RL = 100 Ω, and G = 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TA = 25°C
Input offset voltage
Average offset voltage drift
VCM = 0 V
VCM = 0 V
Average bias current drift
VCM = 0 V
VCM = 0 V
TA= 25°C
2
TA= 0°C to
70°C
3
TA= –40°C to
85°C
3
TA= 0°C to
70°C
±10
TA= –40°C to
85°C
±10
Average bias current drift
VCM = 0 V
VCM = 0 V
Average offset current drift
VCM = 0 V
VCM = 0 V
mV
2
TA= 25°C
15
TA= 0°C to
70°C
20
TA= –40°C to
85°C
20
TA= 0°C to
70°C
±20
TA= –40°C to
85°C
±20
μA
nA/°C
5
TA= 25°C
15
TA= 0°C to
70°C
20
TA= –40°C to
85°C
20
TA= 0°C to
70°C
±20
TA= –40°C to
85°C
±20
μA
nA/°C
TA = 25°C
Input offset current
UNIT
μV/°C
TA = 25°C
Inverting input bias current
MAX
0.3
TA = 25°C
Noninverting input bias current
TYP
1
TA= 25°C
10
TA= 0°C to
70°C
15
TA= –40°C to
85°C
15
TA= 0°C to
70°C
±20
TA= –40°C to
85°C
±20
μA
nA/°C
INPUT CHARACTERISTICS
TA = 25°C
TA= 25°C
Common-mode input range
±3.6
±3.3
TA= 0°C to
70°C
±3
TA= –40°C to
85°C
±3
TA = 25°C
Common-mode rejection ratio
VCM = ±2.0 V, VO = 0 V
V
66
TA= 25°C
60
TA= 0°C to
70°C
57
TA= –40°C to
85°C
57
dB
Noninverting input resistance
TA = 25°C
1.1
MΩ
Noninverting input capacitance
TA = 25°C
1.2
pF
Inverting input resistance
TA = 25°C
32
Ω
Inverting input capacitance
TA = 25°C
1.5
pF
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Electrical Characteristics THS3095 (continued)
VS = ±5 V, RF = 1.15 kΩ, RL = 100 Ω, and G = 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OUTPUT CHARACTERISTICS
TA = 25°C
RL = 1 kΩ
Output voltage swing
±3.4
TA= 25°C
±3.1
TA= 0°C to
70°C
±2.8
TA= –40°C to
85°C
±2.8
TA = 25°C
RL = 100 Ω
TA= 25°C
±2.7
TA= 0°C to
70°C
±2.5
TA= –40°C to
85°C
±2.5
TA = 25°C
Output current (sourcing)
RL = 10 Ω
180
TA= 25°C
140
TA= 0°C to
70°C
120
TA= –40°C to
85°C
120
TA = 25°C
Output current (sinking)
Output impedance
RL = 10 Ω
f = 1 MHz, Closed loop
V
±3.1
mA
–160
TA= 25°C
–140
TA= 0°C to
70°C
–120
TA= –40°C to
85°C
–120
TA = 25°C
mA
Ω
0.09
POWER SUPPLY
TA = 25°C
Specified operating voltage
±5
TA= 25°C
±4.5
TA= 0°C to 70°C
±4.5
TA= –40°C to 85°C
±4.5
TA = 25°C
Maximum quiescent current
8.2
TA= 25°C
9
TA= 0°C to 70°C
9.5
TA= –40°C to 85°C
8.2
TA= 25°C
7
TA= 0°C to 70°C
6.5
TA = 25°C
VS+ = 5.5 V to 4.5 V, VS– = 5 V
73
TA= 25°C
68
TA= 0°C to
70°C
63
TA= –40°C to
85°C
63
TA = 25°C
Power supply rejection (–PSRR)
10
VS+ = 5 V, VS– = –4.5 V to –5.5 V
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mA
6.5
TA= –40°C to 85°C
Power supply rejection (+PSRR)
mA
9.5
TA = 25°C
Minimum quiescent current
V
dB
71
TA= 25°C
65
TA= 0°C to
70°C
60
TA= –40°C to
85°C
60
dB
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Electrical Characteristics THS3095 (continued)
VS = ±5 V, RF = 1.15 kΩ, RL = 100 Ω, and G = 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
POWER-DOWN CHARACTERISTICS (THS3095 ONLY)
REF voltage range (1)
Power-down voltage level (1)
Power-down quiescent current
TA = 25°C
VS+ –4
TA = 25°C
VS–
Enable
TA = 25°C
PD ≥ REF 2
Disable
TA = 25°C
PD ≤ REF 0.8
TA = 25°C
300
PD = 0V
VPD quiescent current
500
TA= 0°C to
70°C
600
TA= –40°C to
85°C
600
15
TA= 0°C to
70°C
20
TA–40°C to
85°C
20
μA
11
TA= 25°C
15
TA= 0°C to
70°C
20
TA= –40°C to
85°C
20
Turnon time delay
90% of final value
TA = 25°C
60
Turnoff time delay
10% of final value
TA = 25°C
150
(1)
μA
11
TA= 25°C
TA = 25°C
VPD = 3.3 V, REF = 0 V
V
TA= 25°C
TA = 25°C
VPD = 0 V, REF = 0 V,
V
μs
For detailed information on the behavior of the power-down circuit, see the power-down functionality and power-down reference sections
in the Application Information section of this data sheet.
6.7 Dissipation Ratings Table
(1)
(2)
POWER RATING
TJ = 125°C
(2)
PACKAGE
θJC (°C/W)
θJA (°C/W) (1)
TA = 25°C
TA = 85°C
D-8
38.3
97.5
1.02 W
410 mW
DDA-8
9.2
45.8
2.18 W
873 mW
This data was taken using the JEDEC standard High-K test PCB.
Power rating is determined with a junction temperature of 125°C. This is the point where distortion starts to substantially increase.
Thermal management of the final PCB should strive to keep the junction temperature at or below 125°C for best performance and longterm reliability.
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6.8 Typical Characteristics
Table 1. Table Of Graphs
±15-V GRAPHS
FIGURE
Noninverting small-signal frequency response
Figure 1, Figure 2
Inverting small-signal frequency response
Figure 3
0.1-dB gain flatness frequency response
Figure 4
Noninverting large-signal frequency response
Figure 5
Inverting large-signal frequency response
Figure 6
Capacitive load frequency response
Figure 7
Recommended RISO
vs Capacitive load
Figure 8
2nd Harmonic distortion
vs Frequency
Figure 9,
Figure 11
3rd Harmonic distortion
vs Frequency
Figure 10,
Figure 12
2nd Harmonic distortion
vs Frequency
Figure 13
3rd Harmonic distortion
vs Frequency
Figure 14
Harmonic distortion
vs Output voltage swing
Figure 15,
Figure 16
Slew rate
vs Output voltage step
Figure 17,
Figure 18,
Figure 19
Noise
vs Frequency
Figure 20
Figure 21,
Figure 22
Settling time
Quiescent current
vs Supply voltage
Figure 23
Quiescent current
vs Frequency
Figure 24
Output voltage
vs Load resistance
Figure 25
Input bias and offset current
vs Case temperature
Figure 26
Input offset voltage
vs Case temperature
Figure 27
Transimpedance
vs Frequency
Figure 28
Rejection ratio
vs Frequency
Figure 29
Noninverting small-signal transient response
Figure 30
Inverting large-signal transient response
Figure 31,
Figure 32
Overdrive recovery time
Figure 33
Differential gain
vs Number of loads
Figure 34
Differential phase
vs Number of loads
Figure 35
Closed-Loop output impedance
vs Frequency
Figure 36
Power-down quiescent current
vs Supply voltage
Figure 37
Turnon and turnoff time delay
12
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Figure 38
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Table 2. Table Of Graphs (Continued)
±5-V GRAPHS
FIGURE
Noninverting small-signal frequency response
Figure 39
Inverting small-signal frequency response
Figure 40
0.1-dB gain flatness frequency response
Figure 41
Noninverting large-signal frequency response
Figure 42
Inverting large-signal frequency response
Figure 43
Settling time
Figure 44
2nd Harmonic distortion
vs Frequency
Figure 45,
Figure 47
3rd Harmonic distortion
vs Frequency
Figure 46,
Figure 48
Harmonic distortion
vs Output voltage swing
Figure 49,
Figure 50
Slew rate
vs Output voltage step
Figure 51,
Figure 52,
Figure 53
Quiescent current
vs Frequency
Figure 54
Output voltage
vs Load resistance
Figure 55
Input bias and offset current
vs Case temperature
Figure 56
Overdrive recovery time
Rejection ratio
Figure 57
vs Frequency
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Figure 58
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6.9 Typical Characteristics (±15 V)
9
24
22
RF = 750 Ω
G = 10, RF = 866 Ω
20
18
7
Noninverting Gain − dB
Noninverting Gain − dB
8
6
RRFF==1.21
1.21kkΩ
Ω
5
4
RF = 1.5 kΩ
3
Gain = 2,
RL =100 Ω,
VO = 200 mVPP,
VS = ±15 V
2
1
0
1M
10 M
100 M
1G
16
14
G = 5, RF = 1 kΩ
12
10
8
6
4
RL = 100 Ω,
VO = 200 mVPP,
VS = ±15 V
G = 2, RF = 1.21 kΩ
2
0
−2
−4
G = 1, RF = 1.78 kΩ
10 M
1M
f − Frequency − Hz
6.3
G = −10, RF = 866 Ω
Gain = 2,
RF = 1.21 kΩ,
RL = 100 Ω,
VO = 200 mVPP,
VS = ±15 V
6.2
G = −5, RF = 909 Ω
12
10
8
6
4
2
0
−2
−4
RL = 100 Ω,
VO = 200 mVPP,
VS = ±15 V
G = −2, RF = 1 kΩ
6.1
6
5.9
5.8
G = −1, RF = 1.05 kΩ
5.7
1M
10 M
100 M
1G
100 k
f − Frequency − Hz
Figure 3. Inverting Small-Signal Frequency Response
16
G = 5, RF = 1 kΩ
G = −5, RF = 909 Ω
12
12
Inverting Gain − dB
Noninverting Gain − dB
1G
14
14
10
8
G = 2, RF = 1.21 kΩ
6
10
8
G = −2, RF = 1 kΩ
6
4
2
4
−2
−4
1M
10 M
VO = 4 VPP,
RL = 100 Ω,
VS = ±15 V
0
VO = 4 VPP,
RL = 100 Ω,
VS = ±15 V
0
100 M
1G
1M
Figure 5. Noninverting Large-Signal Frequency Response
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10 M
100 M
1G
f − Frequency − Hz
f − Frequency − Hz
14
1M
10 M
100 M
f - Frequency - Hz
Figure 4. 0.1-db Gain Flatness Frequency Response
16
2
1G
Figure 2. Noninverting Small-Signal Frequency Response
Noninverting Gain - dB
Inverting Gain − dB
Figure 1. Noninverting Small-Signal Frequency Response
24
22
20
18
16
14
100 M
f − Frequency − Hz
Figure 6. Inverting Large-Signal Frequency Response
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Typical Characteristics (±15 V) (continued)
16
R(ISO) = 30.9 Ω
CL = 22 pF
10
8
R(ISO) = 22.1 Ω
CL = 47 pF
6
R(ISO) = 15.8 Ω
CL = 100 pF
4
2
0
Gain = 5,
RL = 100 Ω,
VS =±15 V
10 M
25
20
15
10
1G
10
100
CL − Capacitive Load − pF
Figure 8. Recommended RISO vs Capacitive Load
-40
VO = 2 VPP,
RL = 100 Ω,
VS = ±15 V
3rd Harmonic Distortion - dBc
2nd Harmonic Distortion − dBc
30
0
100 M
f − Frequency − Hz
−40
−50
35
5
Figure 7. Capacitive Load Frequency Response
−45
Gain = 5,
RL = 100 Ω,
VS = ±15 V
40
Recommended R
−Ω
ISO
Signal Gain − dB
12
−2
45
R(ISO) = 38.3 Ω
CL = 10 pF
14
−55
−60
G = 1, RF = 1.78 kΩ
−65
−70
−75
G = 2, RF = 1.21 kΩ
−80
VO = 2 VPP,
RL = 100 Ω,
VS = ±15 V
-50
-60
G = 1, RF = 1.78 kΩ
-70
-80
G = 2, RF = 1.21 kΩ
-90
−85
-100
−90
100 k
1M
10 M
100 M
100 k
f − Frequency − Hz
100 M
Figure 10. 3rd Harmonic Distortion vs Frequency
−40
−40
VO = 2 VPP,
RL = 1 kΩ,
VS = ±15 V
−60
−70
VO = 2 VPP,
RL = 1 kΩ,
VS = ±15 V
−45
3rd Harmonic Distortion − dBc
2nd Harmonic Distortion − dBc
10 M
f - Frequency - Hz
Figure 9. 2nd Harmonic Distortion vs Frequency
−50
1M
G = 1, RF = 1.78 kΩ
−80
G = 2, RF = 1.21 kΩ
−90
−50
−55
−60
G = 1, RF = 1.78 kΩ
−65
−70
−75
G = 2, RF = 1.21 kΩ
−80
−85
−100
100 k
−90
1M
10 M
100 M
100 k
1M
10 M
100 M
f − Frequency − Hz
f − Frequency − Hz
Figure 11. 2nd Harmonic Distortion vs Frequency
Figure 12. 3rd Harmonic Distortion vs Frequency
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Typical Characteristics (±15 V) (continued)
−30
G = 5,
RF = 1 kΩ,
RL = 100 Ω,
VS = ±15 V
−40
−50
−60
VO = 10 VPP
−70
−80
G = 5,
RF = 1 kΩ,
RL = 100 Ω,
VS = ±15 V
VO = 20 VPP
3rd Harmonic Distortion − dBc
2nd Harmonic Distortion − dBc
−30
VO = 2 VPP
−40
−50
VO = 20 VPP
−60
−70
VO = 10 VPP
−80
VO = 2 VPP
−90
−90
1M
10 M
100 M
1M
10 M
f − Frequency − Hz
Figure 13. 2nd Harmonic Distortion vs Frequency
Figure 14. 3rd Harmonic Distortion vs Frequency
-40
-60
Gain = 5,
RF = 1 kΩ
RL = 100 Ω,
f= 8 MHz
VS = ±15 V
-65
-70
Harmonic Distortion - dBc
Harmonic Distortion - dBc
-50
HD2
-75
-80
HD3
-85
Gain = 5,
RF = 1 kΩ
RL = 100 Ω,
f= 1 MHz
VS = ±15 V
-90
-95
HD2
-60
-70
-80
HD3
-90
-100
-100
0
2
4
6
8
10 12 14 16
0
18 20
2
4
6
8
10
12 14
16 18 20
VO - Output Voltage Swing - VPP
VO - Output Voltage Swing - VPP
Figure 15. Harmonic Distortion vs Output Voltage Swing
Figure 16. Harmonic Distortion vs Output Voltage Swing
2000
6000
1600
Rise
1400
1200
Gain = 2
RL = 100 Ω
RF = 1.21 kΩ
VS = ±15 V
5000
SR - Slew Rate - V/ µ s
Gain = 1
RL = 100 Ω
RF = 1.78 kΩ
VS = ±15 V
1800
SR − Slew Rate − V/ µ s
100 M
f − Frequency − Hz
Fall
1000
800
600
400
4000
3000
2000
Rise
Fall
1000
200
0
0
0.5
1 1.5 2 2.5 3 3.5 4
VO − Output Voltage − VPP
4.5
5
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0
1
2
3
4
5
6
7
8
9
10
VO - Output Voltage - VPP
Figure 17. Slew Rate vs Output Voltage Step
16
0
Figure 18. Slew Rate vs Output Voltage Step
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Typical Characteristics (±15 V) (continued)
8000
6000
Hz
Vn − Voltage Noise − nV/
SR - Slew Rate - V/ µ s
7000
5000
Rise
4000
Fall
3000
2000
I n − Current Noise − pA/ Hz
1000
Gain = 5
RL = 100 Ω
RF = 1 kΩ
VS = ±15 V
100
In−
In+
10
Vn
1000
1
0
0
2
10
4 6
8 10 12 14 16 18 20
VO - Output Voltage - VPP
1
VO - Output Voltage - V
VO - Output Voltage - V
Rising Edge
0.5
Gain = -2
RL = 100 Ω
RF =1 kΩ
VS = ±15 V
0
-0.25
-0.5
-0.75
Falling Edge
-1
-1.25
0
1
2
3
4
5
6
7
8
9
10
4.5
4
3.5
3
2.5
2
1.5
1
0.5
0
-0.5
-1
-1.5
-2
-2.5
-3
-3.5
-4
-4.5
100 k
Rising Edge
Gain = -2
RL = 100 Ω
RF = 1 kΩ
VS = ±15 V
Falling Edge
0
2
4
t - Time - ns
6
8
10
12
t - Time - ns
Figure 21. Settling Time
Figure 22. Settling Time
10
22
TA = 85 °C
9.5
20
TA = 25 °C
9
8.5
TA = −40 °C
8
7.5
7
6.5
I Q − Quiescent Current − mA
I Q− Quiescent Current − mA
10 k
Figure 20. Noise vs Frequency
1.25
0.25
1k
f − Frequency − Hz
Figure 19. Slew Rate vs Output Voltage Step
0.75
100
18
16
VO = 4VPP
14
12
10
VO = 2VPP
8
6
Gain = 5
RF = 1 kΩ,
RL = 100 Ω,
VS = ±15 V
4
2
6
3
4
5
6
7
8
9 10 11 12 13 14 15
0
100 k
VS − Supply Voltage − ±V
Figure 23. Quiescent Current vs Supply Voltage
100 M
10 M
1M
f − Frequency − Hz
1G
Figure 24. Quiescent Current vs Frequency
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Typical Characteristics (±15 V) (continued)
7
12
6.5
6
I IB - Input Bias Currents - µ A
I OS - Input Offset Currents - µ A
16
VO - Output Voltage - V
8
4
VS = ±15 V
TA = -40 to 85°C
0
-4
-8
-12
-16
10
100
1000
5.5
5
4.5
4
IIB+
3.5
3
2.5
2
1.5
1
IOS
0.5
0
-40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90
RL - Load Resistance - Ω
TC - Case Temperature - °C
Figure 25. Output Voltage vs Load
Resistance
Figure 26. Input Bias and Offset Current vs Case
Temperature
3
100
Transimpedance Gain − dB Ohms
VOS - Input Offset Voltage - mV
VS = ±15 V
IIB-
2.5
2
VS = ±15 V
1.5
1
VS = ±5 V
0.5
90
VS = ±15 V and ±5 V
80
70
60
50
40
30
20
10
0
100 k
0
-40 -30 -20-10 0 10 20 30 40 50 60 70 80 90
1M
10 M
100 M
1G
TC - Case Temperature - °C
f − Frequency − Hz
Figure 27. Input Offset Voltage vs Case Temperature
Figure 28. Transimpedance vs Frequency
70
60
0.25
50
CMRR
40
30
PSRR+
20
Output
0.2
PSRR−
VO - Output Voltage - V
Rejection Ratio − dB
0.3
VS = ±15 V
0.15
Input
0.1
0.05
0
-0.05
Gain = 2
RL = 100 Ω
RF = 1 kΩ
VS = ±15 V
-0.1
-0.15
-0.2
10
-0.25
0
100 k
1M
10 M
100 M
1G
-0.3
0
Figure 29. Rejection Ratio vs Frequency
18
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10
20
30
40
50
60
70
t - Time - ns
f − Frequency − Hz
Figure 30. Noninverting Small-Signal Transient Response
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Typical Characteristics (±15 V) (continued)
12
6
10
5
Output
4
VO - Output Voltage - V
VO − Output Voltage − V
3
2
1
0
Input
−1
−2
Gain = −5
RL = 100 Ω
RF = 909 Ω
VS = ±15 V
−3
−4
−5
Gain = -5
RL = 100 Ω
RF =909 Ω
VS = ±15 V
8
6
4
2
Input
0
-2
-4
-6
-8
Output
-10
-12
−6
0
5
10
15
20
25
30
35
0
40
10
Figure 31. Inverting Large-Signal Transient Response
20
Gain = 5,
RL = 100 Ω,
RF = 1 kΩ,
VS = ±15 V
Differential Gain - %
0.08
2
0.07
1
0
0
−5
−1
−10
−2
−15
−3
0.01
−4
0
−20
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
70
1
PAL
0.05
0.04
0.03
0.02
NTSC
0
1
2
3
4
5
6
7
8
Number of Loads - 150 Ω
Figure 33. Overdrive Recovery Time
Figure 34. Differential Gain vs Number of Loads
0.05
100
Gain = 2
RF = 1.21 kΩ
VS = ±15 V
40 IRE − NTSC and Pal
Worst Case ±100 IRE Ramp
0.04
Closed-Loop Output Impedance − Ω
Differential Phase −
60
0.06
t − Time − µs
0.03
PAL
0.02
NTSC
0.01
0
50
Gain = 2
RF = 1.21 kΩ
VS = ±15 V
40 IRE - NTSC and Pal
Worst Case ±100 IRE Ramp
0.09
3
5
°
40
0.10
VI − Input Voltage − V
VO − Output Voltage − V
10
30
Figure 32. Inverting Large-Signal Transient Response
4
15
20
t - Time - ns
t − Time − ns
Gain = 2,
RISO = 5.11 Ω,
RF = 1.21 KΩ,
VS = ±15 V
10
1
1.21 kΩ
1.21 kΩ
0.1
5.11 Ω VO
−
+
0.01
0
1
2
3
4
5
6
7
8
1M
Number of Loads − 150 Ω
10 M
100 M
1G
f − Frequency − Hz
Figure 35. Differential Phase vs Number of Loads
Figure 36. Closed-Loop Output Impedance vs Frequency
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Typical Characteristics (±15 V) (continued)
TA = 85°C
400
TA = -40°C
300
TA = 25°C
200
5
4
Gain = 2,
VI = 0.1 Vdc
RL = 100 Ω
VS = ±15 V and ±5 V
3
2
1
0
0.3
0.2
0.1
Power-on Pulse − V
500
Output Voltage
100
0
−0.1
0
3
4
5
6
7
8
9
0
10 11 12 13 14 15
VS - Supply Voltage - ±V
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1
2
3
4
5
6
7
t − Time − ms
Figure 37. Power-Down Quiescent Current vs Supply
Voltage
20
6
Power-on Pulse
VO − Output Voltage Level − V
Powerdown Quiescent Current - µ A
600
Figure 38. Turnon and Turnoff Time Delay
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24
22
20
18
16
14
12
10
8
6
4
2
0
-2
-4
24
22
20
18
16
14
12
10
G = 10, RF = 909 Ω
G = 5, RF = 1 kΩ
Inverting Gain − dB
Noninverting Gain - dB
6.10 Typical Characteristics (±5 V)
RL = 100 Ω,
VO = 200 mVPP.
VS = ±5 V
G = 2, RF = 1.15 kΩ
10 M
100 M
G = −5, RF = 909 Ω
RL = 100 Ω,
VO = 200 mVPP.
VS = ±5 V
8
6
4
2
0
−2
−4
G =1, RF = 1.5 kΩ
1M
G = −10, RF = 866 Ω
G = −2, RF = 1 kΩ
G = −1, RF = 1.05 Ω
1M
1G
10 M
100 M
f − Frequency − Hz
f - Frequency - Hz
Figure 39. Noninverting Small-Signal Frequency Response
Figure 40. Inverting Small-Signal Frequency Response
16
6.3
6.1
Gain = 2,
RF = 1.21 kΩ,
RL = 100 Ω,
VO = 200 mVPP,
VS = ±5 V
G = 5, RF = 1 kΩ
14
12
Noninverting Gain − dB
Noninverting Gain - dB
6.2
6
5.9
10
8
G = 2, RF = 1.15 kΩ
6
4
5.8
RL = 100 Ω,
VO = 4 VPP,
VS = ±5 V
2
0
5.7
1M
10 M
1M
100 M
10 M
Figure 41. 0.1-db Gain Flatness Frequency Response
16
1.25
G = −5, RF = 909 Ω
0.75
VO - Output Voltage - V
1
12
10
8
6
G = −2, RF = 1 kΩ
0.5
0.25
Gain = -2
RL = 100 Ω
RF = 1 kΩ
VS = ±5 V
0
-0.5
-0.75
RL = 100 Ω,
VO = 4 VPP,
VS = ±5 V
−2
Rising Edge
-0.25
2
0
1G
Figure 42. Noninverting Large-Signal Frequency Response
14
4
100 M
f − Frequency − Hz
f - Frequency - Hz
Inverting Gain − dB
1G
Falling Edge
-1
−4
-1.25
1M
10 M
100 M
f − Frequency − Hz
1G
0
1
2
3
4
5
6
7
8
9
10
t - Time - ns
Figure 43. Inverting Large-Signal Frequency Response
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Figure 44. Settling Time
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Typical Characteristics (±5 V) (continued)
-40
VO = 2 VPP,
RL = 100 Ω,
VS = ±5 V
−50
3rd Harmonic Distortion - dBc
2nd Harmonic Distortion − dBc
−40
−60
G = 1, RF = 1.78 kΩ
−70
−80
G = 2, RF = 1.15 kΩ
−90
−100
100 k
VO = 2 VPP,
RL = 100 Ω,
VS = ±5 V
-50
-60
-70
G = 1, RF = 1.78 kΩ
-80
G = 2, RF = 1.15 kΩ
-90
-100
1M
10 M
100 k
100 M
1M
f − Frequency − Hz
Figure 45. 2nd Harmonic Distortion vs Frequency
−50
3rd Harmonic Distortion − dBc
2nd Harmonic Distortion − dBc
−40
VO = 2 VPP,
RL = 1 kΩ,
VS = ±5 V
−60
G = 1, RF = 1.78 kΩ
−70
−80
G = 2, RF = 1.15 kΩ
−90
−100
100 k
1M
10 M
VO = 2 VPP,
RL = 1 kΩ,
VS = ±5 V
−50
−60
G = 1, RF = 1.78 kΩ
−70
−80
G = 2, RF = 1.15 kΩ
−90
−100
100 k
100 M
1M
f − Frequency − Hz
10 M
100 M
f − Frequency − Hz
Figure 47. 2nd Harmonic Distortion vs Frequency
Figure 48. 3rd Harmonic Distortion vs Frequency
-20
−20
Gain = 5,
RF = 1 kΩ
RL = 100 Ω,
f= 1 MHz
VS = ±5 V
-40
-50
-60
Gain = 5,
RF = 1 kΩ
RL = 100 Ω,
f= 8 MHz
VS = ±5 V
−30
Harmonic Distortion − dBc
-30
Harmonic Distortion - dBc
100 M
Figure 46. 3rd Harmonic Distortion vs Frequency
−40
HD3
-70
-80
HD2
-90
−40
−50
HD3
−60
HD2
−70
−80
−90
-100
−100
0
1
2
3
4
5
6
0
VO - Output Voltage Swing - VPP
Figure 49. Harmonic Distortion vs Output Voltage Swing
22
10 M
f - Frequency - Hz
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2
3
4
5
6
VO − Output Voltage Swing − VPP
Figure 50. Harmonic Distortion vs Output Voltage Swing
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Typical Characteristics (±5 V) (continued)
1600
1600
Gain = 1
RL = 100 Ω
RF = 1.78 kΩ
VS = ±5 V
1200
1000
Fall
800
600
Rise
400
Gain = 1
RL = 100 Ω
RF = 1.21 kΩ
VS = ±5 V
1400
SR − Slew Rate − V/µ s
SR − Slew Rate − V/ µs
1400
1200
1000
Rise
800
600
400
200
200
0
0
0 0.5
1 1.5
2
2.5
3 3.5
4
4.5
5
0
VO − Output Voltage −VPP
Figure 51. Slew Rate vs Output Voltage Step
5
22
Gain = 5
RL = 100 Ω
RF = 1 kΩ
VS = ±5 V
1600
1400
20
Fall
I Q− Quiescent Current − mA
1800
SR - Slew Rate - V/ µ s
1
2
3
4
VO − Output Voltage −VPP
Figure 52. Slew Rate vs Output Voltage Step
2000
Rise
1200
1000
800
600
400
18
16
14
Gain = 5
RF = 1 kΩ,
RL = 100 Ω,
VS = ±5 V
VO = 4 VPP
12
10
8
VO = 2 VPP
6
4
200
2
0
100 k
0
0
0.5
1 1.5 2 2.5 3 3.5 4
VO - Output Voltage -VPP
4.5
5
7
I IB - Input Bias Current - µ A
I OS - Input Offset Current - µ A
8
3
2.5
VS = ±5 V
TA = -40 to 85°C
0
-0.5
10 M
100 M
1G
Figure 54. Quiescent Current vs Frequency
3.5
2
1.5
1
0.5
1M
f − Frequency − Hz
Figure 53. Slew Rate vs Output Voltage Step
VO - Output Voltage - V
Fall
-1
-1.5
-2
-2.5
-3
VS = ±5 V
IIB-
6
5
IOS
4
3
2
IIB+
1
-3.5
10
100
1000
0
-40 -30 -20 -10 0
10 20 30 40 50 60 70 80 90
TC - Case Temperature - °C
RL - Load Resistance - Ω
Figure 55. Output Voltage vs Load Resistance
Figure 56. Input Bias and Offset Current vs Case
Temperature
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Typical Characteristics (±5 V) (continued)
5
60
0.6
0.4
1
0.2
0
0
-1
-0.2
-2
-0.4
-3
-0.6
-4
-0.8
PSRR-
-1
-5
0
0.2
0.4
0.6
0.8
1
Rejection Ratio - dB
2
VS = ±5 V
0.8
VI - Input Voltage - V
3
VO - Output Voltage - A
70
1
Gain = 5,
RL = 100 Ω,
RF = 1 kΩ,
VS = ±5 V
4
50
40
CMRR
30
PSRR+
20
10
0
100 k
t - Time - µs
Figure 57. Overdrive Recovery Time
24
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1M
10 M
f - Frequency - Hz
100 M
Figure 58. Rejection Ratio vs Frequency
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7 Detailed Description
7.1 Overview
The THS3091 and THS3095 are high-voltage, low-distortion , high-speed, current feedback amplifiers designed
to operate over a wide supply range of ± V to ±15 V for applications requiring large, linear output swings such as
Arbitrary Waveform Generators.
The THS3095 features a power-down pin that puts the amplifier in low power standby mode, and lowers the
quiescent current from 9.5 mA to 500 uA
7.2 Feature Description
7.2.1 Saving Power With Power-Down Functionality and Setting Threshold Levels With the Reference
Pin
The THS3095 features a power-down pin (PD) which lowers the quiescent current from 9.5 mA down to 500 μA,
ideal for reducing system power.
The power-down pin of the amplifier defaults to the positive supply voltage in the absence of an applied voltage,
putting the amplifier in the power-on mode of operation. To turn off the amplifier in an effort to conserve power,
the power-down pin can be driven towards the negative rail. The threshold voltages for power on and power
down are relative to the supply rails and are given Typical Characteristics (±15 V) and Typical Characteristics (±5
V) tables. Above the Enable Threshold Voltage, the device is on. Below the Disable Threshold Voltage, the
device is off. Behavior in between these threshold voltages is not specified.
Note that this power-down functionality is just that; the amplifier consumes less power in power-down mode. The
power-down mode is not intended to provide a high-impedance output. In other words, the power-down
functionality is not intended to allow use as a 3-state bus driver. When in power-down mode, the impedance
looking back into the output of the amplifier is dominated by the feedback and gain-setting resistors, but the
output impedance of the device itself varies depending on the voltage applied to the outputs.
Figure 59 shows the total system output impedance which includes the amplifier output impedance in parallel
with the feedback plus gain resistors, which cumulate to 2380 Ω. Figure 60 shows this circuit configuration for
reference.
ZOPD − Powerdown Output Impedance − Ω
2500
VS = ±15 V and ±5 V
2000
1500
1000
1.21 kΩ
500
1.21 kΩ
−
+
50 Ω VO
1M
10 M
0
100 k
100 M
1G
f − Frequency − Hz
Figure 59. Power-Down Output Impedance vs Frequency
As with most current feedback amplifiers, the internal architecture places some limitations on the system when in
power-down mode. Most notably is the fact that the amplifier actually turns ON if there is a ±0.7 V or greater
difference between the two input nodes (V+ and V–) of the amplifier. If this difference exceeds ±0.7 V, the output
of the amplifier creates an output voltage equal to approximately [(V+ – V–) –0.7 V] × Gain. This also implies that
if a voltage is applied to the output while in power-down mode, the V– node voltage is equal to VO(applied) ×
RG/(RF + RG). For low gain configurations and a large applied voltage at the output, the amplifier may actually
turn ON due to the aforementioned behavior.
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Feature Description (continued)
The time delays associated with turning the device on and off are specified as the time it takes for the amplifier to
reach either 10% or 90% of the final output voltage. The time delays are in the order of microseconds because
the amplifier moves in and out of the linear mode of operation in these transitions.
7.2.2 Power-Down Reference Pin Operation
In addition to the power-down pin, the THS3095 features a reference pin (REF) which allows the user to control
the enable or disable power-down voltage levels applied to the PD pin. In most split-supply applications, the
reference pin is connected to ground. In either case, the user needs to be aware of voltage-level thresholds that
apply to the power-down pin. The tables below show examples and illustrate the relationship between the
reference voltage and the power-down thresholds. In the table, the threshold levels are derived by the following
equations:
PD ≤ REF + 0.8 V for disable
PD ≥ REF + 2.0 V for enable
(1)
(2)
where the usable range at the REF pin is:
VS– ≤ VREF ≤ (VS+ – 4 V).
(3)
The recommended mode of operation is to tie the REF pin to midrail, thus setting the enable/disable thresholds
to Vmidrail + 2 V and Vmidrail + 0.8 V respectively.
Table 3. Power-Down Threshold Voltage Levels
SUPPLY
VOLTAGE (V)
REFERENCE PIN
VOLTAGE (V)
ENABLE
LEVEL (V)
DISABLE
LEVEL (V)
±15, ±5
0
2
0.8
±15
2
4
2.8
±15
–2
0
–1.2
±5
1
3
1.8
±5
–1
1
–0.2
30
15
17
15.8
10
5
7
5.8
Note that if the REF pin is left unterminated, it will float to the positive rail and will fall outside of the
recommended operating range given above (VS– ≤ VREF ≤ VS+ – 4 V). As a result, it will no longer serve as a
reliable reference for the PD pin and the enable/disable thresholds given above will no longer apply. If the PD pin
is also left unterminated, it will also float to the positive rail and the device will be enabled. If balanced, split
supplies are used (±Vs) and the REF and PD pins are grounded, the device will be disabled.
7.3 Device Functional Modes
7.3.1 Wideband, Noninverting Operation
The THS309x are unity gain stable 235-MHz current-feedback operational amplifiers, designed to operate from a
±5-V to ±15-V power supply.
Figure 60 shows the THS3091 in a noninverting gain of 2-V/V configuration typically used to generate the
performance curves. Most of the curves were characterized using signal sources with 50-Ω source impedance,
and with measurement equipment presenting a 50-Ω load impedance.
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Device Functional Modes (continued)
15 V
+VS
+
0.1 µF
50-Ω Source
+
VI
6.8 µF
49.9 Ω
THS3091
49.9 Ω
_
50-Ω LOAD
RF
1.21 kΩ
RG
1.21 kΩ
0.1 µF
6.8 µF
+
−VS
−15 V
Figure 60. Wideband, Noninverting Gain Configuration
Current-feedback amplifiers are highly dependent on the feedback resistor RF for maximum performance and
stability. Table 4 shows the optimal gain-setting resistors RF and RG at different gains to give maximum
bandwidth with minimal peaking in the frequency response. Higher bandwidths can be achieved, at the expense
of added peaking in the frequency response, by using even lower values for RF. Conversely, increasing RF
decreases the bandwidth, but stability is improved.
Table 4. Recommended Resistor Values for Optimum Frequency Response
THS3091 and THS3095 RF and RG values for minimal peaking with RL = 100 Ω
GAIN (V/V)
SUPPLY VOLTAGE (V)
RG (Ω)
RF (Ω)
±15
–
1.78 k
±5
–
1.78 k
±15
1.21 k
1.21 k
±5
1.15 k
1.15 k
±15
249
1k
1
2
5
10
±5
249
1k
±15
95.3
866
±5
95.3
866
–1
±15 and ±5
1.05 k
1.05 k
–2
±15 and ±5
499
1k
–5
±15 and ±5
182
909
–10
±15 and ±5
86.6
866
7.3.2 Wideband, Inverting Operation
Figure 61 shows the THS3091 in a typical inverting gain configuration where the input and output impedances
and signal gain from Figure 60 are retained in an inverting circuit configuration.
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Device Functional Modes (continued)
15 V +VS
+
0.1 µF
+
THS3091
6.8 µF
49.9 Ω
_
50-Ω LOAD
50-Ω Source
VI
RG
RF
499 Ω
RM
56.2 Ω
1 kΩ
0.1 µF
6.8 µF
+
−15 V
−VS
Figure 61. Wideband, Inverting Gain Configuration
7.3.3 Single-Supply Operation
The THS309x have the capability to operate from a single-supply voltage ranging from 10 V to 30 V. When
operating from a single power supply, biasing the input and output at mid-supply allows for the maximum output
voltage swing. The circuits shown in Figure 62 show inverting and noninverting amplifiers configured for singlesupply operations.
+VS
50-Ω Source
+
VI
49.9 Ω
RT
49.9 Ω
THS3091
_
50-Ω LOAD
+VS
2
RF
1.21 kΩ
RG
1.21 kΩ
+VS
2
RF
1 kΩ
50-Ω Source
VI
56.2 Ω
+VS
2
VS
RG
499 Ω
RT
_
49.9 Ω
THS3091
+
50-Ω LOAD
+VS
2
Figure 62. DC-Coupled, Single-Supply Operation
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
8.1.1 Video Distribution
The wide bandwidth, high slew rate, and high output drive current of the THS309x matches the demands for
video distribution for delivering video signals down multiple cables. To ensure high signal quality with minimal
degradation of performance, a 0.1-dB gain flatness should be at least 7x the passband frequency to minimize
group delay variations from the amplifier. A high slew rate minimizes distortion of the video signal, and supports
component video and RGB video signals that require fast transition times and fast settling times for high signal
quality.
1.21 kΩ
1.21 kΩ
15 V
−
+
VI
75 Ω
75-Ω Transmission Line
75 Ω
−15 V
n Lines
75 Ω
VO(1)
VO(n)
75 Ω
75 Ω
Figure 63. Video Distribution Amplifier Application
8.1.2 Driving Capacitive Loads
Applications such as FET line drivers can be highly capacitive and cause stability problems for high-speed
amplifiers.
Figure 64 through Figure 69 show recommended methods for driving capacitive loads. The basic idea is to use a
resistor or ferrite chip to isolate the phase shift at high frequency caused by the capacitive load from the
amplifier’s feedback path. See SLOA013 for recommended resistor values versus capacitive load.
45
Gain = 5,
RL = 100 Ω,
VS = ±15 V
Recommended R
−Ω
ISO
40
35
30
25
20
15
10
5
0
10
100
CL − Capacitive Load − pF
Figure 64. Recommended RISO vs Capacitive Load
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Application Information (continued)
1 kΩ
VS
249 Ω
_
5.11 Ω
+
RISO
−VS
100-Ω LOAD
1 µF
49.9 Ω
VS
Figure 65. Driving a Large Capacitive Load Using an Output Series Isolation Resistor
1 kΩ
249 Ω
VS
_
Ferrite Bead
+
−VS
VS
1 µF
100-Ω LOAD
49.9 Ω
Figure 66. Driving a Large Capacitive Load Using an Output Series Ferrite Bead
Placing a small series resistor, RISO, between the amplifier’s output and the capacitive load, as shown in
Figure 65, is an easy way of isolating the load capacitance.
Using a ferrite chip in place of RISO, as shown in Figure 66, is another approach of isolating the output of the
amplifier. The ferrite's impedance characteristic versus frequency is useful to maintain the low-frequency load
independence of the amplifier while isolating the phase shift caused by the capacitance at high frequency. Use a
ferrite with similar impedance to RISO, 20 Ω to 50 Ω, at 100 MHz and low-impedance at DC.
Figure 67 shows another method used to maintain the low-frequency load independence of the amplifier while
isolating the phase shift caused by the capacitance at high frequency. At low frequency, feedback is mainly from
the load side of RISO. At high frequency, the feedback is mainly via the 27-pF capacitor. The resistor RIN in series
with the negative input is used to stabilize the amplifier and should be equal to the recommended value of RF at
unity gain. Replacing RIN with a ferrite of similar impedance at about 100 MHz as shown in Figure 68 gives
similar results with reduced DC offset and low-frequency noise. (See the Related Documentation section for
expanding the usability of current-feedback amplifiers.)
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Application Information (continued)
RF
1 kΩ
27 pF
RIN
RG
1 kΩ
249 Ω
VS
_
100-Ω LOAD
5.11 Ω
+
1 µF
−VS
49.9 Ω
VS
Figure 67. Driving a Large Capacitive Load Using a Multiple Feedback Loop With Stabilizing Input
Resistor (RIN)
RF
1 kΩ
27 pF
FIN
RG
FB
249 Ω
VS
_
100-Ω LOAD
5.11 Ω
+
1 µF
−VS
49.9 Ω
VS
Figure 68. Driving a Large Capacitive Load Using a Multiple Feedback Loop With Stabilizing Input Ferrite
Bead (FIN)
Figure 69 is shown using two amplifiers in parallel to double the output drive current to larger capacitive loads.
This technique is used when more output current is needed to charge and discharge the load faster like when
driving large FET transistors.
1 kΩ
249 Ω
24.9 Ω
VS
5.11 Ω
_
+
−VS
1 kΩ
VS
249 Ω
24.9 Ω
VS
_
1 nF
5.11 Ω
+
−VS
Figure 69. Driving a Large Capacitive Load Using 2 Parallel Amplifier Channels
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Application Information (continued)
Figure 70 shows a push-pull FET driver circuit typical of ultrasound applications with isolation resistors to isolate
the gate capacitance from the amplifier.
VS
VS
5.11 Ω
+
_
−VS
866 Ω
191 Ω
866 Ω
VS
_
5.11 Ω
+
−VS
−VS
Figure 70. PowerFET Drive Circuit
8.2 Typical Application
The fundamental concept of load sharing is to drive a load using two or more of the same operational amplifiers.
Each amplifier is driven by the same source. Figure 71 shows two THS3091 amplifiers sharing the same load.
This concept effectively reduces the curernt load of each amplifier by 1/N, where N is the number of amplifiers.
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Typical Application (continued)
RG
250 W
RF
1 kW
V-
RS
50 W
RSOURCE
50 W
VOUT
THS3091
U3
RT
50 W
VIN
RF1
1 kW
V+
V1
15 V
V-
VIN
RLOAD
50 W
V+
RG1
250 W
RSOURCE
50 W
TL2
Characteristic
Impedance
50 W
V-
+
+
V2
-15 V
RS1
100 W
TL1
Characteristic
Impedance
50 W
THS3091
U1
RT1
100 W
V+
RG2
250 W
RF2
1 kW
VOUT
RLOAD
50 W
V-
RS2
100 W
THS3091
U2
RT2
100 W
V+
Figure 71. Reference THS3091 and THS3091 Load Sharing Test Configurations
8.2.1 Design Requirements
Use two THS3091 amplifiers in a parallel load-sharing circuit to improve distortion performance.
Table 5. Design Parameters
DESIGN PARAMETER
VALUE
VOPP
20 V
RLOAD
100 Ω
8.2.2 Detailed Design Procedure
In addition to providing higher output current drive to the load, the load sharing configuration can also provide
improved distortion performance. In many cases, an operational amplifier shows better distortion performance as
the load current decreases (that is, for higher resistive loads) until the feedback resistor starts to dominate the
current load. In a load sharing configuration of N amplifiers in parallel, the equivalent current load that each
amplifier drives is 1/N times the total load current. For example, in a two-amplifier load sharing configuration with
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matching resistance (refer to Figure 71) driving a resistive load (RL), each series resistance is 2*RL and each
amplifier drives 2*RL . A convenient indicator of whether an op amp will function well in a load sharing
configuration is the characteristic performance graph of harmonic distortion versus load resistance. Such graphs
can be found in most of TI’s high-speed amplifier data sheets. These graphs can be used to obtain a general
sense of whether or not an amplifier will show improved distortion performance in load sharing configurations.
Two test circuits are shown in Figure 71, one for a single THS3091 amplifier driving a double-terminated, 50-Ω
cable and one with two THS3091 amplifiers in a load sharing configuration. In the load sharing configuration, the
two 100-Ω series output resistors act in parallel to provide 50-Ω back-matching to the 50-Ω cable.
Figure 72 and Figure 73 show the 32-MHz, 18-VPP sine wave output amplitudes for the single THS3091
configuration and the load sharing configuration, respectively, measured using an oscilloscope. An ideal sine
wave is also included as a visual reference (the dashed red line). Figure 72 shows visible distortion in the single
THS3091 output. In the load sharing configuration of Figure 73, however, no obvious degradation is visible.
Figure 74 and Figure 75 show the 64-MHz sine wave outputs of the two configurations from Figure 8. While the
single THS3091 output is clearly distorted in Figure 74, the output of the load sharing configuration in Figure 75
shows only minor deviations from the ideal sine wave.
The improved output waveform as a result of load sharing is quantified in the harmonic distortion versus
frequency graphs shown in Figure 76 and Figure 77 for the single amplifier and load sharing configurations,
respectively. While second-harmonic distortion remains largely the same between the single and load sharing
cases, third-harmonic distortion is improved by approximately 8 dB in the frequency range between 20 MHz to 64
MHz.
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Table 6. Bill of Materials
THS3091DDA and THS3095DDA EVM (1)
ITEM
SMD
SIZE
DESCRIPTION
1
Bead, Ferrite, 3 A, 80 Ω
2
Cap, 6.8 μF, Tantalum, 50 V, 10%
3
REFERENCE
DESIGNATOR
1206
PCB
QTY
MANUFACTURER'S
PART NUMBER
DISTRIBUTOR'S
PART NUMBER
FB1, FB2
2
(Steward) HI1206N800R-00
(Digi-Key) 240-1010-1-ND
D
C3, C6
2
(AVX) TAJD685K050R
(Garrett) TAJD685K050R
Cap, 0.1 μF, ceramic, X7R, 50 V
0805
C9, C10
2 (2)
(AVX) 08055C104KAT2A
(Garrett) 08055C104KAT2A
4
Cap, 0.1 μF, ceramic, X7R, 50 V
0805
C4, C7
2
(AVX) 08055C104KAT2A
(Garrett) 08055C104KAT2A
5
Resistor, 0 Ω, 1/8 W, 1%
0805
R9
1 (2)
(KOA) RK73Z2ALTD
(Garrett) RK73Z2ALTD
6
Resistor, 249 Ω, 1/8 W, 1%
0805
R3
1
(KOA) RK73H2ALTD2490F
(Garrett) RK73H2ALTD2490F
7
Resistor, 1 kΩ, 1/8 W, 1%
0805
R4
1
(KOA) RK73H2ALTD1001F
(Garrett) RK73H2ALTD1001F
8
Open
1206
R8
1
9
Resistor, 0 Ω, 1/4 W, 1%
1206
R1
1
(KOA) RK73Z2BLTD
(Garrett) RK73Z2BLTD
10
Resistor, 49.9 Ω, 1/4 W, 1%
1206
R2, R7
2
(KOA) RK73Z2BLTD49R9F
(Garrett) RK73Z2BLTD49R9F
11
Open
2512
R5, R6
2
12
Header, 0.1-inch (2,54 mm) centers,
0.025-inch (6,35 mm) square pins
JP1, JP2
(Sullins) PZC36SAAN
(Digi-Key) S1011-36-ND
13
Connector, SMA PCB Jack
J1, J2, J3
3
(Amphenol) 901-144-8RFX
(Newark) 01F2208
14
Jack, banana receptacle,
0.25-inch (6,35 mm) dia. hole
J4, J5, J6
3
(SPC) 813
(Newark) 39N867
15
Test point, black
TP1, TP2
2
(Keystone) 5001
(Digi-Key) 5001K-ND
16
Standoff, 4-40 hex,
0.625-inch (15,9 mm) length
4
(Keystone) 1808
(Newark) 89F1934
17
Screw, Phillips, 4-40,
0.25-inch (6,35 mm)
4
SHR-0440-016-SN
18
IC, THS3091(3)
IC, THS3095(2)
1
(TI) THS3091DDA (3)
(TI) THS3095DDA (2)
19
Board, printed-circuit
1
(TI) EDGE # 6446289 Rev. A (3)
(TI) EDGE # 6446290 Rev. A (2)
(1)
(2)
(3)
2
U1
(2)
All items are designated for both the THS3091DDA and THS3095 EVMs unless otherwise noted.
THS3095 EVM only.
THS3091 EVM only.
15
15
10
10
Output Voltage (V)
Output Voltage (V)
8.2.3 Application Curves
5
0
–5
5
0
–5
–10
–10
–15
–15
0
10
20
30
40
50
0
Time (ns)
10
20
30
40
50
Time (ns)
Figure 72. 32-MHz Sine Wave Output (Gain = 5 V/V, Signal
Amplitude Referred to Amplifier Output), Single THS3091
Circuit Configuration
Figure 73. 32-MHz Sine Wave Output (Gain = 5 V/V, Signal
Amplitude Referred to Amplifier Output), Two THS3091
Amplifiers in Load Sharing Configuration
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15
15
10
10
Output Voltage (V)
Output Voltage (V)
SLOS423H – SEPTEMBER 2003 – REVISED DECEMBER 2015
5
0
–5
–10
5
0
–5
–10
–15
–15
0
5
10
15
20
25
0
5
10
Time (ns)
Figure 74. 64-MHz Sine Wave Output (Gain = 5 V/V, Signal
Amplitude Referred to Amplifier Output), Single THS3091
Circuit Configuration
25
–10
VO = 20 VPP (at amplifier output)
–20
RS = 50 Ω
–30
RL = 50 Ω
–40
–50
–60
–70
–80
1
10
VO = 10 VPP (at load)
RS (Each Amplifier) = 100 Ω
–30
RL (Shared) = 50 Ω
–40
–50
–60
–70
–80
Second Harmonic
Third Harmonic
–90
VO = 20 VPP (at amplifier output)
–20
VO = 10 VPP (at load)
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
20
Figure 75. 64-MHz Sine Wave Output (Gain = 5 V/V, Signal
Amplitude Referred to Amplifier Output), Two THS3091
Amplifiers in Load Sharing Configuration
–10
36
15
Time (ns)
Second Harmonic
Third Harmonic
–90
100
1
10
100
Frequency (MHz)
Frequency (MHz)
Figure 76. Harmonic Distortion vs Frequency, Single
THS3091 Circuit Configuration
Figure 77. Harmonic Distortion vs Frequency, Two
THS3091 Amplifiers in Load Sharing Configuration
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9 Power Supply Recommendations
The THS3091 can operate off a single supply or with dual supplies as long as the input CM voltage range
(CMIR) has the required headroom to either supply rail. Operating from a single supply can have numerous
advantages. With the negative supply at ground, the DC errors due to the –PSRR term can be minimized.
Supplies should be decoupled with low inductance, often ceramic, capacitors to ground less than 0.5 inches from
the device pins. The use of ground plane is recommended, and as in most high speed devices, it is advisable to
remove ground plane close to device sensitive pins such as the inputs. An optional supply decoupling capacitor
across the two power supplies (for split supply operation) improves second harmonic distortion performance.
10 Layout
10.1 Layout Guidelines
Achieving optimum performance with a high-frequency amplifier, like the THS309x, requires careful attention to
board layout parasitic and external component types.
Recommendations that optimize performance include:
• Minimize parasitic capacitance to any ac ground for all of the signal I/O pins. Parasitic capacitance on the
output and input pins can cause instability. To reduce unwanted capacitance, a window around the signal I/O
pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and
power planes should be unbroken elsewhere on the board.
• Minimize the distance [< 0.25 inch (6.35 mm)] from the power supply pins to high-frequency 0.1-μF and 100pF decoupling capacitors. At the device pins, the ground and power plane layout should not be in close
proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the
pins and the decoupling capacitors. The power supply connections should always be decoupled with these
capacitors. Larger (6.8 μF or more) tantalum decoupling capacitors, effective at lower frequency, should also
be used on the main supply pins. These may be placed somewhat farther from the device and may be shared
among several devices in the same area of the PC board.
• Careful selection and placement of external components preserve the high-frequency performance of the
THS309x. Resistors should be a low reactance type. Surface-mount resistors work best and allow a tighter
overall layout. Again, keep their leads and PC board trace length as short as possible. Never use wire-bound
type resistors in a high-frequency application. Because the output pin and inverting input pins are the most
sensitive to parasitic capacitance, always position the feedback and series output resistors, if any, as close as
possible to the inverting input pins and output pins. Other network components, such as input termination
resistors, should be placed close to the gain-setting resistors. Even with a low parasitic capacitance shunting
the external resistors, excessively high resistor values can create significant time constants that can degrade
performance. Good axial metal-film or surface-mount resistors have approximately 0.2 pF in shunt with the
resistor. For resistor values > 2 kΩ, this parasitic capacitance can add a pole and/or a zero that can effect
circuit operation. Keep resistor values as low as possible, consistent with load-driving considerations.
• Connections to other wideband devices on the board may be made with short direct traces or through
onboard transmission lines. For short connections, consider the trace and the input to the next device as a
lumped capacitive load. Relatively wide traces [0.05 inch (1.3 mm) to 0.1 inch (2.54 mm)] should be used,
preferably with ground and power planes opened up around them. Estimate the total capacitive load and
determine if isolation resistors on the outputs are necessary. Low parasitic capacitive loads (< 4 pF) may not
need an RS because the THS309x are nominally compensated to operate with a 2-pF parasitic load. Higher
parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded
phase margin). If a long trace is required, and the 6-dB signal loss intrinsic to a doubly terminated
transmission line is acceptable, implement a matched impedance transmission line using microstrip or
stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50-Ω
environment is not necessary onboard, and in fact, a higher impedance environment improves distortion as
shown in the distortion versus load plots. With a characteristic board trace impedance based on board
material and trace dimensions, a matching series resistor into the trace from the output of the THS309x is
used as well as a terminating shunt resistor at the input of the destination device. Remember also that the
terminating impedance is the parallel combination of the shunt resistor and the input impedance of the
destination device; this total effective impedance should be set to match the trace impedance. If the 6-dB
attenuation of a doubly terminated transmission line is unacceptable, a long trace can be series- terminated at
the source end only. Treat the trace as a capacitive load in this case. This does not preserve signal integrity
as well as a doubly terminated line. If the input impedance of the destination device is low, there is some
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Layout Guidelines (continued)
•
signal attenuation due to the voltage divider formed by the series output into the terminating impedance.
Socketing a high-speed part like the THS309x is not recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an extremely troublesome parasitic network which can make
it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering
the THS309x parts directly onto the board.
10.2 Layout Example
Figure 78. Layout Recommendation
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Layout Example (continued)
PIN8
(2)
REF
(2)
JP1
(2)
C9
J4
VS−
FB1
C3
6.8 µF
TP2
(2)
JP2
(2) THS3095 EVM Only
FB2
VS−
VS+
+
C4
0.1 µF
+
(2)
C10
J6
VS+
J5
GND
TP1
R9
C7
0.1 µF
C6
6.8 µF
J1
R1
0Ω
R3
R4
249 Ω
1 kΩ
PIN8
VS+
J2
2
7 8 1
6
3
R2
49.9 Ω
R5
REF
Open
R7
49.9 Ω
4
5
VS−
J3
R8
Open
R6
Open
THS3091DDA or THS3095DDA
Figure 79. THS3091 EVM Circuit Configuration
Figure 80. THS3091 EVM Board Layout (Top Layer)
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Layout Example (continued)
Figure 81. THS3091 EVM Board Layout (Second and Third Layers)
Figure 82. THS3091 EVM Board Layout (Bottom Layer)
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10.3 PowerPAD Design Considerations
The THS309x are available in a thermally-enhanced PowerPAD family of packages. These packages are
constructed using a downset leadframe on which the die is mounted [see Figure 83(a) and Figure 83(b)]. This
arrangement results in the lead frame being exposed as a thermal pad on the underside of the package [see
Figure 83(c)]. Because this thermal pad has direct thermal contact with the die, excellent thermal performance
can be achieved by providing a good thermal path away from the thermal pad. Note that devices such as the
THS309x have no electrical connection between the PowerPAD and the die.
The PowerPAD package allows for both assembly and thermal management in one manufacturing operation.
During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be
soldered to a copper area underneath the package. Through the use of thermal paths within this copper area,
heat can be conducted away from the package into either a ground plane or other heat-dissipating device.
The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of
surface mount with the, heretofore, awkward mechanical methods of heatsinking.
DIE
Thermal
Pad
Side View (a)
DIE
End View (b)
Bottom View (c)
Figure 83. Views of Thermal Enhanced Package
Although there are many ways to properly heatsink the PowerPAD package, the following steps illustrate the
recommended approach.
0.300
(7,62)
0.100
(2,54)
0.035
(0,89)
0.010
(0,254)
0.026
(0,66)
0.030
(0,732)
0.060
(1,52)
0.140
(3,56)
0.176
(4,47)
0.050
(1,27)
0.060
(1,52)
0.010
(0.254)
vias
0.035
(0,89)
0.080
(2,03)
All Units in inches (millimeters)
Figure 84. DDA PowerPAD PCB Etch and via Pattern
10.3.1 PowerPAD Layout Considerations
1. PCB with a top-side etch pattern is shown in Figure 84. There should be etch for the leads as well as etch for
the thermal pad.
2. Place 13 holes in the area of the thermal pad. These holes should be 0.01 inch (0.254 mm) in diameter.
Keep them small so that solder wicking through the holes is not a problem during reflow.
3. Additional vias may be placed anywhere along the thermal plane outside of the thermal pad area. This helps
dissipate the heat generated by the THS309x IC. These additional vias may be larger than the 0.01-inch
(0.254 mm) diameter vias directly under the thermal pad. They can be larger because they are not in the
thermal pad area to be soldered so that wicking is not a problem.
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PowerPAD Design Considerations (continued)
4. Connect all holes to the internal ground plane. Note that the PowerPAD is electrically isolated from the
silicon and all leads. Connecting the PowerPAD to any potential voltage such as VS– is acceptable as there
is no electrical connection to the silicon.
5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection
methodology. Web connections have a high thermal resistance connection that is useful for slowing the heat
transfer during soldering operations. This makes the soldering of vias that have plane connections easier. In
this application, however, low thermal resistance is desired for the most efficient heat transfer. Therefore, the
holes under the THS309x PowerPAD package should make their connection to the internal ground plane
with a complete connection around the entire circumference of the plated-through hole.
6. The top-side solder mask should leave the terminals of the package and the thermal pad area with its 13
holes exposed. The bottom-side solder mask should cover the 13 holes of the thermal pad area. This
prevents solder from being pulled away from the thermal pad area during the reflow process.
7. Apply solder paste to the exposed thermal pad area and all of the IC terminals.
8. With these preparatory steps in place, the IC is simply placed in position and run through the solder reflow
operation as any standard surface-mount component. This results in a part that is properly installed.
10.3.2 Power Dissipation and Thermal Considerations
The THS309x incorporates automatic thermal shutoff protection. This protection circuitry shuts down the amplifier
if the junction temperature exceeds approximately 160°C. When the junction temperature reduces to
approximately 140°C, the amplifier turns on again. But, for maximum performance and reliability, the designer
must ensure that the design does not exceed a junction temperature of 125°C. Between 125°C and 150°C,
damage does not occur, but the performance of the amplifier begins to degrade and long-term reliability suffers.
The thermal characteristics of the device are dictated by the package and the PC board. Maximum power
dissipation for a given package can be calculated using the following formula.
T * TA
P Dmax + max
q JA
where:
PDmax is the maximum power dissipation in the amplifier (W).
Tmax is the absolute maximum junction temperature (°C).
TA is the ambient temperature (°C).
θJA = θJC + θCA
θJC is the thermal coefficient from the silicon junctions to the
case (°C/W).
θCA is the thermal coefficient from the case to ambient air
(°C/W).
(4)
For systems where heat dissipation is more critical, the THS3091 and THS3095 are offered in an 8-pin SOIC
(DDA) with PowerPAD package. The thermal coefficient for the PowerPAD packages are substantially improved
over the traditional SOIC. Maximum power dissipation levels are depicted in the graph for the available
packages. The data for the PowerPAD packages assume a board layout that follows the PowerPAD layout
guidelines referenced above and detailed in the PowerPAD application note (SLMA002). If the PowerPAD is not
soldered to the PCB, the thermal impedance will increase substantially which may cause serious heat and
performance issues. Be sure to always solder the PowerPAD to the PCB for optimum performance.
When determining whether or not the device satisfies the maximum power dissipation requirement, it is important
to consider not only quiescent power dissipation, but also dynamic power dissipation. Often times, this is difficult
to quantify because the signal pattern is inconsistent, but an estimate of the RMS power dissipation can provide
visibility into a possible problem.
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 Evaluation Fixtures, Spice Models, and Application Support
Texas Instruments is committed to providing its customers with the highest quality of applications support. To
support this goal, an evaluation board has been developed for the THS309x operational amplifier. The board is
easy to use, allowing for straightforward evaluation of the device. The evaluation board can be ordered through
the Texas Instruments Web site, www.ti.com, or through your local Texas Instruments sales representative.
Computer simulation of circuit performance using SPICE is often useful when analyzing the performance of
analog circuits and systems. This is particularly true for video and RF-amplifier circuits where parasitic
capacitance and inductance can have a major effect on circuit performance. A SPICE model for the THS309x is
available through the Texas Instruments Web site (www.ti.com). The Product Information Center (PIC) is also
available for design assistance and detailed product information. These models do a good job of predicting smallsignal ac and transient performance under a wide variety of operating conditions. They are not intended to model
the distortion characteristics of the amplifier, nor do they attempt to distinguish between the package types in
their small-signal ac performance. Detailed information about what is and is not modeled is contained in the
model file itself.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation, see the following:
• PowerPAD™ Made Easy, application brief (SLMA004)
• PowerPAD™ Thermally Enhanced Package, technical brief (SLMA002)
• Voltage Feedback vs Current Feedback Amplifiers, (SLVA051)
• Current Feedback Analysis and Compensation (SLOA021)
• Current Feedback Amplifiers: Review, Stability, and Application (SBOA081)
• Effect of Parasitic Capacitance in Op Amp Circuits (SLOA013)
• Expanding the Usability of Current-Feedback Amplifiers, 3Q 2003 Analog Applications Journal.
11.3 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 7. Related Links
PARTS
PRODUCT FOLDER
SAMPLE & BUY
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
THS3091
Click here
Click here
Click here
Click here
Click here
THS3095
Click here
Click here
Click here
Click here
Click here
11.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
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11.5 Trademarks
PowerPAD, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
11.6 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
www.ti.com
13-Aug-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
THS3091D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
3091
THS3091DDA
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
3091
THS3091DDAG3
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
3091
THS3091DDAR
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
3091
THS3091DDARG3
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
3091
THS3091DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
3091
THS3095D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
3095
THS3095DDA
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
3095
THS3095DDAR
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 85
3095
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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