THS3110
THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
LOW-NOISE, HIGH-VOLTAGE, CURRENT-FEEDBACK
OPERATIONAL AMPLIFIERS
Check for Samples: THS3110 THS3111
FEATURES
DESCRIPTION
•
The THS3110 and THS3111 are low-noise,
high-voltage, current-feedback amplifiers designed to
operate over a wide supply range of ±5 V to ±15 V for
today's high performance applications.
1
23
•
•
•
•
•
Low Noise
– 2-pA/√Hz Noninverting Current Noise
– 10-pA/√Hz Inverting Current Noise
– 3-nV/√Hz Voltage Noise
High Output Current Drive: 260 mA
High Slew Rate: 1300 V/μs
– (RL = 100 Ω, VO = 8 VPP)
Wide Bandwidth: 90 MHz (G = 2, RL = 100 Ω)
Wide Supply Range: ±5 V to ±15 V
Power-Down Feature: (THS3110 Only)
The THS3110 features a power-down pin (PD) that
puts the amplifier in low-power standby mode, and
lowers the quiescent current from 4.8 mA to 270 μA.
These
amplifiers
provide
well-regulated
ac
performance
characteristics.
The
unity-gain
bandwidth of 100 MHz allows for good distortion
characteristics below 10 MHz. Coupled with a high
1300-V/μs slew rate, the THS3110 and THS3111
amplifiers allow for high output voltage swings at high
frequencies.
APPLICATIONS
•
•
•
•
Video Distribution
Power FET Driver
Pin Driver
Capacitive Load Driver
The THS3110 and THS3111 are offered in the
SOIC-8 (D) and the MSOP-8 (DGN) packages with
PowerPAD™.
space
space
DIFFERENTIAL PHASE
vs
NUMBER OF LOADS
DIFFERENTIAL GAIN
vs
NUMBER OF LOADS
0.3
0.4
Gain = 2,
RF = 1 kΩ,
VS = ±15 V,
40 IRE − NTSC and PAL,
Worst Case ±100 IRE Ramp
PAL
0.15
NTSC
0.1
1 kΩ
1 kΩ
15 V
5
0.2
0.3
Differential Phase −
Differential Gain − %
0.25
VIDEO DISTRIBUTION AMPLIFIER APPLICATION
Gain = 2,
RF = 1 kΩ,
VS = ±15 V,
40 IRE − NTSC and PAL,
Worst Case ±100 IRE Ramp
0.35
0.25
PAL
−
+
VI
0.2
NTSC
75-Ω Transmission Line
75 Ω
−15 V
0.15
75 Ω
0.1
VO(1)
n Lines
75 Ω
VO(n)
75 Ω
0.05
0.05
0
0
0
1
2
3
4
5
6
7
8
0
Number of 150 Ω Loads
1
2
3
4
5
6
Number of 150 Ω Loads
7
8
75 Ω
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2003–2009, Texas Instruments Incorporated
THS3110
THS3111
SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009........................................................................................................................................ www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
TOP VIEW
D, DGN
TOP VIEW
D, DGN
THS3110
REF
VIN−
VIN+
VS−
THS3111
1
8
PD
2
7
3
6
4
5
VS+
VOUT
NC
NC
VIN −
VIN +
VS−
NC = No Internal Connection
1
8
2
7
3
6
4
5
NC
VS+
VOUT
NC
NC = No Internal Connection
NOTE: The device with the power-down option defaults to the ON state if no signal is applied to the PD pin. Additionally, the REF pin
functional range is from VS- to (VS+ - 4 V).
AVAILABLE OPTIONS (1)
TA
PACKAGED DEVICE
PLASTIC SMALL OUTLINE SOIC (D)
0°C to +70°C
–40°C to +85°C
0°C to +70°C
–40°C to +85°C
(1)
(2)
PLASTIC MSOP (DGN)
THS3110CD
THS3110CDGN
THS3110CDR
THS3110CDGNR
THS3110ID
THS3110IDGN
THS3110IDR
THS3110IDGNR
THS3111CD
THS3111CDGN
THS3111CDR
THS3111CDGNR
THS3111ID
THS3111IDGN
THS3111IDR
THS3111IDGNR
(2)
SYMBOL
BJB
BIR
BJA
BIS
For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
The PowerPAD is electrically isolated from all other pins.
DISSIPATION RATINGS TABLE
(1)
(2)
2
POWER RATING
TJ = +125°C
PACKAGE
θJC (°C/W)
θJA (°C/W)
TA = +25°C
TA = +85°C
D-8 (1)
38.3
95
1.05 W
421 mW
DGN-8 (2)
4.7
58.4
1.71 W
685 mW
These data were taken using the JEDEC standard low-K test PCB. For the JEDEC proposed high-K test PCB, the θJA is 95°C/W with
power rating at TA = +25°C of 1.05 W.
These data were taken using 2 oz. trace and copper pad that is soldered directly to a 3 inch × 3 inch (76,2 mm × 76,2 mm) PCB. For
further information, refer to the Application Information section of this data sheet.
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Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
THS3110
THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
RECOMMENDED OPERATING CONDITIONS
MIN
Supply voltage
MAX
±5
±15
Single supply
10
30
0
+70
Commercial
Operating free-air temperature, TA
NOM
Dual supply
–40
+85
Operating junction temperature, continuous operating temperature, TJ
Industrial
–40
+125
Normal storage temperature, TSTG
–40
+85
UNIT
V
°C
ABSOLUTE MAXIMUM RATINGS (1)
Over operating free-air temperature, unless otherwise noted.
UNIT
Supply voltage, VS– to VS+
33 V
Input voltage, VI
± VS
Differential input voltage, VID
±4V
Output current, IO
(2)
300 mA
Continuous power dissipation
Maximum junction temperature, TJ
See Dissipation Ratings Table
(3)
+150°C
Maximum junction temperature, continuous operation, long term reliability, TJ
(4)
Commercial
Operating free-air temperature, TA
Industrial
Storage temperature, Tstg
+125°C
0°C to +70°C
–40°C to +85°C
–65°C to +125°C
ESD ratings:
(1)
(2)
(3)
(4)
HBM
900
CDM
1500
MM
200
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
The THS3110 and THS3111 may incorporate a PowerPAD on the underside of the chip. This feature acts as a heatsink and must be
connected to a thermally dissipating plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction
temperature which could permanently damage the device. See TI Technical Brief SLMA002 for more information about utilizing the
PowerPAD™ thermally-enhanced package.
The absolute maximum temperature under any condition is limited by the constraints of the silicon process.
The maximum junction temperature for continuous operation is limited by package constraints. Operation above this temperature may
result in reduced reliability and/or lifetime of the device.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
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THS3110
THS3111
SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009........................................................................................................................................ www.ti.com
ELECTRICAL CHARACTERISTICS
VS = ±15 V, RF = 1 k Ω,RL = 100 Ω, and G = 2, unless otherwise noted.
TYP
PARAMETER
TEST CONDITIONS
OVER TEMPERATURE
+25°C
+25°C
0°C to
+70°C
–40°C to
+85°C
UNIT
MIN/TYP/
MAX
MHz
TYP
V/μs
TYP
AC PERFORMANCE
G = 1, RF = 1.5 kΩ, VO = 200 mVPP
100
G = 2, RF = 1 kΩ, VO = 200 mVPP
90
G = 5, RF = 806 Ω, VO = 200 mVPP
87
G = 10, RF = 604 Ω, VO = 200 mVPP
66
0.1-dB bandwidth flatness
G = 2, RF = 1.15 kΩ, VO = 200 mVPP
45
Large-signal bandwidth
G = 5, RF = 806 Ω , VO = 4 VPP
95
G = 1, VO = 4-V step, RF = 1.5 kΩ
800
G = 2, VO = 8-V step, RF = 1 kΩ
1300
Slew rate
Recommended maximum SR for
repetitive signals (1)
900
V/μs
MAX
Rise and fall time
G = –5, VO = 10-V step, RF = 806 Ω
8
ns
TYP
Settling time to 0.1%
G = –2, VO = 2 VPP step
27
Settling time to 0.01%
G = –2, VO = 2 VPP step
250
ns
TYP
dBc
TYP
Small-signal bandwidth, –3 dB
Slew rate (25% to 75% level)
Harmonic distortion
2nd harmonic distortion
RL = 100 Ω
52
RL = 1 kΩ
53
RL = 100 Ω
48
RL = 1 kΩ
68
3rd harmonic distortion
G = 2,
RF = 1 kΩ,
VO = 2 VPP,
f = 10 MHz
Input voltage noise
f > 20 kHz
3
nV/√Hz
TYP
Noninverting input current noise
f > 20 kHz
2
pA/√Hz
TYP
Inverting input current noise
f > 20 kHz
10
pA/√Hz
TYP
Differential gain
Differential phase
G = 2,
RL = 150 Ω,
RF = 1 kΩ
NTSC
0.011%
PAL
0.013%
NTSC
0.029°
PAL
0.033°
TYP
DC PERFORMANCE
Transimpedance
Input offset voltage
Average offset voltage drift
Noninverting input bias current
Average bias current drift
Inverting input bias current
Average bias current drift
Input offset current
Average offset current drift
VO = ±3.75 V, gain = 1
VCM = 0 V
VCM = 0 V
VCM = 0 V
VCM = 0 V
1
0.75
0.5
0.5
MΩ
MIN
3
10
12
12
mV
MAX
±10
±10
μV/°C
TYP
1
4
6
6
μA
MAX
±10
±10
nA/°C
TYP
1.5
15
20
20
μA
MAX
±10
±10
nA/°C
TYP
2.5
15
20
20
μA
MAX
±30
±30
nA/°C
TYP
V
MIN
INPUT CHARACTERISTICS
Input common-mode voltage range
Common-mode rejection ratio
VCM = ±12.5 V
±13.3
±13
±12.5
±12.5
68
62
60
60
dB
MIN
Noninverting input resistance
41
MΩ
TYP
Noninverting input capacitance
0.4
pF
TYP
V
MIN
MIN
OUTPUT CHARACTERISTICS
RL = 1 kΩ
±13.5
±13
±12.5
±12.5
RL = 100 Ω
±13.4
±12.5
±12
±12
Output current (sourcing)
RL = 25 Ω
260
200
175
175
mA
Output current (sinking)
RL = 25 Ω
260
200
175
175
mA
MIN
Output impedance
f = 1 MHz, closed loop
0.15
Ω
TYP
Output voltage swing
(1)
4
For more information, see the Application Information section of this data sheet.
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Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
THS3110
THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
ELECTRICAL CHARACTERISTICS (continued)
VS = ±15 V, RF = 1 k Ω,RL = 100 Ω, and G = 2, unless otherwise noted.
TYP
PARAMETER
TEST CONDITIONS
OVER TEMPERATURE
UNIT
MIN/TYP/
MAX
±16
V
MAX
7.5
mA
MAX
2.5
2.5
mA
MIN
+25°C
+25°C
0°C to
+70°C
–40°C to
+85°C
Specified operating voltage
±15
±16
±16
Maximum quiescent current
4.8
6.5
7.5
Minimum quiescent current
4.8
3.8
POWER SUPPLY
Power-supply rejection (+PSRR)
VS+ = 15.5 V to 14.5 V, VS– = 15 V
75
65
60
60
dB
MIN
Power-supply rejection (–PSRR)
VS+ = 15 V, VS– = –15.5 V to –14.5 V
69
60
55
55
dB
MIN
VS+– 4
V
MAX
VS–
V
MIN
Enable
PD ≤
REF+ 0.8
V
MIN
Disable
PD ≥ REF
+2
V
MAX
μA
MAX
μA
TYP
μs
TYP
kΩ || pF
TYP
POWER-DOWN CHARACTERISTICS (THS3110 Only)
REF voltage range
(2)
Power-down voltage level (2)
PD ≥ REF + 2 V
270
VPD = 0 V, REF = 0 V,
11
VPD = 3.3 V, REF = 0 V
11
Turn-on time delay
90% of final value
4
Turn-off time delay
10% of final value
6
Power-down quiescent current
PD pin bias current
Input impedance
(2)
3.4 || 1.7
450
500
500
For detailed information on the behavior of the power-down circuit, see the Saving Power with Power-Down Functionality and
Power-Down Reference Pin Operation sections in the Application Information section of this data sheet.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
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THS3110
THS3111
SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009........................................................................................................................................ www.ti.com
ELECTRICAL CHARACTERISTICS
VS = ±5 V, RF = 1.15 Ω, RL = 100 Ω, and G = 2, unless otherwise noted.
TYP
PARAMETER
TEST CONDITIONS
+25°C
OVER TEMPERATURE
+25°C
0°C to
+70°C
–40°C to
+85°C
UNIT
MIN/TYP/
MAX
MHz
TYP
V/μs
TYP
AC PERFORMANCE
G = 1, RF = 1.5 kΩ, VO = 200 mVPP
85
G = 2, RF = 1.15 kΩ, VO = 200 mVPP
78
G = 5, RF = 806 Ω, VO = 200 mVPP
80
G = 10, RF = 604 Ω, VO = 200 mVPP
60
0.1-dB bandwidth flatness
G = 2, RF = 1.15 kΩ, VO = 200 mVPP
15
Large-signal bandwidth
G = 5, RF = 806 Ω, VO = 4 VPP
80
G = 1, VO = 4-V step, RF = 1.5 kΩ
640
G = 2, VO = 4-V step, RF = 1 kΩ
700
Slew rate
Recommended maximum SR for
repetitive signals (1)
900
V/μs
MAX
Rise and fall time
G = –5, VO = 5-V step, RF = 806 Ω
7
ns
TYP
Settling time to 0.1%
G = –2, VO = 2 VPP step
20
Settling time to 0.01%
G = –2, VO = 2 VPP step
200
ns
TYP
dBc
TYP
Small-signal bandwidth, –3 dB
Slew rate (25% to 75% level)
Harmonic distortion
2nd harmonic distortion
RL = 100 Ω
55
RL = 1 kΩ
56
RL = 100 Ω
45
RL = 1 kΩ
62
3rd harmonic distortion
G = 2,
RF = 1 kΩ,
VO = 2 VPP,
f = 10 MHz
Input voltage noise
f > 20 kHz
3
nV/√Hz
TYP
Noninverting input current noise
f > 20 kHz
2
pA/√Hz
TYP
Inverting input current noise
f > 20 kHz
10
pA/√Hz
TYP
Differential gain
Differential phase
G = 2,
RL = 150 Ω,
RF = 1 kΩ
NTSC
0.011%
PAL
0.015%
NTSC
0.020°
PAL
0.033°
TYP
DC PERFORMANCE
Transimpedance
Input offset voltage
Average offset voltage drift
Noninverting input bias current
Average bias current drift
Inverting input bias current
Average bias current drift
Input offset current
Average offset current drift
VO = ±1.25 V, gain = 1
VCM = 0 V
VCM = 0 V
VCM = 0 V
VCM = 0 V
1
0.75
0.5
0.5
MΩ
MIN
6
10
12
12
mV
MAX
±10
±10
μV/°C
TYP
1
4
6
6
μA
MAX
±10
±10
nA/°C
TYP
1
8
10
10
μA
MAX
±10
±10
nA/°C
TYP
1
6
8
8
μA
MAX
±20
±20
nA/°C
TYP
V
MIN
INPUT CHARACTERISTICS
Input common-mode voltage range
Common-mode rejection ratio
VCM = ±2.5 V
±3.2
±2.9
±2.8
±2.8
65
62
58
58
dB
MIN
Noninverting input resistance
35
MΩ
TYP
Noninverting input capacitance
0.5
pF
TYP
V
MIN
MIN
OUTPUT CHARACTERISTICS
±4
±3.8
±3.6
±3.6
RL = 100 Ω
±3.8
±3.7
±3.5
±3.5
Output current (sourcing)
RL = 10 Ω
220
150
125
125
mA
Output current (sinking)
RL = 10 Ω
220
150
125
125
mA
MIN
Output impedance
f = 1 MHz, closed loop
0.15
Ω
TYP
Output voltage swing
(1)
6
RL = 1 kΩ
For more information, see the Application Information section of this data sheet.
Submit Documentation Feedback
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
THS3110
THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
ELECTRICAL CHARACTERISTICS (continued)
VS = ±5 V, RF = 1.15 Ω, RL = 100 Ω, and G = 2, unless otherwise noted.
TYP
PARAMETER
TEST CONDITIONS
OVER TEMPERATURE
UNIT
MIN/TYP/
MAX
+25°C
+25°C
0°C to
+70°C
–40°C to
+85°C
Specified operating voltage
±5
±4.5
±4.5
±4.5
V
MIN
Maximum quiescent current
4
6
7
7
mA
MAX
POWER SUPPLY
Minimum quiescent current
4
3.2
2
2
mA
MIN
Power-supply rejection (+PSRR)
VS+ = 5.5 V to 4.5 V, VS– = 5 V
71
62
57
57
dB
MIN
Power-supply rejection (–PSRR)
VS+ = 5 V, VS– = –5.5 V to –4.5 V
66
57
52
52
dB
MIN
VS+ –4
V
MAX
VS–
V
MIN
Enable
PD ≤ REF
+ 0.8
V
MIN
Disable
PD ≥ REF
+2
V
MAX
μA
MAX
μA
TYP
μs
TYP
kΩ || pF
TYP
POWER-DOWN CHARACTERISTICS (THS3110 Only)
REF voltage range (2)
Power-down voltage level (2)
PD ≥ REF + 2 V
200
VPD = 0 V, REF = 0 V,
11
VPD = 3.3 V, REF = 0 V
11
Turn-on time delay
90% of final value
4
Turn-off time delay
10% of final value
6
Power-down quiescent current
PD pin bias current
Input impedance
(2)
450
3.4 || 1.7
500
500
For detailed information on the behavior of the power-down circuit, see the Power-Down and Power-down Reference sections in the
Application Information section of this data sheet.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
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7
THS3110
THS3111
SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009........................................................................................................................................ www.ti.com
TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
FIGURE
±15-V Graphs
Noninverting small-signal gain frequency response
1, 2
Inverting small-signal gain frequency response
3
0.1-dB flatness
4
Noninverting large-signal gain frequency response
5
Inverting large-signal gain frequency response
6
Frequency response capacitive load
7
Recommended RISO
vs Capacitive load
8
2nd harmonic distortion
vs Frequency
9
3rd harmonic distortion
vs Frequency
Harmonic distortion
vs Output voltage swing
Slew rate
vs Output voltage step
Noise
vs Frequency
10
11, 12
13, 14, 15, 16
17
Settling time
18, 19
Quiescent current
vs Supply voltage
20
Output voltage
vs Load resistance
21
Input bias and offset current
vs Case temperature
22
Input offset voltage
vs Case temperature
23
Transimpedance
vs Frequency
24
Rejection ratio
vs Frequency
25
Noninverting small-signal transient response
26
Inverting large signal transient response
27
Overdrive recovery time
28
Differential gain
vs Number of loads
29
Differential phase
vs Number of loads
30
Closed loop output impedance
vs Frequency
31
Power-down quiescent current
vs Supply voltage
32
Turn-on and turn-off time delay
33
±5-V Graphs
Noninverting small-signal gain frequency response
34
Inverting small-signal gain frequency response
35
0.1-dB flatness
36
Noninverting large-signal gain frequency response
37
Inverting large-signal gain frequency response
38
Slew rate
vs Output voltage step
2nd harmonic distortion
vs Frequency
3rd harmonic distortion
vs Frequency
Harmonic distortion
vs Output voltage swing
39, 40, 41, 42
43
44
45, 46
Noninverting small-signal transient response
47
Inverting small-signal transient response
48
Overdrive recovery time
Rejection ratio
8
Submit Documentation Feedback
49
vs Frequency
50
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
THS3110
THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
TYPICAL CHARACTERISTICS (±15 V)
space
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
9
RF = 649 Ω
7
Noninverting Gain - dB
6
5
RF = 1.15 kΩ
4
RF = 1.5 kΩ
3
Gain = 2,
RL = 100 Ω,
VO = 0.2 VPP,
VS = ±15 V
2
1
0
1M
10 M
100 M
f - Frequency - Hz
1G
RL = 100 Ω,
VO = 0.2 VPP,
VS = ±15 V
10
8
G = 2, RF = 1.15 kΩ
6
4
2
G = 1, RF = 1.5 kΩ
0
-2
-4
100 k
1M
10 M
100 M
f - Frequency - Hz
1G
G = -1, RF = 1 kΩ
1M
10 M
100 M
16
16
G = 5, RF = 806 Ω
14
12
10
8
2
5.6
0
1M
10 M
f - Frequency - Hz
G = 2, RF = 1 kΩ
4
5.7
100 M
G = -5, RF = 806 Ω
10
8
6
4
2
G =-1, RF = 1 kΩ
0
RL = 100 Ω,
VO = 4 VPP,
VS = ±15 V
1M
RL = 100 Ω,
VO = 2 VPP,
VS = ±15 V
14
12
6
-2
-4
10 M
100 M
f - Frequency - Hz
1M
1G
10 M
100 M
f - Frequency - Hz
Figure 4.
Figure 5.
Figure 6.
FREQUENCY RESPONSE
CAPACITIVE LOAD
RECOMMENDED RISO
vs
CAPACITIVE LOAD
2nd HARMONIC DISTORTION
vs
FREQUENCY
60
16
Gain = 5,
RL = 100 Ω,
VS = ±15 V
14
Recommended RISO - Ω
50
8
R(ISO) = 54.9 Ω
CL = 10 pF
6
‘
R(ISO) = 39.2 Ω
CL = 47 pF
4
2
40
30
20
10
R(ISO) = 28 Ω
CL = 100 pF
0
0
-2
10 M
100 M
100
CL - Capacitive Load - pF
Capacitive Load - MHz
Figure 7.
G = 5, RF = 806 Ω
-40
-50
G = 2, RF = 1 kΩ
-60
-70
-80
G = -2, RF = 1 kΩ
RL = 1 kΩ,
-90
VO = 2 VPP,
RL = 100 Ω,
VS = ±15 V
-100
10
200 M
1G
-30
R(ISO) = 54.9 Ω, CL = 22 pF
Gain = 5,
RL = 100 Ω
VS = ±15 V
1G
f - Frequency - Hz
INVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
5.8
Signal Gain - dB
G = -2, RF = 1.1 kΩ
8
6
4
2
NONINVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
5.9
10
10
0.1-dB FLATNESS
6
12
G = -5, RF = 909 Ω
Figure 3.
Gain = 2,
RF = 1.15 kΩ,
RL = 100 Ω,
VO = 0.2 VPP,
VS = ±15 V
100 k
18
16
14
12
0
-2
-4
Inverting Gain - dB
6.1
G = 5, RF = 806 Ω
RL = 100 Ω,
VO = 0.2 VPP,
VS = ±15 V
G = -10, RF = 649 Ω
Figure 2.
Noninverting Gain - dB
Noninverting Gain - dB
6.2
24
22
20
G = 10, RF = 604 Ω
Figure 1.
6.4
6.3
24
22
20
18
16
14
12
2nd Harmonic Destortion - dBc
Noninverting Gain - dB
8
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
Inverting Gain - dB
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
Figure 8.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
100 k
1M
10 M
100 M
f - Frequency - Hz
Figure 9.
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THS3111
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TYPICAL CHARACTERISTICS (±15 V) (continued)
space
3rd HARMONIC DISTORTION
vs
FREQUENCY
-50
G = 2,
RF = 1 kΩ
-70
-45
-75
G = 5, RF = 806 Ω
-60
-40
HD3
-80
G = -2,
RF = 1 kΩ
RL = 1 kΩ,
-90
-80
HD2
-85
-90
Gain = 2,
RF = 1 kΩ,
RL = 100Ω,
f= 1 MHz
VS = ±15 V
-95
100 k
1M
10 M
-50
-55
HD2
-60
Gain = 2,
RF = 1 kΩ,
RL = 100 Ω,
f = 8 MHz
VS = ±15 V
-70
0
100 M
HD3
-65
-100
-100
1
2
3
4
5
6
7
8
9
10
0
1
VO - Output Voltage Swing - VPP
f - Frequency - Hz
2
3
4
5
6
7
8
Figure 11.
Figure 12.
SLEW RATE
vs
OUTPUT VOLTAGE STEP
SLEW RATE
vs
OUTPUT VOLTAGE STEP
SLEW RATE
vs
OUTPUT VOLTAGE STEP
Gain = 1
RL = 1 kΩ
RF = 1.5 kΩ
VS = ±15 V
Rise
600
400
1000
Gain = 2
RL =100 Ω
RF =1 kΩ
VS = ±15 V
1200
Fall
800
SR - Slew Rate - V/ µ s
1200
SR - Slew Rate - V/ µs
800
Fall
10
9
10
1400
1400
Gain = 1
RL = 100 Ω
RF = 1.5 kΩ
VS = ±15 V
9
VO - Output Voltage Swing - VPP
Figure 10.
1000
SR - Slew Rate - V/ µ s
Harmonic Distortion - dBc
-40
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
-70
VO = 2 VPP,
RL = 100 Ω,
VS = ±15 V
Harmonic Distortion - dBc
2nd Harmonic Destortion - dBc
-30
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
Rise
600
400
1000
Fall
Rise
800
600
400
200
200
200
0
0
0
0.5 1
1.5 2
2.5 3
3.5 4
0
4.5 5
VO - Output Voltage -VPP
1.5 2
2.5
3
3.5 4
Figure 13.
Figure 14.
SLEW RATE
vs
OUTPUT VOLTAGE STEP
NOISE
vs
FREQUENCY
0
Gain = 2
RL =1 kΩ
RF =1 kΩ
VS = ±15 V
400
6
5
6
8
1
In-
10
Vn
In+
7
8
9
VO - Output Voltage -VPP
Figure 16.
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10
0.5
Gain = -2
RL = 100 Ω
RF = 1.1 kΩ
VS = ±15 V
0
-0.5
Falling Edge
-1
1
0
4
7
SETTLING TIME
200
3
5
Figure 15.
VO - Output Voltage - V
Hz
600
2
4
1.5
I n - Current Noise - pA/ Hz
Rise
800
1
3
Rising Edge
1000
0
2
Fall
V n - Voltage Noise - nV/
1200
1
VO - Output Voltage -VPP
100
1400
10
0
4.5 5
VO - Output Voltage -VPP
1600
SR - Slew Rate - V/ µ s
0.5 1
-1.5
10
100
1k
10 k
100 k
f - Frequency - Hz
0
2
4
6
8
10
12
14
16
18
t - Time - ns
Figure 17.
Figure 18.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
THS3110
THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
TYPICAL CHARACTERISTICS (±15 V) (continued)
space
QUIESCENT CURRENT
vs
SUPPLY VOLTAGE
SETTLING TIME
6
3
2.5
TA = 85 °C
1
0.5
Gain = -2
RL = 100 Ω
RF = 1.1 kΩ
VS = ±15 V
0
-0.5
-1
-1.5
-2
5
TA = 25 °C
4
TA = -40 °C
3
2
VO - Output Voltage - V
1.5
I Q - Quiescent Current - mA
Rising Edge
2
VO - Output Voltage - V
OUTPUT VOLTAGE
vs
LOAD RESISTANCE
1
Falling Edge
-2.5
-3
0
2
4
6
8
10 12 14 16 18 20
2
9 10 11 12 13 14 15
10
100
INPUT OFFSET VOLTAGE
vs
CASE TEMPERATURE
TRANSIMPEDANCE
vs
FREQUENCY
IIB-
2
IOS
1
IIB+
4
110
3.5
100
3
2.5
VS = ±5 V
2
1.5
VS = ±15 V
1
0.5
1000
RL - Load Resistance - Ω
INPUT BIAS AND
OFFSET CURRENT
vs
CASE TEMPERATURE
VOS - Input Offset Voltage - mV
I IB - Input Bias Current - µ A
I OS - Input Offset Current - µ A
8
Figure 21.
3
0.5
7
Figure 20.
VS = ±15 V
1.5
5 6
Figure 19.
3.5
2.5
3 4
VS = ±15 V
TA = -40° to 85°C
VS - Supply Voltage - ±V
t - Time - ns
Transimpedance Gain - dB
0
16
14
12
10
8
6
4
2
0
-2
-4
-6
-8
-10
-12
-14
-16
VS = ±15 V and ±5 V
90
80
70
60
50
40
30
20
10
0
-40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90
0
-40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90
TC - Case Temperature - °C
TC - Case Temperature - °C
Figure 22.
Figure 23.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
0
0.1
1
10
100
Frequency - MHz
1000
Figure 24.
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THS3110
THS3111
SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009........................................................................................................................................ www.ti.com
TYPICAL CHARACTERISTICS (±15 V) (continued)
space
REJECTION RATIO
vs
FREQUENCY
NONINVERTING SMALL-SIGNAL
TRANSIENT RESPONSE
VS = ±15 V
VO - Output Voltage - V
50
PSRR+
40
30
Output
4
0.1
0.05
Input
0
-0.05
PSRR20
10
Gain = 2,
RL = 100 Ω,
RF = 1 kΩ,
VS = ±15 V
-0.1
-0.15
0
10 M
1M
100 M
0
10 20
f - Frequency - Hz
2
1
Input
0
-1
-2
-3
Output
-5
-6
30 40 50 60
0
70 80
10 20 30 40 50 60
t - Time - ns
Figure 26.
Figure 27.
OVERDRIVE RECOVERY TIME
DIFFERENTIAL GAIN
vs
NUMBER OF LOADS
DIFFERENTIAL PHASE
vs
NUMBER OF LOADS
0.3
5
0
0
-5
-2.5
-10
-15
Differential Gain - %
0.25
2.5
VI - Input Voltage - V
10
0.4
Gain = 2,
RF = 1 kΩ,
VS = ±15 V,
40 IRE - NTSC and PAL,
Worst Case ±100 IRE Ramp
0.2
0.3
PAL
0.15
NTSC
0.1
Gain = 2,
RF = 1 kΩ,
VS = ±15 V,
40 IRE - NTSC and PAL,
Worst Case ±100 IRE Ramp
0.35
Differential Phase - 5
5
Gain = 4,
RL = 100 Ω,
RF = 681 Ω,
VS = ±15 V
15
70 80
t - Time - ns
Figure 25.
20
0.25
PAL
0.2
NTSC
0.15
0.1
0.05
0.05
-5
-20
0
0.2
0.4
0.6
0.8
1
t - Time - µs
0
0
0
1
2
3
4
5
6
7
8
Figure 28.
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0
1
2
3
4
5
6
7
8
Number of 150 Ω Loads
Number of 150 Ω Loads
12
3
-4
-0.2
100 k
Gain = -5,
RL = 100 Ω,
RF = 909 Ω,
VS = ±15 V
5
0.15
CMRR
VO - Output Voltage - V
60
Rejection Ratio - dB
6
0.2
70
VO - Output Voltage - V
INVERTING LARGE-SIGNAL
TRANSIENT RESPONSE
Figure 29.
Figure 30.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
THS3110
THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
TYPICAL CHARACTERISTICS (±15 V) (continued)
space
POWER-DOWN QUIESCENT
CURRENT
vs
SUPPLY VOLTAGE
1
0.1
0.01
100 k
1.5
1
300
TA = 85°C
250
10 M
100 M
TA = -40°C
150
TA = 25°C
100
5
7
9
11
13
15
VS - Supply Voltage - ±V
f - Frequency - Hz
Figure 31.
Powerdown Pulse
Figure 32.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
6
5
4
3
Gain = 5,
VI = 0.1 Vdc
RL = 100 Ω
VS = ±15 V and ±5 V
50
3
1G
0
−0.5
200
0
1M
Output Voltage
0.5
0
0.1 0.2
0.3
2
PowerDown Pulse − V
Gain = 2,
RF = 1 kΩ,
RF = 100 Ω,
VS = ±15 V
VO − Output Voltage Level − V
10
TURN-ON AND TURN-OFF
TIME DELAY
350
100
Powerdown Quiescent Current - µ A
ZO - Closed-Loop Output Impedance -
Ω
CLOSED-LOOP OUTPUT
IMPEDANCE
vs
FREQUENCY
1
0
−1
0.4 0.5 0.6 0.7
t − Time − ms
Figure 33.
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THS3110
THS3111
SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009........................................................................................................................................ www.ti.com
TYPICAL CHARACTERISTICS (±5 V)
space
G = 10, RF = 604 Ω
RL = 100 Ω,
VO = 0.2 VPP,
VS = ±5 V
G = 5, RF = 806 Ω
10
8
G = 2, RF = 1.15 kΩ
6
4
2
0
G = 1, RF = 1.5 kΩ
−2
−4
1M
10 M
24
22
20
18
16
14
RL = 100 Ω,
VO = 0.2 VPP,
VS = ±5 V
G = -2, RF = 1.1 kΩ
6
4
2
0
-2
-4
G = -1, RF = 1 kΩ
1M
6
5.9
5.8
100 M
10 M
5.6
100 k
1G
10 M
1M
f - Frequency - Hz
100 M
Figure 34.
Figure 35.
Figure 36.
NONINVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE-SIGNAL
FREQUENCY RESPONSE
SLEW RATE
vs
OUTPUT VOLTAGE STEP
16
G = 5, RF = 806 Ω
800
G = -5, RF = 909 Ω
14
14
10
8
G = 2, RF = 1.15 kΩ
6
10
VO = 2 VPP,
RL = 100 Ω,
VS = ±5 V
8
6
4
2
4
VO = 4 VPP,
RL = 100 Ω,
VS = ±5 V
2
SR - Slew Rate - V/ µ s
Inverting Gain - dB
12
G =-12, RF = 1 kΩ
Rise
500
400
300
200
100
-2
-4
10 M
600
Fall
0
0
1M
Gain = 1
RL = 100 Ω
RF = 1.5 kΩ
VS = ±5 V
700
12
100 M
1G
10 M
1
100 M
0
1G
0
0.5 1
f - Frequency - Hz
f - Frequency - Hz
1.5 2
2.5 3
3.5 4
Figure 37.
Figure 38.
Figure 39.
SLEW RATE
vs
OUTPUT VOLTAGE STEP
SLEW RATE
vs
OUTPUT VOLTAGE STEP
SLEW RATE
vs
OUTPUT VOLTAGE STEP
800
800
800
700
600
Rise
Gain = 2
RL = 100 Ω
RF = 1 kΩ
VS = ±5 V
700
SR - Slew Rate - V/ µ s
Gain = 1
RL = 1 kΩ
RF = 1.5 kΩ
VS = ±5 V
Fall
500
400
300
200
100
0
600
0.5 1 1.5 2 2.5 3 3.5 4
VO - Output Voltage -VPP
4.5 5
Figure 40.
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Gain = 2
RL = 1 kΩ
RF = 1 kΩ
VS = ±5 V
Fall
700
Rise
500
400
300
600
Rise
Fall
500
400
300
200
200
100
100
0
0
4.5 5
VO - Output Voltage -VPP
SR - Slew Rate - V/ µ s
Noninverting Gain - dB
6.1
f - Frequency - Hz
16
SR - Slew Rate - V/ µ s
6.2
5.7
f − Frequency − Hz
14
Gain = 2,
RF = 1.15 kΩ,
RL = 100 Ω,
VO = 0.2 VPP,
VS = ±5 V
6.3
G = -5, RF = 909 Ω
12
10
8
1G
100 M
G = -10, RF = 649 Ω
0.1-dB FLATNESS
6.4
Noninverting Gain - dB
24
22
20
18
16
14
12
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
Inverting Gain - dB
Noninverting Gain − dB
NONINVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
0
0
1
2
3
4
5
6
VO - Output Voltage -VPP
Figure 41.
0
1
2
3
4
VO - Output Voltage -VPP
5
6
Figure 42.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
THS3110
THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
TYPICAL CHARACTERISTICS (±5 V) (continued)
space
3rd HARMONIC DISTORTION
vs
FREQUENCY
-30
G = 5, RF = 681 Ω
-40
G = 2, RF = 681 Ω
-50
-60
-70
G = -2, RF = 1 kΩ
RL = 1 kΩ,
-80
VO = 2 VPP,
RL = 100 Ω,
VS = ±5 V
-90
-70
-50
G = 5,
RF = 681 Ω
-60
-70
G = 2,
RF = 681 Ω
-80
G = -2, RF = 1 kΩ
RL = 1 kΩ,
-90
-100
1M
10 M
100 M
HD3
-75
-80
HD2
-85
Gain = 2,
RF = 1.15 kΩ
RL = 100 Ω,
f= 1 MHz
VS = ±5 V
-90
-95
-100
-100
100 k
1M
100 k
f - Frequency - Hz
10 M
0
100 M
1
2
3
4
5
Figure 43.
Figure 44.
Figure 45.
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
NONINVERTING SMALL-SIGNAL
TRANSIENT RESPONSE
INVERTING LARGE-SIGNAL
TRANSIENT RESPONSE
-70
Output
2
0.1
VO - Output Voltage - V
HD2
-60
Input
0.05
0
-0.05
Gain = 2,
RF = 1 kΩ
RL = 100 Ω,
f= 8 MHz
VS = ±5 V
-80
Gain = 2
RL = 100 Ω
RF = 1.15 kΩ
VS = ±5 V
-0.1
-0.15
0
-1.5
-2
Output
0
10 20 30 40 50 60 70 80
REJECTION RATIO
vs
FREQUENCY
70
0.75
0.5
1
0.25
0
0
-1
-0.25
-2
-0.5
-3
-0.75
-4
-1
-1.25
0.4
0.6
60
0.8
1
CMRR
Rejection Ratio - dB
2
VS = ±5 V
1
VI - Input Voltage - V
3
50 60 70 80
Figure 48.
1.25
Gain = 4,
RL = 100 Ω,
RF = 909 Ω,
VS = ±5 V
4
10 20 30 40
t - Time - ns
OVERDRIVE RECOVERY TIME
VO - Output Voltage - V
-0.5
-1
Figure 47.
5
0.2
Input
0
t - Time - ns
Figure 46.
0
1
0.5
-3
0
0.5 1 1.5 2 2.5 3 3.5 4 4.5 5
VO - Output Voltage Swing - VPP
-5
1.5
-2.5
-0.2
-90
Gain = -5,
RL = 100 Ω,
RF = 909 Ω,
VS = ±5 V
2.5
0.15
VO - Output Voltage - V
-50
7
3
0.2
HD3
6
VO - Output Voltage Swing - VPP
f - Frequency - Hz
-40
Harmonic Distortion - dBc
-65
VO = 2 VPP,
RL = 100 Ω,
VS = ±5 V
-40
2nd Harmonic Destortion - dBc
2nd Harmonic Destortion - dBc
-30
HARMONIC DISTORTION
vs
OUTPUT VOLTAGE SWING
Harmonic Distortion - dBc
2nd HARMONIC DISTORTION
vs
FREQUENCY
50
PSRR+
40
30
20
10
PSRR-
0
100 k
t - Time - µs
1M
10 M
100 M
f - Frequency - Hz
Figure 49.
Figure 50.
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
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THS3110
THS3111
SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009........................................................................................................................................ www.ti.com
APPLICATION INFORMATION
MAXIMUM SLEW RATE FOR REPETITIVE
SIGNALS
The THS3110 and THS3111 are recommended for
high slew rate pulsed applications where the internal
nodes of the amplifier have time to stabilize between
pulses. It is recommended to have at least 20-ns
delay between pulses.
The THS3110 and THS3111 are not recommended
for applications with repetitive signals (sine, square,
sawtooth, or other) that exceed 900 V/μs. Using the
part in these applications results in excessive current
draw from the power supply and possible device
damage.
For applications with high slew rate, repetitive signals,
the THS3091 and THS3095 (single), or THS3092 and
THS3096 (dual) are recommended.
WIDEBAND, NONINVERTING OPERATION
The THS3110 and THS3111 are unity-gain stable,
100-MHz, current-feedback operational amplifiers,
designed to operate from a ±5-V to ±15-V power
supply.
Figure 51 shows the THS3111 in a noninverting gain
of 2-V/V configuration typically used to generate the
performance curves. Most of the curves were
characterized using signal sources with 50-Ω source
impedance, and with measurement equipment
presenting a 50-Ω load impedance.
15 V
+VS
+
0.1 µF
50 Ω Source
+
VI
6.8 µF
49.9 Ω
THS3110
49.9 Ω
_
50 Ω LOAD
RF
1 kΩ
1 kΩ
RG
0.1 µF
6.8 µF
+
-VS
-15 V
Figure 51. Wideband, Noninverting Gain
Configuration
Current-feedback amplifiers are highly dependent on
the feedback resistor RF for maximum performance
and stability. Table 1 shows the optimal gain setting
resistors RF and RG at different gains to give
maximum bandwidth with minimal peaking in the
frequency response. Higher bandwidths can be
achieved, at the expense of added peaking in the
frequency response, by using even lower values for
RF. Conversely, increasing RF decreases the
bandwidth, but stability is improved.
Table 1. Recommended Resistor Values for
Optimum Frequency Response
THS3110 AND THS3111 RF AND RG VALUES FOR MINIMAL PEAKING WITH RL = 100 Ω
GAIN (V/V)
1
2
5
RG (Ω)
RF (Ω)
±15
—
1.5 k
±5
—
1.5 k
±15
1k
1k
±5
1.15 k
1.15 k
±15
200
806
±5
200
806
±15
66.5
604
±5
66.5
604
±15
1k
1k
±5
1k
1k
–2
±15 and ±5
549
1.1 k
–5
±15 and ±5
182
909
–10
±15 and ±5
64.9
649
10
–1
16
SUPPLY VOLTAGE (V)
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THS3111
www.ti.com........................................................................................................................................ SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009
WIDEBAND, INVERTING OPERATION
+VS
Figure 52 shows the THS3111 in a typical inverting
gain configuration where the input and output
impedances and signal gain from Figure 51 are
retained in an inverting circuit configuration.
50 Ω Source
+
VI
49.9 Ω
RT
15 V +VS
0.1 µF
THS3110
VI
RG
RF
6.8 µF
RF
50 Ω LOAD
1.1 kΩ
VS
50 Ω Source
1.1 kΩ
RG
VI
0.1 µF
6.8 µF
_
549 Ω
RT
56.2 Ω
+
-15 V
1 kΩ
+VS
2
RF
549 Ω
RM
56.2 Ω
50 Ω LOAD
RG
1 kΩ
49.9 Ω
_
50 Ω Source
_
+VS
2
+
+
49.9 Ω
THS3110
Figure 52. Wideband, Inverting Gain
Configuration
+
50 Ω LOAD
+VS
2
+VS
2
-VS
49.9 Ω
THS3110
Figure 53. DC-Coupled, Single-Supply Operation
Video Distribution
SINGLE-SUPPLY OPERATION
The THS3110 and THS3111 have the capability to
operate from a single-supply voltage ranging from
10 V to 30 V. When operating from a single power
supply, biasing the input and output at mid-supply
allows for the maximum output voltage swing. The
circuits shown in Figure 53 shows inverting and
noninverting amplifiers configured for single supply
operations.
The wide bandwidth, high slew rate, and high output
drive current of the THS3110 and THS3111 match
the demands for video distribution for delivering video
signals down multiple cables. To ensure high signal
quality with minimal degradation of performance, a
0.1-dB gain flatness should be at least 7x the
passband frequency to minimize group delay
variations from the amplifier. A high slew rate
minimizes distortion of the video signal, and supports
component video and RGB video signals that require
fast transition times and fast settling times for high
signal quality.
1 kΩ
1 kΩ
15 V
+
VI
75 Ω
75-Ω Transmission Line
-15 V
75 Ω
n Lines
VO(1)
75 Ω
VO(n)
75 Ω
75 Ω
Figure 54. Video Distribution Amplifier
Application
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THS3111
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Driving Capacitive Loads
Applications such as FET drivers and line drivers can
be highly capacitive and cause stability problems for
high-speed amplifiers.
Figure 55 through Figure 61 show recommended
methods for driving capacitive loads. The basic idea
is to use a resistor or ferrite chip to isolate the phase
shift at high frequency caused by the capacitive load
from the amplifier feedback path. See Figure 55 for
recommended resistor values versus capacitive load.
frequency load independence of the amplifier while
isolating the phase shift caused by the capacitance at
high frequency. Use a ferrite chip with similar
impedance to RISO, 20 Ω to 50 Ω, at 100 MHz and
low impedance at dc.
806 Ω
VS
200 Ω
Ferrite Bead
_
+
60
Recommended RISO - Ω
50
40
30
Figure 57. Ferrite Bead to Isolate Capacitive Load
20
0
10
100
CL - Capacitive Load - pF
Figure 55. Recommended RISO vs Capacitive
Load
Placing a small series resistor, RISO, between the
amplifier output and the capacitive load, as shown in
Figure 56, is an easy way of isolating the load
capacitance.
806 Ω
Figure 58 shows another method used to maintain
the low frequency load independence of the amplifier
while isolating the phase shift caused by the
capacitance at high frequency. At low frequency,
feedback is mainly from the load side of RISO. At high
frequency, the feedback is mainly via the 27-pF
capacitor. The resistor RIN in series with the negative
input is used to stabilize the amplifier and should be
equal to the recommended value of RF at unity gain.
Replacing RIN with a ferrite of similar impedance at
about 100 MHz as shown in Figure 59 gives similar
results with reduced dc offset and low frequency
noise. (See the Additional Reference Material section
for expanding the usability of current-feedback
amplifiers.)
RF
VS
_
5.11 Ω
+
RISO
-VS
VS
100 Ω LOAD
27 pF
750 Ω
RG
49.9 Ω
200 Ω
Figure 56. Resistor to Isolate Capacitive Load
Using a ferrite chip in place of RISO, as shown in
Figure 57, is another approach of isolating the output
of the amplifier. The ferrite impedance characteristic
versus frequency is useful to maintain the low
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806 Ω
RIN
1 µF
VS
_
100 Ω LOAD
5.11 Ω
+
-VS
18
100 Ω LOAD
49.9 Ω
VS
10
200 Ω
1 µF
-VS
Gain = 5,
RL = 100 Ω,
VS = ±15 V
1 µF
49.9 Ω
VS
Figure 58. Feedback Technique with Input
Resistor for Capacitive Load
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Figure 61 shows a push-pull FET driver circuit typical
of ultrasound applications with isolation resistors to
isolate the gate capacitance from the amplifier.
RF
27 pF
806 Ω
FIN
RG
FB
200 Ω
VS
VS
_
5.11 Ω
+
_
+
1 µF
-VS
VS
100 Ω LOAD
5.11 Ω
VS
-VS
49.9 Ω
301 Ω
66.5 Ω
301 Ω
Figure 59. Feedback Technique with Input Ferrite
Bead for Capacitive Load
VS
_
Figure 60 shows how to use two amplifiers in parallel
to double the output drive current to larger capacitive
loads. This technique is used when more output
current is needed to charge and discharge the load
faster like when driving large FET transistors.
806 Ω
200 Ω
24.9 Ω
5.11 Ω
+
806 Ω
200 Ω
24.9 Ω
VS
_
-VS
-VS
Figure 61. PowerFET Drive Circuit
The THS3110 features a power-down pin (PD) which
lowers the quiescent current from 4.8 mA down to
270 μA, ideal for reducing system power.
-VS
VS
+
SAVING POWER WITH POWER-DOWN
FUNCTIONALITY AND SETTING
THRESHOLD LEVELS WITH THE
REFERENCE PIN
VS
_
5.11 Ω
1 nF
5.11 Ω
+
-VS
Figure 60. Parallel Amplifiers for Higher Output
Drive
The power-down pin of the amplifier defaults to the
REF pin voltage in the absence of an applied voltage,
putting the amplifier in the normal on mode of
operation. To turn off the amplifier in an effort to
conserve power, the power-down pin can be driven
towards the positive rail. The threshold voltages for
power-on and power-down are relative to the supply
rails and are given in the specification tables. Below
the Enable Threshold Voltage, the device is on.
Above the Disable Threshold Voltage, the device is
off. Behavior in between these threshold voltages is
not specified.
Note that this power-down functionality is just that;
the amplifier consumes less power in power-down
mode. The power-down mode is not intended to
provide a high-impedance output. In other words, the
power-down functionality is not intended to allow use
as a 3-state bus driver. When in power-down mode,
the impedance looking back into the output of the
amplifier is dominated by the feedback and gain
setting resistors, but the output impedance of the
device itself varies depending on the voltage applied
to the outputs.
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Figure 62 shows the total system output impedance
which includes the amplifier output impedance in
parallel with the feedback plus gain resistors, which
cumulate to 1870 Ω. Figure 51 shows this circuit
configuration for reference.
Powerdown Output Impedance - Ω
2000
1800
1600
1400
1200
1000
800
600
400
200
Gain = 2
RF = 1 kΩ
VS = ±15 V and ±5 V
0
100 k
1M
10 M
100 M
f - Frequency - Hz
1G
Figure 62. Power-Down Output Impedance vs
Frequency
As with most current feedback amplifiers, the internal
architecture places some limitations on the system
when in power-down mode. Most notably is the fact
that the amplifier actually turns ON if there is a ±0.7 V
or greater difference between the two input nodes
(V+ and V–) of the amplifier. If this difference
exceeds ±0.7 V, the output of the amplifier creates an
output voltage equal to approximately [(V+ – V–) –
0.7 V] × Gain. Also, if a voltage is applied to the
output while in power-down mode, the V– node
voltage is equal to VO(applied) × RG/(RF + RG). For low
gain configurations and a large applied voltage at the
output, the amplifier may actually turn ON due to the
aforementioned behavior.
The time delays associated with turning the device on
and off are specified as the time it takes for the
amplifier to reach either 10% or 90% of the final
output voltage. The time delays are in the order of
microseconds because the amplifier moves in and out
of the linear mode of operation in these transitions.
20
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POWER-DOWN REFERENCE PIN
OPERATION
In addition to the power-down pin, the THS3110
features a reference pin (REF) which allows the user
to control the enable or disable power-down voltage
levels applied to the PD pin. In most split-supply
applications, the reference pin is connected to
ground. In either case, the user needs to be aware of
voltage-level thresholds that apply to the power-down
pin. The tables below show examples and illustrate
the relationship between the reference voltage and
the power-down thresholds. In the table, the threshold
levels are derived by the following equations:
PD ≤ REF + 0.8 V for enable
PD ≥ REF + 2.0 V for disable
where the usable range at the REF pin is
VS– ≤ VREF ≤ (VS+ – 4 V).
The recommended mode of operation is to tie the
REF pin to midrail, thus setting the enable/disable
thresholds to Vmidrail + 0.8 V and Vmidrail + 2 V
respectively.
POWER-DOWN THRESHOLD VOLTAGE LEVELS
SUPPLY
VOLTAGE (V)
REFERENCE PIN
VOLTAGE (V)
ENABLE
LEVEL (V)
DISABLE
LEVEL (V)
±15, ±5
0.0
0.8
2.0
±15
2.0
2.8
4
±15
–2.0
–1.2
0
±5
1.0
1.8
3
±5
–1.0
–0.2
1
+30
15
15.8
17
+10
5.0
5.8
7
Note that if the REF pin is left unterminated, it floats
to the positive rail and falls outside of the
recommended operating range given above (VS– ≤
VREF ≤ VS+ – 4 V). As a result, it no longer serves as
a reliable reference for the PD pin and the
enable/disable thresholds given above no longer
apply. If the PD pin is also left unterminated, it also
floats to the positive rail and the device is disabled. If
balanced, split supplies are used (±VS) and the REF
and PD pins are grounded, the device is enabled.
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PRINTED-CIRCUIT BOARD LAYOUT
TECHNIQUES FOR OPTIMAL
PERFORMANCE
Achieving
optimum
performance
with
a
high-frequency amplifier, such as the THS3110 and
THS3111, requires careful attention to board layout
parasitic
and
external
component
types.
Recommendations that optimize performance include:
• Minimize parasitic capacitance to any ac ground
for all of the signal I/O pins. Parasitic capacitance
on the output and input pins can cause instability.
To reduce unwanted capacitance, a window
around the signal I/O pins should be opened in all
of the ground and power planes around those
pins. Otherwise, ground and power planes should
be unbroken elsewhere on the board.
• Minimize the distance [< 0.25 inch (6,35 mm)]
from the power-supply pins to high frequency
0.1-μF and 100-pF decoupling capacitors. At the
device pins, the ground and power plane layout
should not be in close proximity to the signal I/O
pins. Avoid narrow power and ground traces to
minimize inductance between the pins and the
decoupling
capacitors.
The
power-supply
connections should always be decoupled with
these capacitors. Larger (6.8 μF or more)
tantalum decoupling capacitors, effective at lower
frequency, should also be used on the main
supply pins. These may be placed somewhat
farther from the device and may be shared among
several devices in the same area of the PC board.
• Careful selection and placement of external
components
preserve
the
high-frequency
performance of the THS3110 and THS3111.
Resistors should be a very low reactance type.
Surface-mount resistors work best and allow a
tighter overall layout. Again, keep their leads and
PC board trace length as short as possible. Never
use wirewound-type resistors in a high-frequency
application. Because the output pin and inverting
input pins are the most sensitive to parasitic
capacitance, always position the feedback and
series output resistors, if any, as close as possible
to the inverting input pins and output pins. Other
network components, such as input termination
resistors, should be placed close to the
gain-setting resistors. Even with a low parasitic
capacitance shunting the external resistors,
excessively high resistor values can create
significant time constants that can degrade
performance.
Good
axial
metal-film
or
surface-mount resistors have approximately 0.2
pF in shunt with the resistor. For resistor values
greater than 2.0 kΩ, this parasitic capacitance can
add a pole and/or a zero that can affect circuit
operation. Keep resistor values as low as
possible,
consistent
with
load
driving
considerations.
•
•
Connections to other wideband devices on the
board may be made with short direct traces or
through onboard transmission lines. For short
connections, consider the trace and the input to
the next device as a lumped capacitive load.
Relatively wide traces [0.05 inch (1,3 mm) to 0.1
inch (2,54 mm)] should be used, preferably with
ground and power planes opened up around
them. Estimate the total capacitive load and
determine if isolation resistors on the outputs are
necessary. Low parasitic capacitive loads (< 4 pF)
may not need an RS since the THS3110 and
THS3111 are nominally compensated to operate
with a 2-pF parasitic load. Higher parasitic
capacitive loads without an RS are allowed as the
signal gain increases (increasing the unloaded
phase margin). If a long trace is required, and the
6-dB signal loss intrinsic to a doubly-terminated
transmission line is acceptable, implement a
matched impedance transmission line using
microstrip or stripline techniques (consult an ECL
design handbook for microstrip and stripline layout
techniques). A 50-Ω environment is not necessary
onboard, and in fact, a higher impedance
environment improves distortion as shown in the
distortion versus load plots. With a characteristic
board trace impedance based on board material
and trace dimensions, a matching series resistor
into the trace from the output of the
THS3110/THS3111 is used as well as a
terminating shunt resistor at the input of the
destination device. Remember also that the
terminating impedance is the parallel combination
of the shunt resistor and the input impedance of
the destination device: this total effective
impedance should be set to match the trace
impedance. If the 6-dB attenuation of a
doubly-terminated
transmission
line
is
unacceptable,
a
long
trace
can
be
series-terminated at the source end only. Treat
the trace as a capacitive load in this case. This
does not preserve signal integrity as well as a
doubly-terminated line. If the input impedance of
the destination device is low, there is some signal
attenuation due to the voltage divider formed by
the series output into the terminating impedance.
Socketing a high-speed part like the THS3110 and
THS3111 is not recommended. The additional
lead length and pin-to-pin capacitance introduced
by the socket can create an extremely
troublesome parasitic network which can make it
almost impossible to achieve a smooth, stable
frequency response. Best results are obtained by
soldering the THS3110/THS3111 parts directly
onto the board.
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THS3111
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PowerPAD DESIGN CONSIDERATIONS
PowerPAD LAYOUT CONSIDERATIONS
The THS3110 and THS3111 are available in a
thermally-enhanced PowerPAD family of packages.
These packages are constructed using a downset
leadframe upon which the die is mounted (see
Figure 63a and Figure 63b). This arrangement results
in the lead frame being exposed as a thermal pad on
the underside of the package (see Figure 63c).
Because this thermal pad has direct thermal contact
with the die, excellent thermal performance can be
achieved by providing a good thermal path away from
the thermal pad. Note that devices such as the
THS311x have no electrical connection between the
PowerPAD and the die.
1. PCB with a top side etch pattern as shown in
Figure 64. There should be etch for the leads as
well as etch for the thermal pad.
The PowerPAD package allows for both assembly
and thermal management in one manufacturing
operation. During the surface-mount solder operation
(when the leads are being soldered), the thermal pad
can also be soldered to a copper area underneath the
package. Through the use of thermal paths within this
copper area, heat can be conducted away from the
package into either a ground plane or other heat
dissipating device.
The PowerPAD package represents a breakthrough
in combining the small area and ease of assembly of
surface mount with the, heretofore, awkward
mechanical methods of heatsinking.
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
Bottom View (c)
Figure 63. Views of Thermal Enhanced Package
Although there are many ways to properly heatsink
the PowerPAD package, the following steps illustrate
the recommended approach.
space
space
space
space
space
space
space
space
space
space
22
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0.205
(5,21)
0.060
(1,52)
0.017
(0,432)
0.013
(0,33)
0.075
(1,91)
0.025
(0,64)
0.030
(0,76)
0.010
(0,254)
vias
0.035
(0,89)
0.094
(2,39)
0.040
(1,01)
Dimensions are in inches (mm).
Figure 64. DGN PowerPAD PCB Etch and
Via Pattern
2. Place five holes in the area of the thermal pad.
These holes should be 0.01 inch (0,254 mm) in
diameter. Keep them small so that solder wicking
through the holes is not a problem during reflow.
3. Additional vias may be placed anywhere along
the thermal plane outside of the thermal pad
area. This helps dissipate the heat generated by
the THS3110/THS3111 IC. These additional vias
may be larger than the 0.01-inch (0,254 mm)
diameter vias directly under the thermal pad.
They can be larger because they are not in the
thermal pad area to be soldered so that wicking
is not a problem.
4. Connect all holes to the internal ground plane.
Note that the PowerPAD is electrically isolated
from the silicon and all leads. Connecting the
PowerPAD to any potential voltage such as VS–,
is acceptable as there is no electrical connection
to the silicon.
5. When connecting these holes to the ground
plane, do not use the typical web or spoke via
connection methodology. Web connections have
a high thermal resistance connection that is
useful for slowing the heat transfer during
soldering operations. This makes the soldering of
vias that have plane connections easier. In this
application, however, low thermal resistance is
desired for the most efficient heat transfer.
Therefore,
the
holes
under
the
THS3110/THS3111 PowerPAD package should
make their connection to the internal ground
plane with a complete connection around the
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entire circumference of the plated-through hole.
6. The top-side solder mask should leave the
terminals of the package and the thermal pad
area with its five holes exposed. The bottom-side
solder mask should cover the five holes of the
thermal pad area. This prevents solder from
being pulled away from the thermal pad area
during the reflow process.
7. Apply solder paste to the exposed thermal pad
area and all of the IC terminals.
8. With these preparatory steps in place, the IC is
simply placed in position and run through the
solder reflow operation as any standard
surface-mount component. This results in a part
that is properly installed.
Maximum power dissipation levels are depicted in
Figure 65 for the available packages. The data for the
PowerPAD packages assume a board layout that
follows the PowerPAD layout guidelines referenced
above and detailed in the PowerPAD application note
(literature number SLMA002). Figure 65 also
illustrates the effect of not soldering the PowerPAD to
a PCB. The thermal impedance increases
substantially which may cause serious heat and
performance issues. Be sure to always solder the
PowerPAD to the PCB for optimum performance.
TJ = 125°C
POWER DISSIPATION AND THERMAL
CONSIDERATIONS
The THS3110 and THS3111 incorporate automatic
thermal shutoff protection. This protection circuitry
shuts down the amplifier if the junction temperature
exceeds approximately +160°C. When the junction
temperature reduces to approximately +140°C, the
amplifier turns on again. But, for maximum
performance and reliability, the designer must take
care to ensure that the design does not exceed a
junction temperature of +125°C. Between +125°C
and +150°C, damage does not occur, but the
performance of the amplifier begins to degrade and
long
term
reliability
suffers.
The
thermal
characteristics of the device are dictated by the
package and the PC board. Maximum power
dissipation for a given package can be calculated
using the following formula.
TMax - TA
PDMax =
qJA
(1)
Where:
•
•
•
•
•
•
PDMax is the maximum power dissipation in the
amplifier (W)
TMax is the absolute maximum junction
temperature (°C)
TA is the ambient temperature (°C)
θJA = θJC + θCA
θJC is the thermal coefficient from the silicon
junctions to the case (°C/W)
θCA is the thermal coefficient from the case to
ambient air (°C/W)
For systems where heat dissipation is more critical,
the THS3110 and THS3111 are offered in an
MSOP-8 with PowerPAD package offering even
better thermal performance. The thermal coefficient
for the PowerPAD packages are substantially
improved over the traditional SOIC.
TA - Free-Air Temperature - °C
Results are with no airflow and PCB size = 3 in × 3 in (7,62 mm ×
7,62 mm); θJA = 58.4°C/W for MSOP-8 with PowerPAD (DGN); θJA
= 95°C/W for SOIC-8 High-K Test PCB (D); θJA = 158°C/W for
MSOP-8 with PowerPAD, without solder.
Figure 65. Maximum Power Distribution
vs Ambient Temperature
When determining whether or not the device satisfies
the maximum power dissipation requirement, it is
important to not only consider quiescent power
dissipation, but also dynamic power dissipation. Often
times, this is difficult to quantify because the signal
pattern is inconsistent, but an estimate of the RMS
power dissipation can provide visibility into a possible
problem.
DESIGN TOOLS
Evaluation Fixtures, Spice Models, and
Application Support
Texas Instruments is committed to providing its
customers with the highest quality of applications
support. To support this goal an evaluation board has
been developed for the THS3110 and THS3111
operational amplifiers. The board is easy to use,
allowing for straightforward evaluation of the device.
The evaluation board can be ordered through the
Texas Instruments web site, www.ti.com, or through
your local Texas Instruments sales representative.
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THS3111
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Computer simulation of circuit performance using
SPICE is often useful when analyzing the
performance of analog circuits and systems. This is
particularly true for video and RF-amplifier circuits
where parasitic capacitance and inductance can have
a major effect on circuit performance. A SPICE model
for the THS3111 is available through the Texas
Instruments web site (www.ti.com). The product
information center (PIC) is also available for design
assistance and detailed product information. These
models do a good job of predicting small-signal ac
and transient performance under a wide variety of
operating conditions. They are not intended to model
the distortion characteristics of the amplifier, nor do
they attempt to distinguish between the package
types in their small-signal ac performance. Detailed
information about what is and is not modeled is
contained in the model file itself.
J2
GND
TP2
J1
VS+
J7
VS−
FB2
FB1
VS+
+
C3
C5
C4
C2
C1
VS−
C6
+
PD
NOTE: The Edge number for the THS3111 is
6445587.
J7
R5
Z1
R6
0
R4
TP1
Vs+
R3
J5
Vin −
Figure 67. THS3110 EVM Board Layout
(Top Layer)
R8B
7
8
2 _
R1
R8A
6
3 +
1
4
R7A
R7B
Z2
J6
Vout
Vs −
J4
Vin+
R2
REF
J8
1
Figure 66. THS3110 EVM Circuit Configuration
Figure 68. THS3110 EVM Board Layout
(Bottom Layer)
24
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Table 2. Bill of Materials
THS3110DGN and THS3111DGN EVM
DESCRIPTION
SMD SIZE
REFERENCE
DESIGNATOR
PCB
QTY
MANUFACTURER'S
PART NUMBER (1)
1
Bead, ferrite, 3 A, 80 Ω
1206
FB1, FB2
2
(Steward) HI1206N800R-00
2
Capacitor 6.8 μF, tantalum,
35 V, 10%
D
C1, C2
2
(AVX) TAJD685K035R
3
Open
0805
R5, Z1
2
4
Capacitor 0.1 μF, ceramic, X7R, 50
V
0805
C3, C4
2
(AVX) 08055C104KAT2A
5
Capacitor 100 pF, ceramic, NPO,
100 V
0805
C5, C6
2
(AVX) 08051A101JAT2A
ITEM
(1)
(2)
6
Resistor, 0 Ω, 1/8 W, 1%
0805
7
Resistor, 750 Ω, 1/8 W, 1%
0805
8
Open
9
Resistor, 49.9 Ω, 1/4 W, 1%
10
R6
(2)
1
(Phycomp) 9C08052A0R00JLHFT
R3, R4
2
(Phycomp) 9C08052A7500FKHFT
1206
R7A, Z2
2
1206
R2, R8A
2
(Phycomp) 9C12063A49R9FKRFT
Resistor, 53.6 Ω, 1/4 W, 1%
1206
R1
1
(Phycomp) 9C12063A53R6FKRFT
11
Open
2512
R7B, R8B
2
12
Header, 0.1" (2,54 mm) CTRS,
0.025" (6,35 mm) SQ pins
3 Pos.
JP1 (2)
1
(Sullins) PZC36SAAN
(2)
1
(Sullins) SSC02SYAN
13
Shunts
14
Jack, banana receptance,
0.25" (6,35 mm) dia. hole
J1, J2, J3
3
(SPC) 813
15
Test point, red
J7 (2), J8 (2), TP1
3
(Keystone) 5000
16
Test point, black
TP2
1
(Keystone) 5001
17
Connector, SMA PCB jack
J4, J5, J6
3
(Amphenol) 901-144-8RFX
18
Standoff, 4-40 hex,
0.625" (15,875 mm) length
4
(Keystone) 1808
19
Screw, Phillips, 4-40,
0.250" (6,35 mm)
4
SHR-0440-016-SN
20
IC, THS3110
21
Board, printed-circuit (THS3110)
22
IC, THS3111
23
Board, printed-circuit (THS3111)
JP1
U1
U1
1
(TI) THS3110DGN
1
(TI) EDGE # 6445586
1
(TI) THS3111DGN
1
(TI) EDGE # 6445587
Manufacturer part numbers are used for test purposes only.
Applies to the THS3110DGN EVM only.
ADDITIONAL REFERENCE MATERIAL
•
•
•
•
•
•
•
PowerPAD Made Easy, application brief (SLMA004)
PowerPAD Thermally-Enhanced Package, technical brief (SLMA002)
Voltage Feedback vs Current Feedback Amplifiers, (SLVA051)
Current Feedback Analysis and Compensation (SLOA021)
Current Feedback Amplifiers: Review, Stability, and Application (SBOA081)
Effect of Parasitic Capacitance in Op Amp Circuits (SLOA013)
Expanding the Usability of Current-Feedback Amplifiers, by Randy Stephens, 3Q 2003 Analog Applications
Journal www.ti.com/sc/analogapps).
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
Submit Documentation Feedback
25
THS3110
THS3111
SLOS422E – SEPTEMBER 2003 – REVISED OCTOBER 2009........................................................................................................................................ www.ti.com
REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision D (May 2008) to Revision E ...................................................................................................... Page
•
Changed Power-Down Characteristics, Power-down quiescent current test conditions of VS = ±15 V Electrical
Characteristics ...................................................................................................................................................................... 5
•
Changed Power-Down Characteristics, PD pin bias current parameter of VS = ±15 V Electrical Characteristics ............... 5
•
Changed Power-Down Characteristics, Power-down quiescent current test conditions of VS = ±5 V Electrical
Characteristics ...................................................................................................................................................................... 7
•
Changed Power-Down Characteristics, PD pin bias current parameter of VS = ±5 V Electrical Characteristics ................. 7
•
Added caption title to Figure 56 .......................................................................................................................................... 18
•
Added caption title to Figure 57 .......................................................................................................................................... 18
•
Added caption title to Figure 58 .......................................................................................................................................... 18
•
Added caption title to Figure 59 .......................................................................................................................................... 19
•
Added caption title to Figure 60 .......................................................................................................................................... 19
•
Changed the first sentence of the second paragraph of Saving Power with Power-Down Functionality section .............. 19
Changes from Revision C (February, 2007) to Revision D ............................................................................................ Page
•
Changed VS = ±15 V Transimpedance specifications from 1.5 MΩ (typ) to 1 MΩ (typ); 1 MΩ (at +25°C) to 0.75 MΩ;
0.7 MΩ (over temperature) to 0.5 MΩ .................................................................................................................................. 4
•
Changed VS = ±15 V Input offset voltage specifications from 1.5 mV (typ) to 3 mV (typ); 6 mV (at +25°C) to 10 mV;
8 mV (over temperature) to 12 mV ....................................................................................................................................... 4
•
Changed VS = ±15 V +PSRR specifications from 83 dB to 75 dB (typ); from 75 dB to 65 dB (at +25°C); from 70 dB
(over temperature) to 60 dB .................................................................................................................................................. 5
•
Changed VS = ±15 V –PSRR specifications from 78 dB to 69 dB (typ); from 70 dB to 60 dB (at +25°C); from 66 dB
(over temperature) to 55 dB .................................................................................................................................................. 5
•
Changed VS = ±5 V Transimpedance specifications from 1.6 MΩ (typ) to 1 MΩ (typ); 1 MΩ (at +25°C) to 0.75 MΩ;
0.7 MΩ (over temperature) to 0.5 MΩ .................................................................................................................................. 6
•
Changed VS = ±5 V Input offset voltage specifications from 3 mV (typ) to 6 mV (typ); 6 mV (at +25°C) to 10 mV; 8
mV (over temperature) to 12 mV .......................................................................................................................................... 6
•
Changed VS = ±5 V +PSRR specifications from 80 dB to 71 dB (typ); from 72 dB to 62 dB (at +25°C); from 67 dB
(over temperature) to 57 dB .................................................................................................................................................. 7
•
Changed VS = ±5 V –PSRR specifications from 75 dB to 66 dB (typ); from 67 dB to 57 dB (at +25°C); from 62 dB
(over temperature) to 52 dB .................................................................................................................................................. 7
•
Corrected Typical Characteristic figure numbering errors from previous version ................................................................ 9
•
Updated ±15 V Transimpedance vs Frequency characteristic graph ................................................................................. 11
26
Submit Documentation Feedback
Copyright © 2003–2009, Texas Instruments Incorporated
Product Folder Link(s): THS3110 THS3111
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
THS3110ID
ACTIVE
SOIC
D
8
75
RoHS & Green
THS3110IDGN
ACTIVE
HVSSOP
DGN
8
80
THS3110IDGNR
ACTIVE
HVSSOP
DGN
8
THS3110IDR
ACTIVE
SOIC
D
THS3111CD
ACTIVE
SOIC
THS3111ID
ACTIVE
THS3111IDG4
NIPDAU
Level-1-260C-UNLIM
-40 to 85
3110I
Samples
RoHS & Green NIPDAU | NIPDAUAG
Level-1-260C-UNLIM
-40 to 85
BIR
Samples
2500
RoHS & Green NIPDAU | NIPDAUAG
Level-1-260C-UNLIM
-40 to 85
BIR
Samples
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
3110I
Samples
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
3111C
Samples
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
3111I
Samples
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
3111I
Samples
THS3111IDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green NIPDAU | NIPDAUAG
Level-1-260C-UNLIM
-40 to 85
BIS
Samples
THS3111IDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green NIPDAU | NIPDAUAG
Level-1-260C-UNLIM
-40 to 85
BIS
Samples
THS3111IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
Level-1-260C-UNLIM
-40 to 85
3111I
Samples
NIPDAU
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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