Amplifiers: Op Amps
Texas Instruments Incorporated
Expanding the usability of
current-feedback amplifiers
By Randy Stephens (Email: r-stephens@ti.com)
Systems Specialist, Member Group Technical Staff
Introduction
Figure 1. VFB test circuit
Although current-feedback (CFB) amplifiers have been
around as long as the widely utilized voltage-feedback (VFB)
amplifiers, their acceptance has been sporadic. One of the
reasons for this is quite simple—they have a different
name and therefore must be difficult and very hard to use.
This is simply not true. There are numerous papers1, 2, 3
comparing the differences between the two amplifier
types that show they are more similar to each other than
different. In fact, for numerous circuits, a CFB amplifier
may actually yield better results due to its inherent slewrate advantage, lack of a gain-bandwidth product, and
reasonably low noise for the performance.
Almost every paper written about CFB amplifiers cautions
readers that placing a capacitor directly in the feedback path,
without any resistance in series, will cause the CFB amplifier to oscillate. This is true, as the compensation of the
amplifier is tied directly to the feedback impedance. Since a
capacitor has low impedance at high frequencies, this essentially places a short in the feedback path that inadvertently
defeats amplifier compensation, resulting in instability.
Because of this limitation, there are a handful of common
circuits that are not recommended for use with a CFB
amplifier. These include integrators, some types of filters,
and special feedback-compensation techniques. But what
if there was a way to make these circuits work? And what
if the solution was as simple as adding a single component?
This would make it feasible to implement a CFB amplifier
for just about every application for which a VFB amplifier
could be used, with the benefits of the CFB amplifier.
CF = 220 pF
RF = 750 Ω
RG = 187 Ω
+15 V
VOUT
VIN
THS4012
RTerm
50 Ω
RL
100 Ω
–15 V
Figure 2. CFB test circuit with simple
modification
CF = 220 pF
RF = 750 Ω
RG = 187 Ω
+15 V
Compensation
Z
This article does not explain the compensation theory of
VFB and CFB amplifiers, as there are many papers written
on this topic. The only thing that is important is that
there must be resistance, or impedance, in the feedback
path at the open-loop intersection point to make the CFB
amplifier stable.
Figure 1 shows a traditional VFB amplifier, a THS4012,
configured in a noninverting gain of +5 with a simple lowpass gain filter set at approximately 1 MHz by the straightforward 1/(2πRFCF) formula.
If a CFB amplifier like the THS3112 is simply dropped
into this circuit, it will oscillate and the circuit will
become useless. A method of compensating the CFB
amplifier in this circuit is to insert a resistance, or impedance (Z), in the feedback path as shown in Figure 2.
It can easily be seen that regardless of the impedance
of the feedback path represented by RF and CF, the
impedance Z is in the amplifier’s feedback loop dictating
the compensation of the amplifier. The interesting thing
about this configuration is that the feedback resistance
(RF), which normally dictates the compensation of the
VOUT
VIN
RTerm
50 Ω
THS3112
RL
100 Ω
–15 V
amplifier, can now be essentially any resistance desired.
The reader should keep in mind that this is still a highspeed amplifier with speeds over 100 MHz; so the feedback
resistance should always be kept less than a few kilohms
to minimize the effects of parasitic capacitances on the
overall circuit. Conversely, minimizing the resistance too
much will place too much of a load on the amplifier,
typically degrading performance.
One of the drawbacks of adding the impedance Z in this
manner is that the summing node at the inverting terminal
is now separated from the virtual summing node. This can
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Texas Instruments Incorporated
VOUT (dB)
introduce errors into the system due to
Figure 3. Frequency responses with resistors (gain = +5)
the bias current and the dynamic signal
current flowing through this impedance;
but these effects are reasonably small as
25
long as the impedance is minimized.
Adding impedance Z can affect input
Z = 200 Ω
20
offset voltage due to the dc input bias
current, which is typically 1 to 10 µA,
15
multiplied by the impedance Z. This
resulting voltage gets multiplied by the
10
noise gain of the circuit. Additionally, when
a signal appears at the output, the CFB
Z = 475 Ω
5
amplifier (as the name implies) relies on
Z = 681 Ω
an error current flowing through the
0
inverting node through the impedance Z,
producing a signal error. However, since
–5
Z = 1 kΩ
the transimpedance of most CFB amplifiers is well over 100 kΩ and sometimes as
–10
THS4012
high as several megohms, this error is also
–15
minimized if the impedance is kept low. The
10 k
100 k
1M
10 M
100 M
1G
drift of this circuit now also relies on the
Frequency (Hz)
temperature characteristics of impedance
Z and should not be used as a precision
amplifier; but most CFB amplifiers are not
used as precision amplifiers anyway due to
stated previously. This shows that there is a reasonably wide
their inherent topology limitations. Overall, these issues
range of acceptable values for Z and does not imply that the
are minimal and, for most systems, can be effectively
selection for Z is highly critical. Figure 3 also illustrates a
ignored in favor of the CFB amplifier’s advantages as
common trait for current-feedback amplifiers—as the feedpreviously stated.
back impedance is decreased, the peaking will increase. If
Testing with different Z values
the impedance is too low, there is a good chance that the
The easiest way to see if the circuit is stable is to use a
circuit will become unstable and oscillate, as illustrated by
network analyzer frequency sweep. Instability can typicalthe response when Z = 200 Ω.
ly be seen as sharp rises in the frequency response at the
Output noise
amplifier’s bandwidth limitations. If the peaking is smooth,
One element that may be very important in a system is the
or there is no peak, then the amplifier should be stable.
output noise. Adding a resistance in the manner discussed
Figure 3 shows the frequency response of the system with
only makes the output noise worse. The inverting current
different values of resistors for the variable Z.
noise of the amplifier goes through the resistance at Z and
The response of the THS4012 is also shown for reference
creates a voltage noise. This noise then becomes multiplied
to easily compare the performance of the two systems. It
by the circuit’s gain, which is frequency-dependent.
is interesting that no matter what resistance is used for Z,
For a CFB amplifier, the inverting current noise is typithe responses below 20 MHz look identical to each other.
cally the highest noise component of the amplifier. Although
This is the ultimate goal of this configuration—no differthe CFB amplifier voltage noise is inherently very low,
ences in signal performance. For the stability part of the
—
—
typically less than 3 nV/√Hz , the inverting current noise of
circuit, the area above 20 MHz must be examined.
—
—
most CFB amplifiers is generally around 15 to 20 pA/√Hz .
Examining the circuits in Figures 1 and 2 shows us that
The noninverting current noise is only noticeable if the
the feedback impedance is dictated by the capacitor CF.
source impedance is high. Using a 50-Ω environment
Above 20 MHz, this impedance is very small—essentially
minimizes the noninverting current noise.
creating a short from the output to the summing node. This
The THS3112 was designed to have very low noise. The
configuration is commonly referred to as a unity buffer with
—
—
voltage noise is 2.2 nV/√Hz , the noninverting current noise
the signal gain set to 1. The data sheet for the THS3112 4
—
—
is 2.9 pA/√Hz , and the critical inverting current noise is a
recommends that, in a gain of +1 under the circuit condi—
—
low 10.8 pA/√Hz. However, multiplying the inverting current
tions utilized, the feedback resistance be 1 kΩ. Thus, it is
noise by 1 kΩ and then multiplying by the gain can alone
no surprise to see that when Z = 1 kΩ, the response looks
—
—
produce a very substantial output noise of about 54 nV/√Hz
very smooth and well behaved, indicating a very stable
in the pass band. To quantify the output noise of the system,
system. However, when Z = 681 Ω, the response also looks
the circuits shown in Figures 1 and 2 were tested for output
very reasonable and helps minimize the potential issues
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Amplifiers: Op Amps
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noise (see Figure 4). For comparison, the THS4012, with a
—
—
respectable voltage noise of 7.5 nV/√Hz and both current
—
—
noises of 1 pA/√Hz , is also shown in Figure 4.
Note that the output noise of the THS4012 is the same
as when using the THS3112 with Z = 475 Ω. Again, these
responses are just like those of a VFB amplifier in the traditional configuration, showing that the basic functionality is
sound—there are no differences between a VFB amplifier
and this configuration. Figure 4 shows that although using
Z = 1 kΩ produces a very stable amplifier, the output
—
—
noise is 20 nV/√Hz higher than that of the THS4012.
Figure 4. Output noise (gain = +5)
70
—–
Output Noise (nV/√ Hz)
60
Z = 1 kΩ
50
40
Z = 681 Ω
Keep in mind that the THS3112 has very low overall
noise but that many other CFB amplifiers will probably
produce much higher noise. The only way to get around
this is if the unity-gain stability of the amplifier requires a
very small resistor of, say, only 500 Ω or less. But what if
there was another way to make the CFB amplifier stable
and have low noise at the same time?
Fundamentally speaking, the circuit needs high impedance
within the feedback path only at the amplifier’s bandwidth
limit. At frequencies below this point, it really does not
matter what the impedance is, and the amplifier will work
fine. The issues stated previously are also
minimized, resulting in an even better
system than one using pure resistors.
The first solution that comes to mind is
to use an inductor. Inductors have low
impedance at low frequencies and high
impedance at high frequencies—exactly
what is desired; but their relatively large
size and high cost are generally considered
prohibitive. An alternative component
that minimizes these disadvantages and
still functions the same is the ferrite chip.
THS4012; also Z = 475 Ω
Testing with ferrite chips
used for Z
30
Z = 200 Ω
20
10
0
10 k
100 k
1M
10 M
Frequency (Hz)
Figure 5. Frequency responses above 10 MHz with
ferrite chips (gain = +5)
35
30
25
Z = BLM18HD601SN1
VOUT (dB)
20
15
Z = BLM18HG601SN1
10
5
0
–5
Z = 681 Ω
–10
–15
Z = BLM18AG601SN1
–20
10 M
100 M
Frequency (Hz)
1G
Ferrite chips have been available for several
years, are relatively low-cost, and are
available in very small sizes—0402 and
larger. Although several manufacturers
produce ferrite chips, testing was done
with what was available in the test lab—
ferrite chips from Murata’s BLM series.
Examining the impedance characteristics
of these ferrites revealed several possible
components that could be utilized.
The first factor in determining the proper
component was the ferrite’s impedance at
the amplifier’s bandwidth limit. For the
THS3112, this implied an impedance of
at least 600 Ω at about 150 MHz to meet
stability. This can vary, as the first test
results showed (see Figure 3).
Additionally, the Q of the ferrite chips
varies from grade to grade. Some have a
low Q with a fairly smooth rise to the
resonance point that then subsides due to
inherent properties and parasitics, while
other chips have a relatively high Q with a
sharp rise and fall in impedance associated
with them. Although either style may
meet the impedance requirements, testing
was required to see if this Q had an effect
on the circuit. Again, the best way to show
the results was to graph the frequency
response of the system, as shown in
Figure 5. The responses below 10 MHz
were all identical to the original configuration. This figure concentrates on the
stability portion of the responses above
10 MHz. For comparison purposes, the
681-Ω, pure-resistance response is shown.
25
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Analog and Mixed-Signal Products
Amplifiers: Op Amps
Texas Instruments Incorporated
—–
Output Noise (nV/√ Hz)
VOUT (dB)
Although all of these ferrite chips have
Figure 6. Responses with AG series ferrite chips (gain = +5)
the same impedance at 100 MHz (600 Ω),
they produced different results. The HD
series high-Q chip shows a very narrow
30
and large peak that will most likely result
Z = Ferrite Chip
in instability and oscillations. The AG and
25
BLM18AGxxxSN1 Series
xxx = 221
HG series low-Q chips both performed
20
about the same, and either one would
probably produce acceptable results. The
15
only difference is that the HG series has
impedance at higher frequencies and
10
xxx = 471
would probably be better suited for use
with very high-speed CFB amplifiers such
5
as the OPA685 or the THS3202.
0
Notice that the pure resistance has a
xxx = 601
lower response peak than the ferrite chips.
–5
Coupled with the fact that the HD series
xxx = 102
has a high Q and a high peak, this implies
–10
that the slope of the impedance at the
–15
amplifier’s bandwidth is a factor for stabil10 k
100 k
1M
10 M
100 M
1G
ity. This makes a lot of sense; as it is well
known that for any amplifier, if a zero
Frequency (Hz)
intersects the amplifier’s open-loop
response at a rate of closure of 40 dB/
decade, large peaking and oscillations will
most likely result.5 For this circuit configuration, if the impedance of Z has a large
Figure 7. Output noise comparison (gain = +5)
slope that intersects the transimpedance
curve at essentially a rate of closure of
40 dB/decade, peaking and oscillations
50
also will most likely occur. By comparison,
Z = 681 Ω
45
a resistor intersects the transimpedance
curve at a rate of closure of 20 dB/decade,
40
resulting in a stable response. Even though
THS4012
35
the low-Q ferrite beads have some slope
related to their impedance, the rate of
30
Z = 332 Ω
closure is much lower than 40 dB/decade,
25
providing improved stability. Nevertheless,
minimizing this intersection rate of closure
20
as much as possible should produce
Z = All Ferrite Chips
acceptable results.
15
To further expand on the usefulness of
10
the ferrite chips, more testing was done
utilizing the AG series in the circuit, as
5
shown in Figure 6.
0
This figure shows that, just like the
10 k
100 k
1M
10 M
results for the pure resistor, the higher
Frequency
(Hz)
the impedance is, the lower the peaking.
How does this affect the output noise of
the system? Figure 7 shows the output
noise when the ferrite chips were used,
along with the output noise of the THS4012
and some of the original resistor configurations.
Inverting gain configuration
As expected, due to the low frequency impedance of the
All of the testing discussed so far was done with the nonferrite chips, the noise is extremely low. This noise was the
inverting gain configuration. This configuration forces the
same regardless of which ferrite was used. If noise above
inverting node voltage to move proportionally to the input
10 MHz was important, the impedance of these ferrite
voltage applied. So how does the system work in the
chips would start to increase the output noise to the same
inverting gain configuration where the inverting node is
extent as resistors. These tests show that there are several
held at a virtual ground? The easy answer is that it works
advantages of using ferrite chips over resistors.
26
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Texas Instruments Incorporated
Figure 8. Inverting gain of 5 VFB configuration
Figure 9. Inverting gain of 5 CFB configuration
CF = 220 pF
CF = 220 pF
RF = 750 Ω
RF = 750 Ω
+15 V
RG
150 Ω
VIN
THS4012
Z
VIN
VOUT
RTerm
75 Ω
+15 V
RG
150 Ω
VOUT
RTerm
75 Ω
RL
100 Ω
THS3112
–15 V
RL
100 Ω
–15 V
exactly the same as before. Figures 8 and 9 show the test
circuits for this configuration. The signal gain was kept at
a gain of 5.
The same concepts apply for this CFB configuration as
for the noninverting configuration. The advantage of this
circuit is that the attenuation is not limited to unity gain,
or 0 dB, like the noninverting gain circuit. Figure 10 shows
the frequency responses of this configuration with varying
pure resistor values for Z. The THS4012 response is shown
for comparison purposes.
As expected, the responses all look comparable to each
other below 10 MHz. Additionally, the resistance values
affect the stability and again show that the higher the
resistance is, the better the stability. Using a resistance as
low as 475 Ω actually shows respectable performance in this
configuration. Remember that for oscillations to occur, the
gain must be above unity gain, or 0 dB. As long as the peak
is below 0 dB, oscillations should not occur. As in the noninverting case, using 200 Ω shows a large narrow peak that
will most likely result in stability issues and/or oscillations.
However, notice that above 10 MHz the same general
shape occurs for both the CFB and VFB amplifiers. This is
caused by the amplifiers’ input and output impedances
becoming very high above their bandwidth limit. When
this occurs, there is a path for the input signal to flow
through RG, through CF, and then to feed forward to the
load. Of course, the amplifiers’ own input and output
capacitances also affect the amount of feed-through in the
circuit; but it is important to remember that this occurs
above the amplifiers’ usable bandwidths.
Just as for the noninverting configuration, using ferrite
chips has several advantages for the inverting configuration.
Figure 10. Frequency responses with resistors (gain = –5)
15
10
VOUT (dB)
5
Z = 200 Ω
0
–5
Z = 475 Ω
THS4012
–10
–15
–20
Z = 1 kΩ
–25
10 k
100 k
1M
10 M
Frequency (Hz)
100 M
1G
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Analog Applications Journal
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Analog and Mixed-Signal Products
Amplifiers: Op Amps
Figure 11. Frequency responses above 10 MHz
with ferrite chips (gain = –5)
10
5
Z = BLM18HD601SN1
0
VOUT (dB)
Figure 11 shows the frequency responses
of several of these chips. Figure 12 shows
the results of using various ferrite chips
from the same AG family.
As expected, all of these graphs show
the same type of results obtained with
the noninverting configuration. Using a
low-Q ferrite chip with high impedance
will result in a stable system. Although the
noise plots for this configuration are not
presented here, they will show the same
type of results as the noninverting configuration; using ferrite chips will have the
lowest output noise of any configuration.
Texas Instruments Incorporated
Conclusion
References
For more information related to this article,
you can download an Acrobat Reader file
at www-s.ti.com/sc/techlit/litnumber and
replace “litnumber” with the TI Lit. #
for the materials listed below.
Z = BLM18HG601SN1
Z = BLM18AG601SN1
–10
–15
Z = 681 Ω
–20
–25
10 M
100 M
Frequency (Hz)
1G
Figure 12. Frequency responses with AG series
ferrite chips (gain = –5)
15
10
xxx = 221
5
VOUT (dB)
Although this article shows only two configurations with capacitors in the feedback
path, it shows the fundamental feasibility
of this compensation technique. While
resistors do work very well, producing the
most stable responses, the drawbacks of
the output noise coupled with the dc and
ac errors may limit some of the applications.
Using ferrite chips helps alleviate many
of these issues, producing the lowest noise
of all with no dc errors or in-band ac signal errors; and stability is almost as good
as when utilizing resistors. It is important
to choose the proper ferrite chip with the
amplifier; but this is considered normal
procedure for any circuit design and is no
more difficult than selecting the right
amplifier for the system.
This simple technique helps eliminate
one of the major drawbacks of using the
CFB amplifier while allowing any system
to enjoy many of its benefits. Designers of
multiple feedback filters, for example, once
limited to the use of VFB amplifiers, can
now take advantage of the superior slew
rates and lack of gain-bandwidth product
characteristics found in the CFB amplifier.
–5
0
xxx = 471
–5
xxx = 601
–10
–15
Z = Ferrite Chip
BLM18AGxxxSN1 Series
–20
xxx = 102
–25
10 k
100 k
1M
10 M
100 M
1G
Frequency (Hz)
Document Title
TI Lit. #
1. “Voltage Feedback Vs. Current Feedback
Op Amps,” Application Report . . . . . . . . . . . . . .slva051
2. “The Current-Feedback Op Amp: A HighSpeed Building Block,” Application Bulletin . . .sboa076
3. “Current Feedback Amplifiers: Review,
Stability Analysis, and Applications,”
Application Bulletin . . . . . . . . . . . . . . . . . . . . . . .sboa081
4. “Low-Noise, High-Speed Current Feedback
Amplifiers,” Data Sheet . . . . . . . . . . . . . . . . . . . .slos385
5. “Effect of Parasitic Capacitance in Op Amp
Circuits,” Application Report . . . . . . . . . . . . . . .sloa013
Related Web sites
analog.ti.com
www.ti.com/sc/device/partnumber
Replace partnumber with OPA685, THS3112, THS3202 or
THS4012
28
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