SLOS238D − MAY 1999 − REVISED AUGUST 2008
THS4051
D, DGN, AND JG PACKAGES
(TOP VIEW)
FEATURES
D High Speed:
− 70 MHz Bandwidth (G = 1, −3 dB)
− 240 V/µs Slew Rate
− 60-ns Settling Time (0.1%)
NULL
IN−
IN+
VCC−
D High Output Drive, IO = 100 mA (typ)
D Excellent Video Performance:
1OUT
1IN−
1IN+
−VCC
1
8
2
7
3
6
4
5
VCC+
2OUT
2IN−
2IN+
Cross Section View Showing
PowerPAD Option (DGN)
† This device is in the Product Preview stage of development.
Please contact your local TI sales office for availability.
290-MHz Low Distortion High-Speed Amplifiers
100-MHz Low Noise High-Speed Amplifiers
175-MHz Low Power High-Speed Amplifiers
NULL
1
20 19
NC
NC
2
4
18 NC
IN−
5
17 VCC+
NC
6
16 NC
IN+
7
15 OUT
NC
8
14 NC
9
10 11 12 13
NC
DESCRIPTION
3
NC
NC
RELATED DEVICES
THS4051
FK PACKAGE
(TOP VIEW)
NULL
The THS4051 and THS4052 are general-purpose,
single/dual, high-speed voltage feedback amplifiers ideal
for a wide range of applications including video,
communication, and imaging. The devices offer very good
ac performance with 70-MHz bandwidth, 240-V/µs slew
rate, and 60-ns settling time (0.1%). The THS4051/2 are
stable at all gains for both inverting and noninverting configurations. These amplifiers have a high
output drive capability of 100 mA and draw only 8.5-mA
supply current per channel. Excellent professional video
results can be obtained with the low differential gain/phase
errors of 0.01%/ 0.01° and wide 0.1-dB flatness to 30 MHz.
For applications requiring low distortion, the THS4051/2 is
ideally suited with total harmonic distortion of −82 dBc at
1 MHz.
THS4011/2
THS4031/2
THS4081/2
5
THS4052
D AND DGN† PACKAGES
(TOP VIEW)
DESCRIPTION
DEVICE
6
4
VCC−
NC
D
3
NULL
VCC+
OUT
NC
NC
D
7
NC
D
8
2
NC − No internal connection
− 0.1 dB Bandwidth of 30 MHz (G = 1)
− 0.01% Differential Gain
− 0.01° Differential Phase
Very Low Distortion:
− THD = −82 dBc (f = 1 MHz, RL = 150 Ω)
− THD = −89 dBc (f = 1 MHz, RL = 1 kΩ)
Wide Range of Power Supplies:
− VCC = ±5 V to ±15 V
Available in Standard SOIC, MSOP
PowerPAD, JG, or FK Packages
Evaluation Module Available
D
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
DLP is a registered trademark of Texas Instruments. SMBus is a trademark of Intel Corp.
All other trademarks are the property of their respective owners.
Copyright 1999−2008, Texas Instruments Incorporated
!" # $% &" !# '%()$!" *!"&+ *%$"#
$ " #'&$ $!" # '& ",& "& # &-!#
#" % &"# #"!*! * .! !"/+
*%$" ' $&##0 * &# " &$&##! )/ $)%*& "&#"0 !)) '! ! &"& #+
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
CAUTION: The THS4051 and THS4052 provide ESD protection circuitry. However, permanent damage can still occur if this device
is subjected to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance
degradation or loss of functionality.
ABSOLUTE MAXIMUM RATINGS OVER OPERATING FREE-AIR TEMPERATURE (UNLESS
OTHERWISE NOTED)(1)
Supply voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±16.5 V
Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±VCC
Output current, IO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150 mA
Differential input voltage, VIO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±4 V
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Dissipation Rating Table
Maximum junction temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C
Operating free-air temperature, TA: C-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C
I-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 85°C
M-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −55°C to 125°C
Storage temperature, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C
Lead temperature 1,6 mm (1/16 inch) from case for 60 seconds, JG package . . . . . . . . . . . . . . . . . . . 300°C
Case temperature for 60 seconds, FK package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
D
DGN§
JG
θJA
(°C/W)
167‡
θJC
(°C/W)
TA = 25
25°C
C
POWER RATING
38.3
740 mW
58.4
4.7
2.14 W
119
28
1050 mW
FK
87.7
20
1375 mW
‡ This data was taken using the JEDEC standard Low-K test PCB. For the JEDEC Proposed High-K
test PCB, the θJA is 95°C/W with a power rating at TA = 25°C of 1.32 W.
§ This data was taken using 2 oz. trace and copper pad that is soldered directly to a 3 in. × 3 in. PC.
For further information, refer to Application Information section of this data sheet.
RECOMMENDED OPERATING CONDITIONS
MIN
Supply voltage, VCC+ and VCC−
Operating free-air temperature, TA
2
NOM
MAX
Dual supply
±4.5
±16
Single supply
9
32
C-suffix
0
70
I-suffix
−40
85
M-suffix
−55
125
UNIT
V
°C
C
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
AVAILABLE OPTIONS(1)
PACKAGED DEVICES
NUMBER
OF
CHANNELS
TA
1
0°C to 70°C
−40°C to 85°C
PLASTIC MSOP†
(DGN)
PLASTIC
SMALL
OUTLINE†
(D)
THS4051CD
CERAMIC DIP
(JG)
CHIP
CARRIER
(FK)
EVALUATION
MODULE
DEVICE
SYMBOL
ACQ
—
—
THS4051EVM
2
THS4052CD
THS4051CDGN
THS4052CDGN‡
ACE
—
—
THS4052EVM
1
THS4051ID
THS4051IDGN
ACR
—
—
—
THS4052ID
THS4052IDGN‡
ACF
—
—
—
2
−55°C to 125°C
1
—
—
—
THS4051MJG
THS4051MFK
—
(1) For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI web site
at www.ti.com.
† The D and DGN packages are available taped and reeled. Add an R suffix to the device type (i.e., THS4051CDGN).
‡ This device is in the Product Preview stage of development. Please contact your local TI sales office for availability.
FUNCTIONAL BLOCK DIAGRAM
VCC
1IN−
1OUT
1IN+
Null
2IN−
IN−
IN+
2
2OUT
1
2IN+
8
6
OUT
3
−VCC
Figure 2. THS4052 − Dual Channel
Figure 1. THS4051 − Single Channel
T
THD - Total Harmonic Distortion - dBc
−40
−50
HARMONIC DISTORTION
vs
FREQUENCY
VCC = ± 15 V
Gain = 2
VO(PP) = 2 V
−60
RL = 150 Ω
−70
−80
RL = 1 kΩ
−90
−100
100k
1M
f - Frequency - Hz
10M
20M
3
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
ELECTRICAL CHARACTERISTICS AT TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise
noted)
dynamic performance
MIN
TYP
VCC = ±15 V
VCC = ±5 V
Gain = 1
VCC = ±15 V
VCC = ±5 V
Gain = 2
Bandwidth for 0.1 dB flatness
VCC = ±15 V
VCC = ±5 V
Gain = 1
Full power bandwidth§
VO(pp) = 20 V,
VO(pp) = 5 V,
VCC = ±15 V
VCC = ±5 V
Slew rate‡
VCC = ±15 V,
VCC = ±5 V,
20-V step,
Gain = 5
240
5-V step
Gain = −1
200
VCC = ±15 V,
VCC = ±5 V,
5-V step
Settling time to 0.1%
Settling time to 0.01%
VCC = ±15 V,
VCC = ±5 V,
5-V step
Dynamic performance small-signal bandwidth
(−3 dB)
BW
SR
THS405xC, THS405xI
TEST CONDITIONS†
PARAMETER
ts
MAX
UNIT
70
MHz
70
38
MHz
38
30
2-V step
2-V step
MHz
30
3.8
MHz
12.7
V/ s
V/µs
60
Gain = −1
ns
60
130
Gain = −1
ns
140
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix
‡ Slew rate is measured from an output level range of 25% to 75%.
§ Full power bandwidth = slew rate/2 πVO(Peak).
noise/distortion performance
THD
Vn
In
Total harmonic distortion
VO(pp) = 2 V,
f = 1 MHz, Gain = 2
VCC = ±5 V or ±15 V,
VCC = ±5 V or ±15 V,
Input voltage noise
Input current noise
MIN
RL = 150 Ω
RL = 1 kΩ
−82
VCC = ±5 V
RL = 150 Ω
RL = 1 kΩ
−78
f = 10 kHz
f = 10 kHz
Differential gain error
Gain = 2,
40 IRE modulation,
NTSC,
±100 IRE ramp
Differential phase error
Gain = 2,
40 IRE modulation,
NTSC,
±100 IRE ramp
VCC = ±15 V
VCC = ±5 V
VCC = ±5 V or ±15 V,
f = 1 MHz
crosstalk
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix.
TYP
VCC = ±15 V
VCC = ±15 V
VCC = ±5 V
Channel-to-channel
(THS4052 only)
4
THS405xC, THS405xI
TEST CONDITIONS†
PARAMETER
MAX
UNIT
−89
dBc
−87
14
nV/√Hz
0.9
pA/√Hz
0.01%
0.01°
0.01%
0.03°
−57
dB
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
electrical characteristics at TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise noted) (continued)
dc performance
PARAMETER
THS405xC, THS405xI
TEST CONDITIONS†
TYP
9
VCC = ±15 V, RL = 1 kΩ
VO = ±10 V
TA = 25°C
TA = full range
5
VCC = ±5 V, RL = 250 Ω
VO = ±2.5 V
TA = 25°C
TA = full range
2.5
Open loop gain
Input offset voltage
VCC = ±5 V or ±15 V
Offset voltage drift
VCC = ±5 V or ±15 V
IIB
Input bias current
VCC = ±5 V or ±15 V
IOS
Input offset current
VCC = ±5 V or ±15 V
VOS
MIN
MAX
V/mV
3
6
V/mV
2
TA = 25°C
TA = full range
2.5
TA = full range
TA = 25°C
15
10
12
2.5
TA = full range
TA = 25°C
35
TA = full range
mV
µV/°C
6
8
µA
A
250
400
Offset current drift
TA = full range
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix
UNIT
0.3
nA
nA/°C
input characteristics
PARAMETER
TEST CONDITIONS†
VICR
Common-mode input voltage range
VCC = ±15 V
VCC = ±5 V
CMRR
Common mode rejection ratio
VCC = ±15 V,
VCC = ±5 V,
VICR = ±12 V
VICR = ±2.5 V
THS405xC, THS405xI
TA = full range
MIN
TYP
±13.8
±14.3
± 3.8
± 4.3
70
100
70
100
ri
Input resistance
Ci
Input capacitance
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix
MAX
UNIT
V
dB
1
MΩ
1.5
pF
output characteristics
PARAMETER
VO
Output voltage swing
TEST CONDITIONS†
VCC = ±15 V
VCC = ±5 V
RL = 250 Ω
RL = 150 Ω
VCC = ±15 V
VCC = ±5 V
RL = 1 kΩ
RL = 20 Ω
IO
Output current‡
VCC = ±15 V
VCC = ±5 V
ISC
Short-circuit current‡
VCC = ±15 V
THS405xC, THS405xI
MIN
TYP
±11.5
±13
±3.2
±3.5
±13
±13.6
±3.5
±3.8
80
100
50
75
150
MAX
UNIT
V
V
mA
mA
RO
Output resistance
Open loop
13
W
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix
‡ Observe power dissipation ratings to keep the junction temperature below the absolute maximum rating when the output is heavily loaded or shorted.
See the absolute maximum ratings section of this data sheet for more information.
5
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
electrical characteristics at TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise noted) (continued)
power supply
PARAMETER
VCC
ICC
PSRR
Supply voltage operating range
THS405xC, THS405xI
TEST CONDITIONS†
MIN
MAX
Dual supply
±4.5
±16.5
Single supply
9
33
VCC = ±15 V
TA = 25°C
TA = full range
8.5
VCC = ±5 V
TA = 25°C
TA = full range
7.5
VCC = ±5 V or ±15 V
TA = 25°C
TA = full range
Supply current (per amplifier)
Power supply rejection ratio
TYP
UNIT
V
10.5
11.5
9.5
mA
10.5
70
84
dB
68
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix
ELECTRICAL CHARACTERISTICS AT TA = FULL RANGE, VCC = ±15 V, RL = 1 KΩ (UNLESS
OTHERWISE NOTED)
dynamic performance
PARAMETER
Unity gain bandwidth
Dynamic performance small-signal bandwidth (−3
dB)
BW
Bandwidth for 0.1 dB flatness
Full power bandwidth‡
SR
Slew rate
Settling time to 0.1%
ts
Settling time to 0.01%
† Full range = −55°C to 125°C for the THS4051M.
‡ Full power bandwidth = slew rate/2 πVO(Peak).
§ This parameter is not tested.
6
THS4051M
TEST CONDITIONS†
VCC = ±15 V,
VCC = ±15 V
Closed loop
Gain = 1
70
MAX
UNIT
MHz
70
38
Gain = 2
VCC = ±5 V
VCC = ±15 V
VO(pp) = 5 V,
VCC = ±15 V,
TYP
70
VCC = ±5 V
VCC = ±15 V
VCC = ±5 V
VO(pp) = 20 V,
RL = 1 kΩ
MIN
50§
MHz
38
30
Gain = 1
30
VCC = ±15 V
VCC = ±5 V
VCC = ±5 V,
VCC = ±15 V,
5-V step
VCC = ±5 V,
VCC = ±15 V,
2-V step
VCC = ±5 V,
2-V step
3.8
12.7
RL = 1 kΩ
Gain = −1
5-V step
5-V step
MHz
240§
MHz
300
200
V/ s
V/µs
60
Gain = −1
Gain = −1
60
130
140
ns
ns
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
electrical characteristics at TA = full range, VCC = ±15 V, RL = 1 kΩ (unless otherwise noted)
noise/distortion performance
PARAMETER
TEST CONDITIONS†
THD
VO(pp) = 2 V,
f = 1 MHz, Gain = 2,
TA = 25°C
Total harmonic distortion
THS4051M
MIN
TYP
VCC = ±15 V
RL = 150 Ω
RL = 1 kΩ
−82
VCC = ±5 V
RL = 150 Ω
RL = 1 kΩ
−78
MAX
UNIT
−89
dBc
−87
Vn
Input voltage noise
VCC = ±5 V or ±15 V,
TA = 25°C
f = 10 kHz,
RL = 150 Ω
14
nV/√Hz
In
Input current noise
VCC = ±5 V or ±15 V,
TA = 25°C
f = 10 kHz,
RL = 150 Ω
0.9
pA/√Hz
Gain = 2,
40 IRE modulation,
TA = 25°C,
NTSC,
±100 IRE ramp,
RL = 150 Ω
VCC = ±15 V
0.01%
Differential gain error
VCC = ±5 V
0.01%
Gain = 2,
40 IRE modulation,
TA = 25°C,
† Full range = −55°C to 125°C for the THS4051M.
NTSC,
±100 IRE ramp,
RL = 150 Ω
VCC = ±15 V
0.01°
VCC = ±5 V
0.03°
Differential phase error
dc performance
PARAMETER
THS4051M
TEST CONDITIONS†
TYP
9
VCC = ±15 V, VO = ±10 V
TA = 25°C
TA = full range
5
VCC = ±5 V, VO = ±2.5 V
TA = 25°C
TA = full range
2.5
VCC = ±5 V or ±15 V
TA = 25°C
TA = full range
2.5
Input offset voltage
Offset voltage drift
VCC = ±5 V or ±15 V
TA = full range
TA = 25°C
15
Open loop gain
VIO
MIN
IIB
Input bias current
VCC = ±5 V or ±15 V
IIO
Input offset current
VCC = ±5 V or ±15 V
MAX
V/mV
3
6
V/mV
2
10
13
2.5
TA = full range
TA = 25°C
35
TA = full range
mV
µV/°C
6
8
µA
A
250
400
Offset current drift
TA = full range
† Full range = −55°C to 125°C for the THS4051M.
UNIT
0.3
nA
nA/°C
input characteristics
PARAMETER
TEST CONDITIONS†
VICR
Common-mode input voltage range
VCC = ±15 V
VCC = ±5 V
CMRR
Common mode rejection ratio
VCC = ±15 V,
VCC = ±5 V,
ri
Input resistance
Ci
Input capacitance
† Full range = −55°C to 125°C for the THS4051M.
VICR = ±12 V
VICR = ±2.5 V
THS4051M
TA = full range
MIN
TYP
±13.8
±14.3
± 3.8
± 4.3
70
100
70
100
MAX
UNIT
V
dB
1
MΩ
1.5
pF
7
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
electrical characteristics at TA = full range, VCC = ±15 V, RL = 1 kΩ (unless otherwise noted)
(continued)
output characteristics
TEST CONDITIONS†
PARAMETER
VO
Output voltage swing
VCC = ±15 V
VCC = ±5 V
RL = 250 Ω
RL = 150 Ω
VCC = ±15 V
VCC = ±5 V
RL = 1 kΩ
VCC = ±15 V,
TA = 25°C
IO
Output current‡
ISC
Short-circuit current‡
VCC = ±15 V,
TA = full range
RL = 20 Ω
VCC = ±5 V
VCC = ±15 V
THS4051M
MIN
TYP
±12
±13
±3.2
±3.5
±13
±13.6
±3.5
±3.8
80
100
MAX
V
V
mA
70
50
UNIT
75
150
mA
RO
Output resistance
Open loop
13
W
† Full range = −55°C to 125°C for the THS4051M.
‡ Observe power dissipation ratings to keep the junction temperature below the absolute maximum rating when the output is heavily loaded or shorted.
See the absolute maximum ratings section of this data sheet for more information.
power supply
PARAMETER
VCC
ICC
Supply voltage operating range
8
MIN
TYP
MAX
Dual supply
±4.5
±16.5
Single supply
9
33
VCC = ±15 V
TA = 25°C
TA = full range
8.5
VCC = ±5 V
TA = 25°C
TA = full range
7.5
VCC = ±5 V or ±15 V
TA = full range
Supply current (per amplifier)
PSRR Power supply rejection ratio
† Full range = −55°C to 125°C for the THS4051M.
THS4051M
TEST CONDITIONS†
UNIT
V
10.5
11.5
9.5
mA
10.5
70
84
dB
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS
INPUT OFFSET VOLTAGE
vs
FREE-AIR TEMPERATURE
IB Input Bias Current − µ A
2.7
I
2.3
VCC = ± 5 V
−1.5
−2.0
−2.5
VCC = ± 15 V
−3.0
−3.5
−40
−20
0
20
40
60
80
TA - Free-Air Temperature - °C
2.6
2.5
2.4
7
9
11
13
±VCC - Supply Voltage - V
15
Figure 5
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
11
13.5
11
9
7
10
13
VCC = ± 15 V
RL = 250 Ω
VCC = ± 15 V
RL = 1 kΩ
12.5
12
4.5
VCC = ± 5 V
RL = 1 kΩ
4
3.5
VCC = ± 5 V
RL = 150 Ω
5
3
2.5
−40
3
7
9
11
13
±VCC - Supply Voltage - V
15
VN
10
IN
100k
PSRR - Power Supply Rejection Ratio - dB
100
Figure 9
TA=25°C
7
TA=−40°C
6
5
−20
0
20
40
60
80
100
5
7
9
11
13
± VCC - Supply Voltage - V
0
CMRR
vs
FREQUENCY
VCC = ±15 V & ±5 V
−10
−20
−VCC
−30
+VCC
−40
−50
−60
−70
−80
−90
100k
1M
10M
f - Frequency - Hz
Figure 10
15
Figure 8
POWER-SUPPLY REJECTION
RATIO
vs
FREQUENCY
VCC = ± 15 V and ± 5V
TA = 25°C
1k
10k
f - Frequency - Hz
8
Figure 7
VOLTAGE & CURRENT NOISE
vs
FREQUENCY
100
TA=85°C
9
TA − Free-Air Temperature − _C
Figure 6
1000
I CC − Supply Current − mA
VO - Output Voltage -
V
13
CMRR − Common-Mode Rejection Ratio − dB
±V
V ICR - Common-Mode Input Voltage −
5
14
TA=25°C
V n − Voltage Noise − nV/ Hz
I n − Current Noise − pA/ Hz
100
OUTPUT VOLTAGE
vs
FREE-AIR TEMPERATURE
15
0.10
10
RL = 150 Ω
6
Figure 4
COMMON-MODE INPUT VOLTAGE
vs
SUPPLY VOLTAGE
1
8
2
−20
0
20
40
60
80
TA - Free-Air Temperature - °C
Figure 3
5
RL = 1 kΩ
10
4
2.2
−40
100
TA=25°C
12
V
−0.5
14
VCC = ± 5 V & ± 15 V
VO - Output Voltage -
2.8
V IO − Input Offset Voltage − mV
0.0
−1.0
OUTPUT VOLTAGE
vs
SUPPLY VOLTAGE
INPUT BIAS CURRENT
vs
FREE-AIR TEMPERATURE
100M
−20
VCC = ±15 V or ±5 V
RF = 1 kΩ
VI(PP) = 2 V
−30
−40
−50
−60
−70
−80
−90
−100
10k
100k
1M
10M
100M
f − Frequency − Hz
Figure 11
9
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TYPICAL CHARACTERISTICS
CROSSTALK
vs
FREQUENCY
OPEN LOOP GAIN AND
PHASE RESPONSE
vs
FREQUENCY
−20
30
VCC = ± 5 V & ±15 V
80
−40
Open Loop Gain − dB
Crosstalk − dB
100
−50
−60
−70
0
Gain
60
−30
40
Gain
−60
20
−90
Phase
0
−80
100k
1M
10M
100M
−120
−20
f − Frequency − Hz
1k
Figure 12
Phase
VCC = ± 15 V
Gain = 2
RF = 3.6 kΩ
RL = 150 Ω
−30
10k
100k
1M
f - Frequency - Hz
10M
−150
100M
Figure 13
−50
−50
VCC = ± 15 V
Gain = 2
VO(PP) = 2 V
−70
RL = 150 Ω
−50
VCC = ± 15 V
RL = 1 kΩ
G=5
f = 1 MHz
−55
−60
DISTORTION
vs
OUTPUT VOLTAGE
DISTORTION
vs
OUTPUT VOLTAGE
Distortion − dBc
THD - Total Harmonic Distortion - dBc
−40
HARMONIC DISTORTION
vs
FREQUENCY
−60
−66
−70
VCC = ± 15 V
RL = 150 Ω
G=5
f = 1 MHz
−55
Distortion − dBc
T
2nd Harmonic
−75
−60
−66
2nd
Harmonic
−70
−75
−80
3rd Harmonic
3rd Harmonic
−90
−85
−85
−90
−90
1M
f - Frequency - Hz
10M
0
20M
5
15
0
20
5
VO − Output Voltage − V
Figure 14
−40
VCC = ± 15 V
RL = 1 kΩ
−50 G = 2
VO(PP) = 2 V
Distortion − dBc
−50
−60
−70
2nd Harmonic
−40
VCC = ± 5 V
RL = 1 kΩ
G=2
VO(PP) = 2 V
−50
−70
2nd Harmonic
−80
−90
−90
20
DISTORTION
vs
FREQUENCY
−60
−80
15
Figure 16
DISTORTION
vs
FREQUENCY
−40
10
VO − Output Voltage − V
Figure 15
DISTORTION
vs
FREQUENCY
Distortion − dBc
10
Distortion − dBc
−100
100k
−80
−80
RL = 1 kΩ
VCC = ± 15 V
RL = 150 Ω
G=2
VO(PP) = 2 V
−60
−70
2nd Harmonic
−80
3rd Harmonic
3rd Harmonic
−90
3rd Harmonic
−100
−100
100k
10
1M
10M
100M
−100
100k
1M
10M
100M
100k
1M
10M
f − Frequency − Hz
f − Frequency − Hz
f − Frequency − Hz
Figure 17
Figure 18
Figure 19
100M
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TYPICAL CHARACTERISTICS
OUTPUT AMPLITUDE
vs
FREQUENCY
DISTORTION
vs
FREQUENCY
−70
2nd Harmonic
−80
3rd Harmonic
0
−1
−3
RF = 0 Ω
−90
−5
−100
10M
VCC = ± 15 V
Gain = 1
RL = 150 Ω
VO(PP) = 62 mV
−6
100k
100M
f − Frequency − Hz
Figure 20
0.2
0.3
RF = 750 Ω
Output Amplitude − dB
Output Amplitude − dB
0.3
0.1
−0.0
−0.1
RF = 620 Ω
−0.2
RF = 0 Ω
−0.3
−0.4
100k
0.2
6.3
RF = 3.6 kΩ
Output Amplitude − dB
Output Amplitude − dB
6.4
6
5
RF = 2.7 kΩ
4
RF = 1 kΩ
VCC = ±5 V
Gain = 2
RL = 150 Ω
VO(PP) = 125 mV
1M
10M
f - Frequency - Hz
Figure 26
RF = 3.6 kΩ
RF = 620 Ω
5
3
RF = 1 kΩ
2
1
RF = 0 Ω
1M
10M
f - Frequency - Hz
RF = 2.7 kΩ
4
VCC = ±15 V
Gain = 2
RL = 150 Ω
VO(PP) = 125 mV
0
100k
100M
6.2
VCC = ±15 V
Gain = 2
RL = 150 Ω
VO(PP) = 125 mV
6.3
5.9
RF = 2.7 kΩ
5.7
100M
6.4
RF = 3.6 kΩ
6.0
5.6
100k
100M
OUTPUT AMPLITUDE
vs
FREQUENCY
6.1
5.8
1M
10M
f - Frequency - Hz
Figure 25
OUTPUT AMPLITUDE
vs
FREQUENCY
7
100M
6
Figure 24
8
0
100k
7
RF = 750 Ω
−0.1
−0.4
100k
100M
1M
10M
f - Frequency - Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
0.1
−0.2
RF = 0 Ω
VCC = ± 5 V
Gain = 1
RL = 150 Ω
VO(PP) = 62 mV
Figure 22
−0.0
OUTPUT AMPLITUDE
vs
FREQUENCY
1
−4
8
VCC = ± 5 V
Gain = −1
RL = 150 Ω
VO(PP) = 62 mV
Figure 23
2
−3
−6
100k
100M
−0.3
1M
10M
f - Frequency - Hz
3
RF = 620 Ω
−2
OUTPUT AMPLITUDE
vs
FREQUENCY
0.4
VCC = ± 15 V
Gain = 1
RL = 150 Ω
VO(PP) = 62 mV
−1
Figure 21
OUTPUT AMPLITUDE
vs
FREQUENCY
0.4
1M
10M
f - Frequency - Hz
RF = 750 Ω
0
−5
Output Amplitude − dB
1M
RF = 620 Ω
−2
−4
1
RF = 750 Ω
Output Amplitude − dB
−60
100k
2
1
Output Amplitude − dB
−50
Distortion − dBc
2
VCC = ± 5 V
RL = 150 Ω
G=2
VO(PP) = 2 V
Output Amplitude − dB
−40
OUTPUT AMPLITUDE
vs
FREQUENCY
6.2
VCC = ±5 V
Gain = 2
RL = 150 Ω
VO(PP) = 125 mV
6.1
RF = 3.6 kΩ
6.0
5.9
RF = 2.7 kΩ
5.8
5.7
RF = 1 kΩ
1M
10M
f - Frequency - Hz
Figure 27
100M
5.6
100k
RF = 1 kΩ
1M
10M
f - Frequency - Hz
100M
Figure 28
11
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS
OUTPUT AMPLITUDE
vs
FREQUENCY
OUTPUT AMPLITUDE
vs
FREQUENCY
2
CL= 10 pF
1
6
Output Amplitude − dB
5
RL = 1 kΩ
4
RL = 150 Ω
3
2
VCC = ±15 V
Gain = 2
RL = 2.7 k Ω
VO(PP) = 125 mV
1
0
100k
0
−1
RF = 3.9 kΩ
−2
RF = 1 kΩ
−3
VCC = ± 15 V
Gain = −1
RL = 150 Ω
VO(PP) = 62 mV
−4
−5
1M
10M
f - Frequency - Hz
−6
100k
100M
Figure 29
VO(PP)=0.4 V
−10
−20
VCC = ± 15 V
Gain = 2
RF = 2.7 kΩ
RL = 150 Ω
−25
VCC = ± 5 V
0.01%
100
VCC = ± 15 V
0.1%
80
VCC = ± 5 V
0.1%
0.04
RF = 360 Ω
1M
10M
f - Frequency - Hz
1
100M
2
3
4
VO - Output Step Voltage - V
Differential Phase
VCC = ± 5 V
0
0.6°
Number of 150-Ω Loads
Figure 35
4
Gain = 2
40 IRE-PAL Modulation
Worst Case ± 100 IRE Ramp
0.5°
0.3°
VCC = ± 15 V
0.2°
VCC = ± 5 V
0.4°
VCC = ± 15 V
0.3°
VCC = ± 5 V
0.2°
0.1°
0°
3
4
DIFFERENTIAL PHASE
vs
NUMBER OF 150-Ω LOADS
0.1°
0.04
3
Figure 34
Gain = 2
RF = 1 kΩ
40 IRE-NTSC Modulation
Worst Case ± 100 IRE Ramp
0.4°
0.08
2
Number of 150-Ω Loads
0.5°
VCC = ± 15 V
2
1
DIFFERENTIAL PHASE
vs
NUMBER OF 150-Ω LOADS
0.12
VCC = ± 5 V
0
5
Figure 33
Gain = 2
40 IRE-PAL Modulation
Worst Case ± 100 IRE Ramp
1
VCC = ± 15 V
0.06
0.02
20
DIFFERENTIAL GAIN
vs
NUMBER OF 150-Ω LOADS
0.16
0.08
60
40
100M
Gain = 2
40 IRE-NTSC Modulation
Worst Case ± 100 IRE Ramp
0.10
VCC = ± 15 V
0.01%
120
1M
10M
f - Frequency - Hz
DIFFERENTIAL GAIN
vs
NUMBER OF 150-Ω LOADS
0.12
Figure 32
0.2
VCC = ± 5 V
Gain = −1
RL = 150 Ω
VO(PP) = 62 mV
−4
Figure 31
140
VO(PP)=125 mV
RF = 1 kΩ
−3
−6
100k
100M
Differential Gain − %
0
−30
100k
Differential Gain − %
1M
10M
f - Frequency - Hz
160
Settling Time − ns
V O(PP) - Output Voltage - dBV
VO(PP)=2.25 V
−15
RF = 3.9 kΩ
−2
−5
180
−5
−1
SETTING TIME
vs
OUTPUT STEP
10
5
RF = 5.6 kΩ
0
Figure 30
OUTPUT AMPLITUDE
vs
FREQUENCY
12
1
RF = 5.6 kΩ
Differential Phase
Output Amplitude − dB
7
2
Output Amplitude − dB
8
OUTPUT AMPLITUDE
vs
FREQUENCY
0°
1
2
3
4
1
2
3
Number of 150-Ω Loads
Number of 150-Ω Loads
Figure 36
Figure 37
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TYPICAL CHARACTERISTICS
1-V STEP RESPONSE
5-V STEP RESPONSE
−0.0
−0.2
−0.4
0.6
2
0.4
1
0
−1
VCC = ± 5 V
Gain = −1
RF = 3.9 kΩ
RL = 150 Ω
−2
−0.6
50
100 150 200 250 300 350 400
t - Time - ns
VCC = ± 15 V
Gain = 2
RF = 2.7 kΩ
RL = 150 Ω
0.2
−0.0
−0.2
−0.4
−3
0
−0.6
0
50
100 150 200 250 300 350 400
t - Time - ns
Figure 38
Figure 39
0
50
100 150 200 250 300 350 400
t - Time - ns
Figure 40
20-V STEP RESPONSE
15
VCC = ± 15 V
Gain = 5
RF = 2.7 kΩ
RL = 150 & 1 kΩ
10
V O − Output Voltage − V
V O − Output Voltage − V
0.2
1-V STEP RESPONSE
3
V O − Output Voltage − V
VCC = ± 5 V
Gain = 2
RF = 2.7 kΩ
RL = 150 Ω
0.4
V O − Output Voltage − V
0.6
5
0
−5
−10
−15
0
100
200
300
t - Time - ns
400
500
Figure 41
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
THEORY OF OPERATION
bipolar process with NPN and PNP transistors possessing
fTs of several GHz. This results in an exceptionally high
performance amplifier that has a wide bandwidth, high
slew rate, fast settling time, and low distortion. A simplified
schematic is shown in Figure 42.
The THS405x is a high-speed operational amplifier
configured in a voltage feedback architecture. It is built
using a 30-V, dielectrically isolated, complementary
(7) VCC +
(6) OUT
IN − (2)
IN + (3)
(4) VCC −
NULL (1)
NULL (8)
Figure 42. THS405x Simplified Schematic
NOISE CALCULATIONS AND NOISE FIGURE
Noise can cause errors on very small signals. This is especially true when amplifying small signals, where signal-to-noise
ratio (SNR) is very important. The noise model for the THS405x is shown in Figure 43. This model includes all of the noise
sources as follows:
•
•
•
•
14
en = Amplifier internal voltage noise (nV/√Hz)
IN+ = Noninverting current noise (pA/√Hz)
IN− = Inverting current noise (pA/√Hz)
eRx = Thermal voltage noise associated with each resistor (eRx = 4 kTRx )
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APPLICATION INFORMATION
NOISE CALCULATIONS AND NOISE FIGURE (CONTINUED)
RS
eRs
en
eni
IN+
IN−
Figure 43. Noise Model
The total equivalent input noise density (eni) is calculated by using the following equation:
e
ni
+
Ǹ
ǒenǓ ) ǒIN )
2
R
Ǔ
S
2
ǒ
) IN–
ǒR F ø R G ǓǓ
2
ǒ
) 4 kTRs ) 4 kT R ø R
F
G
Ǔ
Where:
k = Boltzmann’s constant = 1.380658 × 10−23
T = Temperature in degrees Kelvin (273 +°C)
RF || RG = Parallel resistance of RF and RG
To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (eni) by the overall amplifier
gain (AV).
e no + e
ni
A
V
ǒ
+ e ni 1 )
Ǔ
RF
(noninverting case)
RG
As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the closed-loop gain
is increased (by reducing RG), the input noise is reduced considerably because of the parallel resistance term. This leads
to the general conclusion that the most dominant noise sources are the source resistor (RS) and the internal amplifier noise
voltage (en). Because noise is summed in a root-mean-squares method, noise sources smaller than 25% of the largest
noise source can be effectively ignored. This can greatly simplify the formula and make noise calculations much easier to
calculate.
For more information on noise analysis, please refer to the Noise Analysis section in Operational Amplifier Circuits
Applications Report (literature number SLVA043).
15
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
NOISE CALCULATIONS AND NOISE FIGURE (CONTINUED)
This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise figure is a
measure of noise degradation caused by the amplifier. The value of the source resistance must be defined and is typically
50 Ω in RF applications.
NF +
ȱ e 2ȳ
10logȧ ni ȧ
ȧ 2ȧ
ȲǒeRsǓ ȴ
Because the dominant noise components are generally the source resistance and the internal amplifier noise voltage, we
can approximate noise figure as:
NF +
ȱ ȡǒ Ǔ2 ǒ
ȧ en ) IN )
ȧ
ȧ Ȣ
10logȧ1 )
4 kTR
ȧ
S
ȧ
Ȳ
Ǔ ȣȳ
S ȧȧ
Ȥȧ
ȧ
ȧ
ȧ
ȴ
2
R
Figure 44 shows the noise figure graph for the THS405x.
NOISE FIGURE
vs
SOURCE RESISTANCE
40
f = 10 kHz
TA = 25°C
35
Noise Figure (dB)
30
25
20
15
10
5
0
10
100
1k
10k
Source Resistance - Ω
100k
Figure 44. Noise Figure vs Source Resistance
16
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
DRIVING A CAPACITIVE LOAD
Driving capacitive loads with high performance amplifiers is not a problem as long as certain precautions are taken. The
first is to realize that the THS405x has been internally compensated to maximize its bandwidth and slew rate performance.
When the amplifier is compensated in this manner, capacitive loading directly on the output will decrease the device’s phase
margin leading to high frequency ringing or oscillations. Therefore, for capacitive loads of greater than 10 pF, it is
recommended that a resistor be placed in series with the output of the amplifier, as shown in Figure 45. A minimum value
of 20 Ω should work well for most applications. For example, in 75-Ω transmission systems, setting the series resistor value
to 75 Ω both isolates any capacitance loading and provides the proper line impedance matching at the source end.
1 kΩ
1 kΩ
Input
_
20 Ω
Output
THS405x
+
CLOAD
Figure 45. Driving a Capacitive Load
OFFSET NULLING
The THS405x has very low input offset voltage for a high-speed amplifier. However, if additional correction is required, an
offset nulling function has been provided on the THS4051. The input offset can be adjusted by placing a potentiometer
between terminals 1 and 8 of the device and tying the wiper to the negative supply. This is shown in Figure 46.
VCC+
0.1 µF
+
THS4051
_
10 kΩ
0.1 µF
VCC −
Figure 46. Offset Nulling Schematic
17
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
OFFSET VOLTAGE
The output offset voltage, (VOO) is the sum of the input offset voltage (VIO) and both input bias currents (IIB) times the
corresponding gains. The following schematic and formula can be used to calculate the output offset voltage:
RF
RG
+IIB−
−
−
+IIB+
V
OO
+V
IO
ǒ
1)
R
R
VO
+
RS
F
G
Ǔ
"I
IB)
R
+
VIO
S
ǒ
1)
R
R
F
G
Ǔ
"I
IB*
R
F
Figure 47. Output Offset Voltage Model
OPTIMIZING UNITY GAIN RESPONSE
Internal frequency compensation of the THS405x was selected to provide very wideband performance yet still maintain
stability when operated in a noninverting unity gain configuration. When amplifiers are compensated in this manner there
is usually peaking in the closed loop response and some ringing in the step response for very fast input edges, depending
upon the application. This is because a minimum phase margin is maintained for the G=+1 configuration. For optimum
settling time and minimum ringing, a feedback resistor of 620 Ω should be used as shown in Figure 48. Additional
capacitance can also be used in parallel with the feedback resistance if even finer optimization is required.
Input
+
THS405x
Output
_
620 Ω
Figure 48. Noninverting, Unity Gain Schematic
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APPLICATION INFORMATION
GENERAL CONFIGURATIONS
When receiving low-level signals, limiting the bandwidth of the incoming signals into the system is often required. The
simplest way to accomplish this is to place an RC filter at the noninverting terminal of the amplifier (see Figure 49).
RG
RF
−
+
VI
R1
C1
f
–3dB
+
Figure 49. Single-Pole Low-Pass Filter
If even more attenuation is needed, a multiple pole filter is required. The Sallen-Key filter can be used for this task. For best
results, the amplifier should have a bandwidth that is 8 to 10 times the filter frequency bandwidth. Failure to do this can
result in phase shift of the amplifier.
C
VI
R1
R2
C2
RG
Figure 50. 2-Pole Low-Pass Sallen-Key Filter
19
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
CIRCUIT LAYOUT CONSIDERATIONS
To achieve the levels of high frequency performance of the THS405x, follow proper printed-circuit board high frequency
design techniques. A general set of guidelines is given below. In addition, a THS405x evaluation board is available to use
as a guide for layout or for evaluating the device performance.
D Ground planes − It is highly recommended that a ground plane be used on the board to provide all components
with a low inductive ground connection. However, in the areas of the amplifier inputs and output, the ground plane
can be removed to minimize the stray capacitance.
D Proper power supply decoupling − Use a 6.8-µF tantalum capacitor in parallel with a 0.1-µF ceramic capacitor
on each supply terminal. It may be possible to share the tantalum among several amplifiers depending on the
application, but a 0.1-µF ceramic capacitor should always be used on the supply terminal of every amplifier. In
addition, the 0.1-µF capacitor should be placed as close as possible to the supply terminal. As this distance
increases, the inductance in the connecting trace makes the capacitor less effective. The designer should strive
for distances of less than 0.1 inches between the device power terminals and the ceramic capacitors.
D Sockets − Sockets are not recommended for high-speed operational amplifiers. The additional lead inductance
in the socket pins will often lead to stability problems. Surface-mount packages soldered directly to the
printed-circuit board is the best implementation.
D Short trace runs/compact part placements − Optimum high frequency performance is achieved when stray series
inductance has been minimized. To realize this, the circuit layout should be made as compact as possible,
thereby minimizing the length of all trace runs. Particular attention should be paid to the inverting input of the
amplifier. Its length should be kept as short as possible. This will help to minimize stray capacitance at the input
of the amplifier.
D Surface-mount passive components − Using surface-mount passive components is recommended for high
frequency amplifier circuits for several reasons. First, because of the extremely low lead inductance of
surface-mount components, the problem with stray series inductance is greatly reduced. Second, the small size
of surface-mount components naturally leads to a more compact layout thereby minimizing both stray inductance
and capacitance. If leaded components are used, it is recommended that the lead lengths be kept as short as
possible.
GENERAL POWERPAD DESIGN CONSIDERATIONS
The THS405x is available packaged in a thermally-enhanced DGN package, which is a member of the PowerPAD family
of packages. This package is constructed using a downset leadframe upon which the die is mounted [see Figure 51(a) and
Figure 51(b)]. This arrangement results in the lead frame being exposed as a thermal pad on the underside of the package
[see Figure 51(c)]. Because this thermal pad has direct thermal contact with the die, excellent thermal performance can
be achieved by providing a good thermal path away from the thermal pad.
The PowerPAD package allows for both assembly and thermal management in one manufacturing operation. During the
surface-mount solder operation (when the leads are being soldered), the thermal pad can also be soldered to a copper area
underneath the package. Through the use of thermal paths within this copper area, heat can be conducted away from the
package into either a ground plane or other heat dissipating device.
The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of the surface
mount with the, heretofore, awkward mechanical methods of heatsinking.
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SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
GENERAL POWERPAD DESIGN CONSIDERATIONS (CONTINUED)
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
NOTE A:
Bottom View (c)
The thermal pad is electrically isolated from all terminals in the package.
Figure 51. Views of Thermally Enhanced DGN Package
Although there are many ways to properly heatsink this device, the following steps illustrate the recommended approach.
Thermal pad area (68 mils x 70 mils) with 5 vias
(Via diameter = 13 mils)
Figure 52. PowerPAD PCB Etch and Via Pattern
1. Prepare the PCB with a top side etch pattern as shown in Figure 52. There should be etch for the leads as well
as etch for the thermal pad.
2. Place five holes in the area of the thermal pad. These holes should be 13 mils in diameter. Keep them small so
that solder wicking through the holes is not a problem during reflow.
3. Additional vias may be placed anywhere along the thermal plane outside of the thermal pad area. This helps
dissipate the heat generated by the THS405xDGN IC. These additional vias may be larger than the 13-mil
diameter vias directly under the thermal pad. They can be larger because they are not in the thermal pad area
to be soldered so that wicking is not a problem.
4. Connect all holes to the internal ground plane.
5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection
methodology. Web connections have a high thermal resistance connection that is useful for slowing the heat
transfer during soldering operations. This makes the soldering of vias that have plane connections easier. In this
application, however, low thermal resistance is desired for the most efficient heat transfer. Therefore, the holes
under the THS405xDGN package should make their connection to the internal ground plane with a complete
connection around the entire circumference of the plated-through hole.
6. The top-side solder mask should leave the terminals of the package and the thermal pad area with its five holes
exposed. The bottom-side solder mask should cover the five holes of the thermal pad area. This prevents solder
from being pulled away from the thermal pad area during the reflow process.
7. Apply solder paste to the exposed thermal pad area and all of the IC terminals.
8. With these preparatory steps in place, the THS405xDGN IC is simply placed in position and run through the
solder reflow operation as any standard surface-mount component. This results in a part that is properly installed.
21
www.ti.com
SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
GENERAL POWERPAD DESIGN CONSIDERATIONS (CONTINUED)
The actual thermal performance achieved with the THS405xDGN in its PowerPAD package depends on the application.
In the example above, if the size of the internal ground plane is approximately 3 inches × 3 inches (or 76.2 mm × 76.2 mm),
then the expected thermal coefficient, θJA, is about 58.4°C/W. For comparison, the non-PowerPAD version of the
THS405x IC (SOIC) is shown. For a given θJA, the maximum power dissipation is shown in Figure 53 and is calculated
by the following formula:
P
Where:
D
+
ǒ
T
–T
MAX A
q
JA
Ǔ
PD = Maximum power dissipation of THS405x IC (watts)
TMAX = Absolute maximum junction temperature (150°C)
TA
= Free-ambient air temperature (°C)
θJA = θJC + θCA
θJC = Thermal coefficient from junction to case
θCA = Thermal coefficient from case to ambient air (°C/W)
MAXIMUM POWER DISSIPATION
vs
FREE-AIR TEMPERATURE
3.5
DGN Package
θJA = 58.4°C/W
2 oz. Trace And Copper Pad
With Solder
Maximum Power Dissipation − W
3
2.5
SOIC Package
High-K Test PCB
θJA = 98°C/W
2
DGN Package
θJA = 158°C/W
2 oz. Trace And
Copper Pad
Without Solder
1.5
1
0.5
SOIC Package
Low-K Test PCB
θJA = 167°C/W
0
−40
NOTE A:
TJ = 150°C
−20
0
20
40
60
80
TA − Free-Air Temperature − °C
100
Results are with no air flow and PCB size = 3”× 3”
Figure 53. Maximum Power Dissipation vs Free-Air Temperature
More complete details of the PowerPAD installation process and thermal management techniques can be found in the
Texas Instruments Technical Brief, PowerPAD Thermally Enhanced Package. This document can be found at the TI web
site (www.ti.com) by searching on the key word PowerPAD. The document can also be ordered through your local TI sales
office. Refer to literature number SLMA002 when ordering.
22
www.ti.com
SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
GENERAL POWERPAD DESIGN CONSIDERATIONS (CONTINUED)
The next consideration is the package constraints. The two sources of heat within an amplifier are quiescent power and
output power. The designer should never forget about the quiescent heat generated within the device, especially devices
with multiple amplifiers. Because these devices have linear output stages (Class A-B), most of the heat dissipation is at
low output voltages with high output currents. Figure 54 to Figure 57 show this effect, along with the quiescent heat, with
an ambient air temperature of 50°C. Obviously, as the ambient temperature increases, the limit lines shown will drop
accordingly. The area under each respective limit line is considered the safe operating area. Any condition above this line
will exceed the amplifier’s limits and failure may result. When using VCC = ±5 V, there is generally not a heat problem, even
with SOIC packages. But, when using VCC = ±15 V, the SOIC package is severely limited in the amount of heat it can
dissipate. The other key factor when looking at these graphs is how the devices are mounted on the PCB. The PowerPAD
devices are extremely useful for heat dissipation. But, the device should always be soldered to a copper plane to fully use
the heat dissipation properties of the PowerPAD. The SOIC package, on the other hand, is highly dependent on how it
is mounted on the PCB. As more trace and copper area is placed around the device, θJA decreases and the heat dissipation
capability increases. The currents and voltages shown in these graphs are for the total package. For the dual amplifier
package (THS4052), the sum of the RMS output currents and voltages should be used to choose the proper package. The
graphs shown assume that both amplifier outputs are identical.
THS4051
MAXIMUM RMS OUTPUT CURRENT
vs
RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS
VCC = ± 5 V
Tj = 150°C
TA = 50°C
180
1000
Maximum Output
Current Limit Line
| IO | − Maximum RMS Output Current − mA
| IO | − Maximum RMS Output Current − mA
200
160
140
Package With
θJA < = 120°C/W
120
100
SO-8 Package
θJA = 167°C/W
Low-K Test PCB
80
60
40
Safe Operating
Area
20
1
2
3
4
| VO | − RMS Output Voltage − V
Figure 54
TJ = 150°C
TA = 50°C
VCC = ± 15 V
Maximum Output
Current Limit Line
DGN Package
θJA = 58.4°C/W
100
SO-8 Package
θJA = 98°C/W
High-K Test PCB
SO-8 Package
θJA = 167°C/W
Low-K Test PCB
Safe Operating
Area
10
0
0
THS4051
MAXIMUM RMS OUTPUT CURRENT
vs
RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS
5
0
3
6
9
12
15
| VO | − RMS Output Voltage − V
Figure 55
23
www.ti.com
SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
GENERAL POWERPAD DESIGN CONSIDERATIONS (CONTINUED)
THS4052
MAXIMUM RMS OUTPUT CURRENT
vs
RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS
180
160
140
120
100
SO-8 Package
θJA = 167°C/W
Low-K Test PCB
80
60
Safe Operating Area
40
SO-8 Package
θJA = 98°C/W
High-K Test PCB
20
0
0
VCC = ± 5 V
TJ = 150°C
TA = 50°C
Both Channels
1
2
3
4
| VO | − RMS Output Voltage − V
Figure 56
24
1000
Maximum Output
Current Limit Line
Package With
θJA ≤ 60°C/W
| IO | − Maximum RMS Output Current − mA
| IO | − Maximum RMS Output Current − mA
200
THS4052
MAXIMUM RMS OUTPUT CURRENT
vs
RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS
VCC = ± 15 V
TJ = 150°C
TA = 50°C
Both Channels
Maximum Output
Current Limit Line
100
SO-8 Package
θJA = 98°C/W
High-K Test PCB
10
DGN Package
θJA = 58.4°C/W
Safe Operating Area
5
1
0
SO-8 Package
θJA = 167°C/W
Low-K Test PCB
3
6
9
12
| VO | − RMS Output Voltage − V
Figure 57
15
www.ti.com
SLOS238D − MAY 1999 − REVISED AUGUST 2008
APPLICATION INFORMATION
EVALUATION BOARD
An evaluation board is available for the THS4051 (literature number SLOP220) and THS4052 (literature number
SLOP234). This board has been configured for very low parasitic capacitance in order to realize the full performance of
the amplifier. A schematic of the evaluation board is shown in Figure 58. The circuitry has been designed so that the
amplifier may be used in either an inverting or noninverting configuration. For more information, please refer to the
THS4051 EVM User’s Guide or the THS4052 EVM User’s Guide. To order the evaluation board, contact your local TI sales
office or distributor.
VCC+
+
C5
0.1 µF
R4
2 kΩ
IN +
C2
6.8 µF
NULL
R5
49.9 Ω
+
R3
49.9 Ω
OUT
THS4051
_
NULL
R2
2 kΩ
+
C4
0.1 µF
C1
6.8 µF
IN −
R1
49.9 Ω
VCC −
Figure 58. THS4051 Evaluation Board Schematic
25
PACKAGE OPTION ADDENDUM
www.ti.com
10-Jun-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
5962-9959901Q2A
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59629959901Q2A
THS4051MFKB
5962-9959901QPA
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
9959901QPA
THS4051M
Samples
THS4051CD
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4051C
Samples
THS4051CDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
ACQ
Samples
THS4051CDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
ACQ
Samples
THS4051CDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4051C
Samples
THS4051ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4051I
Samples
THS4051IDG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4051I
Samples
THS4051IDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ACR
Samples
THS4051IDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ACR
Samples
THS4051IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4051I
Samples
THS4051IDRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4051I
Samples
THS4051MFKB
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59629959901Q2A
THS4051MFKB
THS4051MJG
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
THS4051MJG
Samples
THS4051MJGB
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
9959901QPA
THS4051M
Samples
THS4052CD
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4052C
Samples
THS4052CDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
ACE
Samples
THS4052CDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
ACE
Samples
Addendum-Page 1
Samples
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
10-Jun-2022
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
THS4052CDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4052C
Samples
THS4052ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4052I
Samples
THS4052IDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ACF
Samples
THS4052IDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ACF
Samples
THS4052IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4052I
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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