SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
D Ultralow 3.4 mA Per Channel Quiescent
D
D
D
D
D
D
D
THS4081
D OR DGN PACKAGE
(TOP VIEW)
Current
High Speed
− 175 MHz Bandwidth (−3 dB, G = 1)
− 230 V/µs Slew Rate
− 43 ns Settling Time (0.1%)
High Output Drive, IO = 85 mA (typ)
Excellent Video Performance
− 35 MHz Bandwidth (0.1 dB, G = 1)
− 0.01% Differential Gain
− 0.05° Differential Phase
Very Low Distortion
− THD = −64 dBc (f = 1 MHz, RL = 150 Ω)
− THD = −79 dBc (f = 1 MHz, RL = 1 kΩ)
Wide Range of Power Supplies
− VCC = ±5 V to ±15 V
Available in Standard SOIC or MSOP
PowerPAD Package
Evaluation Module Available
NC
IN −
IN +
1
8
2
7
3
6
VCC−
4
5
NC
VCC+
OUT
NC
NC − No internal connection
THS4082
D OR DGN PACKAGE
(TOP VIEW)
1OUT
1IN −
1IN +
VCC−
1
8
2
7
3
6
4
5
VCC+
2OUT
2IN −
2IN+
Cross Section View Showing
PowerPAD Option (DGN)
description
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
3.8
3.6
I CC − Supply Current − mA
The THS4081 and THS4082 are ultralow-power,
high-speed voltage feedback amplifiers that are
ideal for communication and video applications.
These amplifiers operate off of a very low 3.4-mA
quiescent current per channel and have a high
output drive capability of 85ĂmA. The signalamplifier THS4081 and the dual-amplifier
THS4082 offer very good ac performance with
175-MHz bandwidth, 230-V/µs slew rate, and
43-ns settling time (0.1%). With total harmonic
distortion (THD) of −64 dBc at f = 1 MHz, the
THS4081 and THS4082 are ideally suited for
applications requiring low distortion.
TA=85°C
3.4
3.2
TA=25°C
3.0
2.8
2.6
TA=−40°C
2.4
2.2
5
RELATED DEVICES
DEVICE
DESCRIPTION
THS4011/2
THS4031/2
THS4051/2
290-MHz Low Distortion High-Speed Amplifiers
100-MHz Low Noise High Speed-Amplifiers
70-MHz High-Speed Amplifiers
7
9
11
13
± VCC - Supply Voltage - V
15
CAUTION: The THS4081 and THS4082 provide ESD protection circuitry. However, permanent damage can still occur if this device
is subjected to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance
degradation or loss of functionality.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
Copyright 2001, Texas Instruments Incorporated
!"#$%& " ' ()##* & %' "! +),- (%& " .%&*/
#".)(&' (" !"#$ &" '+*( ! (%& " ' +*# &0* &*#$' "! *1%' '&#)$* &'
'&% .%#. 2%##% &3/ #".)(& " +#"(*'' 4 ."*' "& *(*''%# -3 (-).*
&*'& 4 "! %-- +%#%$*&*#'/
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
1
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
AVAILABLE OPTIONS
PACKAGED DEVICES
TA
0°C to 70°C
NUMBER OF
CHANNELS
PLASTIC
SMALL OUTLINE†
(D)
PLASTIC
MSOP†
(DGN)
MSOP
SYMBOL
EVALUATION
MODULE
1
THS4081CD
THS4081CDGN
AEO
THS4081EVM
2
THS4082CD
THS4082CDGN
AER
THS4082EVM
1
THS4081ID
THS4081IDGN
AEQ
—
−40°C to 85°C
2
THS4082ID
THS4082IDGN
AEP
† The D and DGN packages are available taped and reeled. Add an R suffix to the device type (i.e., THS4081CDGN).
functional block diagram
IN−
IN+
2
6
3
OUT
Figure 1. THS4081 − Single Channel
VCC
1IN−
1OUT
1IN+
2IN−
2OUT
2IN+
−VCC
Figure 2. THS4082 − Dual Channel
2
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
—
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
absolute maximum ratings over operating free-air temperature (unless otherwise noted)†
Supply voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±16.5 V
Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±VCC
Output current, IO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150 mA
Differential input voltage, VIO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±4 V
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Dissipation Rating Table
Maximum junction temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C
Operating free-air temperature, TA: C-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C
I-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 85°C
Storage temperature, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C
† Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
θJA
(°C/W)
θJC
(°C/W)
TA = 25
25°C
C
POWER RATING
D
167‡
38.3
740 mW
DGN§
58.4
4.7
2.14 W
‡ This data was taken using the JEDEC standard Low-K test PCB. For the JEDEC Proposed
High-K test PCB, the θJA is 95°C/W with a power rating at TA = 25°C of 1.32 W.
§ This data was taken using 2 oz. trace and copper pad that is soldered directly to a 3 in. × 3 in.
PC. For further information, refer to Application Information section of this data sheet.
recommended operating conditions
MIN
Supply voltage, VCC+ and VCC−
MAX
±5
±15
Single supply
10
30
0
70
−40
85
C-suffix
Operating free-air temperature, TA
NOM
Dual supply
I-suffix
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
UNIT
V
°C
3
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
electrical characteristics at TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise noted)
dynamic performance
PARAMETER
MIN
TYP
Gain = 1
VCC = ± 15 V
VCC = ± 5 V
Gain = −1
Bandwidth for 0.1 dB flatness
VCC = ± 15 V
VCC = ± 5 V
Gain = 1
Full power bandwidth†
VO(pp) = 20 V,
VO(pp) = 5 V,
VCC = ± 15 V
VCC = ± 5 V
Slew rate‡
VCC = ± 15 V,
VCC = ± 5 V,
20-V step,
Gain = 5
230
5-V step
Gain = 1
170
VCC = ± 15 V,
VCC = ± 5 V,
5-V step
Settling time to 0.1%
VCC = ± 15 V,
VCC = ± 5 V,
5-V step
Settling time to 0.01%
Small-signal bandwidth (−3 dB)
BW
SR
TEST CONDITIONS
VCC = ± 15 V
VCC = ± 5 V
ts
MAX
UNIT
175
MHz
160
70
MHz
65
35
2-V step
2-V step
MHz
35
2.7
MHz
7.1
V/ s
V/µs
43
Gain = −1
ns
30
233
Gain = −1
ns
280
† Slew rate is measured from an output level range of 25% to 75%.
‡ Full power bandwidth = slew rate/2π VO(Peak).
noise/distortion performance
PARAMETER
THD
Vn
In
XT
4
Total harmonic distortion
Input voltage noise
Input current noise
TEST CONDITIONS
VO(pp) = 2 V,
f = 1 MHz, Gain = 2
VCC = ± 5 V or ± 15 V,
VCC = ± 5 V or ± 15 V,
VCC = ± 15 V
VCC = ± 5 V
MIN
RL = 150 Ω
−64
RL = 1 kΩ
−79
RL = 150 Ω
−64
RL = 1 kΩ
−77
f = 10 kHz
f = 10 kHz
Differential gain error
Gain = 2,
40 IRE modulation,
NTSC,
± 100 IRE ramp
VCC = ± 15 V
VCC = ± 5 V
Differential phase error
Gain = 2,
40 IRE modulation,
NTSC,
± 100 IRE ramp
VCC = ± 15 V
VCC = ± 5 V
Channel-to-channel crosstalk
(THS4082 only)
VCC = ± 5 V or ± 15 V,
f = 1 MHz
POST OFFICE BOX 655303
TYP
• DALLAS, TEXAS 75265
MAX
UNIT
dBc
10
nV/√Hz
0.7
pA/√Hz
0.01%
0.05°
0.01%
0.05°
−75
dB
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
electrical characteristics at TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise noted) (continued)
dc performance
PARAMETER
TEST CONDITIONS
VCC = ± 15 V,
VO = ± 10 V, RL = 1 kΩ
TA = 25°C
TA = full range†
VCC = ± 5 V,
VO = ± 2.5 V, RL = 250 Ω
TA = 25°C
TA = full range†
Open loop gain
VOS
Offset voltage drift
Input bias current
IOS
Input offset current
TYP
10
19
8
VCC = ± 5 V or ± 15 V
MAX
16
V/mV
7
1
7
8
TA = 25°C
TA = full range†
1.2
TA = 25°C
TA = full range†
20
mV
µV/°C
15
6
8
µA
A
250
400
Offset current drift
TA = full range†
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix
UNIT
V/mV
9
TA = 25°C
TA = full range†
TA = full range†
Input offset voltage
IIB
MIN
0.3
nA
nA/°C
input characteristics
PARAMETER
TEST CONDITIONS
VICR
Common mode input voltage range
VCC = ± 15 V
VCC = ± 5 V
CMRR
Common mode rejection ratio
VCC = ± 15 V, VICR = ± 12 V,
VCC = ± 5 V, VICR = ± 2 V,
RI
Input resistance
TA = full range†
TA = full range†
MIN
TYP
± 13.8
±14.1
± 3.8
± 3.9
V
78
90
dB
84
93
dB
1
MΩ
1.5
pF
CI
Input capacitance
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix
MAX
UNIT
output characteristics
PARAMETER
VO
Output voltage swing
TEST CONDITIONS
VCC = ± 15 V,
VCC = ± 5 V,
MIN
TYP
RL = 250 Ω
±12
±13.6
RL = 150 Ω
±3.4
± 3.8
±13.5
±13.8
±3.5
± 3.9
65
85
50
70
VCC = ± 15 V
VCC = ± 5 V
RL = 1 kΩ
RL = 20 Ω
IO
Output current
VCC = ± 15 V
VCC = ± 5 V
ISC
Short-circuit current‡
VCC = ± 15 V
100
MAX
UNIT
V
V
mA
mA
RO
Output resistance
Open loop
13
Ω
‡ Observe power dissipation ratings to keep the junction temperature below the absolute maximum rating when the output is heavily loaded or
shorted. See the absolute maximum ratings section of this data sheet for more information.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
5
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
electrical characteristics at TA = 25°C, VCC = ±15 V, RL = 150 Ω (unless otherwise noted) (continued)
power supply
PARAMETER
TEST CONDITIONS
Dual supply
VCC
Supply voltage operating range
Single supply
VCC = ± 15 V
ICC
Supply current (per amplifier)
VCC = ± 5 V
PSRR Power supply rejection ratio
VCC = ± 5 V or ± 15 V
† Full range = 0°C to 70°C for C suffix and − 40°C to 85°C for I suffix
6
POST OFFICE BOX 655303
MIN
TYP
MAX
±4.5
±16.5
9
33
TA = 25°C
TA = full range†
3.4
4.2
TA = 25°C
TA = full range†
TA = full range†
2.9
3.7
• DALLAS, TEXAS 75265
UNIT
V
5
mA
4.5
79
90
dB
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
TYPICAL CHARACTERISTICS
45°
80
0°
60
Gain
40
−45°
90°
Phase
20
135°
VCC = ±15 V
Gain = 1
RF = 0 Ω
RL = 150 Ω
0
−20
−40
−60
180°
0
CROSSTALK
vs
FREQUENCY
20
Crosstalk − dB
100
Phase Responce
Open Loop Gain − dB
OPEN LOOP GAIN
& PHASE RESPONSE
vs
FREQUENCY
VCC = ±5 V and ±15 V
−20
100
1k
10k
100k 1M
10M 100M
−225°
1G
−80
100k
1M
f − Frequency − Hz
TOTAL HARMONIC DISTORTION
vs
FREQUENCY
−60
RL = 150 Ω
−70
RL = 1 kΩ
−90
100.00
1M
f - Frequency - Hz
−50
RL = 150 Ω
−70
RL = 1 kΩ
−80
−90
130
VCC = ±5 V(0.1%)
VCC = ±15 V(0.1%)
2
1000.00
10M
3
4
5
VO − Output Step Voltage − V
Figure 7
DISTORTION
vs
OUTPUT VOLTAGE
DISTORTION
vs
OUTPUT VOLTAGE
−50
−50
2nd Harmonic
2nd Harmonic
−VCC
−40
+VCC
−60
−60
3rd Harmonic
−70
−80
VCC = ± 15 V
RL = 1 kΩ
Gain = 5
f = 1 MHz
−90
−80
Distortion − dBc
−20
Distortion − dBc
PSRR - Power Supply Rejection Ratio - dB
VCC = ±15 V(0.01%)
10
100.00
1M
f - Frequency - Hz
−60
100M
3rd Harmonic
−70
−80
VCC = ± 15 V
RL = 150 Ω
Gain = 5
f = 1 MHz
−90
−100
Figure 8
VCC = ±5 V(0.01%)
170
50
VCC = ± 15 V & ± 5 V
1M
10M
f - Frequency - Hz
210
Figure 6
POWER SUPPLY REJECTION
RATIO
vs
FREQUENCY
−100
100k
250
90
Figure 5
0
290
−60
−100
10.00
100k
1000.00
10M
330
VCC = ± 5 V
Gain = 2
VO(PP) = 2 V
Settling Time − ns
THD - Total Harmonic Distortion - dBc
THD - Total Harmonic Distortion - dBc
−40
VCC = ± 15 V
Gain = 2
VO(PP) = 2 V
−100
10.00
100k
1G
SETTLING
vs
OUTPUT STEP
TOTAL HARMONIC DISTORTION
vs
FREQUENCY
−40
−80
100M
Figure 4
Figure 3
−50
10M
f − Frequency − Hz
−100
0
5
10
15
20
VO − Output Voltage − V
Figure 9
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
0
5
10
15
20
VO − Output Voltage − V
Figure 10
7
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
TYPICAL CHARACTERISTICS
DISTORTION
vs
FREQUENCY
−60
−70
2nd Harmonic
−80
DISTORTION
vs
FREQUENCY
−50
VCC = ± 5 V
RL = 1 kΩ
Gain = 2
VO(PP) = 2 V
−70
VCC = ± 15 V
RL = 150 Ω
Gain = 2
VO(PP) = 2 V
−60
Distortion − dBc
Distortion − dBc
−60
−50
VCC = ± 15 V
RL = 1 kΩ
Gain = 2
VO(PP) = 2 V
Distortion − dBc
−50
DISTORTION
vs
FREQUENCY
2nd Harmonic
−80
3rd Harmonic
−70
2nd Harmonic
−80
3rd Harmonic
−90
−90
−90
3rd Harmonic
−100
10.00
100k
100.00
1M
−100
10.00
100k
1000.00
10M
f − Frequency − Hz
Figure 11
Figure 12
Figure 13
OUTPUT AMPLITUDE
vs
FREQUENCY
3rd Harmonic
−70
2nd Harmonic
−80
−2
−4
−6
10.00
100k
1000.00
10M
Figure 14
OUTPUT AMPLITUDE
vs
FREQUENCY
2
0
RF = 0 Ω
−2
−4
−4
VCC = ± 15 V
Gain = 1
RL = 1 kΩ
VO(PP) = 63 mV
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
Figure 17
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
Figure 16
−6
VCC = ± 5 V
Gain = 1
RL = 1 kΩ
VO(PP) = 63 mV
Figure 18
• DALLAS, TEXAS 75265
RF = 2 kΩ
0
RF = 1 kΩ
−2
−4
−6
−8
10.00
100k 100.001M 1000.00
10M 10000.00
100M100000.00
1G
f - Frequency - Hz
POST OFFICE BOX 655303
RF = 1.3 kΩ
Output Amplitude − dB
−2
VCC = ± 5 V
Gain = 1
RL = 150 Ω
VO(PP) = 63 mV
RF = 51 Ω
Output Amplitude − dB
Output Amplitude − dB
RF = 0 Ω
−2
−6
10.00
100k
2
RF = 51 Ω
0
RF = 0 Ω
−4
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
RF = 130 Ω
0
OUTPUT AMPLITUDE
vs
FREQUENCY
2
8
VCC = ± 15 V
Gain = 1
RL = 150 Ω
VO(PP) = 63 mV
RF = 51 Ω
2
Figure 15
OUTPUT AMPLITUDE
vs
FREQUENCY
−8
10.00
100k
RF = 130 Ω
RF = 0 Ω
f − Frequency − Hz
−6
RF = 51 Ω
0
−90
100.00
1M
4
Output Amplitude − dB
2
1000.00
10M
OUTPUT AMPLITUDE
vs
FREQUENCY
4
VCC = ± 5 V
RL = 150 Ω
Gain = 2
VO(PP) = 2 V
−100
10.00
100k
100.00
1M
f − Frequency − Hz
Output Amplitude − dB
Distortion − dBc
−60
−100
10.00
100k
1000.00
10M
f − Frequency − Hz
DISTORTION
vs
FREQUENCY
−50
100.00
1M
−8
10.00
100k
VCC = ± 15 V
Gain = −1
RL = 150 Ω
VO(PP) = 63 mV
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
Figure 19
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
TYPICAL CHARACTERISTICS
OUTPUT AMPLITUDE
vs
FREQUENCY
OUTPUT AMPLITUDE
vs
FREQUENCY
2
2
2
RF = 1 kΩ
−2
−4
VCC = ± 5 V
Gain = −1
RL = 150 Ω
VO(PP) = 63 mV
RF = 1.3 kΩ
−2
−4
−6
−8
10.00
100k
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
VCC = ± 15 V
Gain = −1
RL = 1 kΩ
VO(PP) = 63 mV
OUTPUT AMPLITUDE
vs
FREQUENCY
8
VCC = ± 15 V
Gain = 2
RL = 150 Ω
VO(PP) = 126 mV
Output Amplitude − dB
2
RF = 1.5 kΩ
RF = 1.5 kΩ
RF = 750 Ω
4
2
0
−2
10.00
100k
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
VCC = ± 5 V
Gain = 2
RL = 150 Ω
VO(PP) = 126 mV
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
Figure 25
5-V STEP RESPONSE
VCC = ± 5 V
Gain = 2
RF = 1.2 kΩ
RL = 150 Ω
2
V O − Output Voltage − V
V O − Output Voltage − V
6
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
VCC = ± 15 V
Gain = 2
RL = 1 kΩ
VO(PP) = 126 mV
3
0.8
VCC = ± 5 V
Gain = 2
RL = 1 kΩ
VO(PP) = 126 mV
2
2-V STEP RESPONSE
RF = 1.2 kΩ
2
4
−2
10.00
100k
1.2
8
4
RF = 1.2 kΩ
Figure 24
OUTPUT AMPLITUDE
vs
FREQUENCY
RF = 1.5 kΩ
6
0
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
Figure 23
Output Amplitude − dB
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
Figure 22
6
Output Amplitude − dB
Output Amplitude − dB
4
Figure 26
VCC = ± 5 V
Gain = −1
RL = 1 kΩ
VO(PP) = 63 mV
RF = 1.2 kΩ
RF = 1.5 kΩ
RF = 750 Ω
−2
10.00
100k
−4
−8
10.00
100k
8
RF = 1.2 kΩ
0
−2
OUTPUT AMPLITUDE
vs
FREQUENCY
8
−2
10.00
100k
RF = 1.3 kΩ
Figure 21
OUTPUT AMPLITUDE
vs
FREQUENCY
6
0
−6
100.00
1000.00
10000.00
1M
10M
100M 100000.00
1G
f - Frequency - Hz
Figure 20
0
RF = 1.5 kΩ
RF = 2 kΩ
0
Output Amplitude − dB
RF = 2 kΩ
0
−8
10.00
100k
RF = 1.5 kΩ
Output Amplitude − dB
Output Amplitude − dB
RF = 1.3 kΩ
−6
OUTPUT AMPLITUDE
vs
FREQUENCY
0.4
0.0
−0.4
−0.8
1
0
−1
VCC = ± 5 V
Gain = −1
RF = 1.3 kΩ
RL = 150 Ω
−2
−1.2
−3
0
200
400
600
t - Time - ns
800
1000
Figure 27
POST OFFICE BOX 655303
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0
200
400
600
t - Time - ns
800
1000
Figure 28
9
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
TYPICAL CHARACTERISTICS
2-V STEP RESPONSE
20-V STEP RESPONSE
12
VCC = ± 15 V
Gain = 2
RF = 1.2 kΩ
RL = 150 Ω
0.8
0.6
1.5
VCC = ± 15 V
Gain = 5
RF = 1.2 kΩ
RL = 150 Ω
10
8
V O − Output Voltage − V
1.0
0.4
0.2
−0.0
−0.2
−0.4
−0.6
6
4
2
0
−2
−4
−6
−0.8
−8
−1.0
−10
−1.2
V IO − Input Offset Voltage − mV
1.2
−12
0
200
400
600
t - Time - ns
800
1000
0
200
Figure 29
400
600
t - Time - ns
800
1000
V
1.8
VO - Output Voltage -
VCC = ±15 V
1.6
1.5
I
VCC = ± 5 V
13
11
RL = 1 kΩ
9
RL = 150 Ω
7
5
3
−20
0
20
40
60
80
TA - Free-Air Temperature - °C
5
100
7
9
11
13
±VCC - Supply Voltage - V
Figure 32
1
−40
VCC = ± 5 V
RL = 1 kΩ
VCC = ± 5 V
RL = 150 Ω
−20
0
20
40
60
80
TA − Free-Air Temperature − _C
Figure 35
10
I CC − Supply Current − mA
VO − Output Voltage − V
7
3
11
9
7
5
3
5
7
9
11
13
±VCC - Supply Voltage - V
15
Figure 34
VOLTAGE & CURRENT NOISE
vs
FREQUENCY
100
3.6
VCC = ± 15 V
RL = 1 kΩ
5
TA=25°C
15
3.8
VCC = ± 15 V
RL = 150 Ω
100
13
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
15
−20
0
20
40
60
80
TA - Free-Air Temperature - °C
15
Figure 33
OUTPUT VOLTAGE
vs
FREE-AIR TEMPERATURE
9
0.3
−40
V n − Voltage Noise − nV/ Hz
I n − Current Noise − pA/ Hz
IB − Input Bias Current − µ A
1.9
11
VCC = ± 5 V
0.5
COMMON-MODE INPUT VOLTAGE
vs
SUPPLY VOLTAGE
15
13
0.7
Figure 31
TA=25°C
1.3
−40
0.9
OUTPUT VOLTAGE
vs
SUPPLY VOLTAGE
2.0
1.4
VCC = ± 15 V
1.1
Figure 30
INPUT BIAS CURRENT
vs
FREE-AIR TEMPERATURE
1.7
1.3
V ICR − Common-Mode Input Voltage − ± V
V O − Output Voltage − V
INPUT OFFSET VOLTAGE
vs
FREE-AIR TEMPERATURE
TA=85°C
3.4
VCC = ± 15 V and ± 5 V
TA = 25°C
VN
10
3.2
TA=25°C
3.0
2.8
TA=−40°C
2.6
IN
1
2.4
100
0.1
2.2
5
7
9
11
13
± VCC - Supply Voltage - V
15
Figure 36
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10
100
1k
10k
f - Frequency - Hz
Figure 37
100k
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
APPLICATION INFORMATION
theory of operation
The THS408x is a high-speed, operational amplifier configured in a voltage feedback architecture. It is built
using a 30-V, dielectrically isolated, complementary bipolar process with NPN and PNP transistors possessing
fTs of several GHz. This results in an exceptionally high performance amplifier that has a wide bandwidth, high
slew rate, fast settling time, and low distortion. A simplified schematic is shown in Figure 38.
(7) VCC +
(6) OUT
IN − (2)
IN + (3)
(4) VCC −
Figure 38. THS4081 Simplified Schematic
noise calculations and noise figure
Noise can cause errors on very small signals. This is especially true when amplifying small signals, where
signal-to-noise ratio (SNR) is very important. The noise model for the THS408x is shown in Figure 39. This
model includes all of the noise sources as follows:
•
•
•
•
en = Amplifier internal voltage noise (nV/√Hz)
IN+ = Noninverting current noise (pA/√Hz)
IN− = Inverting current noise (pA/√Hz)
eRx = Thermal voltage noise associated with each resistor (eRx = 4 kTRx )
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APPLICATION INFORMATION
noise calculations and noise figure (continued)
eRs
RS
en
Noiseless
+
_
eni
IN+
eno
eRf
RF
eRg
IN−
RG
Figure 39. Noise Model
The total equivalent input noise density (eni) is calculated by using the following equation:
e
ni
+
Ǹǒ
ǒ
2
e nǓ ) IN )
R
Ǔ
S
2
ǒ
) IN–
ǒRF ø RGǓǓ
2
ǒ
Ǔ
) 4 kTR s ) 4 kT R ø R
F
G
Where:
k = Boltzmann’s constant = 1.380658 × 10−23
T = Temperature in degrees Kelvin (273 +°C)
RF || RG = Parallel resistance of RF and RG
To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (eni) by the
overall amplifier gain (AV).
e no + e
ǒ
Ǔ
R
A + e ni 1 ) F (noninverting case)
ni V
RG
As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the
closed-loop gain is increased (by reducing RG), the input noise is reduced considerably because of the parallel
resistance term. This leads to the general conclusion that the most dominant noise sources are the source
resistor (RS) and the internal amplifier noise voltage (en). Because noise is summed in a root-mean-squares
method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly
simplify the formula and make noise calculations much easier to calculate.
For more information on noise analysis, please refer to the Noise Analysis section in Operational Amplifier
Circuits Applications Report (literature number SLVA043).
12
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APPLICATION INFORMATION
noise calculations and noise figure (continued)
This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise
figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be
defined and is typically 50 Ω in RF applications.
NF +
ȱ e 2ȳ
10logȧ ni ȧ
2
ǒ
Ǔ
e
Ȳ Rs ȴ
Because the dominant noise components are generally the source resistance and the internal amplifier noise
voltage, we can approximate noise figure as:
NF +
ȱ ȡǒ Ǔ2 ǒ
ȧ en ) IN )
ȧ
Ȣ
ȧ
10logȧ1 )
4 kTR
ȧ
S
ȧ
Ȳ
Ǔ ȣȳ
S ȧ
2
R
Ȥȧ
ȧ
ȧ
ȧ
ȧ
ȴ
Figure 40 shows the noise figure graph for the THS408x.
NOISE FIGURE
vs
SOURCE RESISTANCE
40
35
f = 10 kHz
TA = 25°C
Noise Figure (dB)
30
25
20
15
10
5
0
10
100
1k
10k
100k
Source Resistance − RS (Ω)
Figure 40. Noise Figure vs Source Resistance
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SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
APPLICATION INFORMATION
driving a capacitive load
Driving capacitive loads with high performance amplifiers is not a problem as long as certain precautions are
taken. The first is to realize that the THS408x has been internally compensated to maximize its bandwidth and
slew rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the
output will decrease the device’s phase margin leading to high frequency ringing or oscillations. Therefore, for
capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of
the amplifier, as shown in Figure 41. A minimum value of 20 Ω should work well for most applications. For
example, in 75-Ω transmission systems, setting the series resistor value to 75 Ω both isolates any capacitance
loading and provides the proper line impedance matching at the source end.
1.3 kΩ
1.3 kΩ
_
Input
20 Ω
Output
THS408x
+
CLOAD
Figure 41. Driving a Capacitive Load
offset voltage
The output offset voltage, (VOO) is the sum of the input offset voltage (VIO) and both input bias currents (IIB) times
the corresponding gains. The following schematic and formula can be used to calculate the output offset
voltage:
RF
IIB−
RG
+
−
VI
IIB+
V
OO
+V
IO
ǒ ǒ ǓǓ
1)
R
R
F
G
VO
+
RS
"I
IB)
R
S
ǒ ǒ ǓǓ
1)
R
R
F
G
"I
IB–
Figure 42. Output Offset Voltage Model
14
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R
F
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
APPLICATION INFORMATION
general configurations
When receiving low-level signals, limiting the bandwidth of the incoming signals is often required. The simplest
way to accomplish this is to place an RC filter at the noninverting terminal of the amplifier (see Figure 43).
RG
RF
−
VI
VO
+
R1
C1
f
V
O +
V
I
ǒ
1)
R
R
F
G
Ǔǒ
–3dB
+
1
2pR1C1
Ǔ
1
1 ) sR1C1
Figure 43. Single-Pole Low-Pass Filter
circuit layout considerations
To achieve the levels of high frequency performance of the THS408x, follow proper printed-circuit board high
frequency design techniques. A general set of guidelines is given below. In addition, a THS408x evaluation
board is available to use as a guide for layout or for evaluating the device performance.
D Ground planes − It is highly recommended that a ground plane be used on the board to provide all
components with a low inductive ground connection. However, in the areas of the amplifier inputs and
output, the ground plane can be removed to minimize the stray capacitance.
D Proper power supply decoupling − Use a 6.8-µF tantalum capacitor in parallel with a 0.1-µF ceramic
capacitor on each supply terminal. It may be possible to share the tantalum among several amplifiers
depending on the application, but a 0.1-µF ceramic capacitor should always be used on the supply terminal
of every amplifier. In addition, the 0.1-µF capacitor should be placed as close as possible to the supply
terminal. As this distance increases, the inductance in the connecting trace makes the capacitor less
effective. The designer should strive for distances of less than 0.1 inches between the device power
terminals and the ceramic capacitors.
D Sockets − Sockets are not recommended for high-speed operational amplifiers. The additional lead
inductance in the socket pins will often lead to stability problems. Surface-mount packages soldered directly
to the printed-circuit board is the best implementation.
D Short trace runs/compact part placements − Optimum high frequency performance is achieved when stray
series inductance has been minimized. To realize this, the circuit layout should be made as compact as
possible, thereby minimizing the length of all trace runs. Particular attention should be paid to the inverting
input of the amplifier. Its length should be kept as short as possible. This will help to minimize stray
capacitance at the input of the amplifier.
D Surface-mount passive components − Using surface-mount passive components is recommended for high
frequency amplifier circuits for several reasons. First, because of the extremely low lead inductance of
surface-mount components, the problem with stray series inductance is greatly reduced. Second, the small
size of surface-mount components naturally leads to a more compact layout, thereby minimizing both stray
inductance and capacitance. If leaded components are used, it is recommended that the lead lengths be
kept as short as possible.
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APPLICATION INFORMATION
general PowerPAD design considerations
The THS408x is available packaged in a thermally-enhanced DGN package, which is a member of the
PowerPAD family of packages. This package is constructed using a downset leadframe upon which the die
is mounted [see Figure 44(a) and Figure 44(b)]. This arrangement results in the lead frame being exposed as
a thermal pad on the underside of the package [see Figure 44(c)]. Because this thermal pad has direct thermal
contact with the die, excellent thermal performance can be achieved by providing a good thermal path away
from the thermal pad.
The PowerPAD package allows for both assembly and thermal management in one manufacturing operation.
During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be
soldered to a copper area underneath the package. Through the use of thermal paths within this copper area,
heat can be conducted away from the package into either a ground plane or other heat dissipating device.
The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of the
surface mount with the, heretofore, awkward mechanical methods of heatsinking.
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
Bottom View (c)
NOTE A: The thermal pad is electrically isolated from all terminals in the package.
Figure 44. Views of Thermally Enhanced DGN Package
16
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APPLICATION INFORMATION
general PowerPAD design considerations (continued)
Although there are many ways to properly heatsink this device, the following steps illustrate the recommended
approach.
Thermal pad area (68 mils x 70 mils) with 5 vias
(Via diameter = 13 mils)
Figure 45. PowerPAD PCB Etch and Via Pattern
1. Prepare the PCB with a top side etch pattern as shown in Figure 45. There should be etch for the leads as
well as etch for the thermal pad.
2. Place five holes in the area of the thermal pad. These holes should be 13 mils in diameter. Keep them small
so that solder wicking through the holes is not a problem during reflow.
3. Additional vias may be placed anywhere along the thermal plane outside of the thermal pad area. This helps
dissipate the heat generated by the THS408xDGN IC. These additional vias may be larger than the 13-mil
diameter vias directly under the thermal pad. They can be larger because they are not in the thermal pad
area to be soldered, so wicking is not a problem.
4. Connect all holes to the internal ground plane.
5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection
methodology. Web connections have a high thermal resistance connection that is useful for slowing the heat
transfer during soldering operations. This makes the soldering of vias that have plane connections easier.
In this application, however, low thermal resistance is desired for the most efficient heat transfer. Therefore,
the holes under the THS408xDGN package should make their connection to the internal ground plane with
a complete connection around the entire circumference of the plated-through hole.
6. The top-side solder mask should leave the terminals of the package and the thermal pad area with its five
holes exposed. The bottom-side solder mask should cover the five holes of the thermal pad area. This
prevents solder from being pulled away from the thermal pad area during the reflow process.
7. Apply solder paste to the exposed thermal pad area and all of the IC terminals.
8. With these preparatory steps in place, the THS408xDGN IC is simply placed in position and run through
the solder reflow operation as any standard surface-mount component. This results in a part that is properly
installed.
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SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
APPLICATION INFORMATION
general PowerPAD design considerations (continued)
The actual thermal performance achieved with the THS408xDGN in its PowerPAD package depends on the
application. In the example above, if the size of the internal ground plane is approximately 3 inches × 3 inches,
then the expected thermal coefficient, θJA, is about 58.4_C/W. For comparison, the non-PowerPAD version
of the THS408x IC (SOIC) is shown. For a given θJA, the maximum power dissipation is shown in Figure 46 and
is calculated by the following formula:
P
D
+
Where:
ǒ
T
Ǔ
–T
MAX A
q
JA
PD = Maximum power dissipation of THS408x IC (watts)
TMAX = Absolute maximum junction temperature (150°C)
TA
= Free-ambient air temperature (°C)
θJA = θJC + θCA
θJC = Thermal coefficient from junction to case
θCA = Thermal coefficient from case to ambient air (°C/W)
MAXIMUM POWER DISSIPATION
vs
FREE-AIR TEMPERATURE
Maximum Power Dissipation − W
3.5
DGN Package
θJA = 58.4°C/W
2 oz. Trace And Copper Pad
With Solder
3
DGN Package
θJA = 158°C/W
2 oz. Trace And
Copper Pad
Without Solder
2.5
SOIC Package
High-K Test PCB
θJA = 98°C/W
2
TJ = 150°C
1.5
1
0.5
SOIC Package
Low-K Test PCB
θJA = 167°C/W
0
−40
−20
0
20
40
60
80
100
TA − Free-Air Temperature − °C
NOTE A: Results are with no air flow and PCB size = 3”× 3”
Figure 46. Maximum Power Dissipation vs Free-Air Temperature
More complete details of the PowerPAD installation process and thermal management techniques can be found
in the Texas Instruments Technical Brief, PowerPAD Thermally Enhanced Package. This document can be
found at the TI web site (www.ti.com) by searching on the key word PowerPAD. The document can also be
ordered through your local TI sales office. Refer to literature number SLMA002 when ordering.
18
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SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
APPLICATION INFORMATION
general PowerPAD design considerations (continued)
The next consideration is the package constraints. The two sources of heat within an amplifier are quiescent
power and output power. The designer should never forget about the quiescent heat generated within the
device, especially multiamplifier devices. Because these devices have linear output stages (Class A-B), most
of the heat dissipation is at low output voltages with high output currents. Figure 47 to Figure 50 show this effect,
along with the quiescent heat, with an ambient air temperature of 50°C. Obviously, as the ambient temperature
increases, the limit lines shown will drop accordingly. The area under each respective limit line is considered
the safe operating area. Any condition above this line will exceed the amplifier’s limits and failure may result.
When using VCC = ±5 V, there is generally not a heat problem, even with SOIC packages. But, when using
VCC = ±15 V, the SOIC package is severely limited in the amount of heat it can dissipate. The other key factor
when looking at these graphs is how the devices are mounted on the PCB. The PowerPAD devices are
extremely useful for heat dissipation. But, the device should always be soldered to a copper plane to fully use
the heat dissipation properties of the PowerPAD. The SOIC package, on the other hand, is highly dependent
on how it is mounted on the PCB. As more trace and copper area is placed around the device, θJA decreases
and the heat dissipation capability increases. The currents and voltages shown in these graphs are for the total
package. For the dual amplifier package (THS4082), the sum of the RMS output currents and voltages should
be used to choose the proper package. The graphs shown assume that both amplifier’s outputs are identical.
THS4081
MAXIMUM RMS OUTPUT CURRENT
vs
RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS
VCC = ± 5 V
TJ = 150°C
TA = 50°C
180
1000
Maximum Output
Current Limit Line
| IO | − Maximum RMS Output Current − mA
| IO | − Maximum RMS Output Current − mA
200
160
140
Package With
θJA < = 127°C/W
120
100
SO-8 Package
θJA = 167°C/W
Low-K Test PCB
80
60
40
Safe Operating
Area
20
0
THS4081
MAXIMUM RMS OUTPUT CURRENT
vs
RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS
TJ = 150°C
TA = 50°C
VCC = ± 15 V
Maximum Output
Current Limit Line
DGN Package
θJA = 58.4°C/W
100
SO-8 Package
θJA = 167°C/W
Low-K Test PCB
SO-8 Package
θJA = 98°C/W
High-K Test PCB
Safe Operating
Area
10
0
1
2
3
4
5
0
| VO | − RMS Output Voltage − V
3
6
9
12
15
| VO | − RMS Output Voltage − V
Figure 48
Figure 47
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19
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
APPLICATION INFORMATION
general PowerPAD design considerations (continued)
THS4082
MAXIMUM RMS OUTPUT CURRENT
vs
RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS
180
1000
Maximum Output
Current Limit Line
Package With
θJA ≤ 64°C/W
| IO | − Maximum RMS Output Current − mA
| IO | − Maximum RMS Output Current − mA
200
160
140
120
100
SO-8 Package
θJA = 167°C/W
Low-K Test PCB
80
60
Safe Operating Area
40
VCC = ± 5 V
TJ = 150°C
TA = 50°C
Both Channels
SO-8 Package
θJA = 98°C/W
High-K Test PCB
20
0
0
1
2
3
4
THS4082
MAXIMUM RMS OUTPUT CURRENT
vs
RMS OUTPUT VOLTAGE DUE TO THERMAL LIMITS
VCC = ± 15 V
TJ = 150°C
TA = 50°C
Both Channels
100
SO-8 Package
θJA = 98°C/W
High-K Test PCB
10
DGN Package
θJA = 58.4°C/W
Safe Operating Area
5
1
0
3
SO-8 Package
θJA = 167°C/W
Low-K Test PCB
6
Figure 50
Figure 49
POST OFFICE BOX 655303
9
12
| VO | − RMS Output Voltage − V
| VO | − RMS Output Voltage − V
20
Maximum Output
Current Limit Line
• DALLAS, TEXAS 75265
15
SLOS274D − DECEMBER 1999 − REVISED JUNE 2001
APPLICATION INFORMATION
evaluation board
An evaluation board is available for the THS4081 (literature number SLOP242) and THS4082 (literature number
SLOP239). This board has been configured for very low parasitic capacitance in order to realize the full
performance of the amplifier. A schematic of the evaluation board is shown in Figure 51. The circuitry has been
designed so that the amplifier may be used in either an inverting or noninverting configuration. For more
information, please refer to the THS4081 EVM User’s Guide or the THS4082 EVM User’s Guide. To order the
evaluation board, contact your local TI sales office or distributor.
VCC+
+
C3
0.1 µF
C2
6.8 µF
R4
1.3 kΩ
IN +
R5
49.9 Ω
+
R3
49.9 Ω
OUT
THS4081
_
R2
1.3 kΩ
+
C4
0.1 µF
C1
6.8 µF
IN −
R1
49.9 Ω
VCC −
Figure 51. THS4081 Evaluation Board
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21
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
THS4081CD
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4081C
Samples
THS4081CDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAUAG
Level-1-260C-UNLIM
0 to 70
AEO
Samples
THS4081CDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAUAG
Level-1-260C-UNLIM
0 to 70
AEO
Samples
THS4081CDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4081C
Samples
THS4081ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4081I
Samples
THS4081IDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAUAG
Level-1-260C-UNLIM
-40 to 85
AEQ
Samples
THS4081IDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAUAG
Level-1-260C-UNLIM
-40 to 85
AEQ
Samples
THS4082CD
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4082C
Samples
THS4082CDG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4082C
Samples
THS4082CDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
AER
Samples
THS4082CDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
4082C
Samples
THS4082ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4082I
Samples
THS4082IDG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4082I
Samples
THS4082IDGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
Call TI | NIPDAU
Level-1-260C-UNLIM
-40 to 85
AEP
Samples
THS4082IDGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
Call TI | NIPDAU
Level-1-260C-UNLIM
-40 to 85
AEP
Samples
THS4082IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
4082I
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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