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THS4551
SBOS778C – APRIL 2016 – REVISED JULY 2017
THS4551
Low-Noise, Precision, 150-MHz, Fully Differential Amplifier
1 Features
3 Description
•
•
•
•
The THS4551 fully differential amplifier offers an easy
interface from single-ended sources to the differential
output required by high-precision analog-to-digital
converters (ADCs). Designed for exceptional dc
accuracy, low noise, and robust capacitive load
driving, this device is well suited for data acquisition
systems where high precision is required along with
the best signal-to-noise ratio (SNR) and spurious-free
dynamic range (SFDR) through the amplifier and
ADC combination.
1
•
•
•
•
•
•
•
•
•
•
Bandwidth: 150 MHz (G = 1 V/V)
Differential Output Slew Rate: 220 V/µs
Gain Bandwidth Product: 135 MHz
Negative Rail Input (NRI),
Rail-to-Rail Output (RRO)
Wide Output Common-Mode Control Range
Single-Supply Operating Range: 2.7 V to 5.4 V
Trimmed-Supply Current: 1.37 mA at 5 V
25°C Input Offset: ±175 µV (max)
Input Offset Voltage Drift: ±1.8 µV/°C (max)
Differential Input Voltage Noise: 3.3 nV/√Hz
HD2: –128 dBc at 2 VPP, 100 kHz
HD3: –139 dBc at 2 VPP, 100 kHz
< 50-ns Settling Time: 4-V Step to 0.01%
18-Bit Settling Time: 4-V Step, < 500 ns
The THS4551 features the negative rail input required
when interfacing a dc-coupled, ground-centered,
source signal to a single-supply differential input
ADC. Very low dc error and drift terms support the
emerging 16- to 20-bit successive-approximation
register (SAR) input requirements. A wide-range
output common-mode control supports the ADC
running from 1.8-V to 5-V supplies with ADC
common-mode input requirements from 0.7 V to
greater than 3.0 V.
2 Applications
•
•
•
•
•
The THS4551 device is characterized for operation
over the wide temperature range of –40°C to +125°C,
and is available in 8-pin VSSOP, 16-pin VQFN, and
10-pin WQFN packages.
24-Bit, Delta-Sigma (ΔΣ) ADC Drivers
16- to 20-Bit, Differential, High-Speed SAR
Drivers
Differential Active Filters
Differential Transimpedance Amplifiers
Pin-Compatible Upgrade to the THS4521
(VSSOP-8 only)
Low-Power ADCs Supported by the THS4551
PART
NUMBER
ADC TYPE
RESOLUTION, SPEED
ADS127L01
Delta sigma
24 bits, 0.512 MSPS
ADS8881
SAR
18 bits, 1 MSPS
ADS9110
SAR
18 bits, 2 MSPS
ADC3241
Pipeline
14 bits, 25 MSPS
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
Simplified Schematic: Gain of 1 V/V, Single-Ended Input to Differential Output, 500-kHz,
Multiple Feedback Filter Interface to the ADS127L01
1.2 k
270 pF
1.2 k
1 nF
VOCM
470 pF
330
+
±
3V
5
+
±
1.2 k
THS4551
330
10
AINN
22 nF
ADS127L01
AINP
5
10
270 pF
1.2 k
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
THS4551
SBOS778C – APRIL 2016 – REVISED JULY 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
6.9
7
1
1
1
2
4
5
Absolute Maximum Ratings ...................................... 5
ESD Ratings.............................................................. 5
Recommended Operating Conditions....................... 6
Thermal Information .................................................. 6
Electrical Characteristics: (VS+) – (VS–) = 5 V........... 6
Electrical Characteristics: (VS+) – (VS–) = 3 V........... 9
Typical Characteristics: (VS+) – (VS–) = 5 V............ 13
Typical Characteristics: (VS+) – (VS–) = 3 V............ 16
Typical Characteristics: 3-V to 5-V Supply Range.. 19
Parameter Measurement Information ................ 23
7.1 Example Characterization Circuits .......................... 23
7.2 Output Interface Circuit for DC-Coupled Differential
Testing ..................................................................... 25
7.3 Output Common-Mode Measurements................... 25
7.4 Differential Amplifier Noise Measurements............. 26
7.5 Balanced Split-Supply Versus Single-Supply
Characterization ....................................................... 26
7.6 Simulated Characterization Curves ........................ 26
7.7 Terminology and Application Assumptions ............. 27
8
Detailed Description ............................................ 28
8.1
8.2
8.3
8.4
9
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
28
28
29
38
Application and Implementation ........................ 42
9.1 Application Information............................................ 42
9.2 Typical Applications ................................................ 49
10 Power Supply Recommendations ..................... 56
10.1 Thermal Analysis................................................... 57
11 Layout................................................................... 57
11.1 Layout Guidelines ................................................. 57
11.2 Layout Example .................................................... 58
11.3 EVM Board............................................................ 59
12 Device and Documentation Support ................. 60
12.1
12.2
12.3
12.4
12.5
12.6
12.7
Device Support......................................................
Documentation Support ........................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
60
62
63
63
63
63
63
13 Mechanical, Packaging, and Orderable
Information ........................................................... 63
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (August 2016) to Revision B
Page
•
Changed IQ value in THS4551 row of Device Family Comparison ........................................................................................ 4
•
Added second row and footnote 2 to Voltage parameter of Absolute Maximum Ratings table............................................. 5
•
Added package differences and footnote 3 to ESD Ratings table ......................................................................................... 5
•
Changed footnotes 1 and 2 in 5-V Electrical Characteristics table ........................................................................................ 6
•
Added test conditions to AOL parameter in 5-V Electrical Characteristics table ..................................................................... 7
•
Changed Input offset voltage drift parameter ........................................................................................................................ 7
•
Changed IIB parameter minimum and maximum specifications in last three rows ................................................................ 7
•
Changed Input bias current drift parameter test conditions and specifications ..................................................................... 7
•
Added Input offset current drift parameter test conditions, minimum and maximum specifications, and test level
value to second row................................................................................................................................................................ 7
•
Changed test conditions of Common-mode input, low and Common-mode input, high parameters .................................... 7
•
Changed test conditions of Continuous output current and Linear output current parameters ............................................. 8
•
Changed test conditions of Enable voltage threshold and Disable voltage threshold parameters ........................................ 8
•
Changed specifications of Power-down quiescent current parameter .................................................................................. 8
•
Changed Common-mode loop supply headroom to negative supply parameter test conditions ........................................... 9
•
Changed test conditions and maximum specifications of Common-mode loop supply headroom to positive supply
parameter ............................................................................................................................................................................... 9
•
Added test conditions to DC Performance, AOL parameter .................................................................................................. 10
•
Changed Input offset voltage drift parameter test conditions in first row, added second row.............................................. 10
•
Changed minimum and maximum specifications in last three rows of IIB parameter........................................................... 10
2
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SBOS778C – APRIL 2016 – REVISED JULY 2017
Revision History (continued)
•
Changed Input bias current drift parameter test conditions.................................................................................................. 10
•
Added second row to Input offset current drift parameter ................................................................................................... 10
•
Changed test conditions of Common-mode input, low and Common-mode input, high parameters................................... 10
•
Changed test conditions of Continuous output current and Linear output current parameters ........................................... 11
•
Changed test conditions of Enable voltage threshold and Disable voltage threshold parameters ...................................... 11
•
Changed IQ(PD) parameter specifications .............................................................................................................................. 11
•
Changed Common-mode loop supply headroom to negative supply parameter test conditions ......................................... 12
•
Changed Common-mode loop supply headroom to positive supply parameter test conditions and maximum
specifications ....................................................................................................................................................................... 12
•
Changed conditions of Figure 49 to Figure 54 .................................................................................................................... 21
•
Changed Single-Ended Source to a Differential Gain of a 1-V/V Test Circuit figure ........................................................... 23
•
Changed main Device Functional Modes section: changed value of PD pin voltage ......................................................... 38
•
Changed the minimum value for single-supply operation in the Operating the Power Shutdown Feature section ............. 45
•
Added SBOS476, SBOC466, SBOC463, SBOC467, SBOS460, SBOC477, SBOC472, SLOC341, SBOC469,
SBOC462, SBOC461, SBOC465, SBOC464, SBOC475, SBOC474, SBOC471, SBOC459, SBOC470, SBOC468,
and SBOC473 to Related Documentation section .............................................................................................................. 62
Changes from Revision B (November 2016) to Revision C
•
Page
Changed 47k Ohms , 1.3 pF to 150k Ohms , 7 pF in the Electrical Characteristics: (VS+) – (VS–) = 3 V table .................... 1
Changes from Original (April 2016) to Revision A
•
Page
Released to production .......................................................................................................................................................... 1
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SBOS778C – APRIL 2016 – REVISED JULY 2017
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Table 1. Device Comparison Table
DEVICE
BW, G = 1 (MHz)
IQ, 5 V
(mA)
INPUT NOISE
(nV/√Hz)
THD (dBc) 2 VPP
AT 10 kHz
RAIL-TO-RAIL
DUAL VERSIONS
THS4551
150
1.37
3.3
–138
Negative in, out
THS4552
THS4521
145
1.14
5.6
–120
Negative in, out
THS4522
THS4531A
36
0.25
10
–118
Negative in, out
THS4532
THS4520
620
14.2
2.0
–105
Out
—
THS4541
620
10.1
2.2
–140
Negative in, out
—
5 Pin Configuration and Functions
VS–
VS–
VS–
VS–
RGT Package
16-Pin VQFN With Exposed Thermal Pad
Top View
16
15
14
13
RUN Package
10-Pin WQFN
Top View
VS+
10
OUT–
1
9
OUT+
FB–
1
12
PD
NC
2
8
NC
IN+
2
11
OUT–
PD
3
7
VOCM
IN–
3
10
OUT+
IN+
4
6
IN–
FB+
5
4
9
VOCM
5
6
7
8
VS+
VS+
VS+
VS+
VS–
DGK Package
8-Pin VSSOP
Top View
4
IN-
1
8
IN+
VOCM
2
7
PD
VS+
3
6
VS-
OUT+
4
5
OUT-
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SBOS778C – APRIL 2016 – REVISED JULY 2017
Pin Functions
PIN
NO.
NAME
I/O
DESCRIPTION
RGT (1)
RUN
DGK
FB–
1
—
—
O
Inverting (negative) output feedback
FB+
4
—
—
O
Noninverting (positive) output feedback
IN–
3
6
1
I
Inverting (negative) amplifier input
IN+
2
4
8
I
Noninverting (positive) amplifier input
NC
—
2, 8
—
—
No internal connection
OUT–
11
1
5
O
Inverting (negative) amplifier output
OUT+
10
9
4
O
Noninverting (positive) amplifier output
PD
12
3
7
I
Power down. PD = logic low = power off mode; PD = logic high = normal
operation.
VOCM
9
7
2
I
Common-mode voltage input
VS–
13-16
5
6
I
Negative power-supply input
VS+
5, 6, 7, 8
10
3
I
Positive power-supply input
(1)
Solder the exposed thermal pad (RGT package) to a heat-spreading power or ground plane. This pad is electrically isolated from the
die, but must be connected to a power or ground plane and not floated.
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
Voltage
V
Supply turn-on/off maximum dV/dT (2)
±1
V/µs
(VS–) – 0.5
(VS+) + 0.5
Differential input voltage
±1
Continuous input current
±10
Continuous output current (3)
±20
Continuous power dissipation
(1)
(2)
(3)
V
mA
See the Thermal Information and
Thermal Analysis sections
Maximum junction
Temperature
UNIT
5.5
Input/output voltage range
Current
MAX
Supply voltage, (VS+) – (VS–)
150
Operating free-air, TA
–40
125
Storage, Tstg
–65
150
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
Staying below this ± supply turn-on edge rate ensures that the edge-triggered ESD absorption device across the supply pins remains
off.
Long-term continuous current for electro-migration limits.
6.2 ESD Ratings
VALUE
UNIT
A. THS4551 in DGK, RUN Pacakges
V(ESD)
Electrostatic discharge
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2500
Charged device model (CDM), per JEDEC specification JESD22-C101 (2)
±1250
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1) (3)
±1000
Charged device model (CDM), per JEDEC specification JESD22-C101 (2)
±1250
V
B. THS4551 in RGT Package
V(ESD)
(1)
(2)
(3)
Electrostatic discharge
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
ESD limit of ±1000 V for any pin to thermal pad. Pin-to-pin HBM ESD specifications are rated at ±2500 V.
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6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
NOM
MAX
UNIT
VS+
Single-supply positive voltage
2.7
5
5.4
V
TA
Ambient temperature
–40
25
125
°C
6.4 Thermal Information
THS4551
THERMAL METRIC
(1)
RGT (2)
(VQFN)
RUN
(WQFN)
DGK
(VSSOP)
UNIT
16 PINS
10 PINS
8 PINS
RθJA
Junction-to-ambient thermal resistance
54
142
185
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
72
78
76
°C/W
RθJB
Junction-to-board thermal resistance
28
97
106
°C/W
ψJT
Junction-to-top characterization parameter
3.2
9.7
13
°C/W
ψJB
Junction-to-board characterization parameter
28
97
105
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
12
N/A
N/A
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
Thermal impedance for RGT reported with backside thermal pad soldered to heat spreading plane.
6.5 Electrical Characteristics: (VS+) – (VS–) = 5 V
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL (1)
AC PERFORMANCE
VOUT = 20 mVPP, G = 1, peaking (< 1.0 dB)
SSBW
Small-signal bandwidth
150
C
VOUT = 20 mVPP, G = 2
75
MHz
VOUT = 20 mVPP, G = 10
15
C
135
MHz
C
C
GBP
Gain-bandwidth product
VOUT = 20 mVPP, G = 100
LSBW
Large-signal bandwidth
VOUT = 2 VPP, G = 1
37
MHz
C
Bandwidth for 0.1-dB flatness
VOUT = 2 VPP, G = 1
15
MHz
C
220
V/µs
C
ns
C
(2)
SR
Slew rate
tR, tF
Rise and fall time
tSETTLE
Settling time
Overshoot and undershoot
HD2
HD3
(1)
(2)
6
Second-order harmonic distortion
Third-order harmonic distortion
VOUT = 4 VPP, full-power bandwidth (FPBW),
RL = 1 kΩ
VOUT = 0.5-V step, G = 1, input tR = 2 ns
6
To 0.1%, VOUT = 0.5-V step, input tR = 2 ns, G = 1
30
To 0.01%,VOUT = 0.5-V step, input tR = 2 ns, G = 1
50
VOUT = 0.5-V step G = 1, input tR = 2 ns
ns
8%
f = 100 kHz, VOUT = 2 VPP, G = 1, RL = 1 kΩ
–128
f = 100 kHz, VOUT = 8 VPP, G = 1, RL = 1 kΩ
–124
f = 100 kHz, VOUT = 2 VPP, G = 1, RL = 1 kΩ
–139
f = 100 kHz, VOUT = 8 VPP, G = 1, RL = 1 kΩ
–131
C
C
C
dBc
dBc
C
C
C
C
Input voltage noise
f > 500 Hz, 1/f < 150 Hz
3.3
nV/√Hz
C
Input current noise
f > 20 kHz, 1/f 90-dB CMRR at input
range limits
TA = 25°C
(VS–) – 0.2 (VS–) – 0.1
TA = –40°C to +125°C
(VS–) – 0.1
Common-mode input, high
> 90-dB CMRR at input
range limits
TA = 25°C
(VS+) – 1.2 (VS+) – 1.1
TA = –40°C to +125°C
(VS+) – 1.3 (VS+) – 1.2
Common-mode rejection ratio
Input pins at [(VS+) – (VS–)] / 2
Input impedance differential mode
Input pins at [(VS+) – (VS–)] / 2
Common-mode input, low
CMRR
(3)
(4)
(5)
93
110
100 || 1.2
VS–
V
V
A
B
A
B
dB
A
kΩ || pF
C
Currents out of pin are treated as a positive polarity (with the exception of the power-supply pins).
Trace mismatch measurement is dominated by the variation in contactor resistance. Internal mismatch is less than 0.1 Ω.
Input offset voltage drift, input bias current drift, and input offset current drift are the mean ±1-sigma values calculated by taking
measurements at the maximum-range ambient temperature end points, computing the difference, and dividing by the temperature
range. Maximum drift specifications are set by mean ±4 σ on the device distributions tested over a –40°̊ C to +125°̊ C ambient
temperature range. Drift is not specified by final ATE testing or QA sample test.
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Electrical Characteristics: (VS+) – (VS–) = 5 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
(VS–) +
0.2
(VS–) +
0.23
(VS–) + 0.2
(VS–) +
0.22
UNIT
TEST
LEVEL (1)
OUTPUT
TA = 25°C
Output voltage, low
TA = –40°C to +125°C
(VS+) –
(VS+) – 0.2
0.23
TA = 25°C
Output voltage, high
TA = –40°C to +125°C
Continuous output current
Linear output current
±60
TA = –40°C to +125°C, ±2.1 V, RL= 40 Ω,
VOCM offset < ±20 mV
±50
TA = 25°C, ±2.1 V, RL= 50 Ω, AOL > 80 dB
±40
B
A
V
(VS+) –
(VS+) – 0.2
0.22
TA = 25°C, ±2.5 V, RL= 40 Ω,
VOCM offset < ±20 mV
A
V
B
±65
A
mA
B
±45
A
mA
TA = –40°C to +125°C, ±1.6 V, RL= 50 Ω,
AOL > 80 dB
±30
2.7
5
5.4
TA ≈ 25°C (6), VS+ = 5 V
1.28
1.37
1.44
TA = –40°C to +125°C, VS+ = 5 V
0.97
B
POWER SUPPLY
Specified operating voltage
IQ
Quiescent operating current
dIQ/dT
Quiescent current temperature
coefficient
±PSRR Power-supply rejection ratio
1.92
VS+ = 5 V
2.4
3.9
Either supply pin to differential VOUT
93
110
5.4
V
mA
B
A
B
µA/°C
B
dB
A
POWER-DOWN
Enable voltage threshold
Specified on above (VS–) + 1.15 V
Disable voltage threshold
Specified off below (VS–) + 0.55 V
Disable pin bias current
PD = VS– → VS+
Power-down quiescent current
(VS–) + 1.15
V
A
(VS–) + 0.55
V
A
B
–100
±10
100
nA
–2
1
5
µA
A
tON
Turn-on time delay
Time from PD = low to VOUT = 90% of final value
700
ns
C
tOFF
Turn-off time delay
Time from PD = low to VOUT = 10% of final value
100
ns
C
40
MHz
C
8
MHz
C
OUTPUT COMMON-MODE VOLTAGE (VOCM) CONTROL (7) (See Figure 65)
SSBW
Small-signal bandwidth
VOCM = 100 mVPP at the control pin
LSBW
Large-signal bandwidth
VOCM = 1 VPP at the control pin
SR
Slew rate (2)
From 1-VPP LSBW
18
V/µs
C
Output common-mode noise
(≥ 2 kHz)
VOCM pin driven from low impedance
15
nV/√Hz
C
Gain
VOCM control pin input to output average voltage
(see Figure 65)
Input bias current
DC output balance (differential
mode to common-mode output)
SSBW
Output balance
LSBW
0.997
0.999
1.001
V/V
A
–100
±10
100
nA
A
dB
C
VOUT = ±1 V
85
VOUT = 100 mVPP (output balance drops –3 dB from
the 85-dB dc level)
300
VOUT = 2 VPP (output balance drops –3 dB from the
85-dB dc level)
300
Input impedance
(VOCM pin input)
Default voltage offset from
[(VS+) – (VS–)] / 2
(6)
(7)
8
C
kHz
C
150 || 7
VOCM pin open
VOCM pin open, TA = –40°C to +125°C
kΩ || pF
C
–12
±2
12
mV
A
15
35
55
µA/°C
B
TA = 25°C and ICC ≈ 1.37 mA. The test limit is expanded for the ATE ambient range of 22°C to 32°C with a 4-µA/°C ICC temperature
coefficient considered; see Figure 95.
Specifications are from the input VOCM pin to the differential output average voltage.
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Electrical Characteristics: (VS+) – (VS–) = 5 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
–5.0
±1
5.0
UNIT
TEST
LEVEL (1)
OUTPUT COMMON-MODE VOLTAGE (VOCM) CONTROL (continued)
TA = 25°C
CM
VOS
Common-mode offset voltage
Common-mode offset voltage
drift (5)
VOCM pin driven to
[(VS+) – (VS–)] / 2
TA = 0°C to +70°C
–5.25
5.5
TA = –40°C to +85°C
–5.7
5.6
TA = –40°C to +125°C
–5.7
6.0
TA = –40°C to +125°C
–10
±2
TA = 25°C
Common-mode loop supply
headroom to negative supply
Common-mode loop supply
headroom to positive supply
< ±15-mV shift from
midsupply CM VOS
< ±15-mV shift from
midsupply CM VOS
10
A
B
mV
B
B
µV/°C
B
0.55
TA = 0°C to +70°C
0.6
A
B
V
TA = –40°C to +85°C
0.65
TA = –40°C to +125°C
0.7
B
TA = 25°C
1.2
A
TA = 0°C to 70°C
1.25
TA = –40°C to +85°C
1.3
TA = –40°C to +125°C
1.3
B
B
V
B
B
6.6 Electrical Characteristics: (VS+) – (VS–) = 3 V
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL (1)
AC PERFORMANCE
VOUT = 20 mVPP, G = 1, peaking (< 1.0 dB)
SSBW
Small-signal bandwidth
150
C
VOUT = 20 mVPP, G = 2
80
MHz
VOUT = 20 mVPP, G = 10
14
C
130
MHz
C
C
GPB
Gain-bandwidth product
VOUT = 20 mVPP, G = 100
LSBW
Large-signal bandwidth
VOUT = 1 VPP, G = 1
45
MHz
C
Bandwidth for 0.1-dB flatness
VOUT = 1 VPP, G = 1
14
MHz
C
SR
Slew rate (2)
VOUT = 1 VPP, FPBW, G = 1
110
V/µs
C
tR, tF
Rise and fall time
VOUT = 0.5-V step, G = 1, input tR = 4 ns
7.0
ns
C
To 0.1%, VOUT = 0.5-V step, input tR = 4 ns, G = 1
35
To 0.01%, VOUT = 0.5-V step, input tR = 4 ns, G = 1
55
tSETTLE
Settling time
Overshoot and undershoot
HD2
HD3
(1)
(2)
Second-order harmonic distortion
Third-order harmonic distortion
VOUT = 0.5-V step, G = 1, input tR = 4 ns
7%
f = 100 kHz, VOUT = 2 VPP, G = 1, RL = 1 kΩ
–128
f = 100 kHz, VOUT = 4 VPP, G = 1, RL = 1 kΩ
–127
f = 100 kHz, VOUT = 2 VPP, G = 1, RL = 1 kΩ
–139
f = 100 kHz, VOUT = 4 VPP, G = 1, RL = 1 kΩ
–125
Input voltage noise
f > 500 Hz, 1/f < 150 Hz
Input current noise
Overdrive recovery time
Closed-loop output impedance
ns
C
C
C
dBc
dBc
C
C
C
C
3.4
nV/√Hz
C
f > 20 kHz, 1/f < 10 kHz
0.5
pA/√Hz
C
G = 2, 2X output overdrive, dc coupled
100
ns
C
f = 100 kHz (differential), G = 1
0.02
Ω
C
Test levels (all values set by characterization and simulation): (A) 100% tested at TA ≈ 25°C. (B) Not tested in production; limits set by
characterization and simulation. (C) Typical value only for information.
This slew rate is the average of the rising and falling time estimated from the large-signal bandwidth as: (VPP / √2) × 2π × f–3dB.
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Electrical Characteristics: (VS+) – (VS–) = 3 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL (1)
dB
A
Ω
A
mΩ/°C
B
Ω
A
µΩ/°C
B
DC PERFORMANCE (3)
AOL
Open-loop voltage gain
Internal feedback trace resistance
Internal feedback trace resistance
mismatch
VIO
Input-referred offset voltage
Input offset voltage drift (5)
Input bias current
(positive current out of node)
IIB
Input bias current drift (5)
IOS
Input offset current
Input offset current drift (5)
±2-V differential to 1-kΩ differential load
100
120
TA = 25°C, RGT only (pins 11-1, 10-4)
3.0
3.45
TA = –40°C to +125°C, temperature drift
TA = 25°C, RGT only (pins 11-1, 10-4) (4)
4.7
50
–1
TA = –40°C to +125°C, temperature drift
0.05
1
50
TA = 25°C
–175
±40
175
TA = 0°C to +70°C
–225
265
TA = –40°C to +85°C
–295
295
TA = –40°C to +125°C
–295
375
TA = –40°C to +125°C (DGK package)
–2.0
±0.45
2.0
TA = –40°C to +125°C (RUN package)
–1.7
±0.4
1.7
TA = –40°C to +125°C (RGT package)
–1.8
±0.4
1.8
B
TA = 25°C
0.55
1.0
1.5
A
TA = 0°C to +70°C
0.42
1.73
TA = –40°C to +85°C
0.22
1.80
TA = –40°C to +125°C
0.22
2.0
TA = –40°C to +125°C
2
3.3
5.5
TA = 25°C
–50
±10
50
TA = 0°C to +70°C
–57
63
TA = –40°C to +85°C
–68
67
TA = –40°C to +125°C
–68
78
TA = –40°C to +125°C (DGK package)
–280
±70
280
TA = –40°C to +125°C (RGT and RUN package)
–120
±20
120
A
µV
B
B
B
B
µV/°C
µA
B
B
B
B
nA/°C
B
A
nA
B
B
B
pA/°C
B
B
INPUT
> 87-dB CMRR at input
range limits
TA = 25°C
(VS–) – 0.2 (VS–) – 0.1
TA = –40°C to +125°C
(VS–) – 0.1
Common-mode input, high
> 87-dB CMRR at input
range limits
TA = 25°C
(VS+) – 1.2
(VS+) –1.1
TA = –40°C to +125°C
(VS+) – 1.3
(VS+) –1.2
Common-mode rejection ratio
Input pins at [(VS+) – (VS–)] / 2
90
110
Input impedance differential mode
Input pins at [(VS+) – (VS–)] / 2
Common-mode input, low
CMRR
(3)
(4)
(5)
10
100 || 1.2
VS–
V
V
A
B
A
B
dB
A
kΩ || pF
C
Currents out of pin are treated as a positive polarity (with exception of the power-supply pin currents).
Trace mismatch measurement is dominated by the variation in contactor resistance. Internal mismatch is less than 0.1 Ω.
Input offset voltage drift, input bias current drift, and input offset current drift are the mean ±1-sigma values calculated by taking
measurements at the maximum-range ambient temperature end points, computing the difference, and dividing by the temperature
range. Maximum drift specifications are set by mean ±4 σ on the device distributions tested over a –40°̊ C to +125°̊ C ambient
temperature range. Drift is not specified by final ATE testing or QA sample test.
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Electrical Characteristics: (VS+) – (VS–) = 3 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
(VS–) + 0.2
(VS–) +
0.21
UNIT
TEST
LEVEL (1)
OUTPUT
TA = 25°C
VOL
Output voltage, low
TA = –40°C to +125°C
(VS–) + 0.2
(VS+) –
(VS+) – 0.2
0.21
TA = 25°C
VOH
Output voltage, high
Linear output current
±1.5 V, RL = 40 Ω,
VOCM offset < ±20 mV
TA = 25°C
±1.3 V, RL = 40 Ω,
VOCM offset < ±20 mV
TA = –40°C to +125°C
±30
±1.5 V, RL = 50 Ω,
AOL > 80 dB
TA = 25°C
±28
±1.1 V, RL = 50 Ω,
AOL > 80 dB
TA = –40°C to +125°C
±35
B
A
V
(VS+) –
(VS+) – 0.2
0.22
TA = –40°C to +125°C
Continuous output current
(VS–) +
0.22
A
V
B
±40
A
mA
B
±35
A
mA
±20
B
POWER SUPPLY
Specified operating voltage
IQ
Quiescent operating current
dIQ/dT
Quiescent current temperature
coefficient
±PSRR Power-supply rejection ratio
2.7
3
5.4
TA ≈ 25°C (6), VS+ = 3 V
1.24
1.31
1.40
TA = –40°C to +125°C, VS+ = 3 V
0.96
1.84
VS+ = 3 V
2.0
3.4
Either supply pin to differential VOUT
90
105
5.0
V
mA
B
A
B
µA/°C
B
dB
A
POWER-DOWN
Enable voltage threshold
Specified on above (VS–) + 1.15 V
Disable voltage threshold
Specified off below (VS–) + 0.55 V
Disable pin bias current
PD = VS– → VS+
(VS–) + 1.15
V
A
(VS–) + 0.55
V
A
B
–100
±10
100
nA
–2
1
5
µA
A
IQ(PD)
Power-down quiescent current
tON
Turn-on time delay
Time from PD = low to VOUT = 90% of final value
750
ns
C
tOFF
Turn-off time delay
Time from PD = low to VOUT = 10% of final value
150
ns
C
40
MHz
C
8
MHz
C
C
OUTPUT COMMON-MODE VOLTAGE (VOCM) CONTROL (7) (See Figure 65)
SSBW
Small-signal bandwidth
VOCM = 100 mVPP at the control pin
LSBW
Large-signal bandwidth
VOCM = 1 VPP at the control pin
SR
Slew rate (2)
From 1-VPP LSBW
12
V/µs
Output common-mode noise
VOCM pin driven from low impedance, f ≥ 2 kHz
15
nV/√Hz
Gain
VOCM control pin input to output average voltage
(see Figure 65)
DC output balance (differential
mode to common-mode output)
VOUT = ±1 V
SSBW
Output balance
LSBW
0.997
(7)
85
dB
300
VOUT = 1 VPP (output balance drops –3 dB from the
85-dB dc level)
300
–100
VOCM pin open, TA = –40°C to +125°C
A
C
C
±10
C
100
150 || 7
VOCM pin open
V/V
kHz
Input impedance
(6)
1.001
VOUT = 100 mVPP (output balance drops –3 dB from
the 85-dB dc level)
Input bias current
Default voltage offset from
[(VS+) – (VS–)] / 2
0.999
nA
A
kΩ || pF
C
–12
±2
12
mV
A
15
35
55
µA/°C
B
TA = 25°C and ICC ≈ 1.31 mA. The test limit is expanded for the ATE ambient range of 22°C to 32°C with a 4-µA/°C ICC temperature
coefficient considered; see Figure 95.
Specifications are from input VOCM pin to differential output average voltage.
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Electrical Characteristics: (VS+) – (VS–) = 3 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
–5.0
±1
5.0
UNIT
TEST
LEVEL (1)
OUTPUT COMMON-MODE VOLTAGE (VOCM) CONTROL (continued)
TA = 25°C
CM
VOS
Common-mode offset voltage
Common-mode offset voltage
drift (5)
VOCM input driven to
[(VS+) – (VS–)] / 2
TA = 0°C to +70°C
–5.25
5.5
TA = –40°C to +85°C
–5.7
5.6
TA = –40°C to +125°C
–5.7
6.0
VOCM input driven to [(VS+) – (VS–)] / 2
TA = 25°C
Common-mode loop supply
headroom to negative supply
Common-mode loop supply
headroom to positive supply
12
< ±15-mV shift from
midsupply CM VOS
< ±15-mV shift from
midsupply CM VOS
TA = 0°C to +70°C
–10
±2
10
A
mV
B
B
µV/°C
0.55
0.6
B
B
A
V
B
TA = –40°C to +85°C
0.65
TA = –40°C to +125°C
0.7
B
TA = 25°C
1.2
A
TA = 0°C to +70°C
1.25
TA = –40°C to +85°C
1.3
TA = –40°C to +125°C
1.3
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V
B
B
B
B
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SBOS778C – APRIL 2016 – REVISED JULY 2017
6.7 Typical Characteristics: (VS+) – (VS–) = 5 V
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
9
3
1
3
-1
0
Gain (dB)
Normalized Gain (dB)
2
6
0
-3
-6
-9
100k
-3
-4
VOUT = 20 mVpp
VOUT = 200 mVpp
VOUT = 1 Vpp
VOUT = 2 Vpp
VOUT = 4 Vpp
VOUT = 8 Vpp
-5
G = 0.1 V/V
G = 1 V/V
G = 2 V/V
G = 5 V/V
G = 10 V/V
1M
-2
-6
-7
-8
10M
Frequency (Hz)
-9
100k
100M
1M
10M
Frequency (Hz)
D001
VOUT = 20 mVPP, see Figure 61 and Table 2 for resistor values
100M
D002
See Figure 61
Figure 1. Small-Signal Frequency Response vs Gain
Figure 2. Frequency Response vs VOUT
6
3
2
3
1
0
0
Gain (dB)
Gain (dB)
-1
-2
-3
-4
-5
-6
-7
-8
-9
100k
VOCM = 0.8 V
VOCM = 1 V
VOCM = 1.5 V
VOCM = 2 V
VOCM = 3 V
VOCM = 3.5 V
1M
-6
10M
Frequency (Hz)
-12
100k
100M
D003
100
2
90
1
RO in each output (:)
-1
-2
-3
-4
-8
-9
100k
CL = 10 pF, RO = 97.6 :
CL = 47 pF, RO = 49.9 :
CL = 100 pF, R O = 32.4 :
CL = 470 pF, R O = 10.0 :
CL = 1000 pF, R O = 4.7 :
100M
D004
G = 1 V/V
G = 2 V/V
G = 5 V/V
G = 10 V/V
80
0
-7
10M
Frequency (Hz)
Figure 4. Small-Signal Frequency Response vs RL
3
-6
1M
VOUT = 20 mVPP, see Figure 61 with load resistance (RL) adjusted
Figure 3. Small-Signal Frequency Response vs VOCM
-5
RL = 50 :
RL = 100 :
RL = 200 :
RL = 500 :
RL = 1000 :
-9
VOUT = 20 mVPP , see Figure 61 with VOCM adjusted
Normalized Gain (dB)
-3
70
60
50
40
30
20
10
0
1M
10M
Frequency (Hz)
1
100M
D005
VOUT = 20 mVPP at load, G = 1, two series RO added at output
before capacitive load (CL)
10
100
Differential CL (pF)
1000
D006
Output resistance (RO) is two series output resistors to a
differential CL in parallel with a 1-kΩ load resistance
Figure 5. Small-Signal Frequency Response vs CL
Figure 6. Recommended RO vs CL
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Typical Characteristics: (VS+) – (VS–) = 5 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
1.2
0.5
1
0.4
0.8
0.3
Differential Output (V)
0.6
Differential Output (V)
RO = 0 :
RO = 75 :
0.4
0.2
0
-0.2
-0.4
-0.6
0.2-V step, tR = 1 ns
0.5-V step, tR = 2 ns
1-V step, tR = 4 ns
2-V step, tR = 8 ns
-0.8
-1
0.2
0.1
0
-0.1
-0.2
-0.3
-0.4
-1.2
-0.5
0
20
40
60
80
100 120
Time (ns)
140
160
180
200
0
20
40
60
80
D007
G = 1 V/V, 5-MHz input, single-ended to differential output
100 120
Time (ns)
140
Figure 7. Small- and Large-Signal Step Response
Figure 8. Step Response Into Capacitive Load
0.5
1
RO = 0 :
RO = 46.4 :
0.3
Differential Output (V)
Differential Output (V)
200
D008
0.4
0.5
0
-0.5
0.2-V step, tR = 1 ns
0.5-V step, tR = 1 ns
1-V step, tR = 2 ns
2-V step, tR = 5 ns
-1
0.2
0.1
0
-0.1
-0.2
-0.3
-0.4
-1.5
-0.5
0
20
40
60
80
100 120
Time (ns)
140
160
180
200
0
20
40
60
D009
G = 2 V/V, 5-MHz input, single-ended input to differential output
80
100 120
Time (ns)
140
160
180
200
D010
G = 2 V/V, VOUT = 500-mV step into 22-pF CL, see Figure 64
Figure 9. Small- and Large-Signal Step Response
Figure 10. Step Response Into Capacitive Load
0.2
Input and Differential Output Voltage (V)
10
0.2-V step, tR = 1 ns
1-V step, tR = 4 ns
2-V step, tR = 8 ns
0.15
Error to Final Value (%)
180
G = 1 V/V, VOUT = 500-mV step into 22-pF CL, see Figure 64
1.5
0.1
0.05
0
-0.05
-0.1
-0.15
-0.2
Input
Output
8
6
4
2
0
-2
-4
-6
-8
-10
0
10
20
30
40
50
Time ' from 50% of Input Edge (ns)
60
70
0
0.1
D011
Simulated with G = 1 V/V
0.2
0.3
0.4 0.5 0.6
Time (Ps)
0.7
0.8
0.9
1
D012
Single-ended to differential gain of 2, 2X input overdrive
Figure 11. Small- and Large-Signal Step Settling Time
14
160
Figure 12. Overdrive Recovery Performance
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SBOS778C – APRIL 2016 – REVISED JULY 2017
Typical Characteristics: (VS+) – (VS–) = 5 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
-50
-70
HD2
HD3
-60
HD2, 100 kHz
HD3, 100 kHz
HD2, 1 MHz
HD3, 1 MHz
-80
-70
Distortion (dBc)
Distortion (dBc)
-90
-80
-90
-100
-110
-100
-110
-120
-120
-130
-130
-140
-140
-150
10k
100k
1M
Frequency (Hz)
-150
0.1
10M
1
Differential Output Voltage (Vpp)
D013
G = 1 V/V, VOUT = 2 VPP
D014
G = 1 V/V
Figure 13. Harmonic Distortion vs Frequency
Figure 14. Harmonic Distortion vs Output Swing
-40
-60
Max IMD3
Max IMD2
-45
HD2, 100 kHz
HD3, 100 kHz
HD2, 1 MHz
HD3, 1 MHz
-70
-50
-80
-55
Distortion (dBc)
Spurious Level (dBc)
10
-60
-65
-70
-90
-100
-110
-75
-120
-80
-130
-85
-90
1M
-140
50
10M
Frequency (Hz)
100
Differential Load Resistance (:)
D015
G = 1 V/V, VOUT = 1 VPP each tone
D016
G = 1 V/V, VOUT = 2 VPP, with RL adjusted
Figure 15. Intermodulation Distortion (IMD2 and IMD3)
vs Frequency
Figure 16. Harmonic Distortion vs RL
-20
-105
HD2, 10 kHz
HD2, 100 kHz
HD2, 1 MHz
HD3, 10 kHz
HD3, 100 kHz
HD3, 1 MHz
-60
HD2, 100 kHz
HD3, 100 kHz
-110
-115
Distortion (dBc)
-40
Distortion (dBc)
1000
-80
-100
-120
-120
-125
-130
-135
-140
-140
-160
0.5
-145
1.5
2.5
VOCM - (VS-) (V)
3.5
4.5
1
10
Gain (V/V)
D017
G = 1 V/V, VOUT = 2 VPP, with VOCM adjusted
D018
VOUT = 2 VPP, seeTable 2 for gain setting
Figure 17. Harmonic Distortion vs VOCM
Figure 18. Harmonic Distortion vs Gain
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6.8 Typical Characteristics: (VS+) – (VS–) = 3 V
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
9
3
2
6
1
3
-1
Gain (dB)
Normalized Gain (dB)
0
0
-3
-6
-9
100k
-3
-4
-5
G = 0.1 V/V
G = 1 V/V
G = 2 V/V
G = 5 V/V
G = 10 V/V
1M
-2
-6
-7
-8
10M
Frequency (Hz)
-9
100k
100M
VOUT = 20 mVpp
VOUT = 200 mVpp
VOUT = 1 Vpp
VOUT = 2 Vpp
VOUT = 4 Vpp
1M
10M
Frequency (Hz)
D019
100M
D020
See Figure 61
VOUT = 20 mVPP, see Figure 61 and Table 2 for resistor values
Figure 20. Frequency Response vs VOUT
Figure 19. Small-Signal Frequency Response vs Gain
6
3
2
3
1
0
0
Gain (dB)
Gain (dB)
-1
-2
-3
-4
-3
-6
-5
-6
-7
-8
-9
100k
VOCM = 0.8 V
VOCM = 1 V
VOCM = 1.5 V
1M
-9
10M
Frequency (Hz)
-12
100k
100M
D021
D003
VOUT = 20 mVPP, see Figure 61 with VOCM adjusted
2
110
1
100
0
90
-1
-2
-3
-4
CL = 10 pF, RO = 113 :
CL = 47 pF, RO = 54.9 :
CL = 100 pF, R O = 34.0 :
CL = 470 pF, R O = 10.5 :
CL = 1000 pF, R O = 5.1 :
-7
-8
-9
100k
100M
D022
G = 1 V/V
G = 2 V/V
G = 5 V/V
G = 10 V/V
80
70
60
50
40
30
20
10
0
1M
10M
Frequency (Hz)
1
100M
D023
VOUT = 20 mVPP, G = 1 V/V,
two series RO added at output before CL
10
100
Differential CL (pF)
1000
D024
Two RO at output to differential CL in parallel with a
1-kΩ load resistance
Figure 23. Small-Signal Frequency Response vs CL
16
10M
Frequency (Hz)
Figure 22. Small-Signal Frequency Response vs RL
120
RO in each output (:)
Normalized Gain (dB)
Figure 21. Small-Signal Frequency Response vs VOCM
-6
1M
VOUT = 20 mVPP, see Figure 61 with RL adjusted
3
-5
RL = 50 :
RL = 100 :
RL = 200 :
RL = 500 :
RL = 1000 :
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Figure 24. Recommended RO vs CL
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Typical Characteristics: (VS+) – (VS–) = 3 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
0.5
1.5
RO = 0 :
RO = 82.5 :
0.4
0.3
Differential Output (V)
Differential Output (V)
1
0.5
0
-0.5
0.2-V step, tR = 2 ns
0.5-V step, tR = 4 ns
1-V step, tR = 8 ns
2-V step, tR = 12ns
-1
0.2
0.1
0
-0.1
-0.2
-0.3
-0.4
-1.5
-0.5
0
20
40
60
80
100 120
Time (ns)
140
160
180
200
0
20
40
60
80
D025
G = 1 V/V, 5-MHz input, single-ended input to differential output
100 120
Time (ns)
140
160
180
200
D026
G = 1 V/V, VOUT = 500-mV step into 22-pF CL, see Figure 64
Figure 25. Small- and Large-Signal Step Response
Figure 26. Step Response Into Capacitive Load
0.5
1.5
0.4
0.3
Differential Output (V)
Differential Output (V)
1
0.5
0
-0.5
0.2-V step, tR = 2 ns
0.5-V step, tR = 2 ns
1-V step, tR = 6 ns
2-V step, tR = 12 ns
-1
0.2
0.1
0
-0.1
-0.2
-0.3
RO = 0 :
RO = 51.1 :
-0.4
-1.5
-0.5
0
20
40
60
80
100 120
Time (ns)
140
160
180
200
0
G = 2 V/V, 5-MHz input, single-ended input to differential output.
40
60
80
100 120
Time (ns)
140
160
180
200
D028
G = 2 V/V, VOUT = 500-mV step into 22-pF CL, see Figure 64
Figure 27. Small- and Large-Signal Step Response
Figure 28. Step Response Into Capacitive Load
6
Input and Differential Output Voltage (V)
0.2
0.2-V step, tR = 2 ns
1-V step, tR = 8 ns
2-V step, tR = 12 ns
0.15
Error to Final Value (%)
20
D027
0.1
0.05
0
-0.05
-0.1
-0.15
Input
Output
4
2
0
-2
-4
-6
-0.2
0
10
20
30
40
50
Time ' from 50% of Input Edge (ns)
60
70
0
0.1
D029
Simulated with G = 1 V/V
0.2
0.3
0.4 0.5 0.6
Time (Ps)
0.7
0.8
0.9
1
D030
Single-ended to differential gain of 2, 2X input overdrive
Figure 29. Small- and Large-Signal Step Settling Time
Figure 30. Overdrive Recovery Performance
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Typical Characteristics: (VS+) – (VS–) = 3 V (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
-70
-40
-50
HD2
HD3
HD2, 100 kHz
HD3, 100 kHz
HD2, 1 MHz
HD3, 1 MHz
-80
-60
-90
Distortion (dBc)
Distortion (dBc)
-70
-80
-90
-100
-110
-120
-100
-110
-120
-130
-130
-140
-140
-150
10k
100k
1M
Frequency (Hz)
-150
0.1
10M
D031
G = 1 V/V, VOUT = 2 VPP
Figure 31. Harmonic Distortion vs Frequency
D032
Figure 32. Harmonic Distortion vs Output Swing
-60
Max IMD3
Max IMD2
HD2, 100 kHz
HD3, 100 kHz
HD2, 1 MHz
HD3, 1 MHz
-70
-50
-80
-55
Distortion (dBc)
Spurious Level (dBc)
4
G = 1 V/V
-40
-45
1
Differential Output Voltage (Vpp)
-60
-65
-70
-90
-100
-110
-75
-120
-80
-130
-85
-90
1M
-140
50
10M
Frequency (Hz)
D033
G = 1 V/V, 1 VPP each tone
Figure 33. IMD2 and IMD3 vs Frequency
Figure 34. Harmonic Distortion vs RL
HD2, 10 kHz
HD2, 100 kHz
HD2, 1 MHz
HD3, 10 kHz
HD3, 100 kHz
HD3, 1 MHz
HD2, 100 kHz
HD3, 100 kHz
-115
-90
-110
-130
-150
0.8
-120
-125
-130
-135
-140
1
1.2
1.4
1.6
VOCM - (VS-) (V)
1.8
2
1
10
Gain (V/V)
D035
VOUT = 2-VPP output, with VOCM adjusted
D036
VOUT = 2-VPP output, see Table 2 for gain setting
Figure 35. Harmonic Distortion vs VOCM
18
D034
-110
Distortion (dBc)
Distortion (dBc)
-70
1000
G = 1 V/V, VOUT = 2-VPP output, with RL adjusted
-30
-50
100
Differential Load Resistance (:)
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Figure 36. Harmonic Distortion vs Gain
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6.9 Typical Characteristics: 3-V to 5-V Supply Range
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
Open-Loop Gain (dB)
100
70
-100
40
-150
10
-200
-20
10
100
1k
10k
100k
1M
Frequency (Hz)
10M
100M
100
50
Output Impedance (:)
0
+5 V, Gain
+5 V, Phase
+3 V, Gain
-50
+3 V, Phase
Open-Loop Phase (deg)
130
2
1
0.5
+5 V, G = 1 V/V
+5 V, G = 2 V/V
+5 V, G = 5 V/V
+3 V, G = 1 V/V
+3 V, G = 2 V/V
+3 V, G = 5 V/V
0.2
0.1
0.05
0.02
0.01
0.005
0.002
0.001
10k
-250
1G
100k
1M
Frequency (Hz)
D037
Simulated with a 1-kΩ differential load and 0.6-pF internal
feedback capacitors removed
10M
100M
D038
Simulated closed-loop differential output impedance
Figure 38. Closed-Loop Output Impedance vs Frequency
Figure 37. Main Amplifier Differential Open-Loop Gain and
Phase vs Frequency
90
20
85
10
80
Output Balance (dB)
Input Spot Voltage (nV/—Hz) and
Current (pA/—Hz) Noise
20
10
5
1
+5 V, En
+5 V, In
+3 V, En
+3 V, In
0.1
10
100
75
70
65
60
55
50
45
1k
10k
Frequency (Hz)
100k
+5 V, SSOB
+5 V, 2-VPP OB
+3 V, SSOB
+3 V, 2-VPP OB
40
10k
1M
100k
1M
10M
Frequency (Hz)
D039
D040
Differential mode output to common-mode output,
simulated with G = 1 V/V
Figure 40. Output Balance vs Frequency
Figure 39. Input Spot Noise vs Frequency
120
120
+5 V
+3 V
115
110
100
105
PSRR (dB)
CMRR (dB)
110
100
95
90
80
90
70
85
80
1k
10k
100k
Frequency (Hz)
1M
10M
60
1k
D041
Common-mode input to differential output,
simulated with G = 1 V/V
Figure 41. CMRR vs Frequency
+5 V, VS+
+5 V, VS+3 V, VS+
+3 V, VS10k
100k
Frequency (Hz)
1M
10M
D042
Single-ended to differential gain of 1,
PSRR simulated to differential output
Figure 42. Power-Supply Rejection Ratio vs Frequency
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Typical Characteristics: 3-V to 5-V Supply Range (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
3
0.6
Output Common Mode Voltage (V)
2
1
0
Gain (dB)
-1
-2
-3
-4
-5
-6
+5 V, 100 mVpp
+5 V, 1 Vpp
+3 V, 100 mVpp
+3 V, 1 Vpp
-7
-8
-9
100k
10M
0
-0.2
+5 V, 0.2-V step
+5 V, 1-V step
+3 V, 0.2-V step
+3 V, 1-V step
-0.4
100M
0
0.1
1000
0.4
0.5 0.6
Time(Ps)
0.7
0.8
0.9
1
D044
3
Output CM Voltage Offset (mV)
+3 V, VOCM Pin Driven
+5 V, VOCM Pin Driven
+3 V, VOCM Pin Floating
+5 V, VOCM Pin Floating
100
+3 V
+5 V
2
1
0
-1
1k
10k
100k
Frequency (Hz)
1M
10M
0
0.5
1
D045
1.5
2
2.5
VOCM - (VS-) (V)
3
3.5
4
D046
Average VOCM output offset of 39 units,
standard deviation < 2 mV
Figure 45. Output Common-Mode Noise vs Frequency
Figure 46. VOCM Offset vs VOCM Setting
120
120
+5 V
+3 V
+5 V
+3 V
115
Positive Supply PSRR (dB)
115
Negative Supply PSRR (dB)
0.3
Figure 44. Common-Mode Voltage, Small- and Large-Step
Response (VOCM Pin Driven)
The VOCM pin is either driven to midsupply by low-impedance
source or allowed to float and default to midsupply
110
105
100
95
90
110
105
100
95
90
85
85
80
80
0
1
2
3
VOCM - (VS-) (V)
4
5
0
1
D047
Simulated with single-ended to differential gain of 1 , PSRR for
negative supply to differential output
Figure 47. –PSRR vs VOCM Approaching VS–
20
0.2
D043
Figure 43. Common-Mode Voltage, Small- and Large-Signal
Response (VOCM Pin Driven)
Output Spot Common Mode Noise (nV/—Hz)
0.2
-0.6
1M
Frequency (Hz)
10
100
0.4
2
3
VOCM - (VS-) (V)
4
5
D048
Simulated with single-ended to differential gain of 1, PSRR for
positive supply to differential output
Figure 48. +PSRR vs VOCM Approaching VS+
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Typical Characteristics: 3-V to 5-V Supply Range (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
60
80
+5 V
+3 V
72
50
No. of Units in 5 nA Bins
64
No. of units in 20 PV Bins
+5 V
+3 V
56
48
40
32
24
16
40
30
20
10
8
0
-180
-160
-140
-120
-100
-80
-60
-40
-20
0
20
40
60
80
100
120
140
160
180
-50
-45
-40
-35
-30
-25
-20
-15
-10
-5
0
5
10
15
20
25
30
35
40
45
50
0
Total of 234 DGK units trimmed at a 5-V supply
Total of 234 DGK units trimmed at a 5-V supply
Input Offset Voltage (PV)
Input Offset Current (nA)
D049
Figure 50. Input Offset Current (IOS)
100
50
80
40
60
30
Delta from 25°C I OS (nA)
Delta from 25°C V IO (µV)
Figure 49. Input Offset Voltage (VIO)
40
20
0
-20
-40
-60
20
10
0
-10
-20
-30
-40
-80
-100
-40
-20
0
20
40
60
80
Ambient Temperature (qC)
100
-50
-40
120
-20
D051
5-V and 3-V delta from 25°C VIO, 50 DGK units
0
20
40
60
80
Ambient Temperature (qC)
100
120
D052
5-V and 3-V delta from 25°C IOS, 50 DGK units
Figure 51. Input Offset Voltage vs Temperature
Figure 52. Input Offset Current vs Temperature
40
14
+5 V
+3 V
12
No. of Units in Each 0.05 nA/°C Bin
10
8
6
4
2
+5 V
+3 V
36
32
28
24
20
16
12
8
4
0
-1.8
-1.6
-1.4
-1.2
-1
-0.8
-0.6
-0.4
-0.2
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
1.8
0
Input Offset Voltage Drift (PV/qC)
D053
–40°C to +125°C endpoint drift, total of 62 DGK units
Figure 53. Input Offset Voltage Drift Histogram
-0.5
-0.45
-0.4
-0.35
-0.3
-0.25
-0.2
-0.15
-0.1
-0.05
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
0.45
0.5
No. of Units in Each 0.2 PV/qC Bin
D050
Input Offset Current Drift (nA/qC)
D054
–40°C to +125°C endpoint drift, total of 62 DGK units
Figure 54. Input Offset Current Drift Histogram
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Typical Characteristics: 3-V to 5-V Supply Range (continued)
at TA ≈ 25°C, VOCM pin = open, RF = 1 kΩ, RL = 1 kΩ, VOUT = 2 VPP, 50-Ω input match, G = 1 V/V, PD = VS+, single-ended
input, differential output, and input and output referenced to default midsupply for ac-coupled tests (unless otherwise noted);
see Figure 61 for a gain of 1-V/V test circuit
2.5
1.6
2
1.4
1.5
0.5
r1.5 V, Pos
r1.5 V, Neg
r2.5 V, Pos
r2.5 V, Neg
0
-0.5
1.2
Supply Current (mA)
Maximum VOUT (V)
1
-1
1
0.8
0.6
0.4
-1.5
0.2
-2
-2.5
50
+3 V
+5 V
0
100
Differential Load Resistance (:)
1000
0
1
D055
2
3
Disable Pin Voltage (V)
4
5
D056
Maximum differential output swing, VOCM at midsupply
Figure 55. ±Maximum VOUT vs Differential Load Resistance
Figure 56. Supply Current vs PD Voltage
200
+5 V
+3 V
160
No. of Units in 0.5 mV Bins
No. of Units in 2 mV Bins
180
140
120
100
80
60
40
20
Common Mode Offset Voltage (mV)
-4.5
-4
-3.5
-3
-2.5
-2
-1.5
-1
-0.5
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
VOCM Input driven to midsupply, total of 240 units
7
+3 V, PD
+3 V, VOUT
5
4
3
2
1
0
-1
-2
0
0.2
0.4
0.6
0.8
1
1.2
Time (Ps)
1.4
1.6
1.8
Figure 58. Common-Mode Output Offset from Driven VOCM
Histogram
Disable and Differential Output Voltage (V)
Disable and Differential Output Voltage (V)
Figure 57. Common-Mode Output Offset from VS+ / 2
Default Value Histogram
+5 V, PD
+5 V, VOUT
2
7
+5 V, PD
+5 V, VOUT
+3 V, PD
+3 V, VOUT
6
5
4
3
2
1
0
-1
-2
0
0.2
D059
5 MHz, 2-VPP input, G = 1 V/V, see Figure 61
0.4
0.6
0.8
1
1.2
Time (Ps)
1.4
1.6
1.8
2
D060
5 MHz, 2-VPP input, G = 1 V/V, see Figure 61
Figure 59. PD Turn-On Waveform
22
D058
Common Mode Offset Voltage (mV)
D057
VOCM input floating, total of 240 units
6
+5 V
+3 V
16
14
12
8
10
6
4
2
0
-2
-4
-6
-8
-10
-12
-14
-16
0
70
65
60
55
50
45
40
35
30
25
20
15
10
5
0
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Figure 60. PD Turn-Off Waveform
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7 Parameter Measurement Information
7.1 Example Characterization Circuits
The THS4551 offers the advantages of a fully differential amplifier (FDA) design with the trimmed input offset
voltage and very low drift of a precision op amp. The FDA is an extremely flexible device where the main aim is
to provide a purely differential output signal centered on a user-configurable common-mode voltage usually
matched to the input common-mode voltage required by an analog-to-digital converter (ADC) following this stage.
The primary options revolve around the choices of single-ended or differential inputs, ac-coupled or dc-coupled
signal paths, gain targets, and resistor value selections. The characterizations described in this section focus on
single-ended input to differential output designs as the more challenging application requirement. Differential
sources can certainly be supported and are often simpler to both implement and analyze.
The characterization circuits are typically operated with a single-ended, matched, 50-Ω, input termination to a
differential output at the FDA output pins because most lab equipment is single-ended. The FDA differential
output is then translated back to single-ended through a variety of baluns (or transformers), depending on the
test and frequency range. DC-coupled step response testing used two 50-Ω scope inputs with trace math. Singlesupply operation is most common in end equipment designs. However, using split balanced supplies allows
simple ground referenced testing without adding further blocking capacitors in the signal path beyond those
capacitors already within the test equipment. The starting point for any single-ended input to differential output
measurements (such as any of the frequency response curves) is shown in Figure 61 (available as a TINA-TI™
simulation file).
50- Input Match,
Gain of 1 V/V from RT,
Single-Ended Source to
Differential Output
THS4551 Wideband,
Fully Differential Amplifier
RF1
1k
RO1
487
VS+
RG1
1k
Network
Analyzer,
50- Source
Impedance
ADTL1-4-75+
±
RT1
52.3
VOCM
FDA
RG2
1k
Termination
RS1
50
RT2
52.3
N1
+
VOPP
±
+
50-
31.8-dB
Insertion Loss
from VOPP to a
50- Load
PD
VS-
VS+
RF2
1k
RO2
487
1-k
Differential
Load
RM
52.3
N2
50Single-Ended
Source
Network
Analyzer,
50- Load
Copyright © 2016, Texas Instruments Incorporated
Figure 61. Single-Ended Source to a Differential Gain of a 1-V/V Test Circuit
Most characterization plots fix the RF (RF1 = RF2) value at 1 kΩ, as shown in Figure 61. This element value is
completely flexible in application, but 1 kΩ provides a good compromise for the parasitic issues linked to this
value, specifically:
• Added output loading: the FDA functions similarly to an inverting op amp design with both feedback resistors
appearing as an added load across the outputs (the approximate total differential load in Figure 61 is
1 kΩ || 2 kΩ = 667 Ω). The 1-kΩ value also reduces the power dissipated in the feedback networks.
• Noise contributions resulting from resistor values: these contributions are both the 4kTRF terms and the
current noise times the RF value to the output (see the Noise Analysis section).
• Parasitic feedback pole at the input summing nodes: this pole is created by the feedback resistor (RF) value
and the 1.2-pF differential input capacitance (as well as any board layout parasitic) and introduces a zero in
the noise gain, thus decreasing the phase margin in most situations. This effect must be managed for best
frequency response flatness or step response overshoot. Internal 0.6-pF feedback capacitors on each side
combine with these external feedback resistors to introduce a zero in the noise gain, thereby reducing the
effect of the feedback pole to the differential input capacitance.
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Example Characterization Circuits (continued)
The frequency domain characterization curves start with the selections of Figure 61. Some of the features in this
test circuit include:
• The elements on the non-signal input side exactly match the signal input resistors. This feature has the effect
of more closely matching the divider networks on each side of the FDA. The three resistors on the non-signal
input side can be replaced by a single resistor to ground using a standard E96 value of 1.02 kΩ with some
loss in gain balancing between the two sides; see the Output DC Error and Drift Calculations and the Effect of
Resistor Imbalances section).
• Translating from a 1-kΩ differential load to a 50-Ω environment introduces considerable insertion loss in the
measurements (–31.8 dB in Figure 61). The measurement path insertion loss is normalized out when
reporting the frequency response curves to show the gain response to the FDA output pins.
• In the pass band for the output balun, the network analyzer 50-Ω load reflects to be in parallel with the 52.3-Ω
shunt termination. These elements combine to show a differential 1-kΩ load at the output pins of the
THS4551. The source impedance presented to the balun is a differential 50-Ω source. Figure 62 and
Figure 63 show the TINA-TI™ model (available as a TINA-TI™ simulation file) and resulting response
flatness for this relatively low-frequency balun providing 0.1-dB flatness through 100 MHz.
L1's Inductance : 198.94 uH
L2's Inductance : 198.94 uH
Mutual Inductance : 198.92972 uH
ADTL1-4-75 Model 198.94u
R3 50
+
N1
R1 25
VG1
N2
R2 25
+
V
VM1
Figure 62. Output Measurement Balun Simulation Circuit in TINA-TI™
10
8
-6.02
6
-6.03
4
-6.04
2
-6.05
0
-6.06
-2
-6.07
-4
-6.08
-6.09
-6.1
1k
Phase (deg)
Gain (dB)
-6
-6.01
-6
Gain (dB)
Phase (deg)
10k
-8
100k
1M
Frequency (Hz)
10M
-10
100M
D061
Figure 63. Output Measurement Balun Flatness Test
24
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Example Characterization Circuits (continued)
Starting from the test circuit of Figure 61, various elements are modified to show the effect of these elements
over a range of design targets, specifically:
• The gain setting is changed by adjusting the RT and the two RG elements to provide a 50-Ω input match and
setting the feedback resistors to 1 kΩ.
• Output loading of both resistive and capacitive load testing. Changing to lower resistive loads is accomplished
by adding parallel resistors across the output pins in Figure 61. Changing to capacitive loads adds series
output resistors to a differential capacitance before the 1-kΩ sense path of Figure 61.
• Power-supply settings. Most often, a single 5-V test uses a ±2.5-V supply and a 3-V test uses ±1.5-V supplies
with the VOCM input control at ground.
• The disable control pin (PD) is tied to the positive supply (VS+) for any active channel test.
7.2 Output Interface Circuit for DC-Coupled Differential Testing
The pulse response plots were taken using the output circuit of Figure 64. The two sides of this circuit present a
500-Ω load to ground (for a differential 1-kΩ load) with a 50-Ω source to the two scope inputs. Trace math is
used to combine the two sides into the pulse response plots of Figure 7 to Figure 10 and Figure 25 to Figure 28.
Using balanced bipolar supplies for this test ensures that the THS4551 outputs deliver a ground-centered
differential swing. This setup produces no dc load currents using the circuit of Figure 64.
RO1
475
RM1
56.2
THS4551
Output
RM1
56.2
RO2
475
50Scope
Input
50Scope
Input
Copyright © 2016, Texas Instruments Incorporated
Figure 64. Output Interface for DC-Coupled Differential Outputs
7.3 Output Common-Mode Measurements
The circuit of Figure 65 is a typical setup for common-mode measurements.
THS4551 Wideband,
Fully Differential Amplifier
RG1
1k
RF1
1k
VS+
Signal
Source
100
RS
49.9
VOCM
Input
±
FDA
RT
49.9
+
50Measurement
Equipment
±
+
PD
100
RT
1k
VS-
VS+
RF2
1k
Copyright © 2016, Texas Instruments Incorporated
Figure 65. Output Common-Mode Measurements
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Output Common-Mode Measurements (continued)
In Figure 65, the differential path is simply terminated back to ground on the two 1-kΩ input resistors and the
VOCM control input is driven from a 50-Ω matched source for the frequency response and step response curves
of Figure 43 and Figure 44. The outputs are summed to a center point (to obtain the average, or common-mode,
output) through two 100-Ω resistors. These 100-Ω resistors form an equivalent 50-Ω source to the commonmode output for measurements. This common-mode test circuit is available as a TINA-TI™ simulation file.
Figure 45 illustrates the common-mode output noise measurements with either a ground on the VOCM input pin
or with the VOCM input pin floating. The higher noise in Figure 45 for a floated input can be reduced by including
a capacitor to ground at the VOCM control input pin.
7.4 Differential Amplifier Noise Measurements
To extract out the input-referred noise terms from the total output noise, a measurement of the differential output
noise is required under two external conditions to emphasize the different noise terms. A high-gain, low resistor
value condition is used to emphasize the differential input voltage noise and a higher RF at low gains is used to
emphasize the two input current noise terms. The differential output noise must be converted to single-ended
with added gain before being measured by a spectrum analyzer. At low frequencies, a zero 1/f noise, high-gain,
differential to single-ended instrumentation amplifier (such as the INA188) is used. At higher frequencies, a
differential to single-ended balun is used to drive into a high-gain, low-noise, op amp (such as the LMH6629). In
this case, the THS4551 outputs drive 25-Ω resistors into a 1:1 balun where the balun output is terminated singleendedly at the LMH6629 input with 50 Ω. This termination provides a modest 6-dB insertion loss for the
THS4551 differential output noise that is then followed by a 40-dB gain setting in the very wideband LMH6629.
7.5 Balanced Split-Supply Versus Single-Supply Characterization
Although most end applications use a single-supply implementation, most characterizations are done on a split
balanced supply. Using a split balanced supply keeps the I/O common-mode inputs near midsupply and provides
the most output swing with no dc bias currents for level shifting. These characterizations include the frequency
response, harmonic distortion, and noise plots. The time domain plots are in some cases done via single-supply
characterization to obtain the correct movement of the input common-mode voltage.
7.6 Simulated Characterization Curves
In some cases, a characteristic curve can only be generated through simulation. A good example of this scenario
is the output balance plot of Figure 40. This plot shows the best-case output balance (output differential signal
versus output common-mode signal) using exact matching on the external resistors in simulation using a singleended input to differential output configuration. The actual output balance is set by resistor mismatch at low
frequencies but intersects and follows the high-frequency portion of Figure 40.
The remaining simulated plots include:
• AOL gain and phase, see Figure 37.
• Large- and small-signal settling times, see Figure 11 and Figure 29.
• Closed-loop output impedance versus frequency, see Figure 38.
• CMRR vs frequency, see Figure 41.
• PSRR vs frequency and output common-mode voltage, see Figure 42, Figure 47, and Figure 48.
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7.7 Terminology and Application Assumptions
Numerous common terms that are unique to this type of device exist. This section identifies and explains these
terms.
• Fully differential amplifier (FDA). This term is restricted to devices offering what appears similar to a
differential inverting op amp design element that requires an input resistor (not a high-impedance input) and
includes a second internal control loop that sets the output average voltage (VOCM) to a default or set point.
This second common-mode control loop interacts with the differential loop in certain configurations.
• The desired output signal at the two output pins is a differential signal that swings symmetrically around a
common-mode voltage, which is the average voltage for the two outputs.
• Single-ended to differential. The output must always be used differentially in an FDA; however, the source
signal can be either a single-ended or a differential source with a variety of implementation details for either
source. For an FDA operating in single-ended to differential, only one of the two input signals is applied to
one of the input resistors.
• The common-mode control has limited bandwidth from the input VOCM pin to the common-mode output
voltage. The internal loop bandwidth beyond the input VOCM buffer is a much wider bandwidth than the
reported VOCM bandwidth, but is not directly discernable. A very wide bandwidth in the internal VOCM loop is
required to perform an effective and low-distortion single-ended to differential conversion.
Several features in the application of the THS4551 are not explicitly stated, but are necessary for correct
operation. These features are:
• Good power-supply decoupling is required. Often a larger capacitor (2.2 µF, typical) is used along with a highfrequency, 0.1-µF supply decoupling capacitor at the device supply pins (share this capacitor with the four
supply pins in the RGT package). For single-supply operation, only the positive supply has these capacitors.
Where a split supply is used, connect these capacitors to ground on both sides with the larger capacitor
placed some distance from the package and shared among multiple channels of the THS4551, if used. A
separate 0.1-µF capacitor must be provided to each device at the device power pins. With cascaded or
multiple parallel channels, including ferrite beads from the larger capacitor to the local high-frequency
decoupling capacitor is often useful.
• Although often not stated, the power disable pin (PD) is tied to the positive supply when only an enabled
channel is desired.
• Virtually all ac characterization equipment expects a 50-Ω termination from the 50-Ω source and a 50-Ω,
single-ended source impedance from the device outputs to the 50-Ω sensing termination. This condition is
achieved in all characterizations (often with some insertion loss) but is not necessary for most applications.
Matching impedance is most often required when transmitting over longer distances. Tight layouts from a
source, through the THS4551, and to an ADC input do not require doubly-terminated lines or filter designs.
The only exception is if the source requires a defined termination impedance for correct operation (for
example, mixer outputs).
• The amplifier signal path is flexible for use as single- or split-supply operation. Most applications are intended
to be single supply, but any split-supply design can be used as long as the total supply voltage across the
TH4551 is less than 5.5 V and the required input, output, and common-mode pin headrooms to each supply
are taken into account. When left open, the VOCM pin defaults to near midsupply for any combination of split
or single supplies used. The disable pin (PD) is referenced to the negative rail. Using a negative supply
requires that PD be pulled down to within 0.55 V of the negative supply to disable the amplifier.
• External element values are normally assumed to be accurate and matched. In an FDA, this assumption
translates to equal feedback resistor values and a matched impedance from each input summing junction to
either a signal source or a dc bias reference on each side of the inputs. Unbalancing these values introduces
non-idealities in the signal path. For the signal path, imbalanced resistor ratios on the two sides creates a
common-mode to differential conversion. Furthermore, mismatched RF values and feedback ratios create
additional differential output error terms from any common-mode dc or ac signal or noise terms. Using
standard 1% resistor values is a typical approach and generally leads to some nominal feedback ratio
mismatch. Modestly mismatched resistors or ratios do not by themselves degrade harmonic distortion. Where
there is a meaningful common-mode noise or distortion coming in that gets converted to differential via an
element or ratio mismatch. For the best dc precision, use 0.1% accuracy resistors that are readily available in
E96 values (1% steps).
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8 Detailed Description
8.1 Overview
In addition to the core differential I/O voltage feedback gain block, there are two 5.2-kΩ resistors internally across
the outputs to sense the average voltage at the outputs. These resistors feed the average voltage back into a
VCM error amplifier where the voltage is compared to either a default voltage divider across the supplies or an
externally set VOCM target voltage. When the amplifier is disabled, the default midsupply bias string is disabled to
save power.
To achieve the very-low noise at the low power provided by the THS4551, the input stage transistors are
relatively large, thus resulting in a higher differential input capacitance (1.2 pF in the Functional Block Diagram).
As a default compensation for the 1.2-pF differential input capacitance and the 1-kΩ feedback resistors used in
characterization, internal 0.6-pF capacitors are placed between the two output and input pins. Adjust any desired
external feedback capacitor value to account for these 0.6-pF internal elements. When using the 16-pin WQFN
package and the internal feedback traces to the input side of the package, include the nominal trace impedance
of 3.3 Ω in the design. These elements are not included in the TINA-TI™ model and must be added externally to
a design intending to use the RGT package.
8.2 Functional Block Diagram
VS+
3.3
(RGT Package) FB+
0.6 pF
OUT+
IN±
±
5.2 k
High-AOL +
Differential I/O
Amplifier ±
1.2 pF
IN+
5.2 k
+
OUT±
0.6 pF
VS+
3.3
(RGT Package) FB±
300 k
VS±
±
VCM
Error
Amplifier
+
PD
VOCM
CMOS
Buffer
300 k
VS±
28
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8.3 Feature Description
8.3.1 Differential Open-Loop Gain and Output Impedance
The most important elements to the closed-loop performance are the open-loop gain and open-loop output
impedance. Figure 66 and Figure 67 show the simulated differential open-loop gain and phase from the
differential inputs to the differential outputs with no load and with a 100-Ω load. Operating with no load removes
any effect introduced by the open-loop output impedance to a finite load. This AOL simulation removes the 0.6-pF
internal feedback capacitors to isolate the forward path gain and phase (see Figure 99). The 0.6-pF capacitance
becomes part of the feedback network that sets the noise gain and phase combined with the external elements.
The simulated differential open-loop output impedance is shown in Figure 68.
80
-90
100 : Load
No Load
70
-100
-110
Open-Loop Phase (deg)
Open-Loop Gain (dB)
60
50
40
30
20
10
0
-120
-130
-140
-150
-160
-170
-180
-190
-10
No Load
100 : Load
-200
-20
100k
1M
10M
Frequency (Hz)
100M
-210
100k
1G
1M
10M
Frequency (Hz)
D063
Figure 66. No-Load and 100-Ω Loaded AOL Gain
100M
1G
D064
Figure 67. No-Load and 100-Ω AOL Phase
Differential Output Impedance (:)
10000
1000
100
10
1
10
100
1k
10k
100k
1M
Frequency (Hz)
10M
100M
1G
D062
Figure 68. Differential Open-Loop Output Impedance
This impedance combines with the load to shift the apparent open-loop gain and phase to the output pins when
the load changes. The rail-to-rail output stage shows a very high impedance at low frequencies that reduces with
frequency to a lower midrange value and then peaks again at higher frequencies. The maximum value at low
frequencies is set by the common-mode sensing resistors to be a 10.5-kΩ dc value (see the Functional Block
Diagram section). This high impedance at a low frequency is significantly reduced in closed-loop operation by the
loop gain, as shown in the closed-loop output impedance of Figure 38. Figure 66 compares the no load AOL gain
to the AOL gain driving a 100-Ω load that shows the effect of the output impedance. The heavier loads pull the
AOL gain down faster to lower crossovers with more phase shift at the lower frequencies.
The much faster phase rolloff for the 100-Ω differential load explains the greater peaked response illustrated in
Figure 4 and Figure 22 when the load decreases. This same effect happens for the RC loads common with
converter interface designs. Use the TINA-TI™ model to verify loop phase margin in any design.
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Feature Description (continued)
8.3.2 Setting Resistor Values Versus Gain
The THS4551 offers considerable flexibility in the configuration and selection of resistor values. The design starts
with the selection of the feedback resistor value. The 1-kΩ feedback resistor value used for the characterization
curves is a good compromise between power, noise, and phase margin considerations. With the feedback
resistor values selected (and set equal on each side) the input resistors are set to obtain the desired gain with
input impedance also set with these input resistors. Differential I/O designs provide an input impedance that is
the sum of the two input resistors. Single-ended input to differential output designs present a more complicated
input impedance. Most characteristic curves implement the single-ended to differential design as the more
challenging requirement over differential-to-differential I/O.
For single-ended, matched, input impedance designs, Table 2 illustrates the suggested standard resistors set to
approximately a 1-kΩ feedback. This table assumes a 50-Ω source and a 50-Ω input match and uses a single
resistor on the non-signal input side for gain matching. Better matching is possible using the same three resistors
on the non-signal input side as on the input side. Figure 69 shows the element values and naming convention for
the gain of 1-V/V configuration where the gain is defined from the matched input at RT to the differential output.
50- Input Match,
Gain of 1 V/V from RT,
Single-Ended Source to
Differential Output
50Source
Impedance
THS4551 Wideband,
Fully Differential Amplifier
RF1
1k
VS+
RG1
1k
RS1
50
VOPP
±
RT
52.3
VOCM
FDA
+
+
RG2
1.02 k
RL
1k
±
PD
VS-
VS+
RF2
1k
Copyright © 2016, Texas Instruments Incorporated
Figure 69. Single-Ended to Differential Gain of 1 V/V with Input Matching Using Standard Resistor Values
Starting from a target feedback resistor value, the desired input matching impedance, and the target gain (AV),
the required input RT value is given by solving the quadratic of Equation 1.
RS
§
·
2RS ¨ 2RF
A V2 ¸
2RFRS2 A V
2
©
¹
RT 2 RT
0
2RF 2 A V
RS A V (4 A V ) 2RF 2 A V
RS A V (4 A V )
(1)
When this value is derived, the required input side gain resistor is given by Equation 2 and then the single value
for RG2 on the non-signal input side is given by Equation 3:
R
2 F RS
AV
RG1
RS
1
RT
(2)
RF
AV
RS
1
RT
2
RG2
30
(3)
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Feature Description (continued)
Using these expressions to generate a swept gain table of values results in Table 2, where the best standard 1%
resistor values are shown to minimize input impedance and gain error to target.
Table 2. Swept Gain 50-Ω Input Match with RF = 1-kΩ (±1 Standard Values)
GAIN (V/V)
RF
RG1
RT
RG2
ZIN
AV
0.1
1000
10000
49.9
10000
49.66
0.09965
1
1000
976
51.1
1000
49.2
1.0096
2
1020
499
52.3
523
48.9
1.988
5
1000
187
59
215
50.2
5.057
10
1020
88.7
69.8
118
50.6
10.09
Where an input impedance match is not required, simply set the input resistor to obtain the desired gain without
an additional resistor to ground (remove RT in Figure 69). This scenario is common when coming from the output
of another single-ended op amp (such as the OPA192). This single-ended to differential stage shows a higher
input impedance than the physical RG as given by the expression for ZA (active input impedance) shown as
Equation 4.
ZA
RG1
§
¨1
©
RG1 ·§
RF ·
¸¨ 1
¸
RG2 ¹©
RG1 ¹
RF
2
RG2
(4)
Using Equation 4 for the gain of 1 V/V with all resistors equal to 1-kΩ shows an input impedance of 1.33 kΩ. The
increased input impedance comes from the common-mode input voltage at the amplifier pins moving in the same
direction as the input signal. The common-mode input voltage must move to create the current in the non-signal
input RG resistor to produce the inverted output. The current flow into the signal-side input resistor is impeded
because the common-mode input voltage moves with the input signal, thus increasing the apparent input
impedance in the signal input path.
8.3.3 I/O Headroom Considerations
The starting point for most designs is to assign an output common-mode voltage for the THS4551. For accoupled signal paths, this voltage is often the default midsupply voltage to retain the most available output swing
around the voltage centered at the VOCM voltage. For dc-coupled designs, set this voltage with consideration to
the required minimum headroom to the supplies as described in the specifications of the Electrical
Characteristics table for the VOCM control. For precision ADC drivers, this output VOCM becomes the input VCM to
the ADC. Often, VCM is set to VREF / 2 to center the differential input on the available input when precision ADCs
are being driven.
From the target output VOCM, the next step is to verify that the desired output differential peak-to-peak voltage
(VOPP) stays within the supplies. For any desired differential VOPP, make sure that the absolute maximum voltage
at the output pins swings with Equation 5 and Equation 6 and confirm that these expressions are within the
supply rails minus the output headroom required for the RRO device.
VOPP
VOmax VOCM
4
(5)
VOPP
VOmin VOCM
4
(6)
For instance, when the THS4551 drives the ADC3223 with a 0.95-VCM control using a single 3.0-V supply, the
negative-going signal sets the maximum output swing from 0.95 VCM to 0.2 V above ground. This 0.75-V, singlesided swing becomes an available 4 × 0.75 V = 3-VPP differential around the nominal 0.95-VCM output commonmode voltage. On the high side, the maximum output is equal to 1.7 V (0.95 V + 0.75 V), which is well within the
allowed maximum range of 2.8 V (3.0 V – 0.2 V). This available 3-VPP maximum differential output is also well
beyond the maximum value required for the 2-VPP input ADS3223.
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With the output headrooms confirmed, the input junctions must also stay within the operating range. The input
range limitations only appear when approaching the positive supply where a maximum 1.3-V headroom is
required over the full temperature range because the input range extends to the negative supply voltage over the
full temperature range.
The input pins operate at voltages set by the external circuit design, the required output VOCM, and the input
signal characteristics. For differential-to-differential designs where there is no signal-related movement in the
input VICM voltages, ac-coupled differential input designs have a VICM equal to the output VOCM. Going towards
the positive supply, the output common-mode can be set to within 1.2 V of the supply. AC-coupled input designs
violate the required 1.3-V headroom on the input pins in this case. Going towards the negative supply on the
VOCM setting requires a minimum of 0.55 V above the supply. This extreme is always in range for the input pins
that require a minimum 0-V headroom to the negative supply.
DC-coupled differential input designs must check the voltage divider from the source common-mode input
voltage to the THS4551 VOCM setting. This result must be equal to an input VICM within the specified range. If the
source VCM can vary over some voltage range, validate this result over that range before proceeding.
For single-ended input to differential output designs, the VICM is nominally at a voltage set by the external
configuration with a small swing around the nominal value because of the common-mode loop. An ac-coupled,
single-ended input to differential output design places an average input VICM equal to the output VOCM for the
FDA with an ac-coupled swing around the VOCM voltage following the input voltage. A dc-coupled, single-ended
input to differential design gets a nominal input VICM set by the source signal common-mode level and the VOCM
output voltage with a small signal-related swing around the nominal VICM voltage.
One approach to deriving the VICM voltage range for any single-ended input to differential output design is to
observe the output voltage swing on the non-signal input side of the FDA outputs and simply take the voltage
division on the input pin to ground or to the dc reference used on that side. An example analysis is shown in
Figure 70 using a Thevenized version of the gain of 2 values listed in Table 2 for a 50-Ω matched impedance,
ac-coupled design.
In this example, a single 3.3-V supply is used with the VOCM defaulted to midsupply or 1.65 V as a commonmode output voltage. This value is also the common-mode voltage on the input pins for the ac-coupled input to
the FDA. Targeting a 4-VPP differential output swing means each output pin swings ±1 V around this 1.65-V
common-mode voltage. This output swing is in range because the full swing is 0.65 V to 2.65 V relative to
ground, which is well within the 0.2-V output headroom requirements on a single 3.3-V supply.
THS4551 Wideband,
Fully Differential Amplifier
VS+
Thevinized
Source
VS-
RS1
25.6
1 µF
RF1
1.02 k
VS+
RG1
499
VOUT = 4-VPP
Differential
RL
1k
±
+
±
3.3 V
+
±
VOCM
0V
FDA
VIN = ±1.022 V
1 µF
+
±
+
1 µF
PD
VSRG2
523
VS+
RF2
1.02 k
Copyright © 2016, Texas Instruments Incorporated
Figure 70. Input Swing Analysis Circuit with AC-Coupled, Single-Ended to Differential Signal Path
The output on the lower side of this design ranges from 0.65 V to 2.65 V. This 2-VPP swing (on just one side, the
other output is an inverted version and gives the 4-VPP differential maximum) is divided back by the RF2 and RG2
divider to the input pins to form a common-mode input swing on top of the 1.65-V input common-mode voltage.
This divider is 0.339 × 2 VPP = 0.678 VPP or ±0.34 V around the 1.65-V input common-mode voltage. The 1.31-V
to 1.99-V common-mode input swing for this design is in range for the 0 V to 2.2 V available input range (the
maximum headroom is 3.3 V – 1.1 V, which is equal to 2.2 V). These voltage swings can be directly observed
using the SBOC460 TINA-TI™ simulation file. Shifting the VOCM down slightly (if allowed by the design
requirements) is a good way to improve the positive-swinging input headroom for this low-voltage design.
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Taking a more complex example by using the THS4551 to attenuate a large bipolar input signal in a dc-coupled
design for an ADC is shown in Figure 71. To remove the peaking for this low-noise gain design, the two CF
elements and an input capacitor are added to shape the noise gain at high frequencies to a capacitive divider, as
described in the Designing Attenuators section. In this example (including the 1.2-pF internal differential capacitor
at the inputs and the 0.6-pF internal feedback capacitors), the high-frequency noise gain is 3 V/V and a flat
frequency response with approximately 45 MHz of –3-dB BW is delivered.
3.1 pF
10 k
VOCM
4.096 V
+
THS4551 Wideband,
Fully Differential Amplifier
RF1
1k
VOUT = 7.2 VPP
Differential
1 µF
10 k
±
Gain of 0.2 V/V,
DC-Coupled,
Single-Ended Source to
Differential Output
VS+
RG1
4.99 k
±
VS+
5V
+
±
VS0V
VOCM
VS = ±18 V
2.6 pF
+
FDA
+
RL
1k
±
+
PD
VS-
VS+
For Attenuator Design:
NG1 = 1.2
NG2 = 3
GBP = 135 MHz
Zo = 14.3 MHz
±
RG2
4.99 k
A 1-k load is very
important to include because
of high-frequency resonance;
no load may oscillate.
RF2
1k
3.1 pF
Copyright © 2016, Texas Instruments Incorporated
Figure 71. DC-Coupled, Single-Ended to Differential Attenuator Design
In this example, the output VOCM is 4.096 V / 2, which equals 2.048 V and the source signal VCM is 0 V. These
values set the nominal input pin VICM to 2.048 V × 4.99 kΩ / (4.99 kΩ + 1 kΩ) = 1.71 V. Applying a ±18-V input at
the 4.99-kΩ input resistor produces a 7.2-VPP differential output. That is, a ±1.8-V swing on the lower output side
around the 2.048-V common-mode voltage. This 0.248-V to 3.84-V relative-to-ground swing at the output is well
within the 0.2-V output headrooms to the 0-V to 5-V supplies used in the example in Figure 71 (with the same
swing inverted on the other output side). That output swing on the lower side produces an attenuated input
common mode swing of (±1.8 V × (4.99 kΩ / (4.99 kΩ + 1 kΩ)) = ±1.5 V around the midscale input bias of
1.71 V. This 0.2-V to 3.2-V input common-mode swing is well within the available 0-V to 3.8-V input range. This
±18-V bipolar input signal is delivered to a SAR ADC with a 7.2-VPP differential output with all I/O nodes
operating in range using a single 5-V supply design. The source must sink the 2.048 V / 5.99 kΩ = 0.34-mA
common-mode level-shifting current to take the input 0-V common-mode voltage up to the midscale 1.71-V VICM
operating voltage. Using the single-ended input impedance of Equation 4, the source must also drive an
apparent input load of 5.44 kΩ.
Most designs do not run into an input range limit. However, using the approach shown in this section can allow a
quick assessment of the input VICM range under the intended full-scale output condition. The TINA-TI™
simulation file for Figure 71 can be used to plot the input voltages under the intended swings and application
circuit to verify that there is no limiting from this effect. Driving the I/O nodes out of range in the TINA-TI™ model
results in convergence problems. Increasing the positive and negative supplies slightly in simulation is an easy
way to discover the simulated swings that might be going out of range.
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As a third example of arriving at the input pin voltage swings, use the design of Figure 93 (the ADC3241 design).
Thevenize the source to just one input resistor to get an expression for the input VICM in terms of the input
voltage to be derived. Figure 72 shows the gain of 5 V/V, dc-coupled, matched input impedance, single-ended to
differential circuit of Figure 93 with both sides reduced to a single input resistor. In Figure 93, the design operates
on a single 3.3-V supply with an output VOCM equal to 0.95 V to directly connect to TI’s line of low-power
ADC3xxx series of 12- and 14-bit ADCs. This family accepts a 2-VPP maximum differential voltage, which (at the
input-terminating resistor of Figure 93) is a ±0.2-V swing. Going back to the source through the matching resistor
is then a ±0.4-V source swing. Thevenizing that source with the RT element provides the ±0.217 V shown in
Figure 72 and the total R2 as the sum of RG1 and 50 Ω || 59 Ω.
THS4551 Wideband,
Fully Differential Amplifier
RF1
1k
VOUT
VS+
VOCM
VS-
Thevinized
Source
VS+
R2
215
±1-V Differential Output
Centered on 0.95 VCM
±
3.3 V
+
±
0V
+
±
0.95 V
+
±
VOCM
VTHEV
±0.217 V
FDA
+
+
R2
215
RL
1k
±
PD
VS-
VS+
RF2
1k
Copyright © 2016, Texas Instruments Incorporated
Figure 72. Input VICM Analysis Circuit From the Design of Figure 93
For an input signal (VTHEV) that swings around ground as ±VTHEV, the input pins are within a range given by
Equation 7, which is a superposition of the output VOCM divided back to the input nodes and half of the input
±VTHEV signal.
RG
V
RF
r THEV u
VICM VOCM
RG RF
2
RF RG
(7)
Using the values from the design of Figure 72, the computed input range for the THS4551 input pins is VICM =
0.168 V ± 0.89 mV or 0.079 V to 0.257 V at the input pins. These values are well within range for the negative
rail input available in the THS4551.
A simpler approach to arriving at the input common mode range for this DC coupled single supply design would
be to take the output voltage swing range on the lower side (non – signal input side) and simply divide it back
through its resistor divider to ground on that side.
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The output pin voltage swing is 0.95 V ± 0.5 V or 0.45 V to 1.45 V. This swing is divided back to the input VICM
by a 215 / (215 + 1000) = 0.177 ratio. This ratio computes the input pin range as 79 mV to 0.256 V, matching the
input source swing results in Equation 7. The TINA-TI™ model for Figure 72 (available as SBOC472) also
provides these input swings as shown in the simplified circuit of Figure 73. The large centered swing is the
differential output voltage at the THS4551 output pins (which is actually the two outputs swinging ±0.5 V around
a 0.95 VCM), the small centered bipolar swing is the input swing for the thevenized source of Figure 73, and the
smallest VPP swing on a dc offset is the input VICM voltage at the non-signal side input for the circuit of Figure 73.
1.2
1
0.8
0.6
Output (V)
0.4
0.2
0
-0.2
-0.4
-0.6
VIN+
VOUT
VS
-0.8
-1
-1.2
0
100
200
300
400 500 600
Time (ns)
700
800
900 1000
D069
Figure 73. I/O Swing Simulation Using the TINA-TI™ Model
8.3.4 Output DC Error and Drift Calculations and the Effect of Resistor Imbalances
The THS4551 offers a trimmed input offset voltage and extremely low offset drift over the full –40°C to +125°C
operating range. This offset voltage combines with several other error contribution terms to produce an initial
25°C output offset error band and then a drift over temperature. For each error term, a gain must be assigned to
that term. For this analysis, only dc-coupled signal paths are considered. One new source of output error (versus
the typical op amp analysis) arises from the effect mismatched resistor values and ratios can have on the two
sides of the FDA. Any common-mode error or drift creates a differential output error through the slight
mismatches arising from the external feedback and gain setting resistor tolerances or standard value constraints.
The error terms (25°C and drift), along with the gain to the output differential voltage, include input offset voltage
and input offset current. Input offset voltage has a gain equal to the noise gain or 1 + RF / RG, where RG is the
total dc impedance from the input pins back to the source or a dc reference (typically ground). Input offset current
has a gain to the differential output through the average feedback resistor value.
The remaining terms arise from an assumed range on both the absolute feedback resistor mismatch and the
mismatch in the divider ratio on each side of the FDA. The first of these resistor mismatch terms is the input bias
current that creates a differential output offset via RF mismatch. For simplicity, the upper RF and RG values are
termed RF1 and RG1 with a ratio of RF1 / RG1 ≡ G1. The lower elements are defined as RF2 and RG2 with a ratio of
RF2 / RG2 ≡ G2. To compute worst-case contributions, a maximum variation in the design resistor tolerance is
used in the absolute and ratio mismatches.
For instance, ±1% tolerance resistors are assumed, giving a worst-case G1 that is 2% higher than nominal and a
G2 that is 2% lower than nominal with a worst-case RF value mismatch of 2% as well. Using a 0.1% precision
resistor reduces the gain for the input bias current, but because these precision resistors are usually only
available in 1% value steps, a gain mismatch term may still need to be considered. For matched impedance
designs with RT and RG1 on a single-ended to differential stage, the standard value constraint imposes a fixed
mismatch in the initial feedback ratios with the tolerance of the resistors around the ratio if the non-signal input
side uses a single resistor for RG2.
Define the selected external resistor tolerance as ±T (so for 1% tolerance resistors, T = 0.01). Input bias current
times the feedback resistor mismatch gain is ±2 × T × RFnom.
Anything that generates an output common-mode level or shift over temperature also generates an output
differential error term if the two feedback ratios, G1 and G2, are not equal. An error trying to produce a shift in
the output common-mode voltage is overridden by the common-mode control loop where the error becomes a
balanced differential error around the output VOCM.
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The terms that create a differential error from a common-mode term and feedback ratio mismatch include the
desired VOCM voltage, any source common-mode voltage, any drift on the reference bias to the VOCM control pin,
any internal offset and drift in the VOCM control path, and the input average bias current and drift.
Considering just the output common-mode control and the source common-mode voltage, the conversion to
output differential offsets is through Equation 8.
VOCM G1 G2
VICM G1 G2
VOD
G1 G2
1
2
(8)
Neglecting any G1 and G2 mismatch because of standard values constraint, the conversion gain for these two
terms can be recast in terms of the nominal RF / RG ≡ G and the tolerance T, as shown in Equation 9. When G
increases, this conversion gain approaches 4T.
VOD
G
4T
‡
(1 G) (1 T 2 )
VOCM
(9)
This conversion gain to differential output error is applied to two error terms: VOCM and the input bias current and
drift. (The source common-mode voltage is assumed to be 0 V. If not, apply this gain to the source commonmode voltage and any resulting shift in application.)
The output error is applied to VOCM, assuming that the input control pin is driven and not floating. The input bias
current and drift are multiplied by the average RF value then by the conversion gain to differential output error to
create an added output differential error.
As an example of using these terms to estimate the worst-case output 25°C error band and then the worst-case
drift (by adding all error terms together independently), use the gain of 1-V/V configuration with RF = 1 kΩ and
assume a ±1% tolerance on the resistors with the standard values used in Figure 74.
50- Input Match,
Gain of 1 V/V from RT,
Single-Ended Source to
Differential Output
50Source
Impedance
VS+
+
±
VS-
5V
+
±
VOCM
0V
+
±
THS4551 Wideband,
Fully Differential Amplifier
RF1
1k
VS+
RG1
1k
RS1
50
VOPP
±
2.5 V
RT1
52.3
VOCM
FDA
+
+
RG2
1.02 k
RL
1k
±
PD
VS-
VS+
RF2
1k
Copyright © 2016, Texas Instruments Incorporated
Figure 74. DC-Coupled Gain of 1 with RF = 1 kΩ and Single-to-Differential Matched Input 50-Ω Impedance
The standard value constraint on the non-signal input side actually produces more gain mismatch than the
resistor tolerances. For Figure 74, G2 = 1000 / 1020 = 0.9804 and G1 = 1000 / 1025.6 = 0.9751 nominally, then
with a ±2% tolerance around the initial gain mismatch resulting from the standard values available if 1% resistors
are used.
Using the maximum 25°C error terms and nominal resistor values with an exact 2.5-V input to the VOCM control
pin gives Table 3, gains to the output differential error (VOD), and then the summed output error band at 25°C.
The output error is clearly dominated by the VOCM voltage and the effect of the nominal feedback dividers being
slightly mismatched. This analysis does not include resistor tolerances but the approach is the same with the
wider error bands on the gain terms. Using 1% tolerance on the resistors setting the gain matching dominates
the output error band through the VOCM input voltage. For the lowest output error, this analysis shows that an
exact match on the feedback dividers with precision resistors is preferred. However, doing so would require
duplicating the exact network on the non-signal input side and the signal input side. Where input impedance
matching is not required, the two RG resistors are simply single equal resistors and the gain mismatch is just
from the tolerance of the resistors.
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Table 3. Worst-Case Output VOD Error Band
ERROR TERM
25°C MAX VALUE
GAIN TO VOD
OUTPUT ERROR
(mV)
GAIN COMMENT
Input VIO
±0.175 mV
1.9777
±0.346
Average noise gain
Input IOS
±50 nA
1000
±0.05
Feedback resistor
Input IBCM
1.5 µA
20 Ω
±0.03
Feedback resistor mismatch
Input IBCM
1.5 µA
1 kΩ × 0.00268
±0.004
Converted to differential by gain mismatch
VOCM input
2.5 V
0.00268
±6.7
VOCM to differential by gain mismatch
Total
±7.13
The 0.00268 conversion gain for the gain ratio mismatch is the worst-case ratio starting from the initially lower
G1 value resulting from the standard value constraint and using a ±1% tolerance on the RF and RG elements of
the ratio. Adding in the resistor tolerances to the gain mismatch term greatly increases the contribution of those
terms.
Normally, the expected drift in the output VOD is of more interest than an initial error band. Table 4 shows these
terms for the RGT package and the summed results by adding all terms independently to obtain a worst-case
drift.
Table 4. Worst-Case Output VOD Drift Band
MAX VALUE
GAIN TO VOD
OUTPUT ERROR
(µV/°C)
GAIN COMMENT
Input VIO
±1.8 µV/°C
1.9777
±3.56
Average noise gain
Input IOS
±120 pA/°C
1000
±0.12
Feedback resistor
Input IBCM
5.0 nA/°C
20 Ω
±0.10
Feedback resistor mismatch
Input IBCM
5.0 nA/°C
1 kΩ × 0.00268
±0.013
Converted to differential by gain mismatch
VOCM input
±10 µV/°C
0.00268
±0.027
VOCM to differential by gain mismatch
Total
±3.82
ERROR TERM
In Table 4, the input offset voltage drift dominates the output drift. For the last term, the drift for the VOCM path is
just for the internal offset drift of the common-mode path with a driven input. Any added external drift on the
source of the VOCM input must also be considered. This type of calculation can be repeated for the exact
application circuit considering each of these terms in the context of a specific design.
The absolute accuracy and drift for the THS4551 are exceptionally good. Mismatched resistor feedback ratios
combined with a high drift in the VOCM control input can actually dominate the output VOD drift. Where the output
differential precision is more important than the input matching accuracy, consider matching the networks on the
two sides of the input to obtain improved nominal G1 to G2 match. The gains for the input bias current error
terms are relatively low when using the 1-kΩ feedback values. Higher RF values provide the input-current-related
drift terms more gain.
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8.4 Device Functional Modes
The wideband FDA requires external resistors for correct signal-path operation. When configured for the desired
input impedance and gain setting with these external resistors, the amplifier can be either on with the PD pin
asserted to a voltage greater than (VS–) + 1.15 V, or turned off by asserting PD low (within 0.55 V of the negative
supply). Disabling the amplifier shuts off the quiescent current and stops correct amplifier operation. The signal
path is still present for the source signal through the external resistors, which provides poor signal isolation from
the input to output in power-down mode.
Internal protection diodes remain present across the input pins in both operating and shutdown mode. Large
input signals during disable can turn on the input differential protection diodes, thus producing a load current in
the supply even in shutdown.
The VOCM control pin sets the output average voltage. Left open, VOCM defaults to an internal midsupply value.
Driving this high-impedance input with a voltage reference within the valid range sets a target for the internal VCM
error amplifier. If floated to obtain a default midsupply reference for VOCM, an external decoupling capacitor is
recommended to be added on the VOCM pin to reduce the otherwise high output noise for the internal highimpedance bias (see Figure 45).
8.4.1 Operation from Single-Ended Sources to Differential Outputs
One of the most useful features supported by the FDA device is an easy conversion from a single-ended input to
a differential output centered on a user-controlled, common-mode level. Although the output side is relatively
straightforward, the device input pins move in a common-mode manner with the input signal. The common-mode
voltage at the input pins, which moves with the input signal, increases the apparent input impedance to be
greater than the RG value. The input active impedance issue applies to both ac- and dc-coupled designs, and
requires somewhat more complex solutions for the resistors to account for this active impedance, as discussed in
the Setting Resistor Values Versus Gain section.
8.4.1.1 AC-Coupled Signal Path Considerations for Single-Ended Input to Differential Output
Conversions
When the signal path can be ac-coupled, the dc biasing for the THS4551 becomes a relatively simple task. In all
designs, start by defining the output common-mode voltage. The ac-coupling issue can be separated for the
input and output sides of an FDA design. The input can be ac-coupled and the output dc-coupled, or the output
can be ac-coupled and the input dc-coupled, or both can be ac-coupled. One situation where the output can be
dc-coupled (for an ac-coupled input), is when driving directly into an ADC where the VOCM control voltage uses
the ADC common-mode reference to directly bias the FDA output common-mode voltage to the required ADC
input common-mode voltage. In any case, the design starts by setting the desired VOCM. When an ac-coupled
path follows the output pins, the best linearity is achieved by operating VOCM at midsupply, which can be easily
delivered by floating the VOCM pin. The VOCM voltage must be within the linear range for the common-mode
loop, as specified in the headroom specifications (approximately 0.7 V greater than the negative supply and
1.3 V less than the positive supply for the full –40°C to +125°C operation). If the output path is also ac-coupled,
simply letting the VOCM control pin float is usually preferred in order to obtain a midsupply default VOCM bias with
minimal elements. To limit noise, place a 0.1-µF decoupling capacitor on the VOCM control pin to ground.
After VOCM is defined, check the target output voltage swing to ensure that the VOCM plus the positive and
negative output swing on each side does not clip into the supplies. If the desired output differential swing is
defined as VOPP, divide by 4 to obtain the ±VP (peak voltage) swing around VOCM at each of the two output pins
(each pin operates 180° out of phase with the other). Check that VOCM ±VP does not exceed the absolute supply
rails for the rail-to-rail output (RRO) device. Common-mode current does not flow from the common-mode output
voltage set by the VOCM pin towards the device input pins side, because both the source and balancing resistor
on the non-signal input side are dc blocked (see Figure 70). The ac-coupled input path sets the input pin
common-mode voltage equal to the output common-mode voltage. The input pin positive headroom requirement
(1.2 V) is less than the VOCM positive headroom (1.3 V). If the VOCM is in range, the input pins are also in range
for the ac-coupled input configuration. This headroom requirement functions similarly for when the output VOCM
voltage approaches the negative supply. The approximate minimum headroom of 0.6 V to the negative supply on
the VOCM voltage is greater than the input pin voltage headroom of approximately 0 V for the negative rail input
design. The input common-mode voltage is also in range if the output common-mode voltage is in range and
above 0.6 V from the negative supply because the input common-mode voltage follows the output VOCM setting
for ac-coupled input designs.
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Device Functional Modes (continued)
The input pin voltages move in a common-mode manner with the input signal, as described in the I/O Headroom
Considerations section. Confirm that the VOCM voltage plus the input VPP common-mode swing also stays in
range for the input pins.
8.4.1.2 DC-Coupled Input Signal Path Considerations for Single-Ended to Differential Conversions
The output considerations remain the same as for the ac-coupled design. Again, the input can be dc-coupled
when the output is ac coupled. A dc-coupled input with an ac-coupled output can have some advantages to
move the input VICM down by adjusting the VOCM down if the source is ground referenced. When the source is dccoupled into the THS4551 (see Figure 69), both sides of the input circuit must be dc-coupled to retain differential
balance. Normally, the non-signal input side has an RG element biased to whatever the source midrange is
expected to be, provided that this midscale reference gives a balanced differential swing around VOCM at the
outputs. Often, RG2 is simply grounded for dc-coupled, bipolar-input applications. This configuration provides a
balanced differential output if the source swings around ground. If the source swings from ground to some
positive voltage, grounding RG2 gives a unipolar output differential swing from both outputs at VOCM (when the
input is at ground) to one polarity of the swing. Biasing RG2 to an expected midpoint for the input signal creates a
differential output swing around VOCM.
One significant consideration for a dc-coupled input is that VOCM sets up a common-mode bias current from the
output back through RF and RG to the source on both sides of the feedback. Without input balancing networks,
the source must sink or source this dc current. After the input signal range and biasing on the other RG element
is set, check that the voltage divider from VOCM to VIN through RF and RG (and possibly RS) establishes an input
VICM at the device input pins that is in range. If the average source is at ground, the negative rail input stage for
the THS4551 is in range for applications using a single positive supply and a positive output VOCM setting
because this dc common-mode current lifts the average FDA input summing junctions up off of ground to a
positive voltage (the average of the V+ and V– input pin voltages on the FDA). TINA-TI™ simulations of the
intended circuit offer a good check for input and output pin voltage swings (see Figure 72).
8.4.2 Operation from a Differential Input to a Differential Output
In many ways, this method is a much simpler way to operate the FDA from a design equations perspective.
Again, assuming that the two sides of the circuit are balanced with equal RF and RG elements, the differential
input impedance is now just the sum of the two RG elements to a differential inverting summing junction. In these
designs, the input common-mode voltage at the summing junctions does not move with the signal but must be dc
biased in the design range for the input pins and must take into account the voltage headroom required to each
supply. Slightly different considerations apply to ac- or dc-coupled differential input to differential output designs,
as described in the following sections.
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Device Functional Modes (continued)
8.4.2.1 AC-Coupled, Differential-Input to Differential-Output Design Issues
The most common way to use the THS4551 with an ac-coupled differential source is to simply couple the input
into the RG resistors through the blocking capacitors. Figure 75 shows a typical blocking capacitor approach to a
differential input. An optional input differential termination resistor (RM) is included in this design. The RM element
allows the input RG resistors to be scaled up and still delivers lower differential input impedance to the source. In
this example, the RG elements sum to show a 1-kΩ differential impedance and the RM element combines in
parallel to provide a net 500-Ω ac differential impedance to the source. Again, the design ideally proceeds by
selecting the RF element values, then the RG to set the differential gain, and then an RM element (if needed) to
achieve a target input impedance. Alternatively, the RM element can be eliminated, with the 2 × RG elements set
to the desired input impedance and RF set to obtain the differential gain (equal to RF / RG).
THS4551 Wideband,
Fully Differential Amplifier
VS+
VS-
+
+
Differential I/O
with AC-Coupled
Input
10 nF
VS+
RG1
499
VOUT
±
VOCM
±
3.3 V
±
RF1
1.02 k
0V
VIN
RF1
1.02 k
1 µF
10 nF
FDA
RL
1k
±
+
PD
VSRG2
499
+
VS+
RF2
1.02 k
Copyright © 2016, Texas Instruments Incorporated
Figure 75. Example AC-Coupled Differential Input Design
The dc biasing for an ac-coupled differential input design is very simple. The output VOCM is set by the input
control voltage and, because there is no dc current path for the output common-mode voltage (as long as RM is
only differential and not split and connected to ground for instance), the dc bias also sets the common-mode
operating points for the input pins. For a purely differential input, the voltages on the input pins remain fixed at
the output VOCM setting and do not move with the input signal (unlike the single-ended input configurations where
the input pin common-mode voltages do move with the input signal). The SLOC341 TINA-TI™ simulation file is
available for Figure 75.
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Device Functional Modes (continued)
8.4.2.2 DC-Coupled, Differential-Input to Differential-Output Design Issues
Operating the THS4551 with a dc-coupled differential input source is very simple and only requires that the input
pins stay in range for the dc common-mode operating voltage. The example in Figure 76 takes the output of a
dual precision op amp (such as the OPA2192) where a high differential input signal is attenuated by the
THS4551 down into the range of an 18-bit SAR ADC such as the 2-MSPS ADS9110. The input stage provides a
differential gain of 21 V/V with a common-mode gain of 1 V/V. This example amplifies a small differential signal
on top of a very-wide range common-mode voltage. The input common-mode voltage appears at the outputs of
the OPA2192. The input common-mode voltage is level shifted by the FDA common-mode control to be at the
required output common-mode voltage to drive the ADS9110 SAR ADC (with a 4.096-V reference, as shown in
Figure 76); the FDA output common-mode voltage must be at the 2.048 V shown in Figure 76. This design offers
a very high CMRR using the common-mode control loop of the FDA to reset the output common-mode voltage
from that delivered to the inputs of the OPA2192. The actual CMRR from the OPA2192 inputs to the FDA
outputs is dominated by the resistor mismatches in the FDA. The feedback and differential input capacitors are
included to shape the noise gain as described in the Designing Attenuators section. This full example circuit is
available as a TINA-TI™ simulation file.
Gain of 0.2 V/V,
DC-Coupled,
Differential Input to
Differential Output
2k
VOCM
4.096 V
+
10 k
±
22 pF
VCC
2.048 V
10 k
THS4551 Wideband,
Fully Differential Amplifier
1 µF
RG1
4.99 k
±
VIN1
RF1
1k
10
+
ADS9110
Inputs
VS+
VEE
200
VS+
VS-
VCC
VEE
±
OPA2192
49 pF
2k
VOCM
FDA
+
5V
+
±
0V
+
±
15 V
+
±
-15 V
+
12 nF
±
+
PD
VCC
±
VS-
±
VS+
10
VIN2
+
VEE
RG2
4.99 k
RF2
1k
22 pF
Copyright © 2016, Texas Instruments Incorporated
Figure 76. Example DC-Coupled Differential I/O Design from a Precision Dual Op Amp to an 18-Bit SAR
8.4.3 Input Overdrive Performance
Figure 12 illustrates a 2X overdrive triangle waveform for the THS4551. The input resistor is driven with a ±9-V
swing for the gain of 2-V/V configuration in the test circuit of Figure 61 using a single 5-V supply. When the
output maximum swing is reached at approximately the supply values, the increasing input voltage turns on the
internal protection diodes across the two input pins. The internal protection diodes are two diodes in series in
both polarities. This feature clamps the maximum differential voltage across the inputs to approximately 1.5 V
when the output is limited at the supplies but the input exceeds the available range. The input resistors on both
sides limit the current flow in the internal diodes under these conditions.
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
Most applications for the THS4551 strive to deliver the best dynamic range in a design that delivers the desired
signal processing along with adequate phase margin for the amplifier itself. The following sections detail some of
the design issues with analysis and guidelines for improved performance.
9.1.1 Noise Analysis
The first step in the output noise analysis is to reduce the application circuit to the simplest form with equal
feedback and gain setting elements to ground. Figure 77 shows the simplest analysis circuit with the FDA and
resistor noise terms to be considered.
enRg2
enRf2
RG
RF
r
r
In+2
+
In±
2
eno2
±
eni2
enRg2
enRf2
RG
RF
r
r
Figure 77. FDA Noise Analysis Circuit
The noise powers are shown in Figure 77 for each term. When the RF and RG (or RI) terms are matched on each
side, the total differential output noise is the root sum squared (RSS) of these separate terms. Using NG ≡ 1 +
RF / RG, the total output noise is given by Equation 10. Each resistor noise term is a 4kT × R power (4kT = 1.6E20J at 290K).
eo
eniNG
2
2 iNRF
2
2 4kTRFNG
(10)
The first term is simply the differential input spot noise times the noise gain, the second term is the input current
noise terms times the feedback resistor (and because there are two uncorrelated current noise terms, the power
is two times one of them), and the last term is the output noise resulting from both the RF and RG resistors, at
again twice the value for the output noise power of each side added together. Running a wide sweep of gains
when holding RF close to 1 kΩ and setting the input up for a 50-Ω match gives the standard values and resulting
noise listed in Table 5.
Note that when the gain increases, the input-referred noise approaches only the gain of the FDA input voltage
noise term at 3.3 nV/√Hz.
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Application Information (continued)
Table 5. Swept Gain of the Output- and Input-Referred Spot Noise Calculations
GAIN (V/V)
RF
RG1
RT
RG2
ZIN
AV
EO (nV/√Hz)
0.1
1000
10000
49.9
10000
49.66
0.09965
7
EI (nV/√Hz)
70
1
1000
976
51.1
1000
49.2
1.0096
10.4
10.4
2
1020
499
52.3
523
48.9
1.988
13.9
6.95
5
1000
187
59
215
50.2
5.057
23
4.6
10
1020
88.7
69.8
118
50.6
10.09
36.4
3.64
9.1.2 Factors Influencing Harmonic Distortion
As illustrated in the swept frequency harmonic distortion plots (Figure 13 and Figure 31), the THS4551 provides
extremely low distortion at lower frequencies. In general, an FDA output harmonic distortion mainly relates to the
open-loop linearity in the output stage corrected by the loop gain at the fundamental frequency. When the total
load impedance decreases, including the effect of the feedback resistor elements in parallel for loading purposes,
the output stage open-loop linearity degrades, thus increasing the harmonic distortion; see Figure 16 and
Figure 34. When the output voltage swings increase, very fine scale open-loop output stage nonlinearities
increase that also degrade the harmonic distortion; see Figure 14 and Figure 32. Conversely, decreasing the
target output voltage swings drops the distortion terms rapidly. A nominal swing of 2 VPP is used for harmonic
distortion testing where Figure 14 illustrates the effect of going up to an 8-VPP differential input that is more
common with SAR converters.
Increasing the noise gain functions to decrease the loop gain resulting in the increasing harmonic distortion
terms; see Figure 18 and Figure 36. One advantage to the capacitive compensation for the attenuator designs is
that the noise gain is shaped up with frequency to achieve a crossover at an acceptable phase margin at higher
frequencies. This technique (see the Designing Attenuators section) holds the loop gain high at frequencies
lower than the noise gain zero, thus improving distortion at lower frequencies.
The THS4551 holds nearly constant distortion when the VOCM operating point is moved in the allowed range; see
Figure 17 and Figure 35. Clipping into the supplies with any combination of VOCM and VOPP rapidly degrades
distortion performance.
The THS4551 does an exceptional job of converting from single-ended inputs to differential outputs with very low
harmonic distortions. External resistors of 1% tolerance are used in characterization with good results.
Unbalancing the feedback divider ratios does not degrade distortion directly. Imbalanced feedback ratios convert
common-mode inputs to a differential mode at the outputs with the gain described in the Output DC Error and
Drift Calculations and the Effect of Resistor Imbalances section.
9.1.3 Driving Capacitive Loads
The capacitive load of an ADC or some other next-stage device is commonly required to be driven. Directly
connecting a capacitive load to the output pins of a closed-loop amplifier such as the THS4551 can lead to an
unstable response; see the step response plots into a capacitive load (Figure 8, Figure 10, Figure 26, and
Figure 28). One typical remedy to this instability is to add two small series resistors (RO) at the outputs of the
THS4551 before the capacitive load. Figure 6 and Figure 24 illustrate parametric plots of recommended RO
values versus differential capacitor load values and gains. Operating at higher noise gains requires lower RO
values to obtain a ±0.5-dB flat response for the same capacitive load. Some direct parasitic loading is acceptable
without a series RO that increases with gain setting (see Figure 8, Figure 10, Figure 26, and Figure 28 where the
RO value is 0 Ω). Even when these plots suggest that a series RO is not required, good practice is to leave a
place for the RO elements in a board layout (a 0-Ω value initially) for later adjustment in case the response
appears unacceptable.
The rail-to-rail output stage of the THS4551 has an inductive characteristic in the open-loop output impedance at
higher frequencies; see Figure 68. This inductive open-loop output impedance introduces added phase shift at
the output pins for direct capacitive loads and feedback capacitors. Larger values of feedback capacitors (greater
than 100 pF) can risk a low phase margin. Including a 10-Ω to 15-Ω series resistor with a feedback capacitor can
be used to reduce this effect.
The TINA-TI™ simulation model does a good job of predicting these issues and illustrating the effect for different
choices of capacitive load isolating resistors (RO) and different feedback capacitor configurations.
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9.1.4 Interfacing to High-Performance Precision ADCs
The THS4551 provides a simple interface to a wide variety of precision SAR and delta-sigma (ΔΣ) ADCs. To
deliver the exceptional distortion at the output pins, considerably wider bandwidth than what is typically required
in the signal path to the ADC inputs is provided by the THS4551. This wide amplifier bandwidth provides the low
broadband, closed-loop output impedance to supply the sampling glitches and to recover quickly for the best
SFDR. A particularly challenging task is to drive the high-frequency modulator sample rates for a precision ΔΣ
converter where the modulator frequency can be far higher than the final output data rate. Figure 78 shows a
tested example circuit using the THS4551 in a 500-kHz, active multiple feedback (MFB) filter driving the 24-bit
ADS127L01. This filter is designed for FO = 500 kHz and Q = 0.63 to give a linear phase response with the –3dB frequency at 443 kHz. This example circuit is available as a TINA-TI™ simulation file.
1.2 k
270 pF
VOCM
330
1 nF
470 pF
1.2 k
+
±
3V
5
+
±
1.2 k
THS4551
10
AINN
ADS127L01
22 nF
AINP
330
5
10
270 pF
1.2 k
Copyright © 2016, Texas Instruments Incorporated
Figure 78. 500-kHz Low-Pass Active Filter
This 3-V supply example provides a low-power interface to the very low-power ADC. This circuit is available on
the ADS127L01EVM board.
The 5-Ω resistors inside the loop at the output pins and the 1-nF differential capacitor across the FDA input pins
are not part of the filter design. These elements function to improve the loop-phase margin with minimal
interaction with the active filter operation To observe the loop gain and phase margin, use the SBOC461 TINATI™ simulation file. Tested performance with the ADS127L01 at a 4-kHz input shows the exceptional THD and
SNR of –114 dBc and 106 dB, respectively. Figure 79 uses the ADS127L01 at a modulator frequency of 16 MHz.
0
-20
-40
Amplitude (dB)
-60
-80
-100
-120
-140
-160
-180
-200
0
20
40
60
80
Frequency (kHz)
100
120
D066
Figure 79. 4-kHz FFT Test for the Gain of 1 V/V Interface in Figure 78
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9.1.5 Operating the Power Shutdown Feature
The CMOS input pin must be asserted to the desired voltage for operation. An internal pullup resistor is not
provided on the PD pin so that off-state quiescent current can be minimized. For applications simply requiring the
device to be powered on when the supplies are present, tie the PD pin to the positive supply voltage.
The disable operation is referenced from the negative supply, normally ground. For split-supply operation, with
the negative supply below ground, a disable control voltage below ground is required to turn the THS4551 off. To
assure an off state condition, the disable control pin must be below a voltage within 0.55 V of the negative
supply.
For single-supply operation, a minimum of 1.15 V above the negative supply (ground in this case) is required to
assure on operation. This logic threshold range allows direct operation from a 1.8-V supply logic when the
THS4551 operates with a single positive supply and ground.
9.1.6 Designing Attenuators
Operating the THS4551 at a low-noise gain (or with higher feedback resistors) can cause a lower phase margin
to exist, thus giving the response peaking illustrated in Figure 1 for the gain of a 0.1 (a 1/10 attenuator) condition.
Although operating the THS4551 as an attenuator is often useful, taking a large input range to a controlled output
common-mode voltage with a purely differential signal around the VOCM voltage, the response peaking illustrated
in Figure 1 is usually undesirable. Several approaches can be used to reduce or eliminate this peaking, usually
at the cost of higher output noise. DC attenuation at the input usually increases the output noise broadband,
whereas using an ac noise gain shaping technique that peaks the noise gain only at higher frequencies is more
desirable. This peaking output noise can then be filtered off with the typical passive RC filters often used after
this stage. Figure 80 shows a simplified schematic for the gain of 0.1-V/V test from Figure 1.
Gain of 0.1 V/V,
DC-Coupled,
Single-Ended Source to
Differential Output
VS+
RG1
10 k
2.5 V
VS+
VS-
+
+
±
-2.5 V
THS4551 Wideband,
Fully Differential Amplifier
RF1
1k
VOUT
±
VOCM
VIN
FDA
+
+
±
RL
1k
±
PD
VS-
VS+
RF2
1k
RG2
10 k
Copyright © 2016, Texas Instruments Incorporated
Figure 80. Divide-by-10 Attenuator Application for the THS4551
A 5-dB peaked response (see Figure 82) results from the configuration of Figure 80, which results from a
nominal 32° phase margin. This peaking can be eliminated by placing two feedback capacitors across the RF
elements and a differential input capacitor. Adding these capacitors provides a transition from a resistively set
noise gain (NG1 = 1.1 in Figure 80) to a capacitive divider at high frequency, and flattening out to a higher noise
gain (NG2). The key for this approach is to target a ZO where the noise gain begins to peak up. Using only the
following terms, and targeting a closed-loop flat (Butterworth) response, gives this solution sequence (from
Equation 11 to Equation 13) for ZO and then the capacitor values. See the OPA847 data sheet (page 12) for a
discussion of this inverting noise gain shaping technique.
• Gain bandwidth product in Hz (135 MHz for the THS4551)
• Low-frequency noise gain, NG1 (equal to 1.1 in the attenuator gain of a 0.1-V/V design)
• The target high-frequency noise gain is selected to be higher than NG1 (NG2 = 5 V/V) in this example
• Feedback resistor value, RF (is assumed balanced for this differential design = 1 kΩ)
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From these elements, for any voltage feedback op amp or FDA, solve for ZO as shown in Equation 11:
ZO
GBP §
¨1
NG12 ¨©
NG1
NG2
1 2
NG1 ·
¸
NG2 ¸¹
(11)
From this target zero frequency in the noise gain, the feedback capacitors can be solved as Equation 12:
1
CF
2S ‡ RF ‡ ZO ‡ NG2
(12)
The next step is to resolve the input capacitance on the summing junction. Equation 13 is for a single-ended op
amp where the capacitor goes to ground. To use the capacitance (CS) resulting from Equation 13 for a voltagefeedback FDA, cut the target value in half and place the resulting CS across the two inputs (reducing the external
value by the specified internal differential capacitance).
CS
NG2 1 CF
(13)
Using the computed capacitor values allows for an estimate of the resulting flat response bandwidth f–3dB
frequency, as shown in Equation 14:
f
3dB
|
GBP ‡ ZO
(14)
Running through these steps for the THS4551 in the attenuator circuit of Figure 80 provides the proposed
compensation of Figure 81, where Equation 14 estimates a bandwidth of 22 MHz (the ZO target is 3.5 MHz). The
solutions for CF gives 9 pF, where this value is reduced to 8.4 pF to account for the internal 0.6-pF feedback.
The single-ended solution for CS gives 36 pF, which is reduced to 18 pF to be differential, and is then further
reduced to 16.8 pF to account for the internal 1.2-pF differential input capacitance of the THS4551.
CF1
8.4 pF
Gain of 0.1 V/V,
DC-Coupled,
Single-Ended Source to
Differential Output
VS+
VS+
RG1
10 k
VS-
+
±
-2.5 V
+
±
VOUT
±
VOCM
2.5 V
THS4551 Wideband,
Fully Differential Amplifier
RF1
1k
Vin
CS
16.8 pF
FDA
+
RL
1k
±
+
PD
VS-
VS+
RF2
1k
RG2
10 k
CF2
8.4 pF
Copyright © 2016, Texas Instruments Incorporated
Figure 81. Compensated Attenuator Circuit Using the THS4551
The 16.8 pF across the inputs is really a total of 36 pF for a single-ended design from Equation 13 reduced by
half and then the 1.2-pF internal capacitance is removed.
These two designs (with and without the compensation capacitors) were both bench tested and simulated using
the THS4551 TINA-TI™ model, which resulted in Figure 82. The TINA-TI™ simulation files used for Figure 82
are available both without the compensation capacitors and with the capacitors in place.
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-12
-14
-16
Gain (dB)
-18
-20
-22
-24
-26
-28
Bench, wo Capacitors
Bench, with Capacitors
TINA-TI•, wo Capacitors
TINA-TI•, with Capacitors
-30
1M
10M
Frequency (Hz)
100M
D067
Figure 82. Attenuator Response Shapes With and Without External Capacitors
This approach does a good job of flattening the response for what starts out as a low phase margin attenuator
application. The simulation model does a very good job of predicting the peaking and showing the same
improvement with the external capacitors (both give a flat, approximately 24-MHz, closed-loop bandwidth for the
gain of 0.1-V/V design). The output noise starts to peak up (because of the noise gain shaping of the capacitors)
above 3.5 MHz in this example. These stages normally drive the RC filter at the input of a SAR ADC that filters
off the noise peaking above 3.5 MHz.
9.1.7 The Effect of Adding a Feedback Capacitor
Adding a feedback capacitor to band-limit the signal path is very common in lower frequency designs. This
approach is very effective for the signal path gain but does create the potential for high-frequency peaking and
oscillation for a wideband device such as the THS4551. The feedback capacitor by itself takes the noise gain to
1 V/V at high frequencies. Depending on the frequency where the noise gain goes to 1V/V, and what added
phase margin reduction may already be in place resulting from the load RC, the feedback capacitors can cause
instability.
Figure 83 shows the starting point for a typical band-limited design. At lower frequencies, this example delivers a
gain of 10 V/V with an intentional band limit in the feedback RC at 320 kHz. This single 5-V design targets a
midsupply output common-mode voltage with only a noise reduction capacitor on the VOCM input control.
CF1
250 pF
High-Gain, Single-Ended
to Differential Output Stage
with Feedback Pole
VS+
RF1
2k
VS+
RG1
200
VS-
+
±
0V
+
10
±
VOCM
5V
THS4551 Wideband,
Fully Differential Amplifier
VIN
10 nF
±
FDA
+
10 nF
±
+
SAR ADC
Input
PD
VS-
VS+
10
RF2
2k
RG2
200
CF2
250 pF
Copyright © 2016, Texas Instruments Incorporated
Figure 83. Single-Ended to Differential Stage with a Feedback Pole
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The response shape must be probed at the FDA output pins before the added RC pole to the SAR input.
Running a wideband sweep with the THS4551 TINA-TI™ model using the SBOC475 simulation file shows a
resonance at 50 MHz in Figure 84 resulting from the feedback capacitor.
200
Gain (dB)
Phase (deg) 175
Gain (dB)
15
10
150
5
125
0
100
-5
75
-10
50
-15
25
-20
0
-25
-25
-30
100k
Phase (deg)
20
-50
1M
10M
Frequency (Hz)
100M
D070
Figure 84. Gain and Phase Plot with a Feedback Pole
One approach to increasing the phase margin when there is a feedback capacitor is to include a differential input
capacitor. This approach increases the noise gain at higher frequencies, thus creating a lower-frequency loop
gain equal to a 0-dB crossover with more phase margin. Figure 85 shows a differential input capacitor equal to
the feedback capacitor in the test circuit. This approach increases the noise gain from 1 V/V at higher
frequencies (with only a feedback capacitor) to a noise gain of 3 V/V at higher frequencies.
CF1
250 pF
High-Gain, Single-Ended
to Differential Output Stage
with Feedback Pole
VS+
VS+
RG1
200
VS-
+
±
0V
+
±
10
±
VOCM
5V
THS4551 Wideband,
Fully Differential Amplifier
RF1
2k
VIN
250 pF
10 nF
FDA
+
10 nF
±
+
SAR ADC
Input
PD
VS-
VS+
10
RF2
2k
RG2
200
CF2
250 pF
Copyright © 2016, Texas Instruments Incorporated
Figure 85. Single-Ended to Differential Stage with a Feedback Pole and Differential Input Capacitor
Re-running the wideband response (using the SBOC474 TINA-TI™ simulation file) simulation illustrates in
Figure 86 that the resonance is greatly reduced with the higher noise gain at the loop gain equal to a 0-dB
crossover at a lower frequency. Although this example is only modestly peaking, good design practice is to
include a place for a differential input capacitor (even if not used) for any design using a feedback capacitor
across the feedback resistors. This recommendation applies to this simple example and to multiple feedback
active filter designs.
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Gain (dB)
20
10
200
Gain (dB)
Phase (deg) 160
0
120
-10
80
-20
40
-30
0
-40
-40
-50
-80
-60
100k
Phase (deg)
www.ti.com
-120
1M
10M
Frequency (Hz)
100M
D071
Figure 86. Gain and Phase Plot with a Differential Input Capacitor
9.2 Typical Applications
9.2.1 An MFB Filter Driving an ADC Application
One common application for the THS4551 is to take a single-ended, high VPP voltage swing (from a high-voltage
precision amplifier such as the OPA192) and deliver that swing to precision SAR ADC as a single-ended to
differential conversion with output common-mode control and implement an active 2nd-order multiple feedback
(MFB) filter design. Designing for a 40-VPP maximum input down to an 8-VPP differential swing requires a gain of
0.2 V/V. Targeting a 100-kHz Butterworth response with the RC elements tilted towards low noise gives the
example design of Figure 87. Note that the VCM control is set to half of a 4.096-V reference, which is typical for
5-V differential SAR applications.
OPA192 Output
VS+
5V
+
2.96 k
VS0V
±
RG1
1.57 k
910 pF
100 pF
VS+
+
±
10
20
VIN
±
1.5 nF
100 fF
VOCM
FDA
+
VS+
±
10
20
2.96 k
RG2
1.57 k
RF2
592
8-VPP Differential
SAR ADC Input
PD
VS-
+
2.2 nF
±
+
VOCM
2.048 V
THS4551 Wideband,
Fully Differential Amplifier
RF1
592
100 pF
910 pF
Copyright © 2016, Texas Instruments Incorporated
Figure 87. Example 100-kHz Butterworth Filter
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Typical Applications (continued)
9.2.1.1 Design Requirements
The requirements for this application are:
• Single-ended to differential conversion
• Attenuation by 0.2-V/V gain
• Active filter set to a Butterworth, 100-kHz response shape
• Output RC elements set by SAR input requirements (not part of the filter design)
• Filter element resistors and capacitors are set to limit added noise over the THS4551 and noise peaking
9.2.1.2 Detailed Design Procedure
The design proceeds using the techniques and tools suggested in the Design Methodology for MFB Filters in
ADC Interface Applications application note (SBOA114). The process includes:
• Scale the resistor values to not meaningfully contribute to the output noise produced by the THS4551 by itself
• Select the RC ratios to hit the filter targets when reducing the noise gain peaking within the filter design
• Set the output resistor to 10 Ω into a 2.2-nF differential capacitor
• Add 100-pF common-mode capacitors to the load capacitor to improve common noise filtering
• Inside the loop, add 20-Ω output resistors after the filter feedback capacitor to increase the isolation to the
load capacitor
• Include a place for a differential input capacitor (illustrated as 100 fF in Figure 87)
9.2.1.3 Application Curves
Probing the response to the output pins by using the THS4551 SBOC471 TINA-TI™ simulation model (before
the RC filter to the SAR ADC) illustrates the expected response plus some peaking at higher frequencies. Any
signal or noise peaking that appears at the output because of this peaking is rolled off by the RC filter between
the FDA and SAR inputs. A place for a differential input capacitor is illustrated in Figure 87 (as 0.1 pF) but is not
used for this simulation. This slight peaking is a combination of low phase margin and feedthrough via the
feedback capacitor to the increasing open-loop output impedance of Figure 68. The loop gain and phase
response are available as a TINA-TI™ simulation file.
Obtaining the SNR to the ADC input pins, and assuming an 8-VPP full scale (2.83 VRMS), gives the result of
Figure 89. The 113-dB SNR shown in Figure 89 does not limit the performance for any SAR application.
-40
150
-60
0
-80
-150
-100
-300
-120
-450
T
Phase (deg)
-20
450
Gain (dB)
Phase (deg) 300
Signal to Noise [dB]
Gain (dB)
0
150.00
140.00
130.00
120.00
-140
10k
100k
1M
Frequency (Hz)
10M
-600
100M
D072
110.00
10k
Figure 88. Gain and Phase Plot for a 100-kHz Butterworth
Filter
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100k
1MEG
10MEG
Frequency (Hz)
100MEG
Figure 89. Signal-to-Noise Ratio Plot
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Typical Applications (continued)
9.2.2 Differential Transimpedance Output to a High-Grade Audio PCM DAC Application
The highest-grade audio digital-to-analog converters (DACs) are a differential current-mode output. These
devices normally suggest a two-amplifier transimpedance stage to hold the DAC output voltages fixed when the
amplifiers produce a differential voltage swing at the outputs. Often, the differential voltage swing is then
converted to single-ended in a differencing amplifier stage to drive headphone loads (see Figure 35 in the
OPA1611). The emerging high-power class D audio amplifiers often require differential inputs. Applying the
THS4551 as a differential transimpedance stage offers a simple solution for very low-distortion, differential-output
audio channels.
Starting with the output specifications for a very high-performance PCM1792A audio DAC, the requirements for
the THS4551 interface can be extracted. The DAC is a current-sourcing device that requires its outputs to be
held at ground when using a transimpedance amplifier. Using the DAC 3.3-V supply and the LM27762 low-noise,
low-dropout (LDO) regulator and inverter provides a ±2.5-V supply to the THS4551. Operating the THS4551 on
±2.5-V supplies places all nodes in range for an input VCM equal to GND (and the DAC output voltages as well)
and an FDA output VOCM also equal to GND.
The center current in Table 6 is a fixed 6.2-mA dc current coming out of the DAC outputs regardless of the DAC
code. This dc common-mode current can be absorbed by the –2.5-V supply at the input pins to hold the DAC
compliance voltage and FDA input pins at ground. The FDA controls the output common-mode voltage, set to
ground in this case, whereas the input pin voltage (which does not move with the DAC output differential current)
is controlled with a resistor to the negative supply.
Table 6. PCM1792A Analog Output Specification
ANALOG OUTPUT
TEST CONDITION
Gain error
Gain mismatch, channel-to-channel
Bipolar zero (BPZ) error
At BPZ
Output current
Full-scale (0 dB)
Center current
At BPZ
MIN
TYP
MAX
–6
±2
6
% of FSR
UNIT
–3
±0.5
3
% of FSR
–2
±0.5
2
% of FSR
7.8
mAPP
–6.2
mA
This bias is provided by the 402-Ω resistors to –2.5 V, as illustrated in Figure 90. This design takes the
differential 7.8 mAPP from the DAC and produces a ±1.46-V swing on each output of the THS4551. This
configuration gives a full-scale differential 5.85 VPP available on the ±2.5-V supply design centered at ground at
both the inputs and outputs. Although the LM27762 provides a very-low noise, –2.5-V supply, using 0.1%
resistors in the current sink path to the –2.5-V supply as well as the feedback resistors limits any common-mode
noise on the –2.5-V supply to differential mode conversion at the FDA outputs.
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CF1
2.4 nF
VS+
4.4
VS-
THS4551 Wideband,
Fully Differential Amplifier
VS2.5 V
+
±
-2.5 V
+
LM27762
Supplies
RF1
750
6.2 mA
±
402
VS+
±
VOCM
PCM1792A
Complementary Current
Output DAC
FDA
±
+
20 nF
PD
VS-
6.2 mA
+
VS+
RF2
750
402
VS-
CF2
2.4 nF
4.4
Copyright © 2016, Texas Instruments Incorporated
Figure 90. PCM1792A DAC Output Driver
9.2.2.1 Design Requirements
To implement a differential transimpedance output interface to the PCM1792A DAC, the following requirements
must be met:
• The center current of the DAC must be considered to hold the DAC output voltage at ground. Using an FDA
controls the output side common-mode voltage, but the input common-mode voltage must also be controlled
to ground.
• A direct means of sinking the center current is to add a pulldown resistor at the DAC outputs to a negative
supply. Generating a ±2.5-V supply for this current sink requirement and the THS4551 is accomplished with
the LM27762.
• The transimpedance gain can be set using the feedback resistors of the THS4551 FDA. These resistors are
very flexible, but when set, the bandwidth in this stage is set to 88 kHz using a feedback capacitor in parallel
with the resistive gain element.
• When the feedback capacitor is set, a differential input capacitor is added to increase the high-frequency
noise gain for the overall loop gain stability.
• These frequency response control capacitors interact with the inductive open-loop output impedance to form
a high-frequency resonance. Adding a small series resistor to the feedback capacitor paths reduces this
effect.
9.2.2.2 Detailed Design Procedure
Proceed with this design using the techniques described in the Design for Wideband Differential Transimpedance
DAC Output application note (SBAA150):
• Generate the bipolar balanced supplies using the LM27662.
• Set the THS4551 output common-mode voltage at midsupply by grounding the VOCM pin.
• Control the input pin operating voltage by sinking the center current out of the DAC to the –2.5-V supply with
precision 402-Ω resistors.
• Set the gain for the complementary current output signal from the DAC by selecting the feedback resistor to
be 750 Ω. Set this resistor to keep the resulting output swing to be less than the available 9-VPP differential
swing.
• Control the bandwidth in this differential transimpedance stage to 88 kHz using the 2.4-nF feedback capacitor
on each side.
• Increase the high-frequency noise gain to 17.7 V/V by adding a differential input capacitor of 20 nF.
• Isolate these feedback capacitors with a series 4.4-Ω resistor in series with the feedback capacitors.
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9.2.2.3 Application Curves
The bandwidth is controlled to 88 kHz by using the 2.4-nF feedback capacitors. Amplifier stability is controlled by
the 20-nF differential capacitor across the DAC outputs. The added 4.4 Ω in series with the feedback 2.2-nF
capacitor isolates this capacitance from the inductive open-loop output impedance. To observe the effect of
adding these small resistors in series with the feedback capacitors, use the TINA-TI™ loop gain simulation
circuit. Include the DAC source capacitance in any final design analysis. Running the frequency response for this
circuit (available as a TINA-TI™ simulation file) provides this result. The 63.5-dBΩ gain is the 1.5-kΩ
transimpedance gain provided in this design.
65
240
Gain (dB)
Phase (deg) 200
60
160
55
120
50
80
45
40
40
0
35
-40
30
-80
25
-120
20
1k
10k
100k
1M
Frequency (Hz)
Phase (deg)
Gain (dB)
70
-160
100M
10M
D074
Figure 91. Gain and Phase Plot of DAC Output Driver
Running a full-scale sine wave at 1 kHz with ±1.95 mA on each output from the DAC at 180° out of phase, and
probing each THS4551 output pin separately results in the expected ±1.46 V on each output pin, as shown in
Figure 92. More output swing is available for the RRO device using the ±2.5-V supplies provided by the
LM27762 by simply increasing the feedback resistor values.
2.5
VOUTVOUT+
2
Output Voltage (V)
1.5
1
0.5
0
-0.5
-1
-1.5
-2
-2.5
0
0.2
0.4
0.6
0.8
1
1.2
Time (ms)
1.4
1.6
1.8
2
D075
Figure 92. Output Waveform of the DAC Output Driver
Although this example is on the audio signal generation side, the THS4551 can also be used to convert a singleended line input to a differential driver into an audio ADC.
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9.2.3 ADC3k Driver with a 2nd-Order RLC Interstage Filter Application
The THS4551 is well suited to low-power, dc-coupled requirements driving low-power pipeline ADCs (such as
the ADC3241 25-MSPS, 14-bit, dual device). Figure 93 shows an example design taking a bipolar input to a
–1-dBFS swing at the ADC input of 1.8 VPP. In this case, a 50-Ω source and input matching is assumed with a
gain of 5 V/V to the output pins with a 2nd-order interstage filter adding a –1-dB insertion loss. Full-scale voltage
at the input of RT and RG1 is then ±0.2 V. The 0.95-V output common-mode voltage is provided by the ADC. The
output filter provides a noise-power bandwidth limit with a low overshoot step response with no common-mode
level shift from the 0.95-V voltage provided by the ADC.
3.3 V
VS+
VS-
+
+
0V
±
ADC Output
Common-Mode
Voltage
50- Input Match,
Gain of 5 V/V from RT,
Single-Ended Source to
Differential Output
RF1
1k
50- Source
Impedance
±
10-MHz,
Second-Order
Bessel Filter
VS+
RG1
187
RS1
50
40.2
390 nH
5.6
VOUT
±
VOCM
VIN
950 mV
THS4551 Wideband,
Fully Differential Amplifier
RT1
59
VOCM
+
FDA
+
+
±
RG2
215
360 pF
±
ADC3241
Inputs
732
PD
VS-
VS+
40.2
390 nH
5.6
RF2
1k
Copyright © 2016, Texas Instruments Incorporated
Figure 93. ADC3k Driver with a 2nd-Order RLC Interstage Filter
9.2.3.1 Design Requirements
For this design example, the requirements include:
• Provide a wideband, 50-Ω input impedance match for a single-ended source centered on ground.
• From the input termination, provide a gain of 5 V/V to the FDA output pins as a differential signal.
• Set the output common-mode operating point using the ADC common-mode output voltage as the VOCM
input to the THS4551 FDA.
• Implement a low-overshoot, noise-band-limiting filter between the FDA and the ADC. Use only differential
shunt elements in the filter to pass the FDA output common-mode voltage to the ADC with no level shifting.
• Design the filter as a –1-dB insertion loss filter with a low series resistor to limit the common-mode level shift
resulting from the ADC input sample-rate-dependent common-mode current.
9.2.3.2 Detailed Design Procedure
The design proceeds as follows:
• Select the feedback resistor to be 1 kΩ and use the values from Table 2 at a gain of 5 V/V to implement a
50-Ω input match with a gain of 5 V/V.
• Use a 3.3-V power supply and apply the ADC output common-mode voltage to the VOCM input pin of the
THS4551.
• Design a –1-dB insertion loss, 2nd-order RLC filter using the approach described in the RLC Filter Design for
ADC Interface Applications application note (SBAA108).
• Adjust the total resistive load target in the filter design to hit the standard value for the filter inductors.
• Convert the filter design to differential with only differential shunt elements. These elements must not be split
and connected to a center-point ground. This technique passes the output common-mode voltage from the
FDA to the ADC with no level shift error.
• Add a small series resistor at the ADC inputs. This resistor is not part of the filter design but spreads out the
sampling glitch energy to provide improved SFDR.
• Check the common-mode level shift from the FDA outputs to the ADC resulting from the clock-rate-dependent
common-mode current. This common-mode current into the ADC shifts the common-mode voltage slightly,
but can easily stay in range with a low series resistor in the filter design.
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9.2.3.3 Application Curve
Driving a 2-MHz ±0.2-V square wave into this circuit (using a TINA-TI™ simulation file for the circuit of Figure 93)
gives the response shown in Figure 94 at the ADC. The red trace is a –1-dBFS, 1.8-VPP square wave at the ADC
input pins. The gray trace is the input signal at the RT termination resistor. The black trace is the common-mode
voltage at the FDA input pins. Note that the input pin voltage swing stays above ground and in range for this
bipolar input, single, 3.3-V supply design.
1.5
1
Voltage (V)
0.5
0
-0.5
VIN+
VOUT
VIN
-1
-1.5
0
100
200
300
400 500 600
Time (ns)
700
800
900 1000
D076
Figure 94. Time-Domain Waveform
Unbuffered pipeline ADCs draw a clock-rate-dependent input common-mode current. For the ADC3241, this
input current is specified as 1.5 µA per MSPS. Operating at 25 MSPS, the common-mode current drops the
common-mode voltage from 0.95 V at the THS4551 outputs by 37.5 µA × 45.8 Ω = 1.7 mV to 0.9483 V. This
value is well within the allowed ±25-mV common-mode deviation from the ADC VCM output. Consider this effect
carefully when using higher resistor values in the interface at the ADC.
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10 Power Supply Recommendations
The THS4551 is principally intended to operate with a nominal single-supply voltage of 3 V to 5 V. Supply
voltage tolerances are supported with the specified operating range of 2.7 V (10% low on a 3-V nominal supply)
and 5.4 V (8% high on a 5-V nominal supply). Supply decoupling is required, as described in the Terminology
and Application Assumptions section. Split (or bipolar) supplies can be used with the THS4551, as long as the
total value across the device remains less than 5.5 V (absolute maximum). The thermal pad on the RGT
package is electrically isolated form the die; connect the thermal pad (RGT package only) to any power or
ground plane for reduced thermal impedance to the junction temperature. This pad must be connected to some
power or ground plane and not floated.
For the best input offset voltage drift, the THS4551 uses a proportional to absolute temperature (PTAT)
quiescent current biasing scheme. This approach gives a positive over temperature variation in supply current.
Figure 95 shows the 5-V supply current over a wide TJ range for a number of tested units. The Electrical
Characteristics tables report the typical and range on this supply current temperature coefficient for both 5-V and
3-V supply operation.
1.8
Supply Current (mA)
1.7
1.6
1.5
1.4
1.3
1.2
1.1
1
-45 -30 -15
0
15 30 45 60 75 90 105 120 135
Junction Temperature (qC)
D068
Figure 95. Linear Temperature Coefficient for Supply Current
Using a negative supply to deliver a true swing to ground output when driving SAR ADCs can be desired.
Although the THS4551 quotes a rail-to-rail output, linear operation requires approximately 200-mV headroom to
the supply rails. One easy option for extending the linear output swing to ground is to provide the small negative
supply voltage required using the LM7705 fixed –230-mV, negative-supply generator. This low-cost, fixed,
negative-supply generator can accept the 3-V to 5-V positive supply input used by the THS4551 and provides a
fixed –230-mV supply for the negative power supply. Using the LM7705 provides an effective solution, as
discussed in the Extending Rail-to-Rail Output Range for Fully Differential Amplifiers to Include True Zero Volts
TI design (TIDU187)
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10.1 Thermal Analysis
The very low internal quiescent power dissipation for the THS4551, combined with the excellent thermal
impedance of the 16-pin VQFN package (RGT), limits the possibility of excessively high internal junction
temperatures. A more detailed analysis may be warranted because the 10-pin WQFN (RUN) package has a
much higher junction-to-ambient thermal impedance (θJA = 163°C/W).
To estimate the internal TJ, an estimate of the maximum internal power dissipation is first required. There are two
pieces to the internal power dissipation: quiescent current power and the power used in the output stage to
deliver load current. To simplify the latter, the worst-case output stage power drives a dc differential voltage
across a load using half the total supply voltage. Also assume a maximum ambient temperature of 125°C, giving
the maximum quiescent current as shown in Figure 95. As an example:
• Assume a maximum operating supply voltage of 5.4 V. This 5.4-V supply with a maximum ICC of 1.75 mA
gives a quiescent power term of 9.45 mW.
• Assume a 200-Ω differential load with a static 2.7-V differential voltage established across the load. The
1.35 mA of dc load current generates a maximum output stage power of (5.4 V – 2.7 V) × 1.35 mA =
3.65 mW.
• From the worst-case total internal PD of 13.1 mW, multiplying the internal PD with a 163°C/W thermal
impedance times the 163°C/W thermal impedance for the very small 10-pin WQFN package results in a 2.1°C
rise from ambient.
Even for this extreme condition and the maximum-rated ambient of 125°C, the junction temperature is a
maximum of 127°C, which is less than the rated absolute maximum of 150°C. Follow this same calculation
sequence for the exact application and package selected to predict the maximum TJ.
11 Layout
11.1 Layout Guidelines
11.1.1 Board Layout Recommendations
Similar to all high-speed devices, best system performance is achieved with close attention to board layout. The
THS4551DGKEVM user guide (SLOU447) shows a good example of high-frequency layout techniques as a
reference. This EVM includes numerous extra elements and features for characterization purposes that may not
apply to some applications. General high-speed signal path layout suggestions include:
• Continuous ground planes are preferred for signal routing with matched impedance traces for longer runs;
however, both ground and power planes must be opened up around the capacitive sensitive input and output
device pins. When the signal goes to a resistor, parasitic capacitance becomes more of a band-limiting issue
and less of a stability issue.
• Good high-frequency decoupling capacitors (0.1 µF) are required to a ground plane at the device power pins.
Additional higher-value capacitors (2.2 µF) are also required but can be placed further from the device power
pins and shared among devices. For best high-frequency decoupling, consider X2Y supply decoupling
capacitors that offer a much higher self-resonance frequency over standard capacitors.
• Differential signal routing over any appreciable distance must use microstrip layout techniques with matched
impedance traces.
• Higher-speed FDAs such as the THS4551 include a duplicate of the output pins on the input feedback side of
the larger 16-pin VQFN (RGT) package. This feature is intended to allow the external feedback resistors to be
connected with virtually no trace length on the input side of the package. This internal feedback trace also
provides a second feedback path for connecting a feedback capacitor on the input pin sides for band-limited
or multiple feedback filter designs. This internal trace shows an approximate 3.3-Ω series resistance that must
be considered in any design using that path. The TINA-TI™ model does not include that element (to be
generally applicable to all package styles) and must be added externally if the RGT package is used. Use this
layout approach without extra trace length on the critical feedback path. The smaller 10-pin WQFN package
lines up the outputs and the required inputs on the same side of the package where the feedback RF resistors
must be placed immediately adjacent to the package with minimal trace length.
• The input summing junctions are very sensitive to parasitic capacitance. Any RG elements must connect into
the summing junction with minimal trace length to the device pin side of the resistor. The other side of the RG
elements can have more trace length if needed to the source or to GND.
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11.2 Layout Example
Figure 96. Example Layout
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11.3 EVM Board
Figure 97 and Figure 98 show the layout of the top and bottom layers of the THS4551DGKEVM evaluation
module, respectively.
Figure 97. THS4551DGKEVM Top Layer
Figure 98. THS4551DGKEVM Bottom Layer
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12 Device and Documentation Support
12.1 Device Support
12.1.1 TINA-TI™ Simulation Model Features
The device model is available on the product folder under www.ti.com in a typical application circuit file. The
model includes numerous features intended to speed designer progress over a wide range of application
requirements. The following list shows the performance parameters included in the model:
• For the small-signal response shape with any external circuit:
– Differential open-loop gain and phase
– Parasitic input capacitance
– Open-loop differential output impedance
• For noise simulations:
– Input differential spot voltage noise and a 100-Hz 1/f corner
– Input current noise on each input with a 6-kHz 1/f corner
• For time-domain, step-response simulations:
– Differential slew rate
– I/O headroom models to predict clipping
– Input stage diodes to predict overdrive limiting
• Fine-scale, dc precision terms:
– PSRR
– CMRR
The Typical Characteristics: 3-V to 5-V Supply Range section provides more detail than the macromodels can
provide; some of the unmodeled features include:
• Harmonic distortion
• Temperature drift in dc error terms (VIO and IOS)
• Overdrive recovery time
• Turn-on and turn-off times using the power-down feature
Some unique simulation considerations come with the THS4551 TINA-TI™ model. This device (and model)
include 0.6-pF internal feedback capacitors. These capacitors are intended to improve phase margin when using
higher external feedback resistor values. Higher feedback resistors generate an in-band pole in the feedback
signal with the differential input capacitance, and the internal 0.6 pF capacitors add a zero to the feedback
response shape to shape the noise gain flat at the loop-gain crossover.
In order to generate an accurate open-loop gain and phase simulation, these components must be removed
because they are feedback elements, not forward path elements. Figure 99 illustrates a typical AOL gain and
phase simulation (available as a TINA-TI™ software file) where external –0.6-pF capacitors cancel out the
internal capacitors in the model (TINA-TI™ supports negative value elements). The inductors inside the loop
close the loop for the dc operating point and open the loop immediately for an ac sweep. The input-coupling
capacitors are open at dc, then couple in the differential input immediately on an ac sweep. The somewhat odd
values help reduce numerical chatter in the simulation. When using the internal feedback traces from the outputs
to the inputs on the RGT package, be sure to add the 3.3-Ω trace impedance to any simulation. This impedance
is not included in the core model.
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Device Support (continued)
VS±
V1 2.5
V2 -2.5
C2 -600 fF
VS+
V4 0
L1 999 H
VS+
VOCM
R1 10 m:
C3 9.97 kF
V
VM1
VOCM
VIN+
VCVS1 1
+
-
-
+
C4 9.97 kF
VG1
R5
10 M:
+
VOCM
-
VS±
+
R4
10 M:
L2 999 H
+
-
+
-
VS+
VIN+
VIN-
R3 100 k:
U1 THS4551
+
C1 -600 fF
VINR2 10 m:
Copyright © 2016, Texas Instruments Incorporated
Figure 99. Open-Loop Gain and Phase TINA-TI™ Simulation Setup
This test is set up with a very light load to isolate the no load AOL curve. Adding a load brings in the open-loop
ZOL response to the overall response of the output pins. Running this simulation gives the gain and phase of
Figure 100 that closely matches the plot of Figure 37.
Gain (dB)
T 200.00
100.00
0.00
-100.00
Phase [deg]
0.00
-100.00
-200.00
-300.00
-400.00
10.00
100.00
1.00k
10.00k
100.00k
1.00MEG
10.00MEG
100.00MEG 1.00G
Frequency (Hz)
Figure 100. Open-Loop Gain and Phase Simulation Result
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12.2 Documentation Support
12.2.1 Related Documentation
For related documentation see the following:
• THS4551 TINA-TI™ model
• THS4551DGKEVM User Guide (SLOU447)
• THS452x Very Low Power, Negative Rail Input, Rail-To-Rail Output, Fully Differential Amplifier (SBOS458)
• LMH6629 Ultra-Low Noise, High-Speed Operational Amplifier with Shutdown (SNOSB18)
• OPA847 Wideband, Ultra-Low Noise, Voltage-Feedback Operational Amplifier with Shutdown (SBOS251)
• INA188 Precision, Zero-Drift, Rail-to-Rail Out, High-Voltage Instrumentation Amplifier (SBOS632)
• OPAx192 36-V, Precision, Rail-to-Rail Input/Output, Low Offset Voltage, Low Input Bias Current Op Amp with
e-trim™ (SBOS620)
• OPA161x SoundPlus™ High-Performance, Bipolar-Input Audio Operational Amplifiers (SBOS450)
• ADC322x Dual-Channel, 12-Bit, 25-MSPS to 125-MSPS, Analog-to-Digital Converters (SBAS672)
• ADC324x Dual-Channel, 14-Bit, 25-MSPS to 125-MSPS, Analog-to-Digital Converters (SBAS671)
• ADS127L01 24-Bit, High-Speed, Wide-Bandwidth Analog-to-Digital Converter (SBAS607)
• ADS127L01EVM User's Guide (SBAU261)
• ADS9110 18-Bit, 2-MSPS, 15-mW, SAR ADC with multiSPI™ Interface (SBAS629)
• REF6025EVM-PDK User's Guide (SBAU258)
• 24-Bit, 192-kHz Sampling, Advanced Segment, Audio Stereo Digital-to-Analog Converter (SLES105)
• LM27762 Low-Noise Regulated Switched-Capacitor Voltage Inverter (SNVSAF7)
• LM7705 Low-Noise Negative Bias Generator (SNVS420)
• Extending Rail-to-Rail Output Range for Fully Differential Amplifiers to Include True Zero Volts (TIDU187)
• Design Methodology for MFB Filters in ADC Interface Applications (SBOA114)
• Design for Wideband Differential Transimpedance DAC Output (SBAA150)
• RLC Filter Design for ADC Interface Applications (SBAA108)
• TINA-TI Open Loop No Load Response (SBOC476)
• TINA-TI Basic Gain of 1 Test Circuit (SBOC466)
• TINA-TI ADTL1-4-75 Model Test (SBOC463)
• TINA-TI Common Mode Test CKT (SBOC467)
• TINA-TI AC Coupled Single to Differentiate Gain of 2 (SBOC460)
• TINA-TI Single to Differential Attenuator (SBOC477)
• TINA-TI Gain of 5 Single to Different Simplified (SBOC472)
• TINA-TI AC coupled different IO (SLOC341)
• TINA-TI Differential IO with OPA2192 to FDA to SAR (SBOC469)
• TINA-TI ADS127L01 MFB Driver (SBOC462)
• TINA-TI ADS127L01 MFB Driver LG Test (SBOC461)
• TINA-TI Attenuator With No Caps Gain of 0.1 (SBOC465)
• TINA-TI Attenuator With a Caps Gain of 0.1 (SBOC464)
• TINA-TI High Gain Single to Different with Feedback Pole (SBOC475)
• TINA-TI High Gain Single to Different with Feedback Pole and Input C (SBOC474)
• TINA-TI Gain of 0.2 100kHz Butterworth MFB Filter (SBOC471)
• TINA-TI 100kHz MFB filter LG test (SBOC459)
• TINA-TI Differential Transimpedance LG Sim (SBOC470)
• TINA-TI Differential Audio DAC ZT Design (SBOC468)
• TINA-TI Gain of 5 Single to Different with 10Mhz Bessel (SBOC473)
62
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Copyright © 2016–2017, Texas Instruments Incorporated
Product Folder Links: THS4551
THS4551
www.ti.com
SBOS778C – APRIL 2016 – REVISED JULY 2017
12.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
12.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.5 Trademarks
TINA-TI, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
12.6 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
12.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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Copyright © 2016–2017, Texas Instruments Incorporated
Product Folder Links: THS4551
63
PACKAGE OPTION ADDENDUM
www.ti.com
6-Feb-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
THS4551IDGKR
ACTIVE
VSSOP
DGK
8
2500
Green (RoHS
& no Sb/Br)
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
4551
THS4551IDGKT
ACTIVE
VSSOP
DGK
8
250
Green (RoHS
& no Sb/Br)
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
4551
THS4551IRGTR
ACTIVE
VQFN
RGT
16
3000
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
HS4551
THS4551IRGTT
ACTIVE
VQFN
RGT
16
250
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
HS4551
THS4551IRUNR
ACTIVE
QFN
RUN
10
3000
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
4551
THS4551IRUNT
ACTIVE
QFN
RUN
10
250
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
4551
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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