THS7001IPWP


             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 D Separate Low Noise Preamp and PGA D D D PGA Features Stages Shutdown Control Preamp Features − Low Voltage Noise . . . 1.7 nV/√Hz − Accessible Output Pin for External Filtering − Voltage Feedback, Gmin = −1, 2 − 100 MHz Bandwidth (−3 dB) D D − Digitally Programmable Gain − −22 dB to 20 dB Gain/Attenuation Range − 6 dB Step Resolution − Output Clamp Protection − 70 MHz Bandwidth (−3 dB) − 175 V/µs Slew Rate Wide Supply Range ±4.5 V to ±16 V PowerPAD Package for Enhanced Thermal Performance description The THS7001 (single) and THS7002 (dual) are high-speed programmable-gain amplifiers, ideal for applications where load impedance can often vary. Each channel on this device consists of a separate low-noise input preamp and a programmable gain amplifier (PGA). The preamp is a voltage-feedback amplifier offering a low 1.7-nV/√Hz voltage noise with a 100-MHz (−3 dB) bandwidth. The output pin of the preamp is accessible so that filters can be easily added to the amplifier. The 3-bit digitally-controlled PGA provides a −22-dB to 20-dB attenuation/gain range with a 6-dB step resolution. In addition, the PGA provides both high and low output clamp protection to prevent the output signal from swinging outside the common-mode input range of an analog-to-digital converter. The PGA provides a wide 70-MHz (−3 dB) bandwidth, which remains relatively constant over the entire gain/attenuation range. Independent shutdown control is also provided for power conservation and multiplexing. These devices operate over a wide ±4.5-V to ±16-V supply voltage range. PREAMP OUT PGA IN− G0 G1 G2 PREAMP VCC+ PREAMP IN− CLAMP+ (VH) _ _ Preamp PREAMP IN+ PGA OUT + + PREAMP VCC− CLAMP− (VL) SHDN PGA REF PGA VCC+ PGA VCC− GND Figure 1. THS7001 Block Diagram CAUTION: The THS7001 and THS7002 provides ESD protection circuitry. However, permanent damage can still occur if this device is subjected to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance degradation or loss of functionality. Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments Incorporated. Copyright  1999, Texas Instruments Incorporated  
 
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             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 THS7001 PWP PACKAGE (TOP VIEW) 1 2 3 4 5 6 7 8 9 10 GND VREFPGA −VINPGA VOUTPre-AMP −VINPre-Amp +VINPre-Amp VCC−Pre-Amp VCC+Pre-Amp Spare/NC Spare/NC 20 19 18 17 16 15 14 13 12 11 G0 G1 G2 SHDN VOUTPGA VLNegative Clamp VCC−PGA VCC+PGA VHPositive Clamp Spare/NC THS7002 PWP PACKAGE (TOP VIEW) 1 2 3 4 5 6 7 8 9 10 11 12 13 14 GND-A PGA-A REF PGA-A IN− PREAMP OUT A PREAMP-A IN− PREAMP-A IN+ PREAMP VCC− PREAMP VCC+ PREAMP-B IN+ PREAMP-B IN− PREAMP OUT B PGA-B IN− PGA-B REF GND-B 28 27 26 25 24 23 22 21 20 19 18 17 16 15 G0-A G1-A G2-A SHDN−A PGA-A OUT CLAMP− (VL) PGA VCC− PGA VCC+ CLAMP+ (VH) PGA-B OUT SHDN−B G2-B G1-B G0-B AVAILABLE OPTIONS PACKAGED DEVICES TA 0°C to 70°C −40°C to 85°C 2 NUMBER OF CHANNELS PowerPAD PLASTIC TSSOP (PWP) EVALUATION MODULE 1 THS7001CPWP THS7001EVM 2 THS7002CPWP THS7002EVM 1 THS7001IPWP — 2 THS7002IPWP — POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 block diagram PREAMP OUT A PGA−A REF PGA−A IN− G0A G1A G2A PGA−A GND PREAMP VCC+ PREAMP A IN− CLAMP+ (VH) _ _ Preamp PREAMP A IN+ PGA−A OUT + + SHDN−A SHDN−B PREAMP B IN+ PREAMP B IN− + + Preamp _ PGA−B OUT _ CLAMP− (VL) PREAMP VCC− PREAMP OUT B PGA−B REF PGA−B IN− G0B G1B G2B PGA PGA PGA−B VCC+ VCC− GND Figure 2. THS7002 Dual Channel PGA input preamp To achieve the minimum input equivalent noise required for very small input signals, the input preamp is configured as a classic voltage feedback amplifier with a minimum gain of 2 or −1. The output of the preamp is accessible, allowing for adjustment of gain using external resistors and for external filtering between the preamp and the PGA. programmable gain amplifier (PGA) The PGA is an inverting, programmable gain amplifier. The gain is digitally programmable using three control bits (TTL-compatible terminals) that are encoded to provide eight distinct levels of gain/attenuation. Nominal gain/attenuation is shown in Table 1. Table 1. Nominal Gain/Attenuation G2 G1 G0 PGA GAIN (dB) PGA GAIN (V/V) 0 0 0 −22 0.08 0 0 1 −16 0.16 0 1 0 −10 0.32 0 1 1 −4 0.63 1 0 0 2 1.26 1 0 1 8 2.52 1 1 0 14 5.01 1 1 1 20 10.0 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 3 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 output clamping Output clamping for both upper (VH) and lower (VL) levels for the PGAs is provided. There is only one terminal for the positive output clamp and one for the negative output clamp for both channels. shutdown control The SHDN terminals allow for powering down the internal circuitry for power conservation or for multiplexing. Separate shutdown controls are available for each channel. The control levels are TTL compatible. absolute maximum ratings over operating free-air temperature (see Notes 1 and 2)† Supply voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±16.5 V Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±VCC Output current, IO (preamp) (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150 mA IO (PGA) (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 85 mA Differential input voltage, VID . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±4 V Total continuous power dissipation at (or below) TA = 25°C (see Note 2): THS7001 . . . . . . . . . . . . . . 3.83 W THS7002 . . . . . . . . . . . . . . 4.48 W Maximum junction temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C Operating free-air temperature, TA:C-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C I-suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 85°C Storage temperature, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 125°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C † Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. NOTES: 1. The THS7001 and THS7002 incorporates a PowerPAD on the underside of the chip. The PowerPAD acts as a heatsink and must be connected to a thermal dissipation plane for proper power dissipation. Failure to do so can result in exceeding the maximum junction temperature, which could permanently damage the device. See the Thermal Information section of this document for more information about PowerPAD technology. 2. For operation above TA = 25°C, derate the THS7001 linearly to 2 W at the rate of 30.6 mW/°C and derate the THS7002 linearly to 2.33 W at the rate of 35.9 mW/°C. recommended operating conditions MIN Preamp supply voltage, VCC+ and VCC− Split supply PGA supply voltage, VCC+ and VCC− Split supply C-suffix Operating free-air temperature, TA I-suffix ‡ PGA minimum supply voltage must be less than or equal to preamp supply voltage. 4 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 MAX UNIT ±4.5 ±4.5‡ NOM ±16 V ±16 V 0 70 °C −40 85 °C 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 preamp electrical characteristics, G = 2, TA = 25°C, RL = 150 Ω, (unless otherwise noted) TEST CONDITIONS† PARAMETER VCC Supply voltage operating range Maximum output voltage swing RL = 150 Ω RL = 250 Ω VIO VCC = ±5 V or ±15 V Input offset voltage VCC = ±5 V VCC = ±15 V ±3.6 ±3.8 ±13 ±13.6 VCC = ±5 V VCC = ±15 V ±3.5 ±3.7 ±11 ±12.6 TA = 25°C TA = full range 1 Common-mode input voltage range VCC = ±5 V VCC = ±15 V IO Output current (see Note 3) RL = 20 Ω IOC Short-circuit output current (see Note 3) VCC = ±15 V VCC = ±5 V VCC = ±15 V ±3.8 ±4.2 ±13.8 ±14 40 70 60 95 IIB Input bias current VCC = ±5 V or ±15 V IIO Input offset current VCC = ±5 V or ±15 V TA = 25°C TA = full range VCC = ±5 V, VIC = ±2.5 V TA = 25°C TA = full range 80 VCC = ±15 V, VIC = ±12 V TA = 25°C TA = full range 80 VCC = ±5 V or ±15 V TA = 25°C TA = full range 85 RI Input resistance CI Input capacitance RO Output resistance ICC V V 5 2.5 30 V mA mA 6 400 nA nA/°C 89 78 dB 88 78 100 dB 80 1 MΩ 1.5 pF 13 Ω VCC = ±5 V TA = 25°C TA = full range 5.5 VCC = ±15 V TA = 25°C TA = full range 7 Quiescent current (per channel) µA A 175 0.3 Open loop mV µV/°C 8 Input offset current drift Power supply rejection ratio ±16.5 120 TA = 25°C TA = full range PSRR UNIT 10 VICR Common-mode rejection ratio MAX 7 Input offset voltage drift CMRR TYP ±4.5 Split supply RL = 1 kΩ VOM MIN 7 8 8 mA 9 † Full range for the THS7001/02C is 0°C to 70°C. Full range for the THS7001/022I is − 40°C to 85°C. NOTE 3: A heatsink may be required to keep the junction temperature below absolute maximum when an output is heavily loaded or shorted. (See absolute maximum ratings and thermal information section.) POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 5 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 preamp operating characteristics, G = 2, TA = 25°C, RL = 150 Ω, (unless otherwise noted) TEST CONDITIONS† PARAMETER SR Slew rate (see Note 4) G = −1 Settling time to 0.1% G = −1, 5 V Step ts Settling time to 0.01% MIN TYP VO = ±2 V, VCC = ±5 V 65 VO = ±10 V, VCC = ±15 V 85 VCC = ±5 V VCC = ±15 V 85 VCC = ±5 V VCC = ±15 V 95 70 ns 90 VCC = ±15 V, VO(PP) = 2V fc = 1 MHz, RL = 250 Ω −88 dBc VCC = ±5 V or ±15 V, VCC = ±5 V or ±15 V, f = 10 kHz 1.7 nV/√Hz f = 10 kHz 0.9 pA/√Hz 100 Total harmonic distortion Vn In Input noise voltage BW Small-signal bandwidth (−3 dB) VO(PP) = 0.4V, G=2 VCC = ±5 V VCC = ±15 V VO(PP) = 0.4V, G=2 VCC = ±5 V VCC = ±15 V 35 Bandwidth for 0.1 dB flatness VCC = ±5 V, VCC = ±15 V, VO = 5 VO(PP) VO = 20 VO(PP) 4.1 Full power bandwidth (see Note 5) Differential gain error G = 2, 100 IRE, NTSC VCC = ±5 V VCC = ±15 V 0.02% AD φD G = 2, NTSC VCC = ±5 V VCC = ±15 V 0.01° Differential phase error Open loop gain UNIT V/ s V/µs THD Input noise current MAX 100 IRE, 85 MHz MHz 45 MHz 1.4 0.02% 0.01° VCC = ±5 V, VO = ±2.5 V, RL = 1 kΩ TA = 25°C 85 TA = full range 83 VCC = ±15 V, VO = ±10 V, RL = 1 kΩ TA = 25°C TA = full range 86 89 dB 91 84 Channel-to-channel crosstalk (THS7002) VCC = ±5 V or ±15 V, f = 1 MHz † Full range for the THS7001/02C is 0°C to 70°C. Full range for the THS7001/02I is − 40°C to 85°C. NOTES: 4. Slew rate is measured from an output level range of 25% to 75%. 5. Full power bandwidth = slew rate/2π V(PP). −85 dB shutdown electrical characteristics PARAMETER ICC(standby) Standby current, disabled (per channel) TEST CONDITIONS Preamp VI(SHDN) = 2.5 V Shutdown voltage for power up IIH(SHDN) Shutdown input current high IIL(SHDN) Shutdown input current low Shutdown voltage for power down VCC = ±5 V VCC = ±15 V VCC = ±5 V or ±15 V PGA VIH(SHDN) VIL(SHDN) MIN VCC = ±5 V or ±15 V, VCC = ±5 V or ±15 V, TYP MAX 0.2 0.3 0.65 0.8 0.8 1.2 0.8 Relative to GND VI(SHDN) = 5 V VI(SHDN) = 0.5 V Disable time† Enable time† 2 UNIT mA V V 300 400 µA 25 50 µA tdis VCC = ±5 V or ±15 V, Preamp and PGA 100 ns ten VCC = ±5 V or ±15 V, Preamp and PGA 1.5 µs † Disable time and enable time are defined as the interval between application of the logic signal to SHDN and the point at which the supply current has reached half its final value. 6 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 PGA electrical characteristics, TA = 25°C, Gain = 2 dB, RL = 1 kΩ, (unless otherwise noted) TEST CONDITIONS† PARAMETER VCC Supply voltage range Split supply VOM Maximum output voltage swing RL = 1 kΩ VCC = ±5 V VCC = ±15 V VIO Input offset voltage VCC = ±5 V or ±15 V TA = 25°C TA = full range MIN ±4.5‡ TYP ±3.6 ±4.1 ±13.2 ±13.8 2 VCC = ±5 V VCC = ±15 V IIB Input bias current (reference terminal) VCC = ±5 V or ±15 V TA = 25°C TA = full range IO IOS Output current RL = 20 Ω VCC = ±5 V 30 ±3.8 ±4.0 ±13.5 ±13.8 TA = 25°C 75 TA = full range 72 1 Input resistance RO Output resistance VCC = ±5 V or ±15 V Gain = 20 dB ICC V V 9 V 2 µA A 50 mA 80 mA 82 dB 0.27 Gain = −22 dB kΩ 3 Open loop Ω 20 VCC = ±5 V TA = 25°C TA = full range 4.8 VCC = ±15 V TA = 25°C TA = full range 5 Quiescent supply current (per channel) mV µV/°C 3 Short-circuit output current RI ±16.5 10 Reference input voltage range Power supply rejection ratio UNIT 11 Input offset voltage drift PSRR MAX 6 7 7 mA 8 † Full range for the THS7001/02C is 0°C to 70°C. Full range for the THS7001/02I is − 40°C to 85°C. ‡ PGA minimum supply voltage must be less than or equal to preamp supply voltage. output limiting characteristics TEST CONDITIONS† PARAMETER Clamp accuracy Clamp overshoot Overdrive recovery time Clamp input bias current MIN TYP MAX ±250 ±300 VCC = ±15 V, VI = ±10 V, Gain = 2 dB VH = 10 V, VL = −10 V, TA = 25°C VCC = ±5 V, VI = ±2.5 V, Gain = 2 dB VH = 2 V, VL = −2 V, TA = 25°C VCC = ±15 V, VI = ±10 V, VH = 10 V, tr and tf = 1 ns VL = −10 V, 0.5% VCC = ±5 V, VI = ±2.5 V, VH = 2 V, tr and tf = 1 ns VL = −2 V, 0.3% VCC = ±15 V, VI = ±10 V VH = 10 V, VL = −10 V, 7 VCC = ±5 V, VI = ±2.5 V VH = 2 V, VL = 2 V, 6 VO = 3.3 V, VH = 3.3 V VL = 3.3 V, TA = 25°C TA = full range UNIT ±350 TA = full range ±50 ±80 mV ±100 TA = full range ns 1 5 8 µA A † Full range for the THS7002C is 0°C to 70°C. Full range for the THS7002I is − 40°C to 85°C. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 7 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 PGA electrical characteristics, TA = 25°C, Gain = 2 dB, RL = 1 kΩ, (unless otherwise noted) (continued) digital gain characteristics PARAMETER VIH VIL High-level input voltage IIH IIL High-level input current TEST CONDITIONS MIN TYP MAX UNIT 2 V Relative to GND Low-level input voltage 0.8 VIH = 5 V VIL = 0.5 V Low-level input current (sink current) td Gain-change delay time† VCC = ±5 V or ±15 V † Gain-change delay time is the time needed to reach 90% of its final gain value. V 20 100 nA 0.9 2 µA µs 2 PGA operating characteristics, TA = 25°C, Gain = 2 dB, RL = 1 kΩ, (unless otherwise noted) TEST CONDITIONS† PARAMETER MIN TYP Slew rate (see Note 4) VCC = ±5 V, VCC = ±15 V, VO = ±2.5 V VO = ±10 V 160 SR Settling time to 0.1% 5 V Step VCC = ±15 V VCC = ±5 V 125 ts THD Total harmonic distortion VCC = ±15 V, fc = 1 MHz, VO(PP)= 2 V, Gain = 8 dB Gain = 20 dB, VO(PP) = 0.4 V VCC = ±15 V VCC = ±5 V 65 Gain = 2 dB, VO(PP) = 0.4 V VCC = ±15 V VCC = ±5 V 75 Gain = −22 dB, VO(PP) = 0.4 V VCC = ±15 V VCC = ±5 V 80 Gain = 2 dB, VO(PP) = 0.4 V 20 Bandwidth for 0.1 dB flatness VCC = ±15 V VCC = ±5 V VO(PP) = 5 V, VO(PP) = 20 V, VCC = ±5 V VCC = ±15 V 10 Full power bandwidth (see Note 5) Differential gain error G = 8 dB, 100 IRE, NTSC, RL = 150 Ω VCC = ±5 V VCC = ±15 V 0.04% AD φD G = 8 dB, ±100 IRE, NTSC, RL = 150 Ω VCC = ±15 V VCC = ±5 V 0.07 Differential phase error Gain = −22 dB to 20 dB, All 8 steps, VCC = ±5 V or ±15 V TA = 25°C −7.5% Gain accuracy (see Note 6) TA = full range −8.5% Channel-to-channel gain accuracy (THS7002 only) (see Note 7) Gain = −22 dB to 20 dB, All 8 steps, VCC = ±5 V or ±15 V TA = 25°C −5.5% TA = full range −6.5% VCC = ±5 V or ±15 V, f = 10 kHz Gain = 20 dB 10 Input referred noise voltage Gain = −22 dB 500 PGA channel-to-channel crosstalk (THS7002 only) VCC = ±5 V or ±15 V, f = 1 MHz −77 BW Vn Small-signal bandwidth (−3 dB) † Full range for the THS7001/02C is 0°C to 70°C. Full range for the THS7001/02I is − 40°C to 85°C. NOTES: 4. Slew rate is measured from an output level range of 25% to 75%. 5. Full power bandwidth = slew rate/2π VPEAK 6. Specified as −100 × (output voltage − (input voltage × gain))/(input voltage × gain) 7. Specified as 100 × (output voltage B− output voltage A)/output voltage A 8 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 MAX UNIT V/ s V/µs 175 ns 120 −69 dBc 60 MHz 70 70 MHz 18 MHz 2.8 0.04% ° 0.09 0% 7.5% 8.5% 0% 5.5% 6.5% nV/√Hz dB 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 TYPICAL CHARACTERISTICS STANDBY SUPPLY CURRENT vs FREE-AIR TEMPERATURE INPUT OFFSET VOLTAGE vs FREE-AIR TEMPERATURE QUIESCENT SUPPLY CURRENT vs FREE-AIR TEMPERATURE 2 3.5 9 Per Channel 3 PGA VCC = ±5 V 2.5 2 PGA VCC = ±15 V 1.5 Preamp VCC = ±15 V Preamp VCC = ±5 V 1 1.5 I CC − Supply Current − mA Standby Supply Current − mA V IO − Input Offset Voltage − mV PGA − ICC Preamp − ICC 1 Preamp − IEE 0.5 PGA − IEE 0 0.5 8 Preamp VCC = ±15 V 7 6 Preamp VCC = ±5 V 5 PGA VCC = ±15 V 4 PGA VCC = ±5 V VCC = ±15 V 0 −40 −20 0 20 40 60 80 −0.5 −40 100 TA − Free-AIR Temperature − _C −20 0 20 80 3 −40 100 80 PGA - VCC + PGA - VCC − 10 100 1k 10k 2 Pre−Amp: VCC = ±5 V 1.5 1 PGA: VCC = ±15 V and ±5 V 100k 1M 10M 100M −20 0 20 Figure 6 40 60 80 PGA: G = −22 dB 0.1 |VO | − Output Voltage Swing − V 13 11 9 7 5 12 RL = 1 kΩ 10 RL = 250 Ω 8 6 4 2 3 13 ±VCC − Supply Voltage − V Figure 9 15 5 7 9 11 13 15 ± VCC − Supply Voltage − V Figure 10 POST OFFICE BOX 655303 10M 100M 500M 10M 100M Figure 8 TA = 25° C 11 1M f − Frequency − Hz 14 9 1 0.01 100k 100 PREAMP OUTPUT VOLTAGE vs SUPPLY VOLTAGE TA = 25 _C 7 PGA: G = +20 dB Figure 7 PREAMP INPUT COMMON-MODE VOLTAGE RANGE vs SUPPLY VOLTAGE 5 10 TA − Free-Air Temperature − _C f − Frequency − Hz 15 100 Preamp: G = +2 0 −40 0 80 VCC = ±15 V & ±5 V V|(PP)= 2 V Pre−Amp: VCC = ±15 V 2.5 0.5 VCC = ±15 V & ±5 V 60 100 CMRR − Common-Mode Rejection Ratio − dB 20 40 CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY Z o − Output Impedance − Ω Iib − Input Bias Current − uA 100 20 Figure 5 3 Preamp - VCC + & VCC − 40 0 TA − Free-Air Temperature − _C INPUT BIAS CURRENT vs FREE-AIR TEMPERATURE 120 60 −20 Figure 4 PSRR vs FREQUENCY PSRR − Power-Supply Rejection Ratio − dB 60 TA − Free-Air Temperature − _C Figure 3 V ICR − Input Common-Mode Range − + −V 40 • DALLAS, TEXAS 75265 PREAMP CMRR vs FREQUENCY 100 80 60 40 20 VCC = ±15 V and ±5 V 0 100 1k 10k 100k 1M f − Frequency − Hz Figure 11 9 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 TYPICAL CHARACTERISTICS PREAMP INPUT REFERRED VOLTAGE NOISE AND CURRENT NOISE vs FREQUENCY PREAMP OPEN LOOP GAIN AND PHASE RESPONSE vs FREQUENCY 100 45_ Gain −45_ Phase 40 0 −20 100 V n − Voltage Noise − nV/ Hz I n − Current Noise − pA/ Hz 60 20 10 0_ −90_ −135_ VCC = ±15 V and ±5 V RL = 250 Ω 1k 10k Phase Open Loop Gain − dB 80 20 −180_ 100k 1M VN 1 IN VCC = ±15 V and ±5 V TA = 25 _C 0.1 10 −225_ 10M 100M 1G 100 Figure 12 PREAMP OUTPUT AMPLITUDE vs FREQUENCY 8 3 2 VCC = ±5 V G=2 RL = 150 Ω VO(PP) = 0.4 V 1 0 1M RF = 499 Ω RF = 100 Ω 4 3 2 VCC = ±15 V G=2 RL = 150 Ω VO(PP) = 0.4 V 1 0 −1 100k −1 100k 6 5 10M 100M 500M f − Frequency − Hz −5 VCC = ±5 V G = −1 RL = 150 Ω VO(PP) = 0.4 V −7 100k 500M 1M 10M 100M 500M Figure 15 Figure 16 PREAMP OUTPUT AMPLITUDE vs FREQUENCY PREAMP OUTPUT AMPLITUDE vs FREQUENCY PREAMP OUTPUT AMPLITUDE vs FREQUENCY 16 0 RF = 499 Ω −1 −2 RF = 100 Ω −3 −4 VCC = ±15 V G = −1 RL = 150 Ω VO(PP) = 0.4 V 1M 10 8 6 4 2 100M 500M VCC = ±5 V G=5 RL = 150 Ω VO(PP) = 0.4 V −1 100k 1M 100M 500M RF = 499 Ω 12 10 8 6 4 2 0 10M RF = 5.1 kΩ 14 RF = 499 Ω 12 0 10M 16 RF = 5.1 kΩ 14 Output Amplitude − dB RF = 1 kΩ Output Amplitude − dB Output Amplitude − dB −4 Figure 14 −7 100k 10 100M RF = 100 Ω −3 −6 10M RF = 499 Ω −2 f − Frequency − Hz 1 −6 0 −1 f − Frequency − Hz 2 −5 1M RF = 1 kΩ 1 Output Amplitude − dB Output Amplitude − dB Output Amplitude − dB RF = 100 Ω 4 2 RF = 1 kΩ 7 RF = 499 Ω 5 100k PREAMP OUTPUT AMPLITUDE vs FREQUENCY 8 RF = 1 kΩ 6 10k Figure 13 PREAMP OUTPUT AMPLITUDE vs FREQUENCY 7 1k f − Frequency − Hz f − Frequency − Hz VCC = ±15 V G=5 RL = 150 Ω VO(PP) = 0.4 V −1 100k 1M 10M f − Frequency − Hz f − Frequency − Hz f − Frequency − Hz Figure 17 Figure 18 Figure 19 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 100M 500M 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 TYPICAL CHARACTERISTICS PREAMP LARGE AND SMALL SIGNAL FREQUENCY RESPONSE 12 VO(PP) = 0.8 V 0 VO(PP) = 0.4 V −6 VO(PP) = 0.2 V −12 −18 100k 1M 10M 100M 6 VO(PP) = 0.4 V −6 VO(PP) = 0.2 V 1M 100M 500M 100k −60 VCC=+/−5 V 2nd Harmonic −80 VCC=+/−15 V 3rd Harmonic VCC=+/−5 V 3rd Harmonic 0.3 80 70 60 VCC = ±5 V VO (PP) = 5 V 50 0.1 0 −0.1 RL= 200 Ω 30 −40 −0.3 −20 0 20 40 60 80 100 0 50 100 TA − Free-Air Temperature − °C G= +2 VCC = ±5 V RL = 200 Ω 0 G = −1 VCC = ±5 V RL = 200 Ω 300 G = +5 VCC = ±15 V RL = 200 Ω 10 VO − Output Voltage − V VO − Output Voltage − V 1 250 PREAMP 20-V STEP RESPONSE 12.5 3 2 200 Figure 25 PREAMP 5-V STEP RESPONSE 2 150 t − Time − ns Figure 24 PREAMP 5-V STEP RESPONSE −2 G = +2 VCC = ±5 V RL = 200 Ω −0.2 40 3 10M PREAMP 400-mV STEP RESPONSE 0.2 Figure 23 −1 1M f − Frequency − Hz Figure 22 VCC = ±15 V VO (PP) = 20 V 90 SR − Slew Rate − V/uS Distortion − dBc VCC=+/−15 V 2nd Harmonic −100 0.0 2.5 5.0 7.5 10.0 12.5 15.0 17.5 20.0 VO(PP) − Peak-To-Peak Output Voltage − V VO − Output Voltage − V 10M PREAMP SLEW RATE vs FREE-AIR TEMPERATURE 100 −90 VCC=+/−15 V 3rd Harmonic −100 Figure 21 −30 −70 VCC=+/−15 V 2nd Harmonic −80 f − Frequency − Hz PREAMP HARMONIC DISTORTION vs OUTPUT VOLTAGE −50 VCC=+/−5 V 2nd Harmonic −70 −90 Figure 20 −40 −60 −12 f − Frequency − Hz Gain=5 RF=300 Ω RL=1 kΩ f=1 MHz −50 VO(PP) = 0.8 V 0 −18 100k 500M VCC=+/−5 V 3rd Harmonic RL= 250 Ω Gain=+8 dB VO(PP)= 2 V −40 VO − Output Voltage − V 6 −30 VCC = ± 15 V RL = 150 Ω G = +2 VO(PP) = 1.6 V Distortion − dBc VCC = ± 5 V RL = 150 Ω G = +2 VO(PP) = 1.6 V V 0− Normalized Output Voltage − dBV V 0− Normalized Output Voltage − dBV 12 PREAMP HARMONIC DISTORTION vs FREQUENCY PREAMP LARGE AND SMALL SIGNAL FREQUENCY RESPONSE 1 0 −1 7.5 5 2.5 0 −2.5 −5 −7.5 −2 −10 −3 −12.5 −3 0 50 100 150 200 t − Time − ns Figure 26 250 300 0 50 100 150 200 250 300 t − Time − ns Figure 27 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 0 200 400 600 800 1000 t − Time − ns Figure 28 11 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 TYPICAL CHARACTERISTICS −60 Preamp−1 Input Preamp−2 Output −70 −60 Preamp − Input PGA − Output −70 PGA − Input Preamp − Output −80 −80 Preamp−2 Input Preamp−1 Output −90 100k 1M 10M −90 100k 100M f − Frequency − Hz 0.4 0.3 0.2 0.2 VCC = ±5 V 0.1 0 −0.1 VCC = ±15 V −0.3 0 5 10 15 8 2 −4 −10 −16 −22 G = 1,1,0 G = 1,0,1 G = 1,0,0 G = 0,1,1 G = 0,1,0 G = 0,0,1 G = 0,0,0 −28 100k 1M 10M 100M f − Frequency − Hz Figure 35 12 20 14 8 2 −4 −10 −0.2 −16 −0.3 −22 −20 0 20 40 60 80 TA − Free-AiirTemperature − °C 1G 18 6 0 −6 −12 G = 1,1,0 G = 1,0,1 G = 1,0,0 G = 0,1,1 G = 0,1,0 G = 0,0,1 G = 0,0,0 1M 1G PGA LARGE AND SMALL SIGNAL FREQUENCY RESPONSE 18 VO(PP) = 0.4 V VO(PP) = 0.2 V 100M 1G f − Frequency − Hz Figure 36 POST OFFICE BOX 655303 100M Figure 34 VO(PP) = 0.8 V 10M 10M f − Frequency − Hz VO(PP) = 1.6 V 1M 100k VCC = ±15 V RL = 1 kΩ VO(PP) = 0.4 V G = 1,1,1 −28 100k 100 VCC = ± 5 V RL = 1 kΩ G = +2 dB VO(PP) = 3.2 V −18 100k 10k PGA FREQUENCY RESPONSE −0.1 12 1k 26 PGA LARGE AND SMALL SIGNAL FREQUENCY RESPONSE V 0− Normalized Output Voltage − dBV Output Level − dB 14 100 Figure 33 VCC = ± 5 V RL = 1 kΩ VO(PP) = 0.4 V G = +14 dB G = +20 dB Figure 31 −0.0 PGA FREQUENCY RESPONSE 20 G = + 8 dB Figure 30 Figure 32 G = 1,1,1 G = + 2 dB f − Frequency − Hz 0.1 Gain Setting − dB 26 10 10 VCC=±5 V ±15 V Typical For All Gains −0.4 −40 20 100 f − Frequency − Hz Output Level − dB VCC = 25_ C 1k 100M 0.4 0.3 −0.4 −25 −20 −15 −10 −5 10M NORMALIZED PGA GAIN ACCURACY vs TEMPERATURE Gain Accuracy − % Channel-To-Channel Gain Accuracy − % PGA CHANNEL-TO-CHANNEL GAIN ACCURACY vs GAIN SETTING G = − 22 dB G = − 16 dB G = − 10 dB G = − 4 dB 1 1M Figure 29 −0.2 10k VCC = ±15 V & ±5 V RL = 1 kΩ Preamp: G=2 RF = 499 Ω PGA: G = +2 dB Input Referred Voltage Noise − nV/ Hz −50 Crosstalk − dB −50 Crosstalk − dB −40 VCC = ±15 V & ±5 V G=2 VO(PP) = 1.3 V RL = 1 kΩ RF = 499 Ω V 0− Normalized Output Voltage − dBV −40 PGA INPUT REFERRED VOLTAGE NOISE vs FREQUENCY PREAMP-TO-PGA CROSSTALK vs FREQUENCY THS7002 PREAMP CHANNEL-TO-CHANNEL CROSSTALK • DALLAS, TEXAS 75265 12 6 0 −6 −12 VCC = ± 15 V RL = 1 kΩ G = +2 dB VO(PP) = 3.2 V VO(PP) = 1.6 V VO(PP) = 0.8 V VO(PP) = 0.4 V VO(PP) = 0.2 V −18 100k 1M 10M 100M f − Frequency − Hz Figure 37 1G 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 TYPICAL CHARACTERISTICS PGA HARMONIC DISTORTION vs FREQUENCY −40 −50 VCC=±15 V 3rd Harmonic VCC=±15 V 2nd Harmonic −70 −80 VCC ± 15 V 2nd Harmonic 2nd Harmonic VCC ± 15V −70 3rd Harmonic −80 VCC=±5 V 3rd Harmonic VCC ± 5 V 3rd Harmonic −100 100k 1M f − Frequency − Hz 0 2.5 5 7.5 15-V Condition: VH, VL = ±10 V VI = ±10 V VH − 5V 0 5-V Condition: VH, VL = ±2 V VI = ±2 V VL − 15V −100 −200 20 3 2 Clamped Output VH = 2 V VL = −2 V 1 0 −1 −2 VCC = ±5 V Gain = 1,0,0 (+ 2 dB) RL = 500 Ω G = +2 dB 0 20 40 60 100 200 TA − Free-Air Temperature − _C Figure 41 Unclamped Output Clamped Output VH = 5 V VCC = +15 V Gain = 1,0,0 (+2 dB) VL = VCC− 100 150 t − Time − ns Figure 44 −50 −1 Unclamped Output −2 VCC = +15 V, +5 V Gain = 1,0,0 (+2 dB) VH = VCC+ 100 200 300 400 500 t − Time − ns Figure 43 PGA SHUTDOWN RESPONSE VSHDN (5 V/Div) Preamp: Forward Iso. −40 Preamp: Reverse Iso. −60 −70 −80 VCC = ±5 V,±15 Gain = 1,0,1 (+8 dB) RL = 500 Ω VOUT (500 mV/Div) PGA: Forward Iso. −90 RL = ∞ PGA: Reverse Iso. −4 50 Shutdown Isolation − dB 6 0 0 500 VCC = ± 5 V & ± 15 V VI(PP) = 2.5 V −30 8 2 400 SHUTDOWN ISOLATION vs FREQUENCY −20 4 300 Figure 42 PGA CLAMP RESPONSE 10 Clamped Output VL = 0 V 1 −3 t − Time − ns 12 2 −4 0 100 80 100 PGA CLAMP RESPONSE −4 −20 −20 0 20 40 60 80 TA − Free-Air Temperature − °C Figure 40 Unclamped Output −3 VL − 5V VO − Output Voltage − V RL=1 k Ω 100 −40 4 3 VO − Output Voltage − V Clamp Accuracy − mV 200 0 120 4 VH − 15V −2 17.5 VCC=+/−5 V VO(P-P)=5 V 140 PGA CLAMP RESPONSE 300 0 15 160 Figure 39 PGA CLAMP ACCURACY vs FREE-AIR TEMPERATURE −300 −40 10 12.5 180 VO(PP) − Peak -To-Peak Output Volage − V Figure 38 100 RL = 1 kΩ G= +8 dB f = 1 MHz −100 10M VCC=+/−15 V VO(P-P)=20 V 200 −60 −90 −90 220 VO − Output Voltage − V −60 Distortion − dBc Distortion − dBc −50 VCC ± 5 V VO − Output Voltage − V −40 VCC=±5 V 2nd Harmonic SR − Slew Rate − V/ µ S −30 RL= 1 kΩ Gain=+8 dB VO(PP)= 2 V PGA SLEW RATE vs FREE-AIR TEMPERATURE PGA HARMONIC DISTORTION vs OUTPUT VOLTAGE 200 250 −100 100k 1M 10M 100M 500M f − Frequency − Hz Figure 45 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 0 100 200 300 400 500 600 700 800 9001000 t − Time − ns Figure 46 13 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 TYPICAL CHARACTERISTICS PGA 5-V STEP RESPONSE PREAMP SHUTDOWN RESPONSE PGA SHUTDOWN RESPONSE 3 Gain = 1,0,1 (+8 dB) RL = 500 Ω VSHDN (5 V/Div) VOUT (500 mV/Div) VCC = ±15 V VO − Output Voltage − V 2 VO − Output Voltage − V VO − Output Voltage − V VSHDN (5 V/Div) VCC = ±5 V VCC = ±15 V RL = ∞ 1 0 −1 VO (500 mV/Div) G = 2 dB VCC = ±5 V RL = 1 kΩ −2 VCC = ±5 V 0 RL = 150 Ω 10 20 30 40 50 60 70 80 90 100 0 1 2 t − Time − ns 3 −3 4 5 6 7 8 9 10 t − Time − ns Figure 47 PGA 20-V STEP RESPONSE G = 8 dB VCC = ±15 V RL = 1 kΩ VO − Output Voltage − V 5.0 2.5 0 −2.5 −5.0 −7.5 −10.0 −12.5 0 100 200 300 400 500 t − Time − ns Figure 50 14 POST OFFICE BOX 655303 100 150 Figure 49 12.5 7.5 50 200 t − Time − ns Figure 48 10.0 0 • DALLAS, TEXAS 75265 250 300 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION theory of operation Each section of the THS7001 and THS7002 consists of a pair of high speed operational amplifiers configured in a voltage feedback architecture. They are built using a 30-V, dielectrically isolated, complementary bipolar process with NPN and PNP transistors possessing fTs of several GHz. This results in exceptionally high performance amplifiers that have a wide bandwidth, high slew rate, fast settling time, and low distortion. A simplified schematic of the preamplifiers are shown in Figure 51. VCC + OUT IN − IN + VCC − Figure 51. Pre-Amp Simplified Schematic noise calculations and noise figure Noise can cause errors on very small signals. This is especially true for the preamplifiers, which typically amplify small signals. The noise model is shown in Figure 52. This model includes all of the noise sources as follows: • • • • en = amplifier internal voltage noise (nV/√Hz) IN+ = noninverting current noise (pA/√Hz) IN− = inverting current noise (pA/√Hz) eRx = thermal voltage noise associated with each resistor (eRx = 4 kTRx ) POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 15 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION noise calculations and noise figure (continued) eRs RS en Noiseless + _ eni IN+ eno eRf RF eRg IN− RG Figure 52. Noise Model The total equivalent input noise density (eni) is calculated by using the following equation: e ni + Ǹǒ ǒ 2 e nǓ ) IN ) R Ǔ S 2 ǒ ) IN– ǒRF ø RGǓǓ 2 ǒ Ǔ ) 4 kTR s ) 4 kT R ø R F G (1) Where: k = Boltzmann’s constant = 1.380658 × 10−23 T = temperature in degrees Kelvin (273 +°C) RF || RG = parallel resistance of RF and RG To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (eni) by the overall amplifier gain (AV). e no + e ǒ Ǔ R A + e ni 1 ) F (Noninverting Case) ni V RG (2) As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the closed-loop gain is increased (by reducing RF + RG), the input noise can be reduced considerably because of the parallel resistance term. This leads to the general conclusion that the most dominant noise sources are the source resistor (RS) and the internal amplifier noise voltage (en). Because noise is summed in a root-mean-squares method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly simplify the formula and make noise calculations much easier to calculate. By using the low noise preamplifiers as the first element in the signal chain, the input signal’s signal-to-noise ratio (SNR) is maintained throughout the entire system. This is because the dominant system noise is due to the first amplifier. This can be seen with the following example: 16 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION noise calculations and noise figure (continued) RF2 RF1 RG2 RG1 + Vin _ A1 + _ eno1 + eno2 eni2 eni1 RF1 AV1= 1+ RG1 RF2 AV2= 1+ RG2 Figure 53. Simplified Composite Amplifier System The noise due to amplifier 1 (A1) is the same as derived in equations 1 and 2. The composite system noise is calculated as follows: e no2 + + Ǹeni2 2 ) e no1 2 A V2 Ǹeni2 ) ǒeni1AV1Ǔ 2 2 A (3) V2 In a typical system, amplifier 1 (A1) has a large gain (AV1). Because the noise is summed in the RMS method, if the A1 output noise is more than 25% larger than the input noise of amplifier 2, the contribution of amplifier 2’s input noise to the composite amplifier output noise can effectively be ignored. This reduces equation 3 down to: e no2 ≅ e (4) A A ni1 V1 V2 Equation 4 shows that the very first amplifier (the preamplifier) is critical in any low-level signal system. This also shows that practically any noisy amplifier can be used after the preamplifier with minimal SNR degradation. For more information on noise analysis, please refer to the Noise Analysis section in Operational Amplifier Circuits Applications Report (literature number SLVA043). This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be defined and is typically 50 Ω in RF applications. NF + ȱ e 2ȳ 10logȧ ni ȧ 2 ǒ Ǔ e Ȳ Rs ȴ POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 17 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION noise calculations and noise figure (continued) Because the dominant noise components are generally the source resistance and the internal amplifier noise voltage, we can approximate noise figure as: NF + ȱ ȡǒ Ǔ2 ǒ ȧ en ) IN ) ȧ Ȣ ȧ 10logȧ1 ) 4 kTR ȧ S ȧ Ȳ Ǔ ȣȳ S ȧ 2 R Ȥȧ ȧ ȧ ȧ ȧ ȴ Figure 54 shows the noise figure graph for the THS7001 and THS7002. 16 Noise Figure − dB 14 PREAMP NOISE FIGURE vs SOURCE RESISTANCE f = 10 kHz TA = 25 _C 12 10 8 6 4 2 0 10 100 1k 10k Source Resistance − Ω Figure 54. Noise Figure vs Source Resistance optimizing frequency response for the preamplifiers Internal frequency compensation of the THS7001 and THS7002 was selected to provide very wide bandwidth performance and still maintain a very low noise floor. In order to meet these performance requirements, the preamplifiers must have a minimum gain of 2 (−1). Because everything is referred to the noninverting terminal of an operational amplifier, the noise gain in a G = −1 configuration is the same as a G = 2 configuration. One of the keys of maintaining a smooth frequency response, and hence, a stable pulse response, is to pay particular attention to the inverting terminal. Any stray capacitance at this node causes peaking in the frequency response. There are two things that can be done to help minimize this effect. The first is to simply remove any ground planes under the inverting terminal of the amplifier. This also includes the trace that connects to this terminal. Additionally, the length of this trace should be minimized. The capacitance at this node causes a lag in the voltage being fed back due to the charging and discharging of the stray capacitance. If this lag becomes too long, the amplifier will not be able to correctly keep the noninverting terminal voltage at the same potential as the inverting terminal’s voltage. Peaking and possibly oscillations can occur if this happens. 18 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION optimizing frequency response for the preamplifiers (continued) The next thing that helps to maintain a smooth frequency response is to keep the feedback resistor (Rf) and the gain resistor (Rg) values fairly low. These two resistors are effectively in parallel when looking at the ac small-signal response. This is why in a configuration with a gain of 5, a feedback resistor of 5.1 kΩ with a gain resistor of 1.2 kΩ only shows a small peaking in the frequency response. The parallel resistance is less than 1 kΩ. This value, in conjunction with a very small stray capacitance test PCB, forms a zero on the edge of the amplifier’s natural frequency response. To eliminate this peaking, all that needs to be done is to reduce the feedback and gain resistances. One other way to compensate for this stray capacitance is to add a small capacitor in parallel with the feedback resistor. This helps to neutralize the effects of the stray capacitance. To keep this zero out of the operating range, the stray capacitance and resistor value’s time constant must be kept low. But, as can be seen in Figures 14 − 19, a value too low starts to reduce the bandwidth of the amplifier. Table 1 shows some recommended feedback resistors to be used with the THS7001 and THS7002 preamplifiers. Table 2. Recommended Feedback Resistors GAIN Rf for VCC = ±15 V and ± 5 V 2 499 Ω −1 499 Ω 5 1 kΩ PGA gain control The PGA section of the THS7001 and THS7002 IC allows for digital control of the gain. There are three digital control pins for each side of the PGA (AG0 – AG2, and BG0 – BG2). Standard TTL or CMOS Logic will control these pins without any difficulties. The applied logic levels are referred to the DGND pins of the THS7002. The gain functions are not latched and therefore always rely on the logic at these pins to maintain the correct gain settings. A 3.3 kΩ resistor to ground is usually applied at each input to ensure a fixed logic state. The gain control acts like break-before-make SPDT switches. Because of this action, the PGA will go into an open-loop condition. This may cause the output to behave unpredictably until the switches closes in less than 1.5 µs. Due to the topology of this circuit, the controlling circuitry must be able to sink up to 2 µA of current when 0-V is applied to the gain control pin. A simplified circuit diagram of the gain control circuitry is shown in Figure 55. +VCC To Internal Bias Circuitry Control Gain −VCC DGND Figure 55. Simplified PGA Gain Control POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 19 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION PGA gain control (continued) One aspect of the THS7001 and THS7002 PGA signal inputs is that there are internal variable resistors (RF and RG), which set the gain. The resistance of RG changes from about 270-Ω (Gain = +20 dB) to about 3-kΩ (Gain = −22 dB). Therefore, any source impedance at the input to the PGA amplifiers will cause a gain error to be seen at the output. A buffer/amplifier is highly recommended to directly drive the input of the PGA section to help minimize this effect. Another thing which should be kept in mind is that when each amplifier’s VREF is connected to ground, the internal RG resistor is connected to a virtual ground. Therefore, if a termination resistor is used on the source side, the total terminating resistance is the parallel combination of the terminating resistance and the internal RG resistor. This, in conjunction with the series impedance problem mentioned previously, can potentially cause a voltage mismatch between the output of a 50-Ω source and the expected PGA output voltage. These points can be easily seen in the simplified diagram of the THS7001 and THS7002 PGA section (see Figure 56). No Source Impedance VIN RSOURCE THS7001 and THS7002 IC G0 G1 G2 PGA −VIN RG RF Positive Clamp VH RTERMINATION − PGA PGA VREF R TOTAL TERMINATION + PGA VOUT Negative Clamp VL + ǒRSOURCE ) RGǓ R ) ǒR )R Ǔ TERMINATION SOURCE G R TERMINATION Figure 56. Simplified PGA Section of the THS7001 and THS7002 voltage reference terminal If a voltage is applied to the PGA’s VREF terminal, then the output of the PGA section will amplify the applied reference voltage by one plus the selected gain. Thus, the output gain strictly due to VREF will be from +0.6 dB to +21 dB according to the following formula: V OUT + 20 X Log ƪ1 ) ǒPGA GainǒVńVǓ Ǔ ƫ 10 V REF For most configurations, it is recommended that this pin be connected to the signal ground. 20 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION output clamping Typically, the output of the PGA will directly drive an analog-to-digital converter (ADC). Because of the limited linear input range and saturation characteristics of most ADCs, the PGA’s outputs incorporate a voltage clamp. Unlike a lot of clamping amplifiers which clamp only at the input, the THS7001 and THS7002 clamps at the output stage. This insures that the output will always be protected regardless of the Gain setting and the input voltage. The clamps activate almost instantaneously and recover from saturation in less than 7 ns. This can be extremely important when the THS7001 and THS7002 is used to drive some ADCs which have a very long overdrive recovery time. It is also recommended to add a pair of high frequency bypass capacitors to the clamp inputs. These capacitors will help eliminate any ringing which may ocur when a large pulse is applied to the amplifier. This pulse will force the clamp diodes to abruptly turn on, drawing current from the reference voltages. Just like a power supply trace, you must minimize the inductance seen by the clamp pins. The bypass capacitors will supply the sudden current demands when the clamps are suddenly turned on. A simplified clamping circuit diagram is shown in Figure 57. +VCC VH Output Transistor Drive To Bias Circuits V1 0.1 µF OUT Output Transistor VL V2 0.1 µF −VCC Figure 57. Simplified THS7001 and THS7002 Clamp Circuit Because the internal clamps utilize the same clamping reference voltages, the outputs of both PGAs on the THS7002 are clamped to the same values. These clamps are typically connected to the power supply pins to allow a full output range. But, they can be connected to any voltage reference desired. The clamping range is limited to +VCC and GND for VH and –VCC and GND for VL. It is possible to go beyond GND for each respective clamp, but it is not recommended. This is because this operation relies on the internal bias currents in the Class AB output stage to maintain their linearity. There may also be a chance to reverse bias the PN junctions and possibly cause internal damage to these junctions. But for reference, the graphs in Figure 58 show the output voltage versus the clamping voltage with different loads. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 21 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION output clamping (continued) OUTPUT VOLTAGE vs CLAMP VOLTAGE (VL) 4 4 VCC=±5 V & ±15 V VI=5 V Gain=+2 dB 2 VCC=±5 V & ±15 V VI=−5 V Gain=+2 dB 3 V O − Output Voltage − V 3 V O − Output Voltage − V OUTPUT VOLTAGE vs CLAMP VOLTAGE (VH) RL=5.1 k Ω 1 RL=500 Ω 0 −1 −2 −3 2 1 0 RL=500 Ω −1 −2 RL=5.1 k Ω −3 −4 −4 −4 −3 −2 −1 0 1 2 Clamp Voltage − V 3 4 5 −5 −4 −3 −2 −1 0 1 Clamp Voltage − V 2 3 4 Figure 58. Output Voltage vs Clamp Voltage The accuracy of this clamp is dependant on the amount of current flowing through the internal clamping diodes. As is typical with all diodes, the voltage drop across this diode increases with current. Therefore, the accuracy of the clamp is highly dependant upon the output voltage, the clamping voltage difference, and the output current. The accuracy of the clamps with different load resistances are shown in Figure 59. NEGATIVE CLAMP ACCURACY (VL) NEGATIVE CLAMP ACCURACY (VL) 80 RL=500 Ω Gain = +2dB RL=1 k Ω Gain = +2dB 40 0 −80 VL=−3 V −120 VL=−1 V 0 V O − VCLAMP V O − VCLAMP VL=−2 V −40 VL=−3 V −80 VL=−4 V VL=−4 V −120 −160 −160 −200 VL=−2 V VL=−3 V −40 VL=−4 V −80 −160 −5 0 0 −120 −4 −3 −2 −1 0 −5 Expected Output Voltage − V POSITIVE CLAMP ACCURACY (VH) POSITIVE CLAMP ACCURACY (VH) 160 120 −3 −2 −1 0 POSITIVE CLAMP ACCURACY (VH) 120 120 80 80 80 V O − VCLAMP VH=4 V VH=3 V 40 VH=2 V 0 VH=4 V 40 VH=3 V 0 VH=2 V −40 40 VH=4 V 0 VH=3 V VH=2 V −40 VH=1 V VH=1 V VH=1 V −40 RL=500 Ω Gain = +2dB 0 1 2 3 4 Expected Output Voltage − V RL=1 k Ω −80 RL=5.1 k Ω −80 Gain = +2dB −80 5 Gain = +2dB −120 −120 0 1 2 3 4 5 Expected Output Voltage − V Figure 59. Clamping Accuracy 22 −4 Expected Output Voltage − V V O − VCLAMP V O − VCLAMP VL=−2 V −4 −3 −2 −1 Expected Output Voltage− V VL=−1 V Gain = +2dB −40 −5 RL=5.1 k Ω 40 VL=−1 V V O − VCLAMP NEGATIVE CLAMP ACCURACY (VL) 80 40 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 0 1 2 3 4 Expected Output Voltage − V 5 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION shutdown control There are two shutdown pins which control the shutdown for each half of the THS7002 and one shutdown pin for the THS7001. When the shutdown pins signals are low, the THS7001 and THS7002 is active. But, when a shutdown pin is high (+5 V), a preamplifier and the respective PGA section is turned off. Just like the Gain controls, the shutdown logic is not latched and should always have a signal applied to them. A 3.3-kΩ resistor to ground is usually applied to ensure a fixed logic state. A simplified circuit can be seen in Figure 60. +VCC To Internal Bias Circuitry Control Gain 53 kΩ DGND −VCC DGND Figure 60. Simplified THS7001 and THS7002 Shutdown Circuit One aspect of the shutdown feature, which is often over-looked, is that the PGA section will still have an output while in shutdown mode. This is due to the internally fixed RF and RG resistors. This effect is true for any amplifier connected as an inverter. The internal circuitry may be powered down and in a high-impedance state, but the resistors are always there. This will then allow the input signal current to flow through these resistors and into the output. The equivalent resistance of RF and RG is approximately 3 kΩ. To minimize this effect, a shunt resistor to ground may be utilized, This will act as a classic voltage divider and will reduce the feed-through voltage seen at the PGA output. The drawback to this is the increased load on the PGA while in the active state. driving a capacitive load Driving capacitive loads with high-performance amplifiers is not a problem as long as certain precautions are taken. The first is to realize that the THS7001 and THS7002 has been internally compensated to maximize its bandwidth and slew rate performance. When an amplifier is compensated in this manner, capacitive loading directly on the output will decrease the device’s phase margin leading to high frequency ringing or oscillations. Therefore, for capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of the amplifier, as shown in Figure 61. A minimum value of 20 Ω should work well for most applications. For example, in 75-Ω transmission systems, setting the series resistor value to 75 Ω both isolates any capacitance loading and provides the proper line impedance matching at the source end. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 23 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION driving a capacitive load (continued) 499 Ω 499 Ω _ Input 20 Ω Output PREAMP + CLOAD Figure 61. Driving a Capacitive Load offset voltage The output offset voltage, (VOO) is the sum of the input offset voltage (VIO) and both input bias currents (IIB) times the corresponding gains. The following schematic and formula can be used to calculate the output offset voltage: RF RG IIB− − VOS + RS + − VIO IIB+ V OS ǒ + "V IO "I IB) R Ǔ S ǒ 1) R R F G Ǔ "I IB* R F Figure 62. Output Offset Voltage Model general configurations When receiving low-level signals, limiting the bandwidth of the incoming signals into the system is often required. The simplest way to accomplish this is to place an RC filter at the noninverting terminal of the THS7001 and THS7002 preamplifier (see Figure 63). RG RF f − VI + R1 –3dB V VO O + V I + ǒ 1) C1 Figure 63. Single-Pole Low-Pass Filter 24 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 1 2pR1C1 R R F G Ǔǒ Ǔ 1 1 ) sR1C1 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION general configurations (continued) If even more attenuation is needed, a multiple-pole filter is required. The Sallen-Key filter can be used for this task. For best results, the THS7001 and THS7002 preamplifier should have a bandwidth that is 8 to 10 times the filter frequency bandwidth. Failure to do this can result in phase shift of the amplifier. C1 + _ VI R1 R1 = R2 = R C1 = C2 = C Q = Peaking Factor (Butterworth Q = 0.707) R2 f C2 RG RF –3dB RG = + ( 1 2pRC RF 1 2− Q ) Figure 64. 2-Pole Low-Pass Sallen-Key Filter ADSL The ADSL receive band consists of up to 255 separate carrier frequencies each with its own modulation and amplitude level. With such an implementation, it is imperative that signals received off the telephone line have as high a signal-to-noise ratio (SNR) as possible. This is because of the numerous sources of interference on the line. The best way to accomplish this high SNR is to have a low-noise preamplifier on the front-end. It is also important to have the lowest distortion possible to help minimize against interference within the ADSL carriers. The THS7001 and THS7002 was designed with these two priorities in mind. By taking advantage of the superb characteristics of the complimentary bipolar process (BICOM), the THS7001 and THS7002 offers extremely low noise and distortion while maintaining a high bandwidth. There are some aspects that help minimize distortion in any amplifier. The first is to extend the bandwidth of the amplifier as high as possible without peaking. This allows the amplifier to eliminate any nonlinearities in the output signal. Another thing that helps to minimize distortion is to increase the load impedance seen by the amplifier, thereby reducing the currents in the output stage. This will help keep the output transistors in their linear amplification range and will also reduce the heating effects. One central-office side terminal circuit implementation, shown in Figure 65, uses a 1:2 transformer ratio. While creating a power and output voltage advantage for the line drivers, the 1:2 transformer ratio reduces the SNR for the received signals. The ADSL standard, ANSI T1.413, stipulates a noise power spectral density of –140 dBm/Hz, which is equivalent to 31.6 nV/√Hz for a 100 Ω system. Although many amplifiers can reach this level of performance, actual ADSL system testing has indicated that the noise power spectral density may typically be ≤ –150 dBm/Hz, or ≤ 10 nV/√Hz. With a transformer ratio of 1:2, this number reduces to less than 5 nV/√Hz. The THS7002 preamplifiers, with an equivalent input noise of 1.7 nV/√Hz, is an excellent choice for this application. Coupled with a very low 0.9 pA/√Hz equivalent input current noise and low value resistors, the THS7001 and THS7002 will ensure that the received signal SNR will be as high as possible. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 25 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION ADSL (continued) 12 V THS6012 Driver 1 VI+ 0.1 µF + 6.8 µF 12.5 Ω + _ 1:2 1 kΩ 100 Ω Telephone Line 1 kΩ 0.1 µF 6.8 µF + −12 V 12 V 0.1 µF THS6012 Driver 2 VI− + 1 kΩ 6.8 µF 12.5 Ω + _ 499 Ω 499 Ω − + VO+ THS7002 Preamp 1 1 kΩ 1 kΩ 0.1 µF 499 Ω 6.8 µF + 12 V −12 V 1 kΩ 0.1 µF Driver Block 499 Ω − + VO− THS7002 Preamp 2 −12 V 0.01 µF Receiver Block Figure 65. THS7002 Central-Office ADSL Application 26 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION ADSL (continued) Typically, the outputs of the preamplifiers are carried into a CODEC, which incorporates an analog-to-digital converter (ADC). The problem with this setup is that it only uses fixed gain elements. But, when the client is close to the central office, the gain must be set to receive a high-level signal; or for the opposite, set to receive a low-level signal. To solve this problem, a programmable-gain amplifier (PGA) should be used. The THS7001 and THS7002 PGAs allow the gain of the receiver signals to be varied from −22 dB to 20 dB. By allowing the gains to be controlled with a TTL-compatible signal, it is very easy to integrate the THS7001 and THS7002 into any system. By having the preamplifier output separate from the PGA input, inserting more amplifiers into the system can be accomplished easily. The functionality of the amplifier is typically as an active fixed gain filter. This is shown in Figure 66. C1 TO DSP 3.3 k Preamp V0+ R1 R2 THS6062 + _ 3.3 k 3.3 k G0 G1 G2 C2 +5 V VH RF RECEIVER BLOCK RG _ To CODEC + PGA OPTIONAL CIRCUIT VL Figure 66. Typical PGA Setup (One Channel) circuit layout considerations In order to achieve the levels of high-frequency performance of the THS7001 and THS7002, it is essential that proper printed-circuit board high-frequency design techniques be followed. A general set of guidelines is given below. In addition, a THS7001 and THS7002 evaluation board is available to use as a guide for layout or for evaluating the device performance. D Ground planes—It is highly recommended that a ground plane be used on the board to provide all components with a low inductive ground connection. However, in the areas of the amplifier inputs and output, the ground plane can be removed to minimize the stray capacitance. D Proper power supply decoupling—Use a 6.8-µF tantalum capacitor in parallel with a 0.1-µF ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several amplifiers depending on the application, but a 0.1-µF ceramic capacitor should always be used on the supply terminal of every amplifier. In addition, the 0.1-µF capacitor should be placed as close as possible to the supply terminal. As this distance increases, the inductance in the connecting trace makes the capacitor less effective. The designer should strive for distances of less than 0.1 inches (2,54 mm) between the device power terminals and the ceramic capacitors. D Sockets—Sockets are not recommended for high-speed operational amplifiers. The additional lead inductance in the socket pins will often lead to stability problems. Surface-mount packages soldered directly to the printed-circuit board is the best implementation. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 27 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION circuit layout considerations (continued) D Short trace runs/compact part placements—Optimum high-frequency performance is achieved when stray series inductance has been minimized. To realize this, the circuit layout should be made as compact as possible thereby minimizing the length of all trace runs. Particular attention should be paid to the inverting input of the amplifier. Its length should be kept as short as possible. This will help to minimize stray capacitance at the input of the amplifier. D Surface-mount passive components—Using surface-mount passive components is recommended for high frequency amplifier circuits for several reasons. First, because of the extremely low lead inductance of surface-mount components, the problem with stray series inductance is greatly reduced. Second, the small size of surface-mount components naturally leads to a more compact layout, thereby minimizing both stray inductance and capacitance. If leaded components are used, it is recommended that the lead lengths be kept as short as possible. thermal information The THS7001 and THS7002 is supplied in a thermally-enhanced PWP package, which is a member of the PowerPAD. This package is constructed using a downset leadframe upon which the die is mounted [see Figure 67(a) and Figure 67(b)]. This arrangement exposes the lead frame as a thermal pad on the underside of the package [see Figure 67(c)]. Because this pad has direct contact with the die, excellent thermal performance can be achieved by providing a good thermal path away from the pad. The PowerPAD package allows for both assembly and thermal management in one manufacturing operation. During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be soldered to a copper area underneath the package. Through the use of thermal paths within this copper area, heat can be conducted away from the package into either a ground plane or other heat dissipating device. The PowerPAD package represents a breakthrough in combining the small area requirement and ease of assembly of surface mount with the heretofore awkward mechanical methods of heatsinking. thermal information (continued) DIE Side View (a) Thermal Pad DIE End View (b) Bottom View (c) NOTE A: The thermal pad is electrically isolated from all terminals in the package. Figure 67. Views of Thermally Enhanced PWP Package 28 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION general PowerPAD design considerations Although there are many ways to properly heatsink this device, the following steps illustrate the recommended approach. THS7001 Thermal pad area (120 mils x 250 mils) (3,05 mm x 6,35 mm) with 8 vias Via diameter = 13 mils (0,33 mm) THS7002 Thermal pad area (120 mils x 300 mils) (3,05 mm x 7,62 mm) with 10 vias Via diameter = 13 mils (0,33 mm) Figure 68. PowerPAD PCB Etch and Via Pattern 1. Prepare the PCB with a top side etch pattern as shown in Figure 68. There should be etch for the leads as well as etch for the thermal pad. 2. Place the thermal transfer holes in the area of the thermal pad. These holes should be 13 mils (0,33 mm) in diameter. They are kept small so that solder wicking through the holes is not a problem during reflow. 3. Additional vias may be placed anywhere along the thermal plane outside of the thermal pad area. This helps dissipate the heat generated by the IC. These additional vias may be larger than the 13-mil (0,33 mm) diameter vias directly under the thermal pad. They can be larger because they are not in the thermal pad area to be soldered so that wicking is not a problem. 4. Connect all holes to the internal ground plane. 5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection methodology. Web connections have a high thermal resistance connection that is useful for slowing the heat transfer during soldering operations. This makes the soldering of vias that have plane connections easier. In this application, however, low thermal resistance is desired for the most efficient heat transfer. Therefore, the holes under the IC package should make their connection to the internal ground plane with a complete connection around the entire circumference of the plated-through hole. 6. The top-side solder mask should leave the terminals of the package and the thermal pad area with its thermal transfer holes exposed. The bottom-side solder mask should cover the thermal transfer holes of the thermal pad area. This prevents solder from being pulled away from the thermal pad area during the reflow process. 7. Apply solder paste to the exposed thermal pad area and all of the IC terminals. 8. With these preparatory steps in place, the THS7001PWP/THS7002PWP IC is simply placed in position and run through the solder reflow operation as any standard surface-mount component. This results in a part that is properly installed. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 29 

             SLOS214C − OCTOBER 1998 − REVISED MARCH 2007 APPLICATION INFORMATION general PowerPAD design considerations (continued)123456 The actual thermal performance achieved with the THS7001PWP/THS7002PWP in its PowerPAD package depends on the application. In the example above, if the size of the internal ground plane is approximately 3 inches × 3 inches (76,2 mm x 76.2 mm), then the expected thermal coefficient, θJA, is about 32.6°C/W for the THS7001 and 27.9_C/W for the THS7002. For a given θJA, the maximum power dissipation is shown in Figure 69 and is calculated by the following formula: P + D ǒ T Where: Ǔ –T MAX A q JA PD = Maximum power dissipation of THS7001 and THS7002 IC (watts) TMAX = Absolute maximum junction temperature (150°C) TA = Free-ambient air temperature (°C) θJA = θJC + θCA θJC = Thermal coefficient from junction to case (THS7001 = 1.4°C/W; THS7002 = 0.72°C/W) θCA = Thermal coefficient from case to ambient air (°C/W) THS7002 MAXIMUM POWER DISSIPATION vs FREE-AIR TEMPERATURE THS7001 MAXIMUM POWER DISSIPATION vs FREE-AIR TEMPERATURE 8 8 θJA = 27.9 _C/W 2 oz. Trace and Copper Pad With Solder TJ = 150 _C 6 5 4 3 2 1 0 −40 θJA = 56.2 _C/W 2 oz. Trace and Copper Pad Without Solder −20 0 20 40 60 80 θJA = 32.6 _C/W 2 oz. Trace and Copper Pad With Solder θJA = 74.4 _C/W 2 oz. Trace and Copper Pad Without Solder 7 Maximum Power Dissipation − W Maximum Power Dissipation − W 7 6 5 4 3 2 1 0 −40 100 TA − Free-Air Temperature − _C TJ = 150 _C −20 0 20 40 60 80 100 TA − Free-Air Temperature − _C NOTE A: Results are with no air flow and PCB size = 3”× 3” (76,2 mm x 76,2 mm) Figure 69. Maximum Power Dissipation vs Free-Air Temperature More complete details of the PowerPAD installation process and thermal management techniques can be found in the Texas Instruments technical brief, PowerPAD Thermally Enhanced Package. This document can be found at the TI web site (www.ti.com) by searching on the key word PowerPAD. The document can also be ordered through your local TI sales office. Refer to literature number SLMA002 when ordering. evaluation board An evaluation board is available both the THS7001 (literature number SLOP250) and for the THS7002 (literature number SLOP136). These boards has been configured for very low parasitic capacitance in order to realize the full performance of the amplifiers. These EVM’s incorporate DIP switches to demonstrate the full capabilities of the THS7001 and THS7002 independent of any digital control circuitry. For more information, please refer to the THS7001 EVM User’s Guide (literature number SLOU057)and the THS7002 EVM User’s Guide (literature number SLOU037). To order a evaluation board contact your local TI sales office or distributor. 30 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 PACKAGE OPTION ADDENDUM www.ti.com 6-Feb-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (°C) Device Marking (4/5) THS7001CPWP ACTIVE HTSSOP PWP 20 70 Green (RoHS & no Sb/Br) NIPDAU Level-2-260C-1 YEAR 0 to 70 THS7001C THS7001CPWPR ACTIVE HTSSOP PWP 20 2000 Green (RoHS & no Sb/Br) NIPDAU Level-2-260C-1 YEAR 0 to 70 THS7001C THS7001IPWP ACTIVE HTSSOP PWP 20 70 Green (RoHS & no Sb/Br) NIPDAU Level-2-260C-1 YEAR -40 to 85 THS7001I THS7002CPWP ACTIVE HTSSOP PWP 28 50 Green (RoHS & no Sb/Br) NIPDAU Level-3-260C-168 HR 0 to 70 THS7002C THS7002IPWP ACTIVE HTSSOP PWP 28 50 Green (RoHS & no Sb/Br) NIPDAU Level-3-260C-168 HR -40 to 85 THS7002I (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of