THS7314
SLOS513A – DECEMBER 2006 – REVISED MARCH 2011
www.ti.com
3-Channel SDTV Video Amplifier With 5th-Order Filters and 6-dB Gain
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FEATURES
1
•
•
•
•
•
•
•
•
•
DESCRIPTION
3 SDTV Video Amplifiers for CVBS, S-Video,
Y'P'BP'R 480i/576i, Y'U'V', or G'B'R' (R'G'B')
Integrated Low-Pass Filters
– 5th-Order 8.5-MHz (–3dB) Butterworth
– –1dB Passband Bandwidth at 7-MHz
– 47dB Attenuation at 27-MHz
Versatile Input Biasing
– DC-Coupled With 285-mV Output Shift
– AC-Coupled with Sync-Tip Clamp
– Allows AC-Coupled With DC-Biasing
Built-in 6dB Gain (2V/V)
3-V to 5-V Single Supply Operation
Rail-to-Rail Output:
– Output Swings Within 100-mV From the
Rails Allowing AC or DC Output Coupling
– Able to Drive up to 2 Video Lines – 75 Ω
Low 16-mA at 3.3-V Total Quiscent Current
Low Differential Gain/Phase of 0.1% / 0.1°
SOIC-8 Package
Fabricated using the Silicon-Germanium (SiGe)
BiCom-III process, the THS7314 is a low power
single-supply 3-V to 5-V 3-channel integrated video
buffer. It incorporates a 5th-order Butterworth filter
which is useful as a DAC reconstruction filter or an
ADC anti-aliasing filter. The 8.5-MHz filter is a perfect
choice for SDTV video which includes Composite
(CVBS), S-Video, Y'U'V', G'B'R' (R'G'B'), and Y'P'BP'R
480i/576i.
As part of the THS7314 flexibility, the input can be
configured for ac or dc coupled inputs. The 285-mV
output level shift to allow for a full sync dynamic
range at the output with 0-V input. The AC coupled
modes include a transparent sync-tip clamp option for
CVBS, Y', and G'B'R' signals with sync. AC-coupled
biasing for C'/P'B/P'R channels is achieved by adding
an external resistor to Vs+.
The THS7314 is the perfect choice for all output
buffer applications. Its rail-to-rail output stage with
6-dB gain allows for both ac and dc line driving. The
ability to drive 2-lines, or 75-Ω loads, allows for
maximum flexibility as a video line driver. The 16-mA
total quiescent current at 3.3-V makes it an excellent
choice for USB powered, portable, or other power
sensitive video applications.
APPLICATIONS
•
•
•
Set Top Box Output Video Buffering
PVR/DVDR Output Buffering
USB/Portable Low Power Video Buffering
The THS7314 is available in a small SOIC-8 package
that is RoHS compliant.
3.3 V
75 W
DAC/
Encoder
(THS8200)
SDTV
CVBS
S-Video Y’
S-Video C’
480i/576i
Y’P’BP’R
G’B’R’
CVBS
THS7314
R
Y’
R
75 W
1
CH.1 IN
CH.1 OUT
8
2
CH.2 IN
CH.2 OUT
7
3
CH.3 IN
CH.3 OUT
6
4
VS+
GND
5
75 W
3.3 V
Y’
Out
S-Video
C’
R
CVBS
Out
75 W
C’
Out
75 W
75 W
Figure 1. 3.3-V Single-Supply DC-Input/DC Output Coupled Video Line Driver
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
© 2006–2011, Texas Instruments Incorporated
THS7314
SLOS513A – DECEMBER 2006 – REVISED MARCH 2011
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
PACKAGING/ORDERING INFORMATION
PACKAGE TYPE (1)
PACKAGED DEVICES
THS7314D
Rails, 75
SOIC-8
THS7314DR
(1)
TRANSPORT MEDIA, QUANTITY
Tape and Reel, 2500
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
website at www.ti.com.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)
(1)
VALUE
UNIT
5.5
V
Supply voltage, VS+ to GND
VI
Input voltage
IO
Output current
–0.4 V to VS+
90
Continuous power dissipation
mA
See Dissipation Rating Table
TJ
Maximum junction temperature, any condition (2)
150
°C
TJ
Maximum junction temperature, continuous operation, long term reliability (3)
125
°C
Tstg
Storage temperature range
–65 to 150
°C
ESD ratings
(1)
(2)
(3)
HBM
2000
CDM
1500
MM
200
V
Stresses above those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute maximum rated conditions for extended periods may degrade device reliability.
The absolute maximum junction temperature under any condition is limited by the constraints of the silicon process.
The absolute maximum junction temperature for continuous operation is limited by the package constraints. Operation above this
temperature may result in reduced reliability and/or lifetime of the device.
DISSIPATION RATINGS
(1)
(2)
PACKAGE
θJC
(°C/W)
θJA
(°C/W)
SOIC-8 (D)
50
130 (2)
POWER RATING (1)
(TJ = 125°C)
TA = 25°C
TA = 85°C
769 mW
308 mW
Power rating is determined with a junction temperature of 125°C. This is the point where performance starts to degrade and long-term
reliability starts to be reduced. Thermal management of the final PCB should strive to keep the junction temperature at or below 125°C
for best performance and reliability.
This data was taken with the JEDEC High-K test PCB. For the JEDEC low-K test PCB, the θJA is 196°C/W.
RECOMMENDED OPERATING CONDITIONS
VS+
Supply voltage
TA
Ambient temperature
2
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MIN
MAX
3
5
UNIT
V
–40
85
°C
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SLOS513A – DECEMBER 2006 – REVISED MARCH 2011
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FUNCTIONAL DIAGRAM
+Vs
+
gm
Channel 1
Input
800 kW
Level
Shift
LPF
5-Pole
8.5-MHz
Sync-Tip
Clamp
(DC Restore)
6dB
Channel 1
Output
6dB
Channel 2
Output
6dB
Channel 3
Output
+Vs
+
gm
Channel 2
Input
800 kW
Level
Shift
LPF
5-Pole
8.5-MHz
Sync-Tip
Clamp
(DC Restore)
+Vs
+
gm
-
Channel 3
Input
800 kW
Level
Shift
Sync-Tip
Clamp
(DC Restore)
LPF
5-Pole
8.5-MHz
3 V to 5 V
PIN CONFIGURATION
SOIC-8 (D)
(TOP VIEW)
THS7314
CH.1 IN 1
8
CH.1 OUT
CH.2 IN 2
7
CH.2 OUT
CH.3 IN 3
6
CH.3 OUT
VS+ 4
5
GND
PIN FUNCTIONS
PIN
NAME
NO. SOIC-8
I/O
DESCRIPTION
CH. 1 – INPUT
1
I
Video Input – Channel 1
CH. 2 – INPUT
2
I
Video Input – Channel 2
CH. 3 – INPUT
3
I
Video Input – Channel 3
+Vs
4
I
Positive Power Supply Pin – connect to 3 V to 5 V.
GND
5
I
Ground Pin for all internal circuitry.
CH. 3 – OUTPUT
6
O
Video Output – Channel 3
CH. 2 – OUTPUT
7
O
Video Output – Channel 2
CH. 1 – OUTPUT
8
O
Video Output – Channel 1
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THS7314
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ELECTRICAL CHARACTERISTICS VS+ = 3.3 V:
RL = 150Ω to GND – Reference Figure 2 and Figure 3 (unless otherwise noted)
TYP
PARAMETER
TEST CONDITIONS
OVER TEMPERATURE
–40°C to
85°C
UNITS
MIN/
MAX
25°C
25°C
0°C to
70°C
8.5
6.7/10.3
6.6/10.5
6.5/10.6
MHz
Min/Max
8.5
6.7/10.3
6.6/10.5
6.5/10.6
MHz
Min/Max
AC PERFORMANCE
Small-signal bandwidth (–3dB)
Large-signal bandwidth (–3dB)
VO – 0.2 VPP
VO – 2 VPP
(1)
(1)
–1dB Passband bandwidth
Attenuation
7
MHz
Typ
Min/Max
f = 6 MHz (2)
0.45
–0.3/2.4
–0.35/2.5
–0.4/2.6
dB
36
35
34
f = 27 MHz (2)
47
dB
Min
Group delay
f = 100 kHz
57
ns
Typ
Group delay variation
with respect to 100kHz
f = 5.1 MHz
10.2
ns
Typ
0.3
ns
Typ
Differential gain
NTSC / PAL
0.1 / 0.1
%
Typ
Differential phase
NTSC / PAL
0.1 / 0.1
°
Typ
Total harmonic distortion
f = 1 MHz; VO = 2 VPP; AC coupled I/O
–66
dB
Typ
Signal to noise ratio
NTC-7 Weighting, 100kHz to 4.2MHz
79.6
dB
Typ
Channel-to-channel crosstalk
f = 1 MHz, Worst Case Channels
–60
dB
Typ
dB
Min/Max
Ω
Typ
With respect to 100kHz
Channel-to-channel delay
AC Gain – All channels
Output Impedance
6
f = 5 MHz
5.7/6.3
5.65/6.35
5.65/6.35
0.63
DC PERFORMANCE
Biased output voltage
VI = 0 V
Input voltage range
DC input, limited by output
Sync tip clamp charge current
VI = –0.1 V
285
Input resistance
Input capacitance
mV
Min/Max
–0.1/1.46
210/370
200/380
190/390
V
Typ
180
μA
Typ
800
kΩ
Typ
2
pF
Typ
V
Typ
OUTPUT CHARACTERISTICS
High output voltage swing
RL = 150 Ω to Midrail
3.15
RL = 150 Ω to GND
3.1
RL = 75 Ω to Midrail
RL = 75 Ω to GND
2.85
2.75
2.75
V
Min
3.1
V
Typ
Typ
3
V
RL = 150 Ω to Midrail (VI = –0.2V)
0.15
V
Typ
RL = 150 Ω to GND (VI = –0.2V)
0.1
V
Max
RL = 75 Ω to Midrail (VI = –0.2V)
0.3
V
Typ
RL = 75 Ω to GND (VI = –0.2V)
0.1
V
Typ
Output current (sourcing)
RL = 10 Ω to Midrail
80
mA
Typ
Output current (sinking)
RL = 10 Ω to Midrail
70
mA
Typ
Max
Low output voltage swing
0.17
0.2
0.21
POWER SUPPLY
Maximum operating voltage
See
(1)
3.3
5.5
5.5
5.5
V
Minimum operating voltage
See
(1)
3.3
2.85
2.85
2.85
V
Min
Maximum quiescent current
VI = 0V
16
20
22
24
mA
Max
Minimum quiescent current
VI = 0V
16
12
11.6
11
mA
Min
dB
Typ
Power Supply Rejection (+PSRR)
(1)
(2)
4
52
The Min/Max values listed for this specification are specified by design and characterization only.
3.3-V Supply Filter specifications are specified by 100% testing at 5-V supply along with design and characterization only.
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THS7314
SLOS513A – DECEMBER 2006 – REVISED MARCH 2011
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ELECTRICAL CHARACTERISTICS VS+ = 5 V:
RL = 150Ω to GND – Reference Figure 2 and Figure 3 (unless otherwise noted)
TYP
PARAMETER
TEST CONDITIONS
OVER TEMPERATURE
–40°C to
85°C
UNITS
MIN/
MAX
25°C
25°C
0°C to
70°C
8.5
6.7/10.3
6.6/10.5
6.5/10.6
MHz
Min/Max
8.5
6.7/10.3
6.6/10.5
6.5/10.6
MHz
Min/Max
AC PERFORMANCE
Small-signal bandwidth (–3dB)
Large-signal bandwidth (–3dB)
VO – 0.2 VPP
VO – 2 VPP
(1)
(1)
–1dB Passband bandwidth
Attenuation
With respect to 100kHz
7
MHz
Typ
Min/Max
f = 6 MHz
0.45
–0.3/2.4
–0.35/2.5
–0.4/2.6
dB
36
35
34
f = 27 MHz
47
dB
Min
Group delay
f = 100 kHz
57
ns
Typ
Group delay variation
with respect to 100kHz
f = 5.1 MHz
10
ns
Typ
0.3
ns
Typ
Differential gain
NTSC / PAL
0.1 / 0.1
%
Typ
Differential phase
NTSC / PAL
0.1 / 0.1
°
Typ
Total harmonic distortion
f = 1 MHz; VO =2 VPP
–66
dB
Typ
Signal to noise ratio
NTC-7 Weighting, 100kHz to 4.2MHz
79.6
dB
Typ
Channel-to-channel crosstalk
f = 1 MHz, Worst Case Channels
–60
dB
Typ
dB
Min/Max
Ω
Typ
Channel-to-channel delay
AC Gain – All channels
Output Impedance
6
f = 5 MHz
5.7/6.3
5.65/6.35
5.65/6.35
0.62
DC PERFORMANCE
Biased output voltage
VI = 0 V
Input voltage range
Limited by output
Sync tip clamp charge current
VI = –0.1V
290
Input resistance
Input capacitance
mV
Min/Max
–0.1/2.3
210/370
200/380
190/390
V
Typ
180
μA
Typ
800
kΩ
Typ
2
pF
Typ
V
Typ
OUTPUT CHARACTERISTICS
RL = 150 Ω to Midrail
4.85
RL = 150 Ω to GND
4.7
RL = 75 Ω to Midrail
V
Min
4.7
V
Typ
RL = 75 Ω to GND
4.5
V
Typ
RL = 150 Ω to Midrail (VI = –0.2V)
0.2
V
Typ
RL = 150 Ω to GND (VI = –0.2V)
0.12
V
Max
RL = 75 Ω to Midrail (VI = –0.2V)
0.35
V
Typ
RL = 75 Ω to GND (VI = –0.2V)
0.1
V
Typ
Output current (sourcing)
RL = 10 Ω to Midrail
90
mA
Typ
Output current (sinking)
RL = 10 Ω to Midrail
85
mA
Typ
Max
High output voltage swing
Low output voltage swing
4.2
0.23
4.1
0.26
4.1
0.27
POWER SUPPLY
Maximum operating voltage
See
(1)
5
5.5
5.5
5.5
V
Minimum operating voltage
See
(1)
5
2.85
2.85
2.85
V
Min
Maximum quiescent current
VI = 0V
17
22
24
25
mA
Max
Minimum quiescent current
VI = 0V
17
12.5
12
11.5
mA
Min
dB
Typ
Power Supply Rejection (+PSRR)
(1)
55
The Min/Max values listed for this specification are specified by design and characterization only.
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THS7314
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DUT
R TERM
RSOURCE
VSOURCE
R LOAD
R TERM
1 CH.1 IN
CH.1 OUT
8
2 CH.2 IN
CH.2 OUT
7
3 CH.3 IN
CH.3 OUT
6
GND
5
4 VS +
R LOAD
0.1 mF
R LOAD
R TERM
+
100 mF
+VS
Figure 2. DC Coupled Input and Output Test Circuit
470 mF
+
C IN
DUT
R LOAD
RTERM
CIN
RSOURCE
VSOURCE
RTERM
CIN
1
CH.1 IN
CH.1 OUT
8
2
CH.2 IN
CH.2 OUT
7
3
CH.3 IN
CH.3 OUT
6
4
VS +
GND
5
0.1 mF
0.1 mF
470 mF
+
R LOAD
0.1 mF
470 mF
+
RTERM
RLOAD
+
+VS
100 mF
0.1 mF
Figure 3. AC Coupled Input and Output Test Circuit
6
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TYPICAL CHARACTERISTICS
SMALL-SIGNAL GAIN vs FREQUENCY
PHASE vs FREQUENCY
10
45
0
−45
−10
−90
Phase − °
−20
RL = 150 Ω || 13 pF
RL = 75 Ω || 13 pF
−30
−135
−180
−225
−40
−270
−50
VS = 3.3 V
VO = 200 mVPP
−60
0.1
−315
1
10
100
−360
0.1
1k
f − Frequency − MHz
VS = 3.3 V
VO = 200 mVPP
RL = 150 Ω || 13 pF
1
6.0
85
80
Group Delay − ns
5.5
RL = 75 Ω || 13 pF
4.5
4.0
3.5
55
1
10
G003
SMALL-SIGNAL PULSE RESPONSE
vs
TIME
LARGE-SIGNAL PULSE RESPONSE
vs
TIME
0.6
Input
tr/tf = 140 ns
Input
tr/tf = 1 ns
0.5
0.4
0.3
100 200 300 400 500 600 700 800 900
tr/tf = 140 ns
3.5
tr/tf = 1 ns
3.0
1.5
0.5
0.0
2.5
−0.5
Input
tr/tf = 140 ns
2.0
1.5
Input
tr/tf = 1 ns
1.0
0.0
−100
VS = 3.3 V
RL = 150 Ω || 13 pF
1.0
4.0
0.5
Output Voltage Waveforms
t − Time − ns
VO − Output Voltage − V
tr/tf = 1 ns
Input Voltage Waveforms
4.5
tr/tf = 140 ns
G004
2.0
5.0
VS = 3.3 V
RL = 150 Ω || 13 pF
0.7
1.9
100
f − Frequency − MHz
0.8
0
60
Figure 7.
Input Voltage Waveforms
1.5
−100
65
Figure 6.
2.0
1.6
70
40
0.1
10
f − Frequency − MHz
1.7
75
45
1
1.8
VS = 3.3 V
VO = 200 mVPP
RL = 150 Ω || 13 pF
50
VS = 3.3 V
VO = 200 mVPP
2.5
0.1
VO − Output Voltage − V
GROUP DELAY vs FREQUENCY
90
VI − Input Voltage − V
Small-Signal Gain − dB
SMALL-SIGNAL GAIN vs FREQUENCY
3.0
G002
Figure 5.
6.5
RL = 150 Ω || 13 pF
100
f − Frequency − MHz
G001
Figure 4.
5.0
10
Output Voltage Waveforms
0
G009
Figure 8.
−1.0
−1.5
VI − Input Voltage − V
Small-Signal Gain − dB
0
−2.0
−2.5
−3.0
100 200 300 400 500 600 700 800 900
t − Time − ns
G011
Figure 9.
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TYPICAL CHARACTERISTICS (continued)
HD2 vs OUTPUT VOLTAGE
HD3 vs OUTPUT VOLTAGE
−45
−50
VS = 3.3 V
RL = 150 Ω || 13 pF
3rd Order Harmonic Distortion − dB
2nd Order Harmonic Distortion − dB
−40
f = 2 MHz
−50
f = 4 MHz
−55
−60
−65
−70
−75
−80
0.0
f = 1 MHz
0.5
1.0
1.5
2.0
2.5
VO − Output Voltage − VPP
−55
VS = 3.3 V
RL = 150 Ω || 13 pF
−60
−65
f = 2 MHz
f = 4 MHz
−70
−75
−80
−85
f = 1 MHz
−90
0.0
3.0
0.5
1.0
G013
SLEW RATE vs OUTPUT VOLTAGE
G014
CROSSTALK vs FREQUENCY
VS = 3.3 V
RL = 150 Ω || 13 pF
−40
Crosstalk − dB
SR − Slew Rate − V/µs
3.0
−30
35
30
25
20
15
10
VS = 3.3 V
VO = 1 VPP
RL = 150 Ω || 13 pF
Ch.1 Ch.2
−50
Ch.1 Ch.3
−60
−70
−80
5
0
0.5
2.5
Figure 11.
50
40
2.0
VO − Output Voltage − VPP
Figure 10.
45
1.5
Ch.2 Ch.3
1.0
1.5
2.0
−90
0.1
2.5
VO − Output Voltage − VPP
1
10
100
f − Frequency − MHz
G017
Figure 12.
1k
G019
Figure 13.
SMALL-SIGNAL GAIN vs FREQUENCY
OUTPUT IMPEDANCE vs FREQUENCY
10
100
VS = 3.3 V
ZO − Output Impedance − Ω
Small-Signal Gain − dB
0
CL = 5 pF
−10
−20
CL = 13 pF
−30
CL = 27 pF
−40
−50
−60
0.1
VS = 3.3 V
VO = 200 mVPP
RL = 150 Ω || CL
1
10
f − Frequency − MHz
100
1k
10
1
0.1
0.1
G021
Figure 14.
8
1
10
f − Frequency − MHz
100
G023
Figure 15.
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TYPICAL CHARACTERISTICS (continued)
SMALL-SIGNAL GAIN vs FREQUENCY
PHASE vs FREQUENCY
10
45
0
−45
−10
−90
Phase − °
−20
RL = 150 Ω || 13 pF
RL = 75 Ω || 13 pF
−30
−135
−180
−225
−40
−270
−50
VS = 5 V
VO = 200 mVPP
−60
0.1
−315
1
10
100
−360
0.1
1k
f − Frequency − MHz
VS = 5 V
VO = 200 mVPP
RL = 150 Ω || 13 pF
1
6.0
85
80
Group Delay − ns
5.5
RL = 75 Ω || 13 pF
4.5
4.0
3.5
55
1
10
G007
SMALL-SIGNAL PULSE RESPONSE
vs
TIME
LARGE-SIGNAL PULSE RESPONSE
vs
TIME
tr/tf = 1 ns
1.0
2.6
Input
tr/tf = 140 ns
Input
tr/tf = 1 ns
0.9
0.8
tr/tf = 140 ns
4.5
tr/tf = 1 ns
4.0
2.0
1.0
0.5
3.5
0.0
3.0
Input
tr/tf = 140 ns
2.5
Input
tr/tf = 1 ns
2.0
1.0
−100
0.7
100 200 300 400 500 600 700 800 900
VS = 5 V
RL = 150 Ω || 13 pF
1.5
5.0
1.5
Output Voltage Waveforms
t − Time − ns
Input Voltage Waveforms
5.5
1.1
G008
2.5
6.0
VS = 5 V
RL = 150 Ω || 13 pF
VO − Output Voltage − V
tr/tf = 140 ns
100
f − Frequency − MHz
1.2
0
60
Figure 19.
Input Voltage Waveforms
2.3
−100
65
Figure 18.
2.8
2.4
70
40
0.1
10
f − Frequency − MHz
2.5
75
45
1
2.7
VS = 5 V
VO = 200 mVPP
RL = 150 Ω || 13 pF
50
VS = 5 V
VO = 200 mVPP
2.5
0.1
VO − Output Voltage − V
GROUP DELAY vs FREQUENCY
90
VI − Input Voltage − V
Small-Signal Gain − dB
SMALL-SIGNAL GAIN vs FREQUENCY
3.0
G006
Figure 17.
6.5
RL = 150 Ω || 13 pF
100
f − Frequency − MHz
G005
Figure 16.
5.0
10
Output Voltage Waveforms
0
G010
Figure 20.
−0.5
−1.0
VI − Input Voltage − V
Small-Signal Gain − dB
0
−1.5
−2.0
−2.5
100 200 300 400 500 600 700 800 900
t − Time − ns
G012
Figure 21.
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TYPICAL CHARACTERISTICS (continued)
HD2 vs OUTPUT VOLTAGE
HD3 vs OUTPUT VOLTAGE
−45
−50
VS = 5 V
RL = 150 Ω || 13 pF
−50
3rd Order Harmonic Distortion − dB
2nd Order Harmonic Distortion − dB
−40
f = 2 MHz
f = 4 MHz
−55
−60
−65
−70
f = 1 MHz
−75
−80
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
VO − Output Voltage − VPP
−55
VS = 5 V
RL = 150 Ω || 13 pF
−60
−65
f = 2 MHz
f = 4 MHz
−70
−75
−80
−85
f = 1 MHz
−90
0.0
4.5
0.5
1.0
1.5
G015
Figure 22.
SLEW RATE vs OUTPUT VOLTAGE
3.5
4.0
4.5
G016
CROSSTALK vs FREQUENCY
VS = 5 V
RL = 150 Ω || 13 pF
−40
Crosstalk − dB
SR − Slew Rate − V/µs
3.0
−30
50
40
30
20
VS = 5 V
VO = 1 VPP
RL = 150 Ω || 13 pF
Ch.1 Ch.2
−50
Ch.1 Ch.3
−60
−70
−80
10
0
0.5
2.5
Figure 23.
70
60
2.0
VO − Output Voltage − VPP
Ch.2 Ch.3
1.0
1.5
2.0
2.5
3.0
VO − Output Voltage − VPP
−90
0.1
3.5
1
10
100
f − Frequency − MHz
G018
Figure 24.
1k
G020
Figure 25.
SMALL-SIGNAL GAIN vs FREQUENCY
OUTPUT IMPEDANCE vs FREQUENCY
10
100
VS = 5 V
ZO − Output Impedance − Ω
Small-Signal Gain − dB
0
CL = 5 pF
−10
−20
CL = 13 pF
−30
CL = 27 pF
−40
−50
−60
0.1
VS = 5 V
VO = 200 mVPP
RL = 150 Ω || CL
1
10
f − Frequency − MHz
100
1k
10
1
0.1
0.1
G022
Figure 26.
10
1
10
f − Frequency − MHz
100
G024
Figure 27.
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APPLICATION INFORMATION
The THS7314 is targeted for standard definition video output buffer applications. Although it can be used for
numerous other applications, the needs and requirements of the video signal is an important design parameter of
the THS7314. Built on the Silicon Germanium (SiGe) BiCom-3 process, the THS7314 incorporates many
features not typically found in integrated video parts while consuming very low power.
The THS7314 has the following features:
• Single-Supply 3-V to 5-V operation with low total quiescent current of 16-mA at 3.3-V and 17-mA at 5-V.
• Input configuration accepting DC + Level shift, AC Sync-Tip Clamp, or AC-Bias.
• AC-Biasing is accomplished with the use of external pull-up resistor to the positive power supply.
• 5th-Order Low Pass Filter for DAC reconstruction or ADC image rejection:
– 8.5-MHz for NTSC, PAL, SECAM, Composite (CVBS), S-Video Y'C', 480i/576i Y'P'BP'R , and G'B'R'
(R'G'B') signals.
• Internal fixed gain of 2 V/V (6 dB) buffer that can drive up to 2 video lines with dc coupling or traditional ac
coupling.
• Signal flow-through configuration using an 8-pin SOIC package that complies with the latest lead-free (RoHS
compatible) and Green manufacturing requirements.
OPERATING VOLTAGE
The THS7314 is designed to operate from 3-V to 5-V over a –40°C to 85°C temperature range. The impact on
performance over the entire temperature range is negligible due to the implementation of thin film resistors and
high quality – low temperature coefficient capacitors. The design of the THS7314 allows operation down to
2.85-V, but for best results the use of a 3-V supply or greater should be used to ensure there are no issues with
headroom or clipping.
The power supply pins should have a 0.1 μF to 0.01 μF capacitor placed as close as possible to these pins.
Failure to do so may result in the THS7314 outputs ringing or have an oscillation. Additionally, a large capacitor,
such as 22 μF to 100 μF, should be placed on the power supply line to minimize interference with 50/60 Hz line
frequencies.
INPUT VOLTAGE
The THS7314 input range allows for an input signal range from –0.2V to about (Vs+ – 1.5V). But, due to the
internal fixed gain of 2 V/V (6 dB) and the internal level shift of nominally 145-mV, the output will generally be the
limiting factor for the allowable linear input range. For example, with a 5-V supply, the linear input range is
from –0.2V to 3.5V. But due to the gain and level shift, the linear output range limits the allowable linear input
range to be from about –0.1V to 2.3V .
INPUT OVERVOLTAGE PROTECTION
The THS7314 is built using a very high-speed complementary bipolar and CMOS process. The internal junction
breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in
the Absolute Maximum Ratings table. All input and output device pins are protected with internal ESD protection
diodes to the power supplies, as shown in Figure 28.
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+ Vs
External
Input/
Output
Pin
Internal
Circuitry
Figure 28. Internal ESD Protection
These diodes provide moderate protection to input overdrive voltages above and below the supplies as well. The
protection diodes can typically support 30-mA of continuous current when overdriven.
TYPICAL CONFIGURATION and VIDEO TERMINOLOGY
A typical application circuit using the THS7314 as a video buffer is shown in Figure 29. It shows a DAC (or
encoder such as the THS8200) driving the three input channels of the THS7314. Although the S-Video Y'C'
channels and the Composite video channel of a Standard Definition video (SD) system are shown, these
channels can easily be the Y'P'BP'R (sometimes labeled Y'U'V' or incorrectly labeled Y'C'BC'R) signals of a 480i or
576i system. These signals can also be G'B'R' (R'G'B') signals or other variations.
Note that the Y' term is used for the luma channels throughout this document rather than the more common
luminance (Y) term. The reason is to account for the definition of luminance as stipulated by the CIE –
International Commission on Illumination. Video departs from true luminance since a nonlinear term, gamma, is
added to the true RGB signals to form R'G'B' signals. These R'G'B' signals are then used to mathematically
create luma (Y'). Thus luminance (Y) is not maintained providing a difference in terminology.
This rationale is also used for the chroma (C') term. Chroma is derived from the non-linear R'G'B' terms and thus
it is nonlinear. Chominance (C) is derived from linear RGB giving the difference between chroma (C') and
chrominance (C). The color difference signals (P'B / P'R / U' / V') are also referenced this way to denote the
nonlinear (gamma corrected) signals.
R'G'B' (commonly mislabeled RGB) is also called G’B’R’ (again commonly mislabeled as GBR) in professional
video systems. The SMPTE component standard stipulates that the luma information is placed on the first
channel, the blue color difference is placed on the second channel, and the red color difference signal is placed
on the third channel. This is consistent with the Y'P'BP'R nomenclature. Because the luma channel (Y') carries the
sync information and the green channel (G') also carries the sync information, it makes logical sense that G' be
placed first in the system. Since the blue color difference channel (P'B) is next and the red color difference
channel (P'R) is last, then it also makes logical sense to place the B' signal on the second channel and the R'
signal on the third channel respectfully. Thus hardware compatibility is better achieved when using G'B'R' rather
than R'G'B'. Note that for many G'B'R' systems sync is embeded on all three channels, but may not always be
the case in all systems.
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3.3 V
DAC/
Encoder
SDTV
CVBS
S-Video Y’
S-Video C’
480i/576i
Y’P’BP’R
G’B’R’
330 mF
+
CVBS
THS 7314
R
Y’
R
75 W
CVBS
Out
75 W
1
CH.1 IN
CH.1 OUT
8
2
CH.2 IN
CH.2 OUT
7
3
CH.3 IN
CH.3 OUT
6
4
VS+
GND
5
C’
330 mF
+
75 W
Y’
Out
S-Video
0.1 mF
0.1 mF
R
75 W
C’
Out
75 W
75 W
+
3 V to 5 V
22 mF
Figure 29. Typical SDTV CVBS/Y'/C' Inputs From DC-Coupled Encoder/DAC
With AC-Coupled Line Driving
INPUT MODE OF OPERATION – DC
The inputs to the THS7314 allows for both ac-coupled and dc-coupled inputs. Many DACs or Video Encoders
can be dc connected to the THS7314. One of the drawbacks to dc coupling is when 0-V is applied to the input.
Although the input of the THS7314 allows for a 0-V input signal with no issues, the output swing of a traditional
amplifier cannot yield a 0-V signal resulting in possible clipping. This is true for any single-supply amplifier due to
the limitations of the output transistors. Both CMOS and bipolar transistors cannot go to 0-V while sinking
current. This trait of a transistor is also the same reason why the highest output voltage is always less than the
power supply voltage when sourcing current.
This output clipping can reduce the sync amplitudes (both horizontal and vertical sync amplitudes) on the video
signal. A problem occurs if the receiver of this video signal uses an AGC loop to account for losses in the
transmission line. Some video AGC circuits derive gain from the horizontal sync amplitude. If clipping occurs on
the sync amplitude, then the AGC circuit can increase the gain too much – resulting in too much luma and/or
chroma amplitude gain correction. This may result in a picture with an overly bright display with too much color
saturation.
Other AGC circuits use the chroma burst amplitude for amplitude control, and a reduction in the sync signals
does not alter the proper gain setting. But, it is good engineering design practice to ensure saturation/clipping
does not take place. Transistors always take a finite amount of time to come out of saturation. This saturation
could possibly result in timing delays or other aberrations on the signals.
To eliminate saturation/clipping problems, the THS7314 has a 145-mV input level shift feature. This feature takes
the input voltage and adds an internal +145-mV shift to the signal. Since the THS7314 also has a gain of 6 dB (2
V/V), the resulting output with a 0-V applied input signal is about 290-mV. The THS7314 rail-to-rail output stage
can create this output level while connected to a typical video load with AC or DC coupling. This ensures that no
saturation / clipping of the sync signals occur. This is a constant shift regardless of the input signal. For example,
if a 1-V input is applied, the output is at 2.29-V.
Because the internal gain is fixed at 6 dB, the gain dictates what the allowable linear input voltage range can be
without clipping concerns. For example, if the power supply is set to 3-V, the maximum output is about 2.9-V
while driving a significant amount of current. Thus, to avoid clipping, the allowable input is ((2.9V / 2) – 0.145V) =
1.305V. This is true for up to the maximum recommended 5-V power supply that allows about a ((4.9V / 2) –
0.145V) = 2.305V input range while avoiding clipping on the output.
The input impedance of the THS7314 in this mode of operation is dictated by the internal 800-kΩ pull-down
resistor. This is shown in Figure 30. Note that the internal voltage shift does not appear at the input pin, only the
output pin.
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+ Vs
Internal
Circuitry
Input
+
800 kW
-
145 mV Level
Shifter
Figure 30. Equivalent DC Input Mode Circuit
INPUT MODE OF OPERATION – AC SYNC TIP CLAMP
Some video DACs or encoders are not referenced to ground but rather to the positive power supply. These
DACs typically only sink current rather than the more traditional current sourcing DAC where the resistor is
referenced to ground. The resulting video signals can be too high of a voltage for a dc-coupled video buffer to
function properly. To account for this scenario the THS7314 incorporates a sync-tip clamp circuit. This function
requires a capacitor (nominally 0.1 μF) to be in series with the input. Note that while the term sync-tip-clamp is
used throughout this document, it should be noted that the THS7314 is be better termed as a dc-restoration
circuit based on how this function is performed. This circuit is an active clamp circuit and not a passive diode
clamp function.
The input to the THS7314 has an internal control loop which sets the lowest input applied voltage to clamp at
ground (0-V). By setting the reference at 0-V, the THS7314 allows a dc-coupled input to also function. Hence the
STC is considered transparent since it does not operate unless the input signal goes below ground. The signal
then goes thru the same 145-mV level shifter resulting in an output voltage low level of 290-mV. If the input
signal tries to go below the 0-V, the internal control loop of the THS7314 will source up to 3-mA of current to
increase the input voltage level on the THS7314 input side of the coupling capacitor. As soon as the voltage
goes above the 0-V level, the loop stops sourcing current and becomes high impedance.
One of the concerns about the sync-tip-clamp level is how the clamp reacts to a sync edge that has
overshoot—common in VCR signals or reflections found in poor PCB layouts. Ideally the STC should not react to
the overshoot voltage of the input signal. Otherwise, this could result in clipping on the rest of the video signal as
it may raise the bias voltage too much.
To help minimize this input signal overshoot problem, the control loop in the THS7314 has an internal low-pass
filter as shown in Figure 31. This filter reduces the response time of the STC circuit. This delay is a function of
how far the voltage is below ground, but in general it is about a 100-ns delay. The effect of this filter is to slow
down the response of the control loop so as not to clamp on the input overshoot voltage but rather the flat portion
of the sync signal.
As a result of this delay, the sync may have an apparent voltage shift. The amount of shift is dependant upon the
amount of droop in the signal as dictated by the input capacitor and the STC current flow. Because the sync is
primarily for timing purposes with syncing occurring on the edge of the sync signal, this shift is transparent in
most systems.
While this feature may not fully eliminate overshoot issues on the input signal for excessive overshoot and/or
ringing, the STC system should help minimize improper clamping levels. As an additional method to help
minimize this issue, an external capacitor (ex: 10 pF to 47 pF) to ground in parallel with the external termination
resistors can help filter overshoot problems.
It should be noted that this STC system is dynamic and does not rely upon timing in any way. It only depends on
the voltage appearing at the input pin at any given point in time. The STC filtering helps minimize level shift
problems associated with switching noises or very short spikes on the signal line. This helps ensure a very
robust STC system.
14
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+Vs
+Vs
Comparator
STC LPF
Internal
Circuitry
+
Input
Pin
-
Input
+
0.1 mF
800 kW
-
145 mV Level
Shifter
Figure 31. Equivalent AC Sync Tip Clamp Input Circuit
When the AC Sync-Tip-Clamp (STC) operation is used, there must also be some finite amount of discharge bias
current. As previously described, if the input signal goes below the 0-V clamp level, the internal loop of the
THS7314 will source current to increase the voltage appearing at the input pin. As the difference between the
signal level and the 0-V reference level increases, the amount of source current increases
proportionally—supplying up to 3 mA of current. Thus the time to re-establish the proper STC voltage can be
fast. If the difference is small, then the source current is also small to account for minor voltage droop.
But, what happens if the input signal goes above the 0-V input level? The problem is the video signal is always
above this level, and must not be altered in any way. But if the Sync level of the input signal is above this 0-V
level, then the internal discharge (sink) current will discharge the ac-coupled bias signal to the proper 0-V level.
This discharge current must not be large enough to alter the video signal appreciably, or picture quality issues
may arise. This is often seen by looking at the tilt (aka droop) of a constant luma signal being applied and looking
at the resulting output level. The associated change in luma level from the beginning of the video line to the end
of the video line is the amount of line tilt (droop).
If the discharge current is small, the amount of tilt is low which is good. But, the amount of time for the system to
capture the sync signal could be too long. This is also termed hum rejection. Hum arises from the AC line voltage
frequency of 50-Hz or 60-Hz. The value of the discharge current and the AC-coupling capacitor combine to
dictate the hum rejection and the amount of line tilt.
To allow for both dc-coupling and ac-coupling in the same part, the THS7314 incorporates an 800-kΩ resistor to
ground. Although a true constant current sink is preferred over a resistor, there are significant issues when the
voltage is near ground. This can cause the current sink transistor to saturate and cause potential problems with
the signal. This resistor is large enough as to not impact a dc-coupled DAC termination. For discharging an
ac-coupled source, Ohm’s Law is used. If the video signal is 1 V, then there is 1 V / 800 kΩ = 1.25-μA of
discharge current. If more hum rejection is desired or there is a loss of sync occurring, then decrease the 0.1-μF
input coupling capacitor. A decrease form 0.1 μF to 0.047 μF increases the hum rejection by a factor of 2.1.
Alternatively an external pull-down resistor to ground may be added which decreases the overall resistance, and
ultimately increases the discharge current.
To ensure proper stability of the AC STC control loop, the source impedance must be less than 1-kΩ with the
input capacitor in place. Otherwise, there is a possibility of the control loop to ring and this ringing may appear on
the output of the THS7314. Because most DACs or encoders use resistors to establish the voltage, which are
typically less than 300-Ω, then meeting the 1 MHz) signals
with much lower impedance than the large capacitors.
Although it is common to use the same capacitor values for all the video lines, the frequency bandwidth of the
chroma signal in a S-Video system are not required to go as low – or as high of a frequency – as the luma
channels. Thus the capacitor values of the chroma line(s) can be smaller – such as 0.1-μF.
330 mf
(Note A) 75 W
Y’
Out 1
+
75 W
330 mf
(Note A) 75 W
3.3 V
DAC/
Encoder
75 W
0.1 mF
330 mf
Y’
3.3 V
P’ B
3.3 V
3.01 MW
0.1 mF
R
SDTV
480i/576i
Y’P’BP’R
G’B’R’
+
3.3 V
R
3.3 V
R
Y’
Out 2
3.3 V
3.01 MW
0.1 mF
(Note A)
THS7314
+
1
CH.1 IN
CH.1 OUT
8
2
CH.2 IN
CH.2 OUT
7
3
CH.3 IN
CH.3 OUT
6
4
VS+
GND
5
P’R
75 W
P’ B
Out 1
75 W
330 mf
(Note A) 75 W
P’B
Out 2
+
75 W
0.1 mF
330 mf
+
3.3 V
(Note A) 75 W
P’ R
Out 1
+
22 mF
75 W
330 mf
(Note A) 75 W
P’R
Out 2
+
75 W
A.
Due to the high frequency content of the video signal, it is recommended, but not required, to add a 0.1-μF or 0.01-μF
capacitor in parallel with these large capators.
B.
Current sinking DAC / Encoder shown. See the application notes.
Figure 34. Typical 480i/576i Y'P' BP' R AC-Input System Driving 2 AC-Coupled Video Lines
18
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LOW PASS FILTER
Each channel of the THS7314 incorporates a 5th-Order Low Pass Filter. These video reconstruction filters
minimize DAC images from being passed onto the video receiver. Depending on the receiver design, failure to
eliminate these DAC images can cause picture quality problems due to aliasing of the ADC. Another benefit of
the filter is to smooth out aberrations in the signal which some DACs can have if their own internal filtering is not
good. This helps with picture quality and helps insure the signal meets video bandwidth requirements.
Each filter has a Butterworth characteristic associated with it. The benefit of the Butterworth response is the
frequency response is flat with a relatively steep initial attenuation at the corner frequency. The problem is that
the group delay rises near the corner frequency. Group delay is defined as the change in phase (radians/second)
divided by a change in frequency. An increase in group delay corresponds to a time domain pulse response that
has overshoot and some possible ringing associated with the overshoot.
The use of other type of filters, such as elliptic or chebyshev, are not recommended for video applications due to
their very large group delay variations near the corner frequency resulting in significant overshoot and ringing.
While these elliptic or chebyshev filters may help meet the video standard specifications with respect to
amplitude attenuation, their group delay is well beyond the standard specifications. Couple this with the fact that
video can go from a white pixel to a black pixel over and over again, ringing can easily occur. Ringing typically
causes a display to have ghosting or fuzziness appear on the edges of a sharp transition. On the other hand, a
Bessel filter has ideal group delay response, but the rate of attenuation is typically too low for acceptable image
rejection. Thus the Butterworth filter is a respectable compromise for both attenuation and group delay.
The THS7314 filter has a slightly lower group delay variation near the corner frequency compared to an ideal
Butterworth filter. This results in a time domain pulse response which still has some overshoot, but not as much
as a true Butterworth filter. Additionally, the initial rate of attenuation in the frequency response is not quite as
fast as an ideal Butterworth response, but it is an acceptable initial rate of attenuation considering the pulse and
group delay characteristic benefits. The THS7314 still achieves 47-dB of attenuation at 27-MHz, which typically
exceeds most SD video requirements.
The THS7314 filters have a nominal corner (-3dB) frequency at 8.5-MHz and a –1 dB passband typically at
7-MHz. This 8.5-MHz filter is ideal for Standard Definition (SD) NTSC, PAL, and SECAM composite video
(CVBS) signals. It is also useful for S-Video signals (Y'C'), 480i/576i Y'P'BP'R, Y'U'V', broadcast G’B’R’ (R’G’B’)
signals, and computer video signals. The 8.5-MHz -3dB corner frequency was designed to allow a maximally flat
video signal while achieving 47-dB of attenuation at 27-MHz – a common sampling frequency between the
DAC/ADC 2nd and 3rd Nyquist zones found in many video systems. This is important because any signal
appearing around this frequency can appear in the baseband due to aliasing effects of an analog to digital
converter found in a receiver.
Keep in mind that images do not stop at 27-MHz, they continue around the sampling frequencies of 54-MHz,
81-MHz, 108-MHz, etc. Because of these multiple images that an ADC can fold down into the baseband signal,
the low pass filter must also eliminate these higher order images. The THS7314 has over 70-dB attenuation at
54-MHz, 68-dB attenuation at 81-MHz, and over 60-dB attenuation at 108-MHz. Attenuation above 108-MHz is at
least 55-dB which makes sure that images do not effect the desired video baseband signal.
The 8.5-MHz filter frequency was chosen to account for process variations in the THS7314. To ensure the
required video frequencies are effectively passed, the filter corner frequency must be high enough to allow
component variations. The other consideration is the attenuation must be large enough to ensure the
anti-aliasing / reconstruction filtering is enough to meet the system demands. Thus, the selection of the filter
frequencies was not arbitrarily selected and is a good compromise that should meet the demands of most
systems.
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BENEFITS OVER PASSIVE FILTERING
Two key benefits of using an integrated filter system, such as the THS7314, over a passive system is PCB area
and filter variations. The small SOIC-8 package for 3-video channels is much smaller over a passive RLC
network, especially a 5-pole passive network. Add in the fact that inductors have at best ±10% tolerances
(normally ±15% to ±20% is common) and capacitors typically have ±10% tolerances. Using a Monte Carlo
analysis shows that the filter corner frequency (–3 dB), flatness (–1 dB), Q factor (or peaking), and
channel-to-channel delay will have wide variations. This can lead to potential performance and quality issues in
mass-production environments. The THS7314 solves most of these problems with only the corner frequency
being essentially the only variable.
One concern about an active filter in an integrated circuit is the variation of the filter characteristics when the
ambient temperature and the subsequent die temperature changes. To minimize temperature effects, the
THS7314 uses low temperature coefficient resistors and high quality – low temperature coefficient capacitors
found in the BiCom-3 process. The filters have been specified by design to account for process variations and
temperature variations to maintain proper filter characteristics. This maintains a low channel-to-channel time
delay which is required for proper video signal performance.
Another benefit of a THS7314 over a passive RLC filter are the input and output impedances. The input
impedance presented to the DAC will vary significantly with a passive network and may cause voltage variations
over frequency. The THS7314 input impedance is 800-kΩ and only the 2-pF input capacitance plus the PCB
trace capacitance impacting the input impedance. As such, the voltage variation appearing at the DAC output is
better controlled with the THS7314.
On the output side of the filter, a passive filter will again have a impedance variation over frequency. The
THS7314 is an op-amp which approxiamates an ideal voltage source. A voltage source is desirable because the
output impedance is very low and can source and sink current. To properly match the transmission line
characteristic impedance of a video line, a 75-Ω series resistor is placed on the output. To minimize reflections
and to maintain a good return loss, this output impedance must maintain a 75-Ω impedance. A passive filter
impedance variation cannot guarantee this while the THS7314 has about 0.6-Ω of output impedance at 5.1-MHz.
Thus, the system is matched much better with a THS7314 compared to a passive filter.
One last benefit of the THS7314 over a passive filter is power dissipation. A DAC driving a video line must be
able to drive a 37.5-Ω load - the reciever 75-Ω resistor and the 75-Ω impedance matching resistor next to the
DAC to maintain the source impedance requirement. This forces the DAC to drive at least 1.25Vpeak (100%
Saturation CVBS) / 37.5Ω = 33.3mA. A DAC is a current steering element and this amount of current flows
internally to the DAC even if the output is 0-V. Thus, power dissipation in the DAC may be very high - especially
when 6-channels are being driven. Using the THS7314, with a high input impedance and the capability to drive
up to 2-video lines, can reduce the DAC power dissipation significantly. This is because the resistance the DAC
is driving can be substantially increased. It is common to set this in a DAC by a current setting resistor on the
DAC. Thus, the resistance can be 300-Ω or more - substantially reducing the current drive demands from the
DAC and saving substantial amount of power. For example, a 3.3-V 6-Channel DAC dissipates 660mW just for
the steering current capability (6ch X 33.3mA X 3.3V) if it needs to drive 37.5-Ω load. With a 300-Ω load, the
DAC power dissipation due to current steering current would only be 82.5mW (6ch X 4.16mA X 3.3V).
20
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EVALUATION MODULE
To evaluate the THS7314, an evaluation module (EVM) is available. This allows for testing of the THS7314 in
many different systems. Inputs and outputs include RCA connectors for consumer grade interconnections, or
BNC connectors for higher level lab grade connections. Several unpopulated component pads are found on the
EVM to allow for different input and output configurations as dictated by the user.
+
Figure 35 shows the schematic of the THS7314EVM. Figure 36 and Figure 37 shows the top layer and bottom
layer of the EVM which incorporates standard high-speed layout practices. The Bill of materials can located at:
SLOR103
+
+
+
Figure 35. THS7314 Schematic
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THS7314
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www.ti.com
Figure 36. Top View
Figure 37. Bottom View
22
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THS7314
SLOS513A – DECEMBER 2006 – REVISED MARCH 2011
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Evaluation Board/Kit Important Notice
Texas Instruments (TI) provides the enclosed product(s) under the following conditions:
This evaluation board/kit is intended for use for ENGINEERING DEVELOPMENT, DEMONSTRATION, OR EVALUATION PURPOSES
ONLY and is not considered by TI to be a finished end-product fit for general consumer use. Persons handling the product(s) must have
electronics training and observe good engineering practice standards. As such, the goods being provided are not intended to be complete
in terms of required design-, marketing-, and/or manufacturing-related protective considerations, including product safety and environmental
measures typically found in end products that incorporate such semiconductor components or circuit boards. This evaluation board/kit does
not fall within the scope of the European Union directives regarding electromagnetic compatibility, restricted substances (RoHS), recycling
(WEEE), FCC, CE or UL, and therefore may not meet the technical requirements of these directives or other related directives.
Should this evaluation board/kit not meet the specifications indicated in the User’s Guide, the board/kit may be returned within 30 days from
the date of delivery for a full refund. THE FOREGOING WARRANTY IS THE EXCLUSIVE WARRANTY MADE BY SELLER TO BUYER
AND IS IN LIEU OF ALL OTHER WARRANTIES, EXPRESSED, IMPLIED, OR STATUTORY, INCLUDING ANY WARRANTY OF
MERCHANTABILITY OR FITNESS FOR ANY PARTICULAR PURPOSE.
The user assumes all responsibility and liability for proper and safe handling of the goods. Further, the user indemnifies TI from all claims
arising from the handling or use of the goods. Due to the open construction of the product, it is the user’s responsibility to take any and all
appropriate precautions with regard to electrostatic discharge.
EXCEPT TO THE EXTENT OF THE INDEMNITY SET FORTH ABOVE, NEITHER PARTY SHALL BE LIABLE TO THE OTHER FOR ANY
INDIRECT, SPECIAL, INCIDENTAL, OR CONSEQUENTIAL DAMAGES.
TI currently deals with a variety of customers for products, and therefore our arrangement with the user is not exclusive.
TI assumes no liability for applications assistance, customer product design, software performance, or infringement of patents or
services described herein.
Please read the User’s Guide and, specifically, the Warnings and Restrictions notice in the User’s Guide prior to handling the product. This
notice contains important safety information about temperatures and voltages. For additional information on TI’s environmental and/or
safety programs, please contact the TI application engineer or visit www.ti.com/esh.
No license is granted under any patent right or other intellectual property right of TI covering or relating to any machine, process, or
combination in which such TI products or services might be or are used.
FCC Warning
This evaluation board/kit is intended for use for ENGINEERING DEVELOPMENT, DEMONSTRATION, OR EVALUATION PURPOSES
ONLY and is not considered by TI to be a finished end-product fit for general consumer use. It generates, uses, and can radiate radio
frequency energy and has not been tested for compliance with the limits of computing devices pursuant to part 15 of FCC rules, which are
designed to provide reasonable protection against radio frequency interference. Operation of this equipment in other environments may
cause interference with radio communications, in which case the user at his own expense will be required to take whatever measures may
be required to correct this interference.
EVM Warnings and Restrictions
It is important to operate this EVM within the input voltage range of 0V to 2.3V and the output voltage range of 0V to 5V.
Exceeding the specified input range may cause unexpected operation and/or irreversible damage to the EVM. If there are questions
concerning the input range, please contact a TI field representative prior to connecting the input power.
Applying loads outside of the specified output range may result in unintended operation and/or possible permanent damage to the EVM.
Please consult the EVM User's Guide prior to connecting any load to the EVM output. If there is uncertainty as to the load specification,
please contact a TI field representative.
During normal operation, some circuit components may have case temperatures greater than 85°C. The EVM is designed to operate
properly with certain components above 85°C as long as the input and output ranges are maintained. These components include but are
not limited to linear regulators, switching transistors, pass transistors, and current sense resistors. These types of devices can be identified
using the EVM schematic located in the EVM User's Guide. When placing measurement probes near these devices during operation,
please be aware that these devices may be very warm to the touch.
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2011, Texas Instruments Incorporated
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THS7314
SLOS513A – DECEMBER 2006 – REVISED MARCH 2011
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REVISION HISTORY
Changes from Original (December 2006) to Revision A
•
24
Page
Added the EVALUATION MODULE section ....................................................................................................................... 21
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PACKAGE OPTION ADDENDUM
www.ti.com
19-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
THS7314D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
7314
Samples
THS7314DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
7314
Samples
THS7314DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
7314
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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