THS7315
www.ti.com
SLOS532 – JUNE 2007
3-Channel SDTV Video Amplifier with 5th-Order Filters and 5.2-V/V Gain
FEATURES
•
•
•
•
•
•
•
•
•
DESCRIPTION
Three SDTV Video Amplifiers for CVBS,
S-Video, Y'P'BP'R 480i/576i, Y'U'V', or G'B'R'
(R'G'B')
Integrated Low-Pass Filters:
– 5th-Order, 8.5-MHz (–3dB) Butterworth
– –1dB Passband Bandwidth at 7 MHz
– 47 dB Attenuation at 27 MHz
Versatile Input Biasing:
– DC-Coupled with 230-mV Output Shift
– AC-Coupled with Sync-Tip Clamp
– Allows AC-Coupled With DC-Biasing
Built-in 5.2 V/V Gain (14.3 dB)
+3-V to +5-V Single Supply Operation
Rail-to-Rail Output:
– Output Swings Within 100 mV of the Rails,
Allowing AC- or DC-Output Coupling
– Supports Driving Two Lines per Channel
Low Total Quiescent Current: 15.6 mA at 3.3 V
Low Differential Gain/Phase of 0.2%/0.3°
Lead-free, Green SOIC-8 Package
Fabricated using the revolutionary complementary
silicon-sermanium (SiGe) BiCom3 process, the
THS7315 is a low-power, single-supply, 3-V to 5-V
3-channel integrated video buffer. It incorporates a
5th-order Butterworth filter that is useful as a
digital-to-analog converter (DAC) reconstruction filter
or an analog-to-digital converter (ADC) anti-aliasing
filter. The 8.5-MHz filter is a perfect choice for SDTV
video, including Composite Video Baseband Signals
(CVBS), S-Video, Y’U’V’, G’B’R’ (R’G’B’), and
Y’P’BP’R 480i/576i.
The THS7315 inputs can be either ac- or dc-coupled.
The 230-mV output level shift allows for a full sync
dynamic range at the output with 0 V input. The
ac-coupled modes include a transparent sync-tip
clamp option for CVBS, Y', and G'B'R' signals with
bottom-level sync. AC-coupled biasing for C'/P'B/P'R
channels can easily be achieved by adding an
external resistor to VS+.
The THS7315 is the perfect choice for all output
buffer applications. Its rail-to-rail output stage with
5.2-V/V gain allows for both ac and dc line driving,
making it a perfect choice for DaVinci™ processors.
The ability for each channel to drive two video lines,
or 75-Ω loading, allows for maximum flexibility as a
video line driver. The 15.6-mA quiescent current at
3.3 V also makes it an excellent choice for
USB-powered, portable, or other power-sensitive
video applications
APPLICATIONS
•
•
•
Set Top Box Output Video Buffering
PVR/DVDR Output Buffering
USB/Portable Low-Power Video Buffering
The THS7315 is available in a small SOIC-8
package that is lead-free and compliant with green
requirements.
+1.8 V
75 W
DAC/
Encoder
TM
(DaVinci
SDTV
CVBS
S-Video Y’
S-Video C’
480i/576i
Y’P’BP’R
G’B’R’
CVBS
OUT
CVBS
THS7315
500 W
)
Y’
500 W
CH.1 OUT 8
2 CH.2 IN
CH.2 OUT 7
3 CH.3 IN
CH.3 OUT 6
4 VS+
C’
500 W
3.3 V
75 W
1 CH.1 IN
75 W
Y’
OUT
S-Video
GND 5
Gain =
5.2 V/V
75 W
C’
OUT
75 W
75 W
3.3-V Single-Supply DC-Input/DC-Output Coupled Video Line Driver
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
DaVinci is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2007, Texas Instruments Incorporated
THS7315
www.ti.com
SLOS532 – JUNE 2007
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
PACKAGING/ORDERING INFORMATION (1)
PACKAGED DEVICES
PACKAGE TYPE
THS7315D
Rails, 75
SOIC-8
THS7315DR
(1)
TRANSPORT MEDIA, QUANTITY
Tape and reel, 2500
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
website at www.ti.com.
ABSOLUTE MAXIMUM RATINGS (1)
Over operating free-air temperature range (unless otherwise noted)
THS7315
UNIT
5.5
V
–0.4 to VS+
V
±90
mA
Supply voltage, VS+ to GND
VI
Input voltage
IO
Output current
Continuous power dissipation
TJ
See Dissipation Ratings Table
TJ
Maximum junction temperature, continuous operation, long term reliability
Tstg
Storage temperature range
ESD ratings
(1)
(2)
(3)
+150
°C
+125
°C
–65 to +150
°C
Maximum junction temperature, any condition (2)
(3)
HBM
2000
CDM
1500
MM
200
V
Stresses above those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute maximum rated conditions for extended periods may degrade device reliability.
The absolute maximum junction temperature under any condition is limited by the constraints of the silicon process.
The absolute maximum junction temperature for continuous operation is limited by the package constraints. Operation above this
temperature may result in reduced reliability and/or lifetime of the device.
DISSIPATION RATINGS
PACKAGE
SOIC-8 (D)
(1)
(2)
θJC
(°C/W)
θJA
(°C/W)
16.8
130
(2)
POWER RATING (1)
(TJ = +125°C)
TA = +25°C
TA = +85°C
769 mW
308 mW
Power rating is determined with a junction temperature of +125°C. This temperature is the point where performance starts to degrade
and long-term reliability starts to be reduced. Thermal management of the final printed circuit board (PCB) should strive to keep the
junction temperature at or below +125°C for best performance and reliability.
This data was taken with the JEDEC High-K test PCB. For the JEDEC low-K test PCB, the θJA is 196°C/W.
RECOMMENDED OPERATING CONDITIONS
MIN
2
VS+
Supply voltage
TA
Ambient temperature
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NOM
MAX
UNIT
3
5
V
–40
+85
°C
THS7315
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SLOS532 – JUNE 2007
ELECTRICAL CHARACTERISTICS: VS+ = 3.3 V
RL = 150Ω to GND, unless otherwise noted. See Figure 1 and Figure 2.
TYP
PARAMETER
TEST CONDITIONS
OVER TEMPERATURE
+25°C
+25°C
0°C to
+70°C
–40°C to
+85°C
UNIT
MIN/MAX/
TYP
8.5
6.8/10.4
6.7/10.5
6.6/10.6
MHz
Min/Max
8.5
6.8/10.4
6.7/10.5
6.6/10.6
MHz
Min/Max
AC PERFORMANCE
Small-signal bandwidth (–3 dB)
VO – 0.2 VPP
Large-signal bandwidth (–3 dB)
(1)
VO – 2 VPP
(1)
Passband bandwidth (–1dB)
7.0
MHz
Typ
Min/Max
f = 6 MHz (2)
0.25
–0.3/2.4
–0.35/2.5
–0.4/2.6
dB
f = 27 MHz (2)
47
36
35
34
dB
Min
Group delay
f = 100 kHz
61
ns
Typ
Group delay variation (with respect to
100 kHz)
f = 5.1 MHz
12
ns
Typ
0.3
ns
Typ
Differential gain
NTSC/PAL
0.2/0.2
%
Typ
Differential phase
NTSC/PAL
0.25/0.35
degrees
Typ
Total harmonic distortion
f = 1 MHz, VO = 2 VPP, ac-coupled I/O
–62
dB
Typ
Signal-to-noise ratio
NTC-7 weighting, 100 kHz to 4.2 MHz
73
dB
Typ
Channel-to-channel crosstalk
f = 1 MHz, output-referred
–65
dB
Typ
dB
Min/Max
Ω
Typ
Attenuation (with respect to 100 kHz)
Channel-to-channel delay
AC gain, all channels
14.3
Output impedance
f = 1 MHz
14/14.6
14/14.6
14/14.6
0.8
DC PERFORMANCE
Biased output voltage
VIN = 0 V
Input voltage range
DC input, limited by output
Sync tip clamp charge current
VIN = –0.1 V
230
Input resistance
Input capacitance
mV
Min/Max
–0.1/0.56
80/390
68/415
48/420
V
Typ
200
μA
Typ
800
kΩ
Typ
2
pF
Typ
V
Typ
OUTPUT CHARACTERISTICS
High output voltage swing
Low output voltage swing
Output current
Sourcing
Sinking
RL = 150 Ω to 1.65 V
3.15
RL = 150 Ω to GND
3.10
RL = 75 Ω to 1.65 V
V
Min
3.10
V
Typ
RL = 75 Ω to GND
3.0
V
Typ
RL = 150 Ω to 1.65 V (VIN = –0.15 V)
0.15
V
Typ
RL = 150 Ω to GND (VIN = –0.15 V)
0.05
V
Max
RL = 75 Ω to 1.65 V (VIN = –0.15 V)
0.26
V
Typ
RL = 75 Ω to GND (VIN = –0.15 V)
0.1
V
Typ
80
mA
Typ
70
mA
Typ
Max
RL = 10 Ω to 1.65 V
2.85
0.13
2.75
0.14
2.75
0.14
POWER SUPPLY
Maximum operating voltage
3.3
5.5
5.5
5.5
V
Minimum operating voltage
3.3
2.85
2.85
2.85
V
Min
Max
Maximum quiescent current
VIN = 0 V
15.6
20
22
24
mA
Minimum quiescent current
VIN = 0 V
15.6
12
11.6
11
mA
Min
dB
Typ
Power supply rejection (+PSRR)
(1)
(2)
43
The Min/Max values listed for this specification are specified by design and characterization only.
3.3-V supply filter specifications are specified by 100% testing at 5-V supply along with design and characterization only.
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THS7315
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SLOS532 – JUNE 2007
ELECTRICAL CHARACTERISTICS: VS+ = 5 V
RL = 150Ω to GND, unless otherwise noted. See Figure 1 and Figure 2.
TYP
PARAMETER
TEST CONDITIONS
OVER TEMPERATURE
+25°C
+25°C
0°C to
+70°C
–40°C to
+85°C
UNIT
MIN/MAX/
TYP
8.5
6.8/10.4
6.7/10.5
6.6/10.6
MHz
Min/Max
8.5
6.8/10.4
6.7/10.5
6.6/10.6
MHz
Min/Max
AC PERFORMANCE
Small-signal bandwidth (–3 dB)
VO – 0.2 VPP
Large-signal bandwidth (–3 dB)
(1)
VO – 2 VPP
(1)
Passband bandwidth (–1dB)
Attenuation (with respect to 100 kHz)
7.0
MHz
Typ
Min/Max
f = 6 MHz
0.25
–0.3/2.4
–0.35/2.5
–0.4/2.6
dB
36
35
34
f = 27 MHz
47
dB
Min
Group delay
f = 100 kHz
61
ns
Typ
Group delay variation (with respect to
100 kHz)
f = 5.1 MHz
11
ns
Typ
0.3
ns
Typ
Differential gain
NTSC/PAL
0.2/0.2
%
Typ
Differential phase
NTSC/PAL
0.3/0.35
degrees
Typ
Total harmonic distortion
f = 1 MHz, VO = 2 VPP, ac-coupled I/O
–61
dB
Typ
Signal-to-noise ratio
NTC-7 weighting, 100 kHz to 4.2 MHz
73
dB
Typ
Channel-to-channel crosstalk
f = 1 MHz, output-referred
–65
dB
Typ
dB
Min/Max
Ω
Typ
Channel-to-channel delay
AC gain, all channels
14.3
Output impedance
f = 1 MHz
14/14.6
14/14.6
14/14.6
0.8
DC PERFORMANCE
Bias output voltage
VIN = 0 V
Input voltage range
Limited by output
Sync tip clamp charge current
VIN = –0.1 V
235
Input resistance
Input capacitance
mV
Min/Max
–0.1/0.9
80/390
68/415
48/420
V
Typ
200
μA
Typ
800
kΩ
Typ
2
pF
Typ
V
Typ
OUTPUT CHARACTERISTICS
High output voltage swing
Low output voltage swing
Output current
Sourcing
Sinking
RL = 150 Ω to 2.5 V
4.85
RL = 150 Ω to GND
4.7
RL = 75 Ω to 2.5 V
V
Min
4.8
V
Typ
RL = 75 Ω to GND
4.5
V
Typ
RL = 150 Ω to 2.5 V (VIN = –0.15 V)
0.2
V
Typ
RL = 150 Ω to GND (VIN = –0.15 V)
0.05
V
Max
RL = 75 Ω to 2.5 V (VIN = –0.15 V)
0.35
V
Typ
RL = 75 Ω to GND (VIN = –0.15 V)
0.07
V
Typ
90
mA
Typ
85
mA
Typ
Max
RL = 10 Ω to 2.5 V
4.4
0.14
4.3
0.16
4.25
0.18
POWER SUPPLY
Maximum operating voltage
5
5.5
5.5
5.5
V
Minimum operating voltage
5
2.85
2.85
2.85
V
Min
Max
Maximum quiescent current
VIN = 0 V
16.5
22
24
25
mA
Minimum quiescent current
VIN = 0 V
16.5
12.5
12
11.5
mA
Min
dB
Typ
Power supply rejection (+PSRR)
(1)
4
44
The Min/Max values listed for this specification are specified by design and characterization only.
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SLOS532 – JUNE 2007
PIN CONFIGURATION
SOIC-8 (D)
(Top View)
THS7315
1 CH.1 IN
CH.1 OUT 8
2 CH.2 IN
CH.2 OUT 7
3 CH.3 IN
CH.3 OUT 6
4 VS+
GND 5
TERMINAL FUNCTIONS
TERMINAL
NO.
(SOIC-8)
I/O
CH. 1 IN
1
I
Video Input, Channel 1
CH. 2 IN
2
I
Video Input, Channel 2
CH. 3 IN
3
I
Video Input, Channel 3
VS+
4
I
Positive Power Supply Pin. Connect to 3 V to 5 V.
GND
5
I
Ground pin for all internal circuitry.
CH. 3
OUT
6
O
Video Output, Channel 3
CH. 2
OUT
7
O
Video Output, Channel 2
CH. 1
OUT
8
O
Video Output, Channel 1
NAME
DESCRIPTION
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FUNCTIONAL BLOCK DIAGRAM
+VS
gm+
Level
Shift
-
Channel 1
Input
LPF
Sync-Tip
Clamp
(DC restore)
800 kW
5.2 V/V
Channel 1
Output
5.2 V/V
Channel 2
Output
5.2 V/V
Channel 3
Output
5-Pole
8.5 MHz
+VS
+
-
Level
Shift
gm
Channel 2
Input
LPF
Sync-Tip
Clamp
(DC restore)
800 kW
5-Pole
8.5 MHz
+VS
+
-
Level
Shift
gm
Channel 3
Input
800 kW
LPF
Sync-Tip
Clamp
(DC restore)
5-Pole
8.5 MHz
+3 V to +5 V
TEST CIRCUITS
470 mF
+
CIN
RLOAD
THS7315
RTERM
RSOURCE
RTERM
3 CH.3 IN CH.3 OUT 6
4 VS+
CIN
1 CH.1 IN CH.1 OUT 8
2 CH.2 IN CH.2 OUT 7
RLOAD
GND 5
CIN
4 VS+
GND 5
0.1 mF
RTERM
VSOURCE
+
+VS
+
RLOAD
0.1 mF
470 mF
+
RTERM
RLOAD
RLOAD
+
0.1 mF
100 mF
+VS
Figure 1. DC-Coupled Input and Output Test Circuit
6
0.1 mF
470 mF
3 CH.3 IN CH.3 OUT 6
RTERM
RSOURCE
0.1 mF
VSOURCE
RLOAD
THS7315
RTERM
1 CH.1 IN CH.1 OUT 8
2 CH.2 IN CH.2 OUT 7
100 mF
Figure 2. AC-Coupled Input and Output Test Circuit
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APPLICATION INFORMATION
The THS7315 is targeted for standard definition (SD) video output buffer applications. Although it can be used
for numerous other applications, the needs and requirements of the video signal are the most important design
parameters of the THS7315. Built on the revolutionary complementary silicon-germanium (SiGe) BiCom3
process, the THS7315 incorporates many features not typically found in integrated video parts while consuming
very low power.
The THS7315 has the following features:
• Single-supply 3-V to 5-V operation with low total quiescent current of 15.6 mA at 3.3 V and 16.5 mA at 5 V.
• Input configuration accepting dc + level-shift, ac sync-tip clamp, or ac bias selection; ac-biasing is
accomplished with the use of an external pull-up resistor to the positive power supply.
• 5th-order low-pass filter for DAC reconstruction or ADC image rejection:
– 8.5-MHz for NTSC, PAL, SECAM, Composite (CVBS), S-Video Y'C', 480i/576i Y'P'BP'R , and G'B'R'
(R'G'B') signals.
• Internal fixed gain of 5.2 V/V (+14.3 dB) buffer that can drive up to two video lines per channel with dc
coupling or traditional ac coupling.
• Signal flow-through configuration using an 8-pin SOIC package that complies with the latest lead-free (RoHS
compatible) and green manufacturing requirements.
OPERATING VOLTAGE
The THS7315 is designed to operate from 3 V to 5 V over a –40°C to +85°C temperature range. The impact on
performance over the entire temperature range is negligible because of the implementation of thin film resistors
and high-quality, low temperature coefficient capacitors. The design of the THS7315 allows operation down to
2.85 V, but for best results, the use of a 3 V or greater supply should be used to ensure there are no issues with
headroom or clipping.
A 0.1-μF to 0.01-μF capacitor should be placed as close as possible to the power-supply pins. Failure to do so
may result in the THS7315 outputs ringing or oscillating. Additionally, a large capacitor, such as 22 μF to
100 μF, should be placed on the power-supply line to minimize interference with 50-Hz/60-Hz line frequencies.
INPUT VOLTAGE
The THS7315 input range allows for an input signal range from –0.3 V to approximately (VS+ – 1.5V). However,
because of the internal fixed gain of 5.2 V/V (+14.3 dB) and the internal level shift that shifts the output by
230 mV, the output is generally the limiting factor for the allowable linear input range. For example, with a 5-V
supply, the linear input range is from –0.3 V to +3.5 V. As a result of the gain and level shift, the linear output
range limits the allowable linear input range from about –0.1 V to +2.3 V.
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APPLICATION INFORMATION (continued)
INPUT OVERVOLTAGE PROTECTION
The THS7315 is built using a very high-speed complementary bipolar and CMOS process. The internal junction
breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in
the Absolute Maximum Ratings table. All input and output device pins are protected with internal ESD protection
diodes to the power supplies, as shown in Figure 3.
VS+
External
Input/
Output
Pin
Internal
Circuitry
Figure 3. Internal ESD Protection
These diodes provide moderate protection to input overdrive voltages above and below the supplies as well. The
protection diodes can typically support 30 mA of continuous current when overdriven.
TYPICAL CONFIGURATION and VIDEO TERMINOLOGY
A typical application circuit using the THS7315 as a video buffer is shown in Figure 4. It shows a video DAC
output, such as the DaVinci, driving the three input channels of the THS7315. Although the S-Video Y’/C’
channels and the composite video (CVBS) channel of an SD video system are shown, these channels can easily
be the Y’P’BP’R (sometimes labeled Y’U’V’ or incorrectly labeled Y’C’BC’R) signals of a 480i or 576i system.
These signals can also be G’B’R’ (R'G'B') signals or other variations. Note that for computer signals, the sync
should be embedded within the signal for a system with only three outputs. This configuration is sometimes
labeled as R’G’sB’ (sync on green) or R’sG’sB’s (sync on all signals).
+1.8 V
DaVinci/
DM2xx/
DM3xx/
OMAP
330 mF
+
CVBS
TM
SDTV
CVBS
S-Video Y’
S-Video C’
480i/576i
Y’P’BP’R
G’B’R’
THS7315
500 W
Y’
500 W
75 W
CVBS
OUT
75 W
1 CH.1 IN
CH.1 OUT 8
2 CH.2 IN
CH.2 OUT 7
3 CH.3 IN
CH.3 OUT 6
4 VS+
C’
330 mF
+
Y’
OUT
S-Video
GND 5
0.1 mF
0.1 mF
500 W
+
+3.3 V
75 W
Gain =
5.2 V/V
75 W
C’
OUT
22 mF
Figure 4. Typical SDTV CVBS/Y'/C' Inputs From DC-Coupled Encoder/DAC
With AC-Coupled Line Driving
8
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75 W
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APPLICATION INFORMATION (continued)
Note that the Y' term is used for the luma channels throughout this document rather than the more common
luminance (Y) term. The reason for this usage is to account for the definition of luminance as stipulated by the
CIE (International Commission on Illumination). Video departs from true luminance because a nonlinear term,
gamma, is added to the true RGB signals to form R'G'B' signals. These R'G'B' signals are then used to
mathematically create luma (Y'). Therefore, true luminance (Y) is not maintained, and thus a difference in
terminology arises.
This rationale is also used for the chroma (C') term. Chroma is derived from the nonlinear R'G'B' terms and
therefore it is also nonlinear. True chominance (C) is derived from linear RGB, and thus the difference between
chroma (C') and chrominance (C) exists. The color difference signals (P'B/ P'R/U'/V') are also referenced this way
to denote the nonlinear (gamma-corrected) signals.
R'G'B' (commonly mislabeled RGB) is also called G'B'R' (again commonly mislabeled as GBR) in professional
video systems. The SMPTE component standard stipulates that the luma information is placed on the first
channel, the blue color difference is placed on the second channel, and the red color difference signal is placed
on the third channel. This approach is consistent with the Y'P'BP'R nomenclature. Because the luma channel (Y')
carries the sync information and the green channel (G') also carries the sync information, it makes logical sense
that G' be placed first in the system. Since the blue color difference channel (P'B) is next and the red color
difference channel (P'R) is last, then it also makes logical sense to place the B' signal on the second channel
and the R' signal on the third channel, respectively. Thus, hardware compatibility is better achieved when using
G'B'R' rather than R'G'B'. Note that for many G'B'R' systems, sync is embedded on all three channels; this
configuration may not always be the case for all systems.
INPUT MODE OF OPERATION—DC
The THS7315 allows for both ac-coupled and dc-coupled inputs. Many DACs or video encoders can be
dc-connected to the THS7315. One of the drawbacks to dc-coupling, however, occurs when 0 V is applied to the
input. Although the THS7315 allows for a 0-V input signal with no issues, the output swing of a traditional
amplifier cannot yield a 0-V signal, resulting in possible clipping. This condition is true for any single-supply
amplifier because of the output transistor limitations. Both CMOS and bipolar transistors cannot go to 0 V while
sinking current. This transistor characteristic is also the same reason why the highest output voltage is always
less than the power-supply voltage when sourcing current.
This output clipping can reduce both the horizontal and vertical sync amplitudes on the video signal. A problem
occurs if the video signal receiver uses an AGC loop to account for losses in the transmission line. Some video
AGC circuits derive gain from the horizontal sync amplitude. If clipping occurs on the sync amplitude, then the
AGC circuit can increase the gain too much—resulting in too much luma and/or chroma amplitude gain
correction. This effect may result in a picture with an overly bright display and too much color saturation.
Other AGC circuits use the chroma burst amplitude for amplitude control, and a reduction in the sync signals
does not alter the proper gain setting. However, it is good engineering design practice to ensure that saturation
and/or clipping does not take place. Transistors always take a finite amount of time to come out of saturation.
This saturation could possibly result in timing delays or other signal aberrations.
To eliminate saturation or clipping problems, the THS7315 has a 230 mV output level shift feature. This feature
takes the input voltage and adds an internal level shift to the signal. The THS7315 rail-to-rail output stage can
create this output level while connected to a typical video load. This process ensures that no saturation or
clipping of the sync signal occurs. This level shift is constant, regardless of the input signal. For example, if a
0.5-V input is applied, the output is at (0.5 V × 5.2 V/V) + 0.23 V = 2.92 V.
The fixed internal gain of 5.2 V/V (14.3 dB) dictates what the allowable linear input voltage range can be without
clipping concerns. For example, if the power supply is set to 3 V, the maximum output is about 2.9 V while
driving a significant amount of current. Thus, to avoid clipping, the allowable input will be [ (3.1 V – 0.23 V) / 5.2
V/V) ] = 0.55 V. This relationship holds true up to the maximum recommended 5 V power supply that allows an
approximate input range of [ (4.9 V – 0.23 V) / 5.2 V/V) ] = 0.9 V while avoiding clipping on the output.
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APPLICATION INFORMATION (continued)
The THS7315 input impedance in this operating mode is dictated by the internal 800-kΩ pull-down resistor, as
shown in Figure 5. Note that the internal voltage shift does not appear at the input pin, but only at the output pin.
Internal
Circuitry
+VS
Input
+
800 kW
-
Internal Level
Shifter
Figure 5. Equivalent DC Input Mode Circuit
INPUT MODE OF OPERATION —AC SYNC-TIP CLAMP
Some video DACs or encoders are not referenced to ground but rather to the positive power supply. These
DACs typically only sink current, rather than the more traditional current-sourcing DAC where the resistor is
referenced to ground. The resulting video signal voltages can be too high for a dc-coupled video buffer to
function properly. To account for this scenario, the THS7315 incorporates a sync-tip clamp (STC) circuit. This
function requires a capacitor (nominally 0.1 μF) to be placed in series with the input. Note that while the term
sync-tip clamp or STC is used throughout this document, it should be noted that the THS7315 is better termed
as a dc-restoration circuit based on how this function is performed. The STC circuit is an active clamp circuit and
not a passive diode clamp function.
The input to the THS7315 has an internal control loop that sets the lowest input-applied voltage to clamp at
ground (0 V). By setting the reference at 0 V, the THS7315 allows a dc-coupled input to also function. Therefore,
the STC is considered transparent because it does not operate unless the input signal goes below ground. The
signal then goes through the same internal level shifter, resulting in an output voltage low level of 230 mV. If the
input signal tries to go below 0 V, the internal control loop of the THS7315 will source up to 2 mA of current to
increase the THS7315 input voltage level on the input side of the coupling capacitor. As soon as the voltage
goes above 0 V, the loop will stop sourcing current and become very high impedance.
One of the concerns about the STC level is how the clamp reacts to a sync edge that has overshoot—a
common effect in VCR signals or reflections found in poor PCB layouts. Ideally, the STC should not react to the
overshoot voltage of the input signal. Otherwise, this effect could result in clipping on the rest of the video signal
because it may raise the bias voltage too much.
To help minimize this input signal overshoot problem, the control loop in the THS7315 has an internal low-pass
filter as shown in Figure 6. This filter reduces the response time of the STC circuit. This delay is a function of
how far the voltage is below ground, but generally, it is about a 100-ns delay. The effect of this filter is to slow
down the response of the control loop so as not to clamp on the input overshoot voltage, but rather the flat
portion of the sync signal.
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APPLICATION INFORMATION (continued)
+VS
+VS
Comparator
Internal
Circuitry
STC LPF
+
Input
Pin
-
Input
+
0.1 m F
800 kW
-
Internal Level
Shifter
Figure 6. Equivalent AC Sync-Tip Clamp Input Circuit
As a result of the delay, the sync may have an apparent voltage shift. The amount of shift depends on the
amount of droop in the signal as dictated by the input capacitor and the STC current flow. Because the sync is
primarily for timing purposes—with syncs occurring on the edge of the sync signal—this shift is transparent in
most systems.
While this feature may not fully eliminate overshoot issues on the input signal in cases of extreme overshoot
and/or ringing, the STC system should help minimize improper clamping levels. As an additional way to
minimize this problem, an external capacitor (for example, 10 pF to 47 pF) to ground in parallel with the external
termination resistors can help filter overshoot problems.
It should be noted that the STC system is dynamic and does not rely upon timing in any way. It only depends on
the voltage appearing at the input pin at any given point in time. The STC filtering helps minimize level shift
problems associated with switching noises or very short spikes on the signal line, ensuring a very robust STC
system.
When using the ac sync-tip clamp operation, there must also be some finite amount of discharge bias current.
As previously discussed, if the input signal goes below the 0 V clamp level, the THS7315 internal loop will
source current to increase the voltage appearing at the input pin. As the difference between the signal level and
the 0 V reference level increases, the amount of source current increases proportionally—supplying up to 2 mA
of current. As a result, the time to re-establish the proper STC voltage can be very short. If this difference is very
small, then the source current will also be very small to account for minor voltage droop.
What happens if the input signal goes above the 0 V input level? The problem is that the video signal will always
be above this level and must not be altered in any way. If the sync level of the input signal is above 0 V,
however, then the internal discharge (sink) current will reduce the ac-coupled bias signal to the proper 0 V level.
This discharge current must not be large enough to significantly alter the video signal, or picture quality issues
may arise. This effect is often seen by looking at the tilt (or droop) of a constant luma signal being applied and
observing the resulting output level. The associated change in luma level from the beginning of the video line to
the end of the video line is the amount of droop.
If the discharge current is very small, the amount of tilt (or droop) is very low, which is a generally a good thing.
Unfortunately, the amount of time for the system to capture the sync signal could be too long. This effect is also
termed hum rejection. Hum arises from the ac line voltage frequency of 50 Hz or 60 Hz. The values of the
discharge current and the ac-coupling capacitor combine to dictate the hum rejection and the amount of line tilt.
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APPLICATION INFORMATION (continued)
To allow for both dc-coupling and ac-coupling in the same part, the THS7315 incorporates an 800-kΩ resistor to
ground. Although a true constant current sink is preferred over a resistor, there are significant issues when the
voltage is near ground. (For example, voltage near ground can cause the current sink transistor to saturate and
produce potential signal problems.) This resistor is large enough to not impact a dc-coupled DAC termination.
For discharging an ac-coupled source, Ohm's Law is utilized. If the video signal is 0.5 V, then there will be 0.5 V
/ 800 kΩ = 0.625 μA of discharge current. If more hum rejection is desired or there is a loss of sync occurring,
simply decrease the 0.1 μF input coupling capacitor. A decrease from 0.1 μF to 0.047 μF increases the hum
rejection by a factor of 2:1. Alternatively, an external pull-down resistor to ground may be added, decreasing the
overall resistance and ultimately increasing the discharge current.
To ensure proper stability of the ac STC control loop, the source impedance must be less than 1 kΩ with the
input capacitor in place. Otherwise, there is a possibility of the control loop ringing. This ringing may appear on
the THS7315 output. Because most DACs or encoders use resistors that are typically ≤ 500 Ω to establish the
voltage, meeting the < 1 kΩ requirement is easily done. However, if the source impedance looking from the
THS7315 input is very high, then simply adding a 1-kΩ resistor to GND will ensure proper operation of the
THS7315.
INPUT MODE OF OPERATION —AC BIAS
Sync-tip clamps work very well for signals that have horizontal and/or vertical syncs associated with them. Some
video signals, on the other hand, do not have a sync embedded within the signal. If ac-coupling of these signals
is desired, then a dc bias is required to properly set the dc operating point within the THS7315. This function is
easily accomplished with the THS7315 by simply adding an external pull-up resistor to the positive power
supply, as shown in Figure 7.
Internal
Circuitry
+3.3 V
+3.3 V
9.31 MW
Input
+
CIN
Input
Pin
800 kW
-
Internal Level
Shifter
Figure 7. AC-Bias Input Mode Circuit Configuration
The dc voltage appearing at the input pin is approximately equal to:
ǒ
V DC + VS
Ǔ
800k
800k ) RPU
(1)
The allowable input range of the THS7315 is very wide: approximately (+VS – 1.5 V). The input range is limited
by the allowable output voltage range and the internal gain. As such, the input dc bias point is very flexible, with
the output dc bias point being the primary factor. For example, if the desired output dc bias point is 1.6 V on a
3.3-V supply, then the input dc bias point should be (1.6 V – 230 mV) / 5.2 = 0.263 V. Consequently, the pull-up
resistor calculates to be about 9.31 MΩ, resulting in 0.261 V. If the desired input dc-bias point is 2.4 V with a 5-V
power supply, then the pull-up resistor calculates to be about 8.66 MΩ.
Keep in mind that the internal 800-kΩ resistor has approximately a ±20% variance. As such, the calculations
should account for this variance. For the 0.261 V example above, using an ideal 9.31-MΩ resistor, the input dc
bias voltage is about 0.261 V ±0.05 V, which translates to an output bias voltage of about 1.64 V ±0.26 V.
12
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APPLICATION INFORMATION (continued)
One other issue that must be taken into account is the dc-bias point. The dc-bias point is a function of the power
supply. As such, there is as a low-pass filter. Additionally, the time to charge the capacitor to the final dc bias
point is also a function of the pull-up resistor and the input capacitor. Lastly, the input capacitor forms a
high-pass filter with the parallel impedance of the pull-up resistor and the 800-kΩ resistor. Generally, it is good to
have this high-pass filter at about 3 Hz to minimize any potential droop on a P'B, P'R, or non-sync B' or R' signal.
A 0.1-μF input capacitor with a 9.31-MΩ pull-up resistor equals about a 2.2-Hz high-pass corner frequency.
This mode of operation is recommended for use with chroma (C'), P’B, P'R, U', V', and non-sync B' and/or R'
signals.
OUTPUT MODE OF OPERATION—DC-COUPLED
The THS7315 incorporates a rail-to-rail output stage that can be utilized to drive the line directly without the
need for large ac-coupling capacitors, as shown in Figure 8. This architecture offers the best line tilt and field tilt
(or droop) performance because no ac coupling occurs. Keep in mind that if the input is ac-coupled, then the
resulting tilt arising from the input ac coupling is still seen on the output, regardless of the output coupling. The
80-mA output current drive capability of the THS7315 was designed to drive two video lines per channel
simultaneously—essentially, a 75-Ω load—while keeping the output dynamic range as wide as possible.
+1.8 V
DAC/Encoder
75 W
TM
(DaVinci
)
CVBS
THS7315
500 W
SDTV
CVBS
S-Video Y’
S-Video C’
480i/576i
Y’P’BP’R
G’B’R’
CVBS
OUT
Y’
500 W
75 W
1 CH.1 IN
CH.1 OUT 8
2 CH.2 IN
CH.2 OUT 7
3 CH.3 IN
CH.3 OUT 6
4 VS+
C’
75 W
S-Video
GND 5
0.1 mF
500 W
+
+3.3 V
Y’
OUT
Gain =
5.2 V/V
75 W
C’
OUT
75 W
75 W
22 mF
Figure 8. Typical SDTV CVBS/Y'/C' System with DC-Coupled Line Driving
One concern about dc coupling arises when the line is terminated to ground. If the ac-bias input configuration is
used, the THS7315 output has a dc bias on the output. With two lines terminated to ground, this configuration
creates a dc current path, resulting in a slightly decreased high output voltage swing as well as an increase in
device power dissipation. While the THS7315 was designed to operate with junction temperatures of up to
+125°C, care must be taken to ensure that the junction temperature does not exceed this level; otherwise,
long-term reliability could suffer. Although this configuration only adds less than 10 mW of power dissipation per
channel, the overall low power dissipation of the THS7315 design minimizes potential thermal issues even when
using the SOIC package at high ambient temperatures.
Note that the THS7315 can drive the line with dc coupling regardless of the input mode of operation. The only
requirement is to verify that the video line has proper termination in series with the output (typically 75 Ω). This
termination also helps isolate capacitive loading effects from the THS7315 output. Failure to isolate capacitive
loads may result in instabilities with the output buffer, potentially causing ringing or oscillating to appear. The
stray capacitance appearing directly at the THS7315 output pins should be kept below 25 pF.
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APPLICATION INFORMATION (continued)
OUTPUT MODE OF OPERATION—AC-COUPLED
The most common method of coupling the video signal to the line is through the use of a large capacitor. This
capacitor is typically between 220 μF and 1000 μF, although 330 μF is very common. The value of this capacitor
must be this large to minimize the line tilt (droop) and/or field tilt associated with ac coupling as discussed
previously in this document. AC coupling is done for several reasons; generally, it is done to ensure full
interoperability with the receiving video system. This coupling eliminates possible ground loops. It also ensures
that regardless of the reference dc voltage used on the transmit side, the receive side will re-establish the dc
reference voltage to its own requirements.
In the same way that the dc output mode of operation is configured (as discussed earlier), each line should have
a 75-Ω source termination resistor in series with the ac-coupling capacitor. If driving two lines, it is best to have
each line use its own capacitor and resistor rather than sharing these components, as Figure 9 shows. This
configuration helps ensure line-to-line dc isolation and avoids the potential problems discussed earlier. Using a
single 1000-μF capacitor for two lines can be done, but there is a chance for ground loops and additional
interference to be created between the two receivers.
Y’
OUT 1
(1)
330 mF
+
75 W
75 W
Y’
OUT 2
(1)
+1.8 V
+1.8 V
R
DAC/
Encoder
330 mF
+
75 W
0.1 mF
P’B
Y’
+1.8 V
R
9.31 MW
OUT 1
(1)
+3.3 V
330 mF
+
THS7315
75 W
1 CH.1 IN CH.1 OUT 8
P’B
SDTV
480i/576i
Y’P’BP’R
G’B’R’
75 W
75 W
2 CH.2 IN CH.2 OUT 7
+1.8 V
R
0.1 mF
9.31 MW
P’B
+3.3 V
3 CH.3 IN CH.3 OUT 6
4 VS+
P’R
GND 5
OUT 2
(1)
330 mF
+
75 W
75 W
0.1 mF
0.1 mF
P’R
OUT 1
(1)
+
+3.3 V
330 mF
+
75 W
22 mF
75 W
P’R
OUT 2
(1)
330 mF
+
75 W
75 W
(1)
As a result of the high frequency content of the video signal, it is recommended, but not required, to add a 0.1-μF or
0.01-μF capacitor in parallel with these large capators.
(2)
Current sinking DAC / Encoder shown. See the application notes.
Figure 9. Typical 480i/576i Y'P'BP'R AC-Input System Driving Two AC-Coupled Video Lines
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APPLICATION INFORMATION (continued)
Lastly, because of the edge rates and frequencies of operation, it is recommended (but not required) to place a
0.1-μF to 0.01-μF capacitor in parallel with the large 220-μF to 1000-μF capacitor. These large-value capacitors
are generally aluminum electrolytic. It is well-known that these types of capacitors have significantly large
equivalent series resistance, or ESR, and the respective impedances at high frequencies is rather large as a
result of the associated inductances involved with the leads and construction. The small 0.1-μF to 0.01-μF
capacitors help to pass these high-frequency ( greater than 1 MHz) signals with much lower impedance than the
large capacitors.
Although it is common to use the same capacitor values for all the video lines, the frequency bandwidth of the
chroma signal in an S-Video system is not required to go to as low (or as high) a frequency as the luma
channels. Therefore, the capacitor values of the chroma line(s) can be smaller, such as 0.1 μF.
LOW-PASS FILTER
Each channel of the THS7315 incorporates a 5th-order low-pass filter. These video reconstruction filters
minimize DAC images from passing on to the video receiver. Depending on the receiver design, failure to
eliminate these DAC images can cause picture quality problems that result from ADC aliasing. Another benefit of
the filter is that it smooths out aberrations in the signal that some DACs can demonstrate if the internal filtering is
not good. These benefits help with picture quality and ensure that the signal meets video bandwidth
requirements.
Each filter has an associated Butterworth characteristic. The benefit of the Butterworth response is that the
frequency response is flat, with a relatively steep initial attenuation at the corner frequency. The problem with the
Butterworth filter, however, is that the group delay also rises near the corner frequency. Group delay is defined
as the change in phase (radians/second) divided by a change in frequency. An increase in group delay
corresponds to a time-domain pulse response that has overshoot (and possible ringing associated with the
overshoot).
Other filter types (such as elliptic or chebyshev) are not recommended for video applications because of the very
large group delay variations that occur near the corner frequency, also resulting in significant overshoot and
ringing. While elliptic or chebyshev filters may help meet the video standard specifications with respect to
amplitude attenuation, the group delay is well beyond the standard specifications. Combined with the fact that
video can switch from a white pixel to a black pixel over and over again, ringing can easily occur. Ringing
typically causes a display to have ghosting or fuzziness on the edges of a sharp transition. On the other hand, a
Bessel filter has ideal group delay response, but the rate of attenuation is typically too low for acceptable image
rejection. Consequently, the Butterworth filter is a respectable compromise for both attenuation and group delay.
The THS7315 filters have a nominal corner (–3 dB) frequency at 8.5 MHz and a –1 dB passband, typically at 7
MHz. This 8.5-MHz filter is ideal for SDTV, NTSC, PAL, and SECAM composite video (CVBS) signals. It is also
useful for S-Video signals (Y'C'), 480i/576i Y'P'BP'R, Y'U'V', broadcast G'B'R' signals, and computer R'G'B' video
signals. The 8.5-MHz, –3-dB corner frequency was designed to allow a maximally flat video signal while
achieving 47 dB of attenuation at 27 MHz—a common sampling frequency between the DAC/ADC 2nd and 3rd
Nyquist zones that is found in many video systems. This feature is important because any signal appearing
around this frequency can appear in the baseband because of aliasing effects of an ADC found in a receiver.
Keep in mind that images do not stop at 27 MHz; they continue around the sampling frequencies of 54 MHz, 81
MHz, 108 MHz, and so forth. Because of these multiple images that an ADC can fold down into the baseband
signal, the low-pass filter must also eliminate these higher-order images. The THS7315 has over 70-dB
attenuation at 54 MHz and 81 MHz, along with over 65-dB attenuation at 108 MHz. Attenuation above 108 MHz
is at least 55 dB, ensuring that images do not affect the desired video baseband signal.
The 8.5-MHz filter frequency was chosen to account for process variations in the THS7315. To ensure the
required video frequencies are effectively passed, the filter corner frequency must be high enough to allow
component variations. The other filter design consideration is the attenuation. It must be large enough to ensure
that anti-aliasing/reconstruction filtering is enough to meet the system demands. Thus, the filter frequency
selection was not arbitrary; it is a good compromise that should meet the demands of most systems.
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APPLICATION INFORMATION (continued)
ADVANTAGES OVER PASSIVE FILTERING
Two key benefits of using an integrated filter system such as the THS7315 over a passive system are PCB area
and filter variations. For overall board area, the small SOIC-8 package for 3-video channels is much smaller over
a passive RLC network, especially a 5-pole passive network. As for filter variations, consider that inductors
generally have 10% tolerances (normally 15% to 20%) and capacitors typically have 10% tolerances. A Monte
Carlo analysis shows that the desired filter corner frequency (–3 dB), flatness (–1 dB), Q-factor (or peaking), and
channel-to-channel delay will have wide variations. These variations can lead to potential performance and
quality issues in mass-production environments. The THS7315 solves most of these problems with the corner
frequency being the only variable.
One concern about using an active filter in an integrated circuit is the variation of the filter characteristics when
the ambient temperature and the subsequent die temperature change. To minimize temperature effects, the
THS7315 uses low temperature coefficient resistors and high quality/low temperature coefficient capacitors
found in the BiCom3 process. The filters have been specified by design to account for process and temperature
variations to maintain proper filter characteristics. This design guideline maintains a low channel-to-channel time
delay that is required for proper video signal performance.
The input and output impedances are another benefit of the THS7315 over a passive RLC filter. The input
impedance presented to the DAC varies significantly with a passive network and may cause voltage variations
over frequency. The THS7315 input impedance is 800 kΩ; only the 2-pF input capacitance plus the PCB trace
capacitance affect this value. As such, the voltage variation appearing at the DAC output is better controlled with
the THS7315.
On the output side of the filter, a passive filter again has an impedance variation over frequency. The THS7315
is an operational amplifier that approximates an ideal voltage source. A voltage source is desirable because the
output impedance is very low and can source and sink current. To properly match the transmission line
characteristic impedance of a video line, a 75-Ω series resistor is placed on the output. To minimize reflections
and to maintain a good return loss, this output impedance must maintain a 75-Ω impedance. A passive filter
impedance variation cannot specify this condition, while the THS7315 has about 0.8 Ω of output impedance at 1
MHz. Thus, the system is matched much better with a THS7315 when compared to a passive filter.
One final benefit of the THS7315 over a passive filter is power dissipation. A DAC driving a video line must be
able to drive a 37.5-Ω load—the receiver 75-Ω resistor and the 75-Ω impedance-matching resistor next to the
DAC to maintain the source impedance requirement. This design requirement forces the DAC to drive at least
1.25 VPP (100% saturation CVBS) / 37.5 Ω = 33.3 mA. A DAC is a current-steering element, and this amount of
current flows internally to the DAC even if the output is 0 V. Thus, power dissipation in the DAC may be very
high, especially when six channels are being driven. Using the THS7315, with a high input impedance and the
capability to drive up to two video lines, can reduce the DAC power dissipation significantly. This reduction
occurs because the resistance that the DAC is driving can be substantially increased. It is common to set this
increase in a DAC by a current-setting resistor on the device. Thus, the resistance can be 300 Ω or
more—significantly reducing the current drive demands from the DAC and saving a substantial amount of power.
For example, a 3.3-V, six-channel DAC dissipates 660 mW just for the steering current capability (6 channels ×
33.3 mA × 3.3 V) if it needs to drive 37.5-Ω load. With a 300-Ω load, the DAC power dissipation as a result of
current steering current would only be 82.5 mW (6 channels × 4.16 mA × 3.3 V).
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EVALUATION MODULE
To evaluate the THS7315, an evaluation module (EVM) is available. The EVM allows for testing the THS7315 in
many different systems. Inputs and outputs include RCA connectors for consumer grade interconnections, or
BNC connectors for higher-level lab grade connections. Several unpopulated component pads are found on the
EVM to allow for different input and output configurations as dictated by the user.
Figure 10 shows the THS7315EVM schematic. Figure 11 and Figure 12 illustrate the top layer and bottom layer
(respectively) of the EVM PCB, incorporating standard high-speed layout practices. Table 1 lists the bill of
materials as supplied from Texas Instruments.
Figure 10. THS7315D EVM Schematic
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EVALUATION MODULE (continued)
Figure 11. THS7315D EVM PCB Top Layer
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EVALUATION MODULE (continued)
Figure 12. THS7315D EVM PCB Bottom Layer
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EVALUATION MODULE (continued)
THS7315EVM Bill of Materials
Table 1. THS7315D EVM
ITEM
REF DES
QTY
DESCRIPTION
1
FB1
1
Bead, Ferrite, 2.5A, 330 Ω
SMD SIZE
0805
MANUFACTURER
PART NUMBER
DISTRIBUTOR
PART NUMBER
(TDK) MPZ2012S331A
(Digi-Key) 445-1569-1-ND
(AVX)
TPSC107K010R0100
(Digi-Key) 478-1765-1-ND
(AVX) 0603YC104KAT2A
(Garrett)
0603YC104KAT2A
2
C16
1
Capacitor, 100μF, Tantalum, 10V, 10%,
Low-ESR
3
C17, C18, C19
3
Open
0603
4
C15
1
Capacitor, 0.1μF, Ceramic, 16V, X7R
0603
5
C1, C2, C3, C12,
C13, C14
6
OPEN
0805
6
C5
1
Capacitor, 0.01μF, Ceramic, 100V, X7R
0805
(AVX) 08051C103KAT2A
(Digi-Key) 478-1358-1-ND
7
C7, C9, C11
3
Capacitor, 0.1μF, Ceramic, 50V, X7R
0805
(AVX) 08055C104KAT2A
(Digi-Key) 478-1395-1-ND
8
C4
1
Capacitor, 1μF, Ceramic, 16V, X7R
0805
(TDK) C2012X7R1C105K
(Digi-Key) 445-1358-1-ND
(Cornell)
AFK477M10F24B
(Newark) 97C7597
C
9
C6, C8, C10
3
Capacitor, Aluminum, 470μF, 10V, 20%
F
10
RX1–RX6
6
Open
0603
11
R4–R9, Z1, Z2, Z3
9
Resistor, 0 Ω
0805
(ROHM) MCR10EZHJ000
(Digi-Key)
RHM0.0ACT-ND
12
R1, R2, R3, R10,
R11, R12
6
Resistor, 75 Ω, 1/8W, 1%
0805
(ROHM) MCR10EZHF75.0
(Digi-Key)
RHM75.0CCT-ND
13
J9, J10
2
Jack, Banana Receptance, 0.25" dia.
hole
(SPC) 813
(Newark) 39N867
14
J1, J2, J3, J6, J7,
J8
6
Connector, BNC, Jack, 75 Ω
(Amphenol)
31-5329-72RFX
(Newark) 93F7554
15
J4, J5
2
Connector, RCA, Jack, R/A
(CUI) RCJ-32265
(Digi-Key) CP-1446-ND
16
TP1, TP2, TP3
3
Test Point, Red
(Keystone) 5000
(Digi-Key) 5000K-ND
17
TP4, TP5
2
Test Point, Black
(Keystone) 5001
(Digi-Key) 5001K-ND
18
U1
1
IC, THS7315
19
4
Standoff, 4-40 Hex, 0.625" Length
(Keystone) 1808
(Newark) 89F1934
20
4
Screw, Phillips, 4-40, .250"
(BF) PMS 440 0031 PH
(Digi-Key) H343-ND
21
1
Printed circuit board
Edge # 6483760 REV. A
D
(TI) THS7315D
EVALUATION BOARD/KIT IMPORTANT NOTICE
Texas Instruments (TI) provides the enclosed product(s) under the following conditions:
This evaluation board/kit is intended for use for ENGINEERING DEVELOPMENT, DEMONSTRATION, OR
EVALUATION PURPOSES ONLY and is not considered by TI to be a finished end-product fit for general
consumer use. Persons handling the product(s) must have electronics training and observe good engineering
practice standards. As such, the goods being provided are not intended to be complete in terms of required
design-, marketing-, and/or manufacturing-related protective considerations, including product safety and
environmental measures typically found in end products that incorporate such semiconductor components or
circuit boards. This evaluation board/kit does not fall within the scope of the European Union directives
regarding electromagnetic compatibility, restricted substances (RoHS), recycling (WEEE), FCC, CE or UL, and
therefore may not meet the technical requirements of these directives or other related directives.
Should this evaluation board/kit not meet the specifications indicated in the User’s Guide, the board/kit may be
returned within 30 days from the date of delivery for a full refund. THE FOREGOING WARRANTY IS THE
EXCLUSIVE WARRANTY MADE BY SELLER TO BUYER AND IS IN LIEU OF ALL OTHER WARRANTIES,
EXPRESSED, IMPLIED, OR STATUTORY, INCLUDING ANY WARRANTY OF MERCHANTABILITY OR
FITNESS FOR ANY PARTICULAR PURPOSE.
20
Submit Documentation Feedback
THS7315
www.ti.com
SLOS532 – JUNE 2007
EVALUATION BOARD/KIT IMPORTANT NOTICE (continued)
The user assumes all responsibility and liability for proper and safe handling of the goods. Further, the user
indemnifies TI from all claims arising from the handling or use of the goods. Due to the open construction of the
product, it is the user’s responsibility to take any and all appropriate precautions with regard to electrostatic
discharge.
EXCEPT TO THE EXTENT OF THE INDEMNITY SET FORTH ABOVE, NEITHER PARTY SHALL BE LIABLE
TO THE OTHER FOR ANY INDIRECT, SPECIAL, INCIDENTAL, OR CONSEQUENTIAL DAMAGES.
TI currently deals with a variety of customers for products, and therefore our arrangement with the user is not
exclusive.
TI assumes no liability for applications assistance, customer product design, software performance, or
infringement of patents or services described herein.
Please read the User’s Guide and, specifically, the Warnings and Restrictions notice in the User’s Guide prior
to handling the product. This notice contains important safety information about temperatures and voltages. For
additional information on TI’s environmental and/or safety programs, please contact the TI application engineer
or visit www.ti.com/esh.
No license is granted under any patent right or other intellectual property right of TI covering or relating to any
machine, process, or combination in which such TI products or services might be or are used.
FCC Warning
This evaluation board/kit is intended for use for ENGINEERING DEVELOPMENT, DEMONSTRATION, OR
EVALUATION PURPOSES ONLY and is not considered by TI to be a finished end-product fit for general
consumer use. It generates, uses, and can radiate radio frequency energy and has not been tested for
compliance with the limits of computing devices pursuant to part 15 of FCC rules, which are designed to
provide reasonable protection against radio frequency interference. Operation of this equipment in other
environments may cause interference with radio communications, in which case the user at his own expense
will be required to take whatever measures may be required to correct this interference.
EVM WARNINGS AND RESTRICTIONS
It is important to operate this EVM within the input voltage range of 2.85 V to 5.5 V single supply and the
output voltage range of 0 V to 5.5 V.
Exceeding the specified input range may cause unexpected operation and/or irreversible damage to the EVM.
If there are questions concerning the input range, please contact a TI field representative prior to connecting
the input power.
Applying loads outside of the specified output range may result in unintended operation and/or possible
permanent damage to the EVM. Please consult the EVM User's Guide prior to connecting any load to the EVM
output. If there is uncertainty as to the load specification, please contact a TI field representative.
During normal operation, some circuit components may have case temperatures greater than +85＝C. The
EVM is designed to operate properly with certain components above +85＝C as long as the input and output
ranges are maintained. These components include but are not limited to linear regulators, switching transistors,
pass transistors, and current sense resistors. These types of devices can be identified using the EVM
schematic located in the EVM User's Guide. When placing measurement probes near these devices during
operation, please be aware that these devices may be very warm to the touch.
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2007, Texas Instruments Incorporated
Submit Documentation Feedback
21
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
THS7315D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
7315
THS7315DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
7315
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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